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The motivation for this work was to develop a srnall aperture, low profile.
dual frequency, dual polarized antenna suitable for portable and mobile satellite
teminals. The design was to incorporate the elevation angle of the satellite for
a given location such that, when in operation, the radiation apemire is
approximately in a horizontal position, while the pointing in the azimuth plane is
achieved mechanically. This thesis is an investigation of travelling-wave long-
slot array antennas wÏth an attempt to meet a set of predefhed specifications
and at the same time, to address these key features. As an effort to prove the
concepts, an experimental model was built and tested. The measured results
compared well with the theoretical calculations.
It is suggested that the work presented in the thesis has fulfilled these
requirements, with some limitations. Duting the progress of this work, however,
and because of the experience gained at UBC on the Advanced
Communications Technology Satellite (ACTS) project, the idea of an antenna
used in an inverted configuration was conceived in order to minirnize the effect
of attenuation due to wet antenna surfaces duhg rain events. This idea would
be applicable in both satellite teminals and Local: Multipoint Distribution System
(LMDS) applications.
University of British Columbia Etectn'cai and Cornputet EngÏneecïng
.. Abstract ................................................................................................................. II ... ................................................................................................. Table of Contents III
List of Tables ........................................................................................................ vi ** List of Figures ...................................................................................................... vii
Acknowledgements ............................................................................................... x
Chapter 1 Introduction ....................................................................................................... 1-1
1.1 Objective ................................................................................................. 1-2 ............................................................................ 1.2 Organization of Thesis 1-5
Chapter 2 Survey of Curren t Technologjc .......................................................................... 2-1
2.1 Suitable Antenna Types and Their Limitations ........................................ 2-2 2 1 1 Parabolic Reflector.. ......................................................................... 2-2
................................................................................... 2.1 -2 Lens Antenna 2-3 ..................................................................................... 2.1.3 Planar Array 2-3
2.2 Cunent Developments .......................................................................... 2 . 6 2.3 Çummaty ...................................................................................... 2 - 7
Chapter 3 Types of Travelling-Wave Slot Amy Antennas ................................................. 3-1
3.1 Single Frequency Horizontally Polarized Anays ...................................... 3-2 3.1 . 1 Longitudinal Slot Elements .............................................................. 3-3 3.1.2 Longitudinal Slat Anay ...................................................................... 3-8 3.1.3 ParaIlel-Plate Waveguide .................................................................. 3-9
3.2 Single Frequency Veitically Polarized Arrays ........................................ 3-10 3.2.1 Transverse Slot Elements ............................................................... 3-10 3.2.2 Transverse Sots Anay .................................................................... 3-13 3.2.3 ParalleGPlate Waveguide .............................................................. 3 - 1 7
3.3 Duaî Frequency Duaf Polarized Amys .................................................. 3-18 33.1 Coupling .......................................................................................... 3-20 3-32 ParaIlel-plate Waveguide ................................................................ 3-22
U ~ v e ~ t y of British Columbia
Chapfer 4 Feed Condderati'ons ......................................................................................... 4-1
4-1 Hom-Lens Combination ........................................................................... 4-2 4.1.1 Hom Design ............................................ .. ........................................ 4-5
.................................................................................... 4.1.2 Lens Design 4-11 4.1 -3 180' Bend ...................................................................................... 4-12
................. 4.1.4 Dual Polarked Dual Frequency Feed ................... ... 4-13 ..................................................................... 4.2 Resonant Slot Array Feed 4-14
....................................................................................... 4.2.1 Slot Types 4-15 ........................................................................... 4.2.2 Design Limitations 4-17
............................................................................... 4.3 Summary of Feeds 4-18
Chapter 5 A Design Example ............................................................................................. 5-1
....................................................................... 5.1 Specifications ..... ............ 5-2 ................................................................................ 5.2 Antenna Description 5-4 ................................................................................ 5.3 Operating Principles 5-7
5.4 ParalleCPlate Waveguide Structure ....................................................... 5-10 ............................................................................. 5.4 Radiating Slot Anays 5 1 1
5.6 Feed Stnicture ....................................................................................... 5-13 5.7 Design Summary .................................................................................. 5-14
Chapter 6 ................................................................................ Expenmental Investigation 6-1
6.1 Test Setup ................................................... ............................................ 6-2 6.1 -1 Aperture Measurement Setup ........................................................... 6-2 6-12 Voltage Standing Wave Ratio Measurement Setup .......................... 6-5
................................................................................ 6.1 -3 Antenna Range 6-7 ............................................................. 6.2 Hom-Lens Aperture Verifkation 6-10 ........................................................... 6.2.1 E-Sectoral Hom Verification 6-10
6.2.2 H-Sectorai Hom Verification ............................................................ 643 .............................................................................. 6.3 Bend Measurements 6-16
6.4 Radiation Testing of the individual Slot Anays ...................................... 6-17 ....................................................... 6.4.1 Longitudinal Slot Array Testing 6-18 ......................................................... 6.4.2 Transverse Slot Anay Testing 6-21
6.5 Integration Test ...................................................................................... 6.24 65.1 Voltage Standing wave Ratio Measurement ..............o................... 6-24 6.5.2 Gain ............................................................................................... 6-25 6.5.3 Radiation Patterns .......................................................................... 6-27 . . 6.5.3.1 Longrtudinal Slot Anay* ............................................................. 6-27
............................................................... 6-5-32 Transverse Slot Anay 6-31 653.3 Beam Sqtiint ............................~................................................ 6-35
6.5.4 Cornpliance with Onginal Specifications ............................................. 6-36
University of British Columbia
Chapter 7 .......................................................... ....... Inverted Configuration ................... 7-1
7.1 Concept ................................................................................................. 7-2 C * .................................................................. 7.2 Experimental Investigation 7 . 5
...................................................................................... 7.2.1 Experiment I 7-5 7.2.1 Experiment Il ..................................................................................... 7-6
7.3 Summaiy .................... .. ..................................................................... 7-12
Chapter 8 Conclusions ................................................................................................... 8 - 1
8.1 Antenna Concept and Design ................................................................ 8-2 8.1.1 Basic Array Design ........................................................................ 8 - 2 8.1.2 Parallebplate Design .................................................................... 8 - 3 8.1.3 Feed Design ..~..~................................................................................ 8-3
.............................. 8.1 -4 lnverted Configuration ................................ 8 4 8.2 Recomrnendations for Further Work ........................................................ 8 4
8.2.1 Resonant Slot Waveguide Feed ......~................................................ 8-5 8.2.2 Alternative Dielectric Materials ......................................................... 8-5 8.2.3 Application to Circular Polarkation .................................................... 8-6
8.3 Summary ................................................................................. ........ 8-7
Append PdSo on Append Elemen Append Elemen Append Append Append Append Append Append Append Append Append
ix A: The Effect of the Slot Length. I . and the Phase Coefficient Ratio .................................................. the Various Radiation Characte ristics A 4
k 6: The Phase Coefficient in a Longitudinally Slotted Waveguide .............................................................................................................. 6-1 k C: The Attenuation Coefficient in a Longitudinally Slotted Waveguide ............................................................................................................. C-1 x D: The Effect of Attenuation on Sidelobe LeveC ................................. D-1 x E: The Effect of Amplitude Taper on Sidelobe Level ....................... €4 x F: Radiation Patterns of Longitudinally Slotted Waveguides ............. F-1 x G: Calculations for Coupling of Slots ................................................. G-1 x H: Design of Comigated Hom ....................................................... H - 1 x I: Design of Lens Profile ..................................................................... 1-1 x J: Matching of the Lens ................................................................~..... J-1 x K: Mathcad Prograrn for Design of Bendç ......................................... K-1 X t: Mounting Configuration of Antenna Range .................................... L-1
University of British Cofumbia B&cd and Computer Engineering
LIST OF TABLES
I
Table 1-1 Antenna Specifications ...................................................................... 1-3
Table 2.2.1 Summary of Current Antenna Types .............................................. 2-9
.................................................... Table 5.1 -1 Antenna Example Specifications 5-3 Table 5.44 Design of Parallel-Plate Waveguide Structure ............................. 5-11 Table 5.5-1 Design of Horizontally Polanzed Array ......................................... 5-12
............................................. Table 5.5-2 Design of Vertically Polarked Array 5-12 Table 5.6-1 Design of Longitudinal Slot Array Hom-lens Feed ................... .... 5-13 Table 5.6.2 Design of Transverse Slot Anay Hom-lens Feed ......................... 5-14
.................................................................... Table 5.6.3 Design of 180' Bend 5-14
Table 6 1 -1 Equipment List for Aperture Measurement ..................................... 6-3 Table 6.1.2 Equipment List for Retum Loss Measurernent ..........................Cc... 6-5 Table 6.1.3 Equipment List for Antenna Range ................................................ 6-8 Table 6.3-1 VSWR Measurements for the E-sectoral Horn-lens Structure with and without the Bend ..............................................................~........................ 6-16 Table 6.54 Voltage Standing Wave Ratio of Antenna .................................... 6-25 Table 6.5.2 Gain of Antenna ................................................................... 6 - 2 6 Table 6.503 Loss in the Antenna ..................................................................... 6-26
......................... Table 6.54 Theoretical and Measured Values of Look Angle 6-35 Table 6.55 Cornpliance Table .....................................................................*.. 6-36
Electricai and Computer Engineering
LIST OF FIGURES
............................................................ Figure 3.1 -1 Longitudinal Slot Elements 3.6 Figure 3.1.2 Surface Currents on a TEl mode ParalleCplate Waveguide ......... 3-7 Figure 3.1.3 Longitudinal Slot Anay ............................................................... 3 . 8
........................................................... Figure 3.2-1 Transverse Slot Elements 3-11 Figure 3.2.2 Surface Currents on a TM1 mode ParalleCplate Waveguide ....... 3-12 Figure 3.293 Transverse Slot Anay ............................................................... 3-15 Figure 3.24 Element Pattern of a Bachard Radiation Transverse Slot ......... 3-16
........................................... Figure 3.3-1 Multi-pipe Model for a Wire Junction 3-21 Figure 3.3.2 Side-loading of ParaIlel-plate Waveguide ................................... 3-24 Figure 3-39 Centre-loading of ParaIlel-plate Waveguide ...................... 3-24
........................................................... Figure 4.1.1 Horn-Lens Feed Structure 4.3 Figure 4.1-2 Radiating Aperture with Horn-Lens Feed Structure Located
B ~ O W ......................................................................................................... 4-4 ........................................................................... Figure 4.1.3 H-Sectoral Hom 4-7
Figure 4.1 4 E-Sectoral Hom ............................................................................ 4-9 Figure 4.105 Diffraction .................................................................................... 4-10 Figure 4.24 Resonant Slot Amy Feed Structures ......................................... 4-16
.................................. Figure 5.2.1 Dual-Frequency DuaCPoiarkatÏon Antenna 5-5 Figure 5.2-2 Dual-Frequency DuaCPolarization Antenna without 1 80' Bends .. 5-6
................................................................ Figure 5.34 High Level Diagram 5 - 9
Figure 6.1.1 Aperture Measuring Setup ..................e......................................... 6-4 Figure 6.1.2 Voltage Standing Wave Ratio Measurement Setup ................... ... 6-6 Figure 6.1 -3 Outdoor Antenna Range ............................................................... 6-9 Figure 6.2-1 Amplitude Measurement for the E-Sectoral Hom Aperture ......... 6-11 Figure 6.292 Phase Measurements for the ESectoraI Hom Aperture ............. 6-12 Figure 6.2.3 Amplitude Measurements for the H-Sectoral Hom Aperture ....... 6-14 Figure 6-24 Phase Measurements for the HSectoraI Hom Aperture ............. 6-1 5
University of Brïcisfr Cohmbta Eiectricd and Cornputer Engineering
Figure 6.4-1 E-plane Experimental Radiation Pattem for the Longitudinal Slot Anay ................................................................................................. 6 - 7 9
Figure 6.4-2 H-plane Experimental Radiation Pattern for the Longitudinal Slot Anay ........................................................................................................ 6-20
Figure 6.4-3 E-plane Experirnental Radiation Pattem for the Transverse Slot A m y ........................................................................................................ 6-22
Figure 6.44 H-plane Experimental Radiation Pattem for the Transverse Slot Anay ........................................................................................................ 6-23
Figure 6.5-1 E-plane Experimental Radiation Pattem for the Longitudinal Slot Anay ........................................................................................................ 6-28
Figure 6.5-2 H-plane Experimental Radiation Pattem for the Longitudinal Slot Array ........................................................................................................ 6-29
Figure 6.5-3 Cross-polar Experimental Radiation Pattem for the Longitudinal Slot Amy ........................................................................................................ 6-30
Figure 6.5-4 E-plane Experimental Radiation Pattem for the Transverse Sot Amy ........................................................................................................ 6-32
Figure 6.5-5 H-plane Experimental Radiation Pattem for the Transverse Slot Anay ........................................................................................................ 6.33
Figure 6.5-6 Cross-polar Experimental Radiation Pattem for the Transverse Slot Array ........................................................................................................ 6-34
................................................... ....... Figure 7.1 -1 lnverted Configuration ... 7-3 Figure 7.102 Slotted Waveguide Anay in lnverted Configuration ................... ... 7-4 Figure 7.24 Experimental Setup with Simu tated Rain Source .....................*.ce. 7-7 Figure 7.2-2 Photograph of Exparimental Setup with Simulated Rain ............... 7-8 Figure 7.203 Sample Attenuation Data for Experiment I ($ = 327 ................... -7-9 Figure 7.24 Sarnple Attenuation Data for ExperÏment I (O, = 320) ................. 7-10
................. Figure 7.2.5 Sample Attenuation Data for Expenment 11 (0, = 60') 7-11
Figure A-1 Radiation Patterns of a 5 L and a 1 O A, Array ................................ A-3 Figure A-2 Angle of Maximum Radiation with Bo as Parameter ......................... A-4
................................................... Figure A-3 Directivity with j3$po as Parameter A-5 Figure A-4 Beamwidth versus P$Bo with Length as Parameter ......................... A.6 Figure A-5 Sidelobe Level for a Unifom Slot Relative to Main Beam Versus Slot
Length with Bo as Parameter .................................................................... A-7
............................. Figure B-1 Theoretical and Experimental Phase Coefficient 8-3
..................... Figure G1 Theoretical and Experimental Attenuation Coetficient Ce3
Figure D-1 Radiation Pattem with Attenuation Coefficient as Parameter ...................................................................................................... 0-2
Figure E-t Radiaüon Pattern with Attanuation Coefficient as Parameter ...................................................................................................... €2
Uectn*ui and Cornputer Engineering
Figure F-1 Comparison of Measured and Predicted Radiation Pattems ...................... at 29.6 GHz for /= 100 mm and w = 0.46 mm ..........,.............. F-2
Figure F-2 Cornparison of Measured and Predicted Radiation Pattems ............................................... at 29.6 GHz for 1= 100 mm and w= 0.89 mm F-3
Figure G-1 Calculated Couplhg Vs Attenuation Coefficient with Slot .............................. Width as Parameter ............,.......,,................................... (3-7
............................................................................ Figure 1-1 Ray Theory of Lens 1-2 ................................................................. Figure 1-2 Collimating Action of Lens 1-4
..................................................... Figure J-1 Matching of Lens by Peiturbation J-3
.......................................... Figure K-1 Retum Loss for the TM wave Bend K - 6
................................................................. Figure L-1 Mounting Configuration I L-1 ................................................................ Figure L-2 Mounting Configuration II L-2
University of British Columbia ElectncaI and Cornputer E%@neerÏng
INVESTIGATION OF TRAVELLING-WAVE MM-WAW ARRAY ANTENNAS
l am very grateful to many people who helped contribute to this thesis. I
cannot Say enough about the enthusiasm of individuals who dedicated so much
of their time and effort to help with this research. Most of al[. 1 would like to thank
my thesis supervisor, Dr. M.M.2 Kharadly, whose guidance, encouragement,
support and patience helped me throughout the research and wnting. I also want
to thank the excellent staff at the university for their skill, diligence and
professionalisrn, with special appreciation to David Fletcher, Anthony Leugner,
Donald Dawson, Leif Kiolby, and David Chu Chong for their support with
machining parts, purchasing of materials, and setting up the antenna range.
Finally, I would like to thank my husband, Chris Chong, to whom I owe a
considerable debt for helping me with radiation measurernents.
Unlvemmty of British Columbia Electrîcd and Cornputer Engineering
Chapter I
Cunently, the frequency spectrum is congested and satellite
communication systems are venturing into higher and higher frequencies. Since
the announcement of the allocation of Ka-band for sateNte services at the 1971
World Administrative Radio Conference (WARC-71), over a dozen geostationary
and non-geostationary systems have been filed with the Federal
Communications Commission (FCC) 111. With these new Ka-band systems, the
developers are faced with a new realm of technology involving srnalter
components, higher tosses, greater min fades, and overall, more uncertainties.
One particular area of interest is the antenna system ni a Ka-band satellite
terminal. With the potential applications in personal cornrnunications systems
involvhg mobile and portable teminafs 121 131, the requirements for small
Ka-band antennas are pushing the lirnits of the existing technology. There is an
increasing demand for small, low profile, power efficient antennas with a high
gain and a specific radiation pattern to accommodate the elevation angle of the
satellite in applications such as in [4] [5] and [6J. With the present day
technology, there are several antenna types that are suitable for satellite
communication in the Ka band m, but for mobile and portable applications, the
requirements have extended beyond that of the electrical performance. The
physical and mechanical aspects are becoming equally important. This thesis is
an investigation of travelling-wave long-slot array antennas that offer a
combination of some of the key features. In the following sections, the design
objectives are described and the organization of the thesis work is presented.
The objective of this thesis is tu investigate a class of antennas known as
travelling-wave long-slot arrays for use in direct reception and ! or transmission
for personal communications, parüculariy in mobile and portable terminal
applications. A set of specifications for a typical Ka-band mobile teminal based
on [61 is given in Table 1-1. Alüiough the specificatims wilI differ €rom system to
system, the antenna design concepts should be flexible enough such that it is
not Iimited to a parüctilar system.
University of 6-h Columbia ElecWxi[ and Cornputer Engineering
Receive Frequency Band (GHz) Transmit Receive
Polarization Transmit Receive
Sidelo be levels 6. Cross-polarkation
Efficiencv 1 VSWR 1
Horizontal Vertical
> 50% 1.51 maximum
Aside from the electrical specifications, other desirable features are:
1) Low profile - A low profile is important for portable terminais such as
[5] in order to facilitate storage; likewise, in a mobile terminal [8], not to
affect the aerodynamics of the ve hicle.
2) Tilted main beam - It is convenient to have the main beam of the
antenna directed at the satellite when the antenna is at a leveled
position as in (31 [6] and 181. Since the satellite look angle is usually at
an elevation angle other than go0, most anays with the main beam on
their central axis must be tilted in order to align the beam with the
satellite, defeating the purpose of a low profile antenna. By havïng
the main beam tiIted off the main ais, a low profile can be maintained.
University of British Columbia Efectn-cal and Cornputer Engineering
3) Cornbined transmit and receive apertures - AIViough most array
antennas require separate anays for the transmission and reception, it
would be desirable to have the anays share a cornmon aperture to
conserve space and for aesthetics as in [8] and [91.
4) Minimal performance degradation in precipitation - It has been found
that various antennas, based on their configuration. have different
seventy of degradation in rain and snow [l O]. One type of antenna, an
array with a tilted main beam, when not needed in a low profile
configuration, can be used ni an inverted configuration to prevent the
wetting of aperture and hence minimize the performance degradation
[ I l ] [12]. This is a desirable attribute for Ka-band systems where the
rain attenuation allowance in the Iink margin can have a significant
cost impact on the terminal design.
5) Low cost - A major determining factor, aside from performance, for the
best type of antenna is often the cost. A suitable terminal antenna
should be low cost such that it is affordable to the consumer.
This work is an attempt to meet the above specifications and desired attributes
through an investigation of travelling-wave long-dot a m y antennas. The main
contributions of this work are the cornbining of two long-dot arrays into a single
apeiture for transmitting and receiving ai dual frequencies and dual poiarizations
and the possibility of ushg the antenna effectively in rain.
University of British Columbia EIectn*caE and Cornputer Engineering
Chapter 1 -fntroduciion
The thesis is divided into eight chapters. A review of cunent mm-wave
antenna technology, and a survey and anaIysis of existing antenna designs with
respect to the demands and constraints of a typical Ka-band satellite terminal
are given in Chapter 2. Vanous types of antennas (including parabolic dishes,
lenses, microstrip patch arrays, and resonant-dot arrays), and latest antenna
developments for terminal applications are discussed.
Chapter 3 gives the theory of travelling-wave long-slot array antennas and
discusses the advantages of these antennas over those in Chapter 2. A high-
level description of the th ree types of t ravelling-wave long-slot arrays is
presentad, together with a qualitative explanation of how each type can achieve
the desired characteristics.
The different feed options for travelling-wave long-slot arrays are
discussed in Chapter 4. The design details for the two types of feeds, namely
the hom-lens structure and the resonant slotted waveguide feed, are compared.
Chapter 5 gkes a design example of a dual-frequency, duai-polarized
anay antenna. kt gives the specifcations of the array and the design details for
the dot elements, the paraIlel-plate waveguide and the feed structure.
Chapter 6 deals with the ïmplementation and the measurements of an
University of 6- h Colurnbla UectricaE and Cornputer Engineerhg
Chapter 1 -Introduction
experimental model based on the design given in Chapter 5. A list of the test
setups, the various steps of implementation, sub-system testing and final test
resultç of the integrated antenna are given.
Chapter 7 is dedicated to the investigation of using the antenna in an
inverted configuration. Experimental results are given for various inverted angles
under simulated-rain conditions.
Discussion and conclusions are given in Chapter 8. Discrepancies
behnreen theoretical predictions and experimental results are discussed and
recommendations for further studies are provided.
University of British Columbia UectrîcaC and Cornputer Engineering
References
[l] William W. Wu, "Satellite Communication," Proceedings of the IEEE, Vol. 85, No. 6, June 1997, pp. 998-1 01 0.
[2] T. Lentsch et al., "Pico-Terminal-A portable Ka-band system," Ka Band Utilization Conference, Florence, Italy, Sept. 24-26, 1 996, pp. 121 -1 28.
[3] H. Wakana. et al., "COMETS Experiments for Ka-band and Millimeter-wave Advanced Mobile Satellite Communications", IEE 1998 International Conference on Universal Personal Communications, pp. 1-6.
[4] Raquet, C., et al.: Ka-Band MMlC Arrays for ACTS Aero Terminal Experiment. Presented at the 43rd Congres of the International Astronautical Federation. Aug. 28 to Sept. 5,1992.
[5] J. L. Fikart, "RF Front End for a 20130 GHz briefcase terminal," Ka Band Utilization Conference, Florence, Italy, Sept. 24-26, 1996, pp. 141-148.
[6] A. Densmore et al., "K- and Ka-band Mobile-Vehicular Satellite-Tracking Reflector Antenna System for the NASA ACTS Mobile Terminal", Proceedings of the Thkd lntemational Mobile Satellite Conference, Pasadena, California, June 16-1 8,1993, pp. 563-568.
m F. K. Schwering, "Millimeter Wave Antennas", Proceedings of the IEEE, Vol. 80, No. 1, January 1992, pp. 92-1 02.
[81 A. Tulintseff, "An Active KJKa-Band Antenna Amy for the NASA ACTS mobile Terminalu, Proceedings of the Third International Mobile Satellite Conference, Pasadena, Califo mia, June 1 6-1 8,1993.
[91 C. Pike, 'tEHF Planar Anay Measurements". Communications Research Centre, Communications Canada.
[IO] M. Kharadly and R. Ross, ' Performance of Soma Conventional Ka-band Antennas in (Shulated) Rain : A P-2000 Millennium Con ference on Antennas & Propagation, Davos, Switzerland, 9-14 Apn'l, 2000, P0472.
[I I] 1 M. M. 2. Kharadly and A. Y. Chan, uMm-wave Antenna Anays with Minimal Degradation of Performance in Precipitation", 21" ESTEC Anfenna Workshop on A m y Antenna Technoiogy, ESTEC, Noordwijk, The Netheriands, May 64,1998.
University of British Columbia EiectricaI and Cornputer Engineering
[12] M. Kharadly and A. Chan, "A Mm-Wave Antenna with 'Non-Degradable' Performance in Rain, * A P-2WO Miilemium Conference on Antennas & Propagation, Davos, Switzeriand, 9-14 April, 2000. P0375.
Unketswof British Columbla Electrlca[ and Cornputer Engineering
Chapter 2
To date, there are various types of antennas available for Ka-band
applications. It Ïs useful to survey these antennas with respect to their performance.
ability for duabband operation, mechanics, cost, and suitability for mobile and
portable satellite terminal applications. The set of specifications given in Table 1-1
is used as a guideline for comparing the various types of antennas. The desirable
attributes described ni Chapter 1 must also be considered. In the first section of this
chapter, the various types of antenna technologies available on the market for Ka-
band terminal applications are brieffy discussed. Then, an investigation of the latest
emerghg antenna technologies is preçented, where representative examples of
antenna systems are described. Finalfy, a summary of cornparisons between
previously discussed anteanas is provided in ternis of the advantages and
disadvantages for Ka-band teminal applications.
Chapter 2 - S u ~ e y of Cunent Techndogy
2.1 SOITABLE ANTENNA TYPES AND THEIR L~MITAT~ONS
With the present day technology, there are several antenna types that are
suitable for satellite communication in the Ka Band. Each antenna type has its
unique characteristics that !end itself well to a particular application. However, not
many of these address the specific requirements and desirable features of a mobile
or portable terminal. Several antenna types, including the parabolic reflector, the
horn-lens antenna and the planar anay, are described briefly to provide general
background information.
At the present time, the most congenial antenna for very srnall aperture
temiinal (VSAT) applications is still the parabolic reffector. Because of its wide-band
characteristics, usually the same reflector can be used for both the up-link and
downlink. A parabolic antenna is usually specified in ternis of its focal length to
diameter ratio (Kt?). Larger Vd ratios usually result in better performance; the trade-
offs are the aesthetics of a more cumbersome system, and more importantly, the
additional rigidity required for the boom to support the transceiver on the end. In
general. parabolic dishes are too bulky to be considered for mobile temiinals.
Chapter 2 - b e y of Cunent Technology
Another possible type of antenna is the hom-lens antenna, which is widely
used in the Ka band for Local Multi-point Distribution System (LMDS) applications.
Hom-lens antennas do not suffer from aperture blockage as in the case of reflector
antennas; thus better radiation pattern characteristics are to be expected. They are
however bulkier and have slightly higher loss than reflector antennas. Mechanical
tolerances for lens antennas are not as critical as for parabolic dishes, but the bulk
of lens material, and hence its cost, usually increase rapidly with the size of the
antenna. This antenna is generally considered to be too bulky for mobiIe or portable
applications.
A planar anay consists of arrays of radiating elernents excited with
predetemiined amplitudes and phases to achieve a specific pattern. Its planar
characteristics lend itself well to mobile and portable applications. In general,
separate antennas are required for the transmit and receive functions because of
the limited bandwidth associated with most array type antennas. Potential problems
hcfude poor efficiency and bearn squht, which are highly dependent on the feed
netwok Two common types of planar arrays are discussed below.
2.1.3J Common Types of Arrays
UNWERSEWOF ~RITIsH COWMBIA 2-3
Chapier 2 - Sunrey of Cment Technofogy
Two common types of array antennas are microstrip patch anays and slotted
wâveguide anays. They are similar in that they share similar advantages and
disadvantages of planar anays. Their primary dwerences are in the type of feed
network and the type of radiating elements. A microstrip anay consists of patches,
dots or other types of microstrip elements, fed by a microstrip network, while a
siotted waveguide array may have either resonant or leaky-wave slots. which are fed
by a waveguide network. These are discussed in some detail below.
2.1.3.1 .1 Miciostrip A m y
Microstrip arrays have the advantage of conformability, low cost, light weight.
and facility of utilizhg a prÎnted architecture. There is a lot of flexibility in the design
of the feed network and radiating elements, hence a tilt in the main beam can be
easily incorporated in most cases. Another advantage of microstrip arrays is that
both the transml and the receive radiating elements and feed networks can share
the same aperture to form a single antenna. This works well in dual polanzation
systems where the interaction between the transmit and receive elements is
minimized. Two examples of this are delineated in the next section. In addition,
active elements such as high power amplifies (HPAs) or low noise amplifien
(LNAs) can be directly combined with the printed circuit to obtain an integrated
antenna system.
At first glance, microstrip arrays appear to be suitable candidates but they
have an inherent disadvantage: poor efficiency. For high gain antennas. as the
number of elements in the army hcreases, the losses Ri the feed network increase
Chapter 2 - Survey of Cunent Technology
proportionally. The efficiency of a high-gain microstrip antenna anay in the Ka band
is generally below 50%. With the current technology, it is difficult, if not impossible
to meet the gain requirements for the present application with the given aperture
size.
2.1 5.1 -2 Slotted Waveguide Array
There are two types of slotted waveguide arrays. The best known of the
slotted waveguide anays consists of a planar anay of resonant slots fed by a series
of waveguides [Il [2]. The feed network is usually centre-fed using slot-type
aperture couplers. For large arrays, the feed network can be divided into several
subanays to reduce beam squint. Generally. the main beam is located directly on
the central axis, hence the planar structure must be üIted to achieve the desired
elevation angle. Slotted waveguide arrays are less complicated to design than the
microstrip anays and their losses are significantly lower; the typicat antenna
efficiency is about 80% [2]. The main disadvantage of resonant slotted waveguide
anays is that the waveguide feed network and the dot elements are expansive to
manufacture since precision rnilng is required. Furthemore, with resonant slot
anays, it is difficult to combine two arrays into a single aperture.
The other type of slotted waveguide anays consists of travelling-wave slots.
These slots and feed networks are simpler and requke less precision. In addition,
these anays lend themselves well ta combining two perpendicular arrays into a
common aperture. The slots can be resonant [31 or non-resonant. The latter is the
topic of the thesis and will be covered in detail in the following chapters.
Within the last few years, considerable attention has centred on the use of
small Ka-band antennas for satellite terminal applications (41 [5]. Of the many
important advances made, a few are mentioned briefly here as representative
examples. Jet Propulsion Laboratory (JPL) has developed a mobile vehicular
satellite-tracking parabolic reflector antenna system to be on top of a vehicle Ri
NASA's ACTS mobile terminal experiments [6]. The antenna system is an elliptical
parabolic dish positioned on a rotary platfom that mechanically tracks the satellite in
the azimuthal direction. In the elevation direction, the main beam is tilted 46Ooff the
main axis and is fan-shaped such that no tracking is needed up to +/- 6" Although
the antenna meets the performance specifications of the terminal, the bulkiness is
nevertheless an issue,
Anolher example is a low profile planar microstrip antenna developed by
Communications Research Centre for use in a portable suitcase satellite teminal
application for the Olym pus satellite in the 28/ 1 9 GHz band m. It uses interleaving
transmit and receive sub-anays of microstrip quarter-wave stub elements. Although
the antenna achieved its initial obiective of combinhg two perpendicularly polarized
arrays into a cornmon aperture, it has not overcome the inherent loss problems
assaciated with microstrip arrays.
JPL is working on an active K- and Ka-band low profile antenna which is a
Chopter 2 - Surdey of Current Techndogy
rnulti-layered assem bly interieaving a receive anay of radiating dots and a transmit
array of microstrip dipoles such that they share a cornmon apeiture [8]. Unlike the
CRC antenna anay, JPL reduces the cornplexity of the divider network by
integrating the low noise amplifiers (LNAs) and high power amplifiers (HPAs) directly
inio the sub-arrays. The problem, however, is maintaining the tracking of the
individual active devices such that the sub-arrays are excited predictably. It is found
that, given the curent state of K- and Ka-band active devices, the amplitudes and
phases do not necessarily change the same amount from device to device over tima
and temperature. The resuft is a degradation of the radiation pattern and gain.
Of the profusion of antenna types and designs existing today, none really
satisfies al1 the demands of the Ka-band teminal specifications in Table 1-1, as well
as the desired features for a mobile and portable terminal. Reflector and horn-lens
type antennas are too bulky compared to the convenient low profile antennas
envisioned by system designers. Flat planar arrays are suitable for portable and
mobile applications. but they , too, have their drawbacks. Microstrip anays. despite
their design flexibility, are too lossy at high frequencies. The waveguide array of
resonant slots is capable of meeting the specifications, but the feed network as well
as the resonant slot radiating elements themselves are bandwidth limiteci and
difficult to manufacture. In addition, the resonant slot array approach requires two
separate antennas tu perfom the transmit and receive functions. Table 2.2-1
surnmarÏzes the advantages and disadvantages of these various antenna types.
Chapter 2 - Swey of Cunent Technology
Travelling-wave slot anayç are promising since they are less sensitive to tolerances,
and are suited tu hivo amys sharîng a common aperture. This is the topic of the
next chapter.
ADVANTAGES
most widely used only one antenna needed for transmit and receive highefficiency(60%-70%)
good antenna pattern low tolerances
- - - -
a compact and can be made conformal lightweight low manufacturing costs both transmit and receive arrays can share same apeiture ability to incorporate active devices
high effkiency(80 %) low profile
only one antenna needed for transmit and receive compact and can be made conformai
* light weight low manufacturing costs both transmit and receive arrays share same apecture
bulky, not low profile
bulky lens material can be expensive for large antennas
poor efficiency (40%) difticult to track active devices over time and temperature
beam squint costiy two antennas are required for transmit and receive bulky
poor efkiency (c 50')
Chapter 2 - Swey of Cunent Techndo~y
ANTENNA TYPE JPCs arrays compact and can be made
conformal iight weight low manufacturing costs both transmit and receive arrays share same aperture ability to incorporate active devices
DISADVANTAGES difficuit to track active devices over time and temperature
Table 2.2-1 Summary of Current Antenna Types
Chapter 2 - Survey of Curent Technology
References
[1] J. Hirokawa et al., "A Low-profile single-layer, leaky-wave slotted waveg uide anay for mobile DBS Receptionn, Antennas and Propagation, International Symposium Digest, 1993, pp. 132-1 35.
[2] H. P. Muhs, "Mm-Wave Antenna", Microwave Journal, Vol. 28, No.7, July 1985. Cover Story.
[3] J. Hirokawa, "A Single-Layer Slotted Leaky Waveguide Anay Antenna for Mobile Reception of Direct Broadcast from Satellite", IEEE Trans. On Vehicular Tech. Vol. 46, N0.4, Novernber, 1995, pp. 749-755.
[4] M-Takahashi, "A Slot Design for Uniform Aperture Field Distribution in Single- Layered Radial Line Slot Anay", l€€€ Tms. On Anfennas and Propagation. Vol. 39, No.7, July, 1991, pp. 954-959.
[a C. Raquet, et al.: Ka-Band MMlC Amys for ACTS Aero Terminal Experïrnent, Presented at the 43rd Congress of the International AstronauticaI Federation. Aug. 28 to Sept. 5, 1992.
[6] A. Densrnoie et al., "K- and Ka-band Mobile-Vehicular Satellite-Tracking Reflector Antenna S ystern for the NASA ACTS Mo bile Terminal : Proceedings of the Third International Mobile Sa telfie Con fer ence, Pasadena, Califamia. June 16-18,1993, pp. 563-568.
m C. Pike, '€HF Planar Array Measurements". Communications Research Centre, Communications Canada,
[8] A. Tulintseff, "An Active WKa-Band Antenna Anay for the NASA ACTS mobile Terminal", Proceedings of the Third Intemaüonal Mobile Satellie Conference, Pasadena, California, June 16-1 8, 1993.
Chapter 3
Travelling-wave long-dot arrays have been used for decades [Il [2] [3J [4] but
with the increasing damand for antennas in the mm-wave or higher frequency
spectrums, they are fînding their way into new applications. Unlike traditional
resonant dots used in slotted waveguides [51[6] described in the fast chapter, these
anays use long slots which have wider bandwidths and are easier to manufacture
due to more relaxed tolerances. They can be made relatively compact and since
they are basicalfy a waveguide structure, they are relatively power efficient
compared to microstrip patch arrays.
In general, travelling-wave long-dot arrays are composed of long slots that
radiate as the wave propagates. tesulting in a main beam that is tilted off the centrai
axis m. Depending on the propagation coefficient of the wave in the transmission
medium, the main beam c m be designed for a specik look angle. Although the
Chapter 3 - Types of Tlilvelîing-Slot Amy Anterinas
transmission medium can be of varioos m e s [8] Ic)], for the scope of the thesis, only
dots on a parallei-plate waveguide will be investigated.
Travelling-wave long-slot anay antennas can be divided into three types:
single frequency horizontal polarization, single frequency veitical polarization and
single or dual frequency and dual polarization. The first uses longitudinal dot
elements where the slot length is dong the direction of propagation; the second
uses transverse slot elements where the slot length is perpendicular to the direction
of propagation and the last uses a combination of the two.
Because travelling-wave slotted waveguide amays have an inherent tilt in the
main beam, they are particulariy suited for use in satellite terminal applications. In
its usual configuration, such an array can be designed for the look angle of the
satellite and thus achieve a relatively low profile. When used in an inverted
configuration [l O], the radiating aperture can be shielded from precipitation hence
improving its performance in rain and snow.
This chapter gives the analysis for the three types of arrays in ternis of the
radiating elements, the anay and the parallel-plate waveguide design.
Horizontally polarked arrays are longitudinal slot arrays with the slot length
along the direction of propagation. The longitudinal dots pmduce a main beam in
Vie direction of propagation in the 0 plane while the array is used to achieve
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Chapter 3 -Types of Tmvelling-Slot Amy Antennas
directivity in bruadside direction of the 4 plane. This is iilustrated in Figure 3.1-1.
The dots can be excited by intermpting the surface currents of one of the TE,,
modes in the paraltel-plate waveguide, where n 2 1. This is shown in Figure 3.1-2
for the TE1 mode. The following sections describe the theory for the longitudinal slot
elements. the array, and the paraltel-plate waveguide for a single frequency
horîrontally poladzed slot anay.
A longitudinal dot in a waveguide is a member of a broadband family of
travelling-wave antennas that radiates as the wave propagates down the
transmission guide resulting in a main beam tilted off the central a i s in the direction
of propagation. This family traces its roots to the days of amateur radio where a
simple piece of wire was stretched out between two poles for reception [Il]. Since
then, a similar concept has been applied to waveguides where Goldstone and Oliner
[12] introduced the longitudinally slotted rectangular waveguide. Shiiar to its wire
counterpart, such a structure has an inherent advantage of easily incorporating a tilt
angle to the main beam.
3.1.1.1 Radiation Pattern
The theoiy of the longitudinal slot is essenüal the same as that of the
fong itudinal ly slotted rectangular waveguide in [12] 1131. Furihemo re, the theoiy is
applicable regardkss of which of the TEn modes is used to excite the dots. To
University of BntTsh Columbia Electn*caI and Cornputer Engineering
Chapter 3 -Types of TravellingSlot Amy Antennas
elucidate the factors that affect the radiation pattern of a leaky-wave slot, one must
examine the equation that approximatas the far-zone E-field for an infinitely namw
slot in a Ieaky-wave rectangufar waveguide given by [14]:
sin( Bo l( cos O - cos O,, )) E@ = 2 E,, sine ,
j?,l( cos e - cos O,, )
where Bg is the phase coefficient of the wave propagating in the waveguide and Bo is
that of free space. There are essentially two design parameters used to control the
radiation pattern: the slot length, I, and the ratio of which can be adjusted
either by vaiying the dimensions of the waveguide or by dielectric loading. These
parameters have an impact on the directivity, the angle of maximum radiation, the
beamwidth and the sidelobe levels, See Appendix A. As 1 increases, the angle of
maximum radiation approaches go. For applications where i > > h, 80 is
synonyrnous with the angle of maximum radiation. Note that because p9 < Bo, 9, is
within the Iimits of o0 and 90' for horizontally polarïzed dot arrays.
3.1.1.2 Attenuation Coefficient
In principle, the dot is infhitely thin. In practice, however, the widlh of the slot
also influences the design in that it aiters the propagaüon coefficient, and
specifically the affenuaüon coefficient, a, of the waveguide. (See Appendix B and C
for theoretid and experhental resuks.) This attenuation, which cm be shown to
University of Briaih Caîumbia 3-4 EtectrÎÎiai and Computer Engineering
Chapter 3 -Types of Travelling-Slot Anay Aotennas
be proportional to the slot width, w, has several important consequences. First.
attenuation along the waveguide further compounds the complexity of the design,
since the field strength in the slot aperture is no longer constant along its length. but
rather experiences an exponential decay in the direction of propagation. This
inherent exponential taper generally degrades the sidelobe performance of the slot
anay. Appendix D shows how the sidelobe leveis increase with an increase in a.
Secondly, the power efficiency of the anay, q, is given by the following equation:
-d ' q = l - ( e )*. (3.1 -3)
By increasing w and thus a, the power efficiency can be improved. But conversely,
for a given slot width, W. there is a point where the field contribution from an
additional increase in length, I, becomes negligible. Hence the aperture efficiency of
the dot array decreases with an increase in W. Finally, any residual power at the
end of the slot that is not absorbed will form a reflected wave to generate a sidelobe
at 180' - a. Thus the optimal design of w is a balance between the sidelobe
performance and the aperture efficiency of the anay.
Variation of the slot width along the dot length introduces another degree of
freedom. This additional design parameter can be varied to achieve a taper in the
amplitude excitation of the slot for the purpose of improving the poor sidelobe
performance caused by the inherent exponential taper. Since the amplitude of
excitation varies proporüonally with the width of the slot. the slot can assume various
shapes to achieve the desired taper. The computed patterns for a 30 Â. anay for
some common amplitude tapen are given in Appendix E. Cornparisons of predicted
and measured radiation patterns of long dots are given in Appendix F.
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Chapter 3 -Types of Travefllng-Slot Amy Antennas
- TOPVIEW
/ Slots
Pg Direction of b
Propagation
Figure 3.1-1 Longitudinal Slot Elements
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, Slots
I I
Currents
TE1 mode direction of propagation
O Electrical Field Lines
Magnetic Field Lines
Figure 3.1-2 Sunace Currents on a TE? moâe ParalIeGplate Waveguide
University of British Columbia UectriCai and Cornputer Engineering
The longitudinal dot anay can be described by the following equation:
where T, is the excitation of the Ah element in the array. This is shown in Figure 3.1-
3. The tapers in Appendbc E are also applicable here. Since the slot anay elements
are excited in phase, the radiation is broadside if the spacing between the elements,
6, is limited to less than M2.
O Direction of
O Propagation
Figure 3.1-3 Longitudinal Slot Array
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ln the last section, the tilt of the main beam has been shown to be dependent
on f)dpo, hence the phase coefficient of the paraIlel-plate waveguide for the desired
TEn mode is an important part of the design. The phase coefficient, p,, of a parallel-
plate waveguide propagating a TEn mode is given by [15]:
fl, = J d j i ~ , , ~ , - ( n x / a ) ' , (3.1 -5)
where ais the separation of the parallel-plates. Theoretically the appropriate pg can
be selected such that e0 can take on values between 0' to 90*. In practice, however,
this range is limited. For instance, for small values of p$po, the tolerance in a is
more critical, thus smaller values of eo rnay be restricted by the manufacturnig
tolerances. Similarly, for larger values of a more cornpiex design rnay be
required to suppress higher order modes since the value of a may have to be rather
large to obtain large pg. For most instances, 20' c go c 70' is the recommended
range for practical design. Furthemore, unless it is restricted by the application, the
lowest order mode, the TE1 mode. is recommended to minimize the Iikelihood of
exciüng other undesired modes. Although the theory is given for a homogenous
waveguide, the general theoiy applies to a partially loaded waveguide as well.
Equaüon (3.14) just needs to be modified to refled the desired loading
configuration.
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Chapter 3 - Types of TliiveJlingSlot Anay Antannas
3.2 SINGLE FREQUENCY VERTICALLY POLARIZED ARRAYS
Vertically polarized anays consist of transverse dot elements whose Iengths
are perpendicular to the direction of propagation. The elements of the anay yield a
main beam in the broadside direction of the plane. Here, the travelling-wave effect
is used to achieve directivity in the e plane. This is shown in Figure 3.2-1. Unlike
horkontally polarked arrays, the anay is discrete along the direction of propagation
as shown in Figure 3.2-2 for the TM1 mode. It will be shown that this gives the
advantage of having a wider choice of eo than its horizontalty polarized counterparts,
which are Iimited to O < 00 c 180'. These dots can be excited by intenupting the
surface currents of any one of the TMn modes in the paraltekplate waveguide, where
n 20.
Since a transverse slot is an aperture radiator whose electrical field has a
uniform phase along the length of the slot, the following equation can be used for
the radiation pattern of the slots in the plane:
where T(yl is the taper applied to the excitation. Similar to the longitudinal dot, it
can be shown that the amplitude of excitation is proportional to the width of the dot,
W. provided w « Equation (3.24) yietds a broadside radiation in the 6 plane.
The various types of tapers given in Appendk E are afso applicable.
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Chapter 3 -Types of Travelling-Slot Amy Antemas
TOP VIEW
SlDE VIEW
Slots
Ps Direction of F
Propagation
Figure 3.24 Transverse Slot Elements
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Para! le!-plate Waveguide
TM1 mode direction of propagation
O Magnetic Field Lines
Figure 3.2-2 Surface Curnrnts on a TM1 mode Parallekplate Waveguide
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Chapter 3 -Types of TraveiHng-Sfot Amy Antannas
-
Transverse slot arrays have been used 1161 [l fl for end-fire radiation. but in this
application, the slot anay must be excited in a manner to yield a main beam in the
desired direction in the e plane. Because the array is discrete, go can be designed
for angles between 0' and 180'.
3.2.2.1 Radiation Pattern
The equation for the array pattern is given by [18]:
where 5 is the progressive phase shift between the elernents, d is the spacing
between elements and T;. is the excitation of the hh elernent. This is shown in Figure
3.2-3. Various tapers are given in Appandix E. Since the array is in the direction of
propagation,
5 = B#. (3.2-3)
In order to achieve a tilted main beam in the 0, direction, the atray excitation must
have a progressive phase shift detemined by:
flOdcos(0,)= j9,d + ha, (3.24)
where d is the spacing of the anay and rn is an integer. Since d must be kept s M2
so that only one main lobe is generated, it can be show that m 5 0. Note that for m
Universîtyof British Columbia 3-1 3 ffectricaî and Cornputer Eitgineeri'ng
Chapter 3 -Types of Traveiiïng-Slot Amy Antennas
= O, Equation (3.2-4) becomes Equation (3.1 -2) and the main beam would be in the
direction of propagation. As for backwards radiation where 90' < 0 c 180~. cos (a)
is negative, and therefore, m must be negative and it can be show that P, s Bo
shce d must be kept I hd2. This implies a loaded waveguide configuration.
3.2.2.2 Attenuation Coeff Ment
Similar to the longitudinally slotted arrays, the attenuation coefficient, a, will
have the sarne effects on the array pattern. Fust, the exponential taper generally
degrades the sidelobe performance of the slot array. Secondly, the aperture
efficiency of the slot anay decreases while the power effciency increases with an
increase in W. Any residual power at the end of the dot anay that is not absorbed
will fom a reflected wave to generate a sidelobe at 180' - go. An addition to the last
point is that, for the cases with backward radiation, the width of the slot fuither
enhances the magnitude of the sidelobe caused by the refiected wave. This is
because for p, > Po, transverse dot elements are inherently end-fire radiators in the
0 plane, and thus if the slot width is not kept relatively small compared to the
waveguide wavelength, na the element pattern would assume an end-fire pattern
which would degrade the array pattern. The followhg equation gives the radiation
pattern for a transverse dot in the 0 plane [tg]:
University of British Columbia Efecîrid and Cornputer Engineering
Chapier 3 -TF of T~vellingSlot Anay Antenna~
The plot in Figure 3.24 shows that as the slot width. w, increases, the element
pattern becomes less omni-directional and more directive in the end-fire direction.
Since the array pattern is Ri the backward direction. an increase in w will result in a
decrease in the directivity of the main lobe but increase in the directivity of the
sidelobe fomed by the reflected wave. This imposes a lirnit in the design such that
the power efficiency of the antenna can only be increased at the expense of higher
sidelobe ievels. Equation (3.1-3) is applicable for calculating the power efficiency.
d Direction of Propagation
O bd 2P,d 3P,d
Figure 3.2-3 Transverse Slot Array
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Chapter 3 -Types of TravellingSlot Amy Antennas
hglc 0 (Di?grrt) - w - 0 5 waveguide wavelengths - - w - 1.5 waveguide wavelengihs - w - 10 waveguide waveIengths
Figure 3.2-4 Element Pattern of a Backward Radiation Transverse Slot
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Chapter 3 -Types of Travelling-Slot Anay Antennas
Depending on the value of m in Equation (3.2-4), the design of the parallet-
plate waveguide is divided into two cases: m = 0, and m < O. In the first case. and
simplest, it is basically the same as the longitudinally slotted anay where Equation
(3.2-4) is reduced to:
Here, pg c Pa hence this would imply that only TMn modes with n 1 are applicable.
The phase coefficient, Pg, of a paraltekplate waveguide propagating a TMn is given
by [15]:
8, = J o z p ~ U ~ r - ( n a / u ) 2 , (3.2-7)
where a is the separation of the parallei-plates. Again 20' < go c 70' is the
recommended range for practical design for the same reasons as for the
longitudinally slotted array. Furthemore, unless it is restricted by the application,
the lowest order mode, the TM1 mode, is recommended to minimize the likelihood of
excitnig other undesired modes. This is applicable to fully or partially loaded
waveguides as long as 8, c po.
In the second case, m c O. thus p, r Bo. implying that the parallekplate
waveguide is either loaded or a is >> k such that = Bo and &, = 180~. Here, the
mode cm be TMn with n ;r 0. where TMo is the TEM mode. For a fully loaded
University of British Columbia 3-1 7 Eiectrid and Cornputer Engineering
Chapter 3 - Types of TraveUing-Slot Array Antennas
waveguide, the phase coefficient pg is given by (1 51:
where 0 is the relative pemittivity of the dielectric in the waveguide. This is
applicable to fully or partially loaded waveguides as long as pg 2 Bo.
3.3 DUAL FREQUENCY DUAL POLARlZED ARRAYS
By combining a horizontally polarized array with a vertically polarized array
onto a common aperture, the result is a dual polarized array. Since each array is
excited by a different mode within the parallel-plate waveguide, different frequencies
can be used. If both anays are used at the same frequency, one anay can be
phased at 90' to the other to give circular polarization. Looking in the direction of
propagation, if the vertïcally polarized array is leading in phase relative to the
horizontally polarized wave, the result is left hand polarization. If it is lagging, right
hand polarization results. Regardless of circular or linear polarization, in general,
most applications would require the beams of both anays to be aligned in space.
The main beam of the horizontally polarïzed array is always in the direction of
propagation. That of the vertically polarized array, however, can be in the forward
direction or the backward direction. The result is two possible configurations for the
parallei-plate waveguide excitation: the waves in the waveg uide for both modes
propagaüng in the same direction, or in opposite directions. For the design of the
first configuration, the waveguide must be fed from the same end to generate the
corresponding TE and f M modes at the respective frequencies. A load is placeci at
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Chapter 3 -Types of Travelling-Slot Amy Anter
the other end to absorb the residual power not radiated at the end of the travelling-
wave anays. The second configuration requires the paraltekplate waveguide to be
fed from opposite ends; isolators are required at the feeds to provide loads for
residual power. These isolators must be positioned at the feed before the wave
reaches the cutoff regions, if there are any, in order to provide effective loads.
The selection between the hnro configurations will depend on the frequencies
of operation, the feed mechanism for generating the desired modes, the bandwidth
of the application, and most importantiy, the feasibility of simultaneously achieving
the appropriate Pdpo ratios to obtain the desired look angle. Secondary issues
include econornics such as the costs of loads versus isolators or the use of one
complex feed versus two simple feeds.
The element and anay design of a duai frequency dual polarked array is
essentially the same as that of the individual arrays. However, there are two
aspects that increase the complexity of the design. The first is that the coupling
between the arrays must be nivestigated. Coupling degrades the antenna
performance in ternis of cross-polarkation, and isolation between the transml and
teceive ports. For a circulariy poiarïzed array, the axial ratio is degraded. The
second area is the design of the parallei-plate structure. In single frequency and
single polarkation designs, the waveguide can be air-filled or totally filfed with
dielectric. In some cases, in order to achieve the appropriate $$Bo ratio for two
modes and N o frequencies, parüally filled waveguides are required. These topics
are covered in the folfowing sections.
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Chapter 3 - Types of Travelling-Slot A m y Antennas
The coupling of the slots can be investigated by reducing the problern to
examining the coupling of a single cross junction of slots. For the ease of
calculation, wires are used to replace the dots by the duality theorem [20]. By using
the Method of Moments [2t], each wire is divided into segments, also known as
pipes, along its length. Several pipes pIaced side by side approximate the width of
the wire. This configuration, known as the multi-pipe model is shown Figure 3.3-1.
The ends of the wires must be 2 M4 away from the cross-junction to ensure that the
end effects are negligible. A two-dimensional mutual impedance matrix is fomed
for the segments. To obtain the coupling, the excitation is applied to one wire and
the current matrix is obtained for all the segments. The ratio of the cunents
summed up for the excited wire to that of the other wire is the coupling. The Method
of Moments program and the results are shown in Appendix G. It was found that
the coupling increases with the width of the slot. Furthemore, asymmetttcal
excitation across the junction also hcreases the coupling. In a typical sIot anay,
there are usualty two causes of asymmetrical excitation. First, any taper added to
the sbt for reducing sidelobe levels would introduce asymmetry. Secondly, since
the dot is radiating as the wave propagates, the exponential decay of the excitation
wouM also add to this. Consequently, the design of a duaI frequency dual polarized
anay must consist of tradeoffs between power efficiency, sidelobe levels, and cross-
polarizaüon performance.
University of British Columbia 3-20 Eiectnkaf and Cornputer Erigineerlng
Pipe Wire
/ Junction
Figure 3.34 Multi-pipe Modal for a Wim Junction
University of Britrs h CoCumbia EImcai anci Cornputer Engineering
Chapter 3 -Types of Travelling-Slot Anay Antennas
One of the most difficult facets of the dual frequency dual polarkation array
design is to obtain the appropriate P$po ratios simultaneously for both operating
frequencies and waveguide modes in order to collocate the main beams in space.
To meet this constraint, the parallei-plate waveguide can be loaded with dielectric
slabs. First, in order to study the effects of the vanous parameters associated with
dielectric loading, the transverse resonance technique is used to reduce the problem
to that of a system of equations to find the propagation constants. Secondly. a
study of the attenuation in the parallei-plate waveguide structure must be conducted
to detemine the losses before the slats are incorporated. This will niclude both the
dielectric and conductor tosses. FÏnally, to complete the analysis, the limitations of
this approach are identified.
3.3.2.1 Propagation Coeff Ment
For an air-filled waveguide, if a horizontally polarized array and a vertically
polarized anay of diÎerent frequencies were to be cornbined, it is unlikely that the
$deo ratios would yield the same tilt in the main beam. If one were to start loading
the wavaguide parüally wiai dielectric, both ~ s / @ ratks would increase. Dependhg
on the loading configuration, the p$po ratio of one mode would nicrease more
University of Bntlsh Columbia 3-22 Eiectrical and Cornputer Engineering
Chapter 3 - Types of Traveiüng-Stot Amy Antannas
significantly than that of the other. The two loading configurations, side loading and
centre loading, are shown in Figures 3.3-2 and 3.3-3 respectively. In Figure 3.3-2,
the side-loading configuration. the dielectric is along the plates where the electric
field is relatively weak for a TE mode but concentrated for the TM mode.
Consequently, the p$po for the TM mode will increase more than the TE mode with
this loading configuration. In Figure 3.3-3, the centre-loading configuration, the
dielectric is positioned for maximum influence for the TE mode and minimal
influence for the TM mode hence the pdpo ratio for the TE mode will increase more
than the TM mode.
The eigenvalue equation for the inhomogeneous waveguide, as shown in
Figure 3.3-2, can be derived by using the transverse resonance technique [22]. It is
given by Equations (3.3-l ), and (3.3-2) for the TE mode. The variables ka and kd
represent the transverse propagation coefficients in air and in the dielectric regions
of the waveguide, respectively. Solving for these equations simultaneously, given
the thickness of the dielectric Risert, 2 will yield p,, the phase coeficient in the
waveguide and ad, the attenuation coefficient due to the dielectric losses.
Similady, Equations (3.303)~ and (3.3-4) give the eigenvalue equation for the
inhomogeneous parallel-plate waveguide propagating the TM mode.
University of British Columbia 3-23 Uectnd and Cornputer Engineering
Chapter 3 - Types of Travelling-Sfot Amy Antennas
Figure 3.3-2 Side-loading of ParaIlel-plate Waveguide
Figure 3.33 Centre-loading of Parallet-plate Waveguide
University of British CMumbia Bactrical and Cornputer Englieering
Chapter 3 -Types of Travefling-Siot Amy Antennas
The eigenvalue equation for the inhomogeneous waveguide as shown in
Figure (3.3-3) is given by Equations (3.3-5), and (3.3-6) for the TE mode. Solving
these equations simultaneously, given the thickness of the dielectric insert, t, will
yield pg, the phase coefficient in the waveguide and m, the attenuation coefficient
due to the dielectric tosses.
Similady, Equations (3.3-7), and (3.3-8) give the eigenvalue equation for the
inhomogeneous parallel-plate waveguide propagating the TM mode.
in both sets of eigenvalue equations, there are Rifinite solutions. The first solution is
for the first mode and the nth solution is for the nth mode,
University of British Columbia Uecfn'cal and Cornputer Engineering
Chapter 3 -Types of Traveiling-Slot Amy Antennas
To minimize losses at high frequencies, low-loss matedals with tight
permittivity tolerances mus€ be used. These requirements preclude the PTFMbre
glas substrates commonly used in lower frequency applications. Possible
candidates include polystyrene (0 = 2.54), polyethylene (a = 2.25) and Teflon (er =
2-08).
3.3.2.2 Attenuation Coefficient
The attenuation of the parallel-plate waveguide, neglecting radiation losses,
can be decomposed into that contributed by the dielectric and that by conducting
plates. The fîrst iç calculated using the transverse resonance technique. As for the
conductor losses, they are different for the TE and TM modes due to the different
surface cuvent configurations. From Pozar [5], Equation (3.3-9) gives the
attenuation due to two lossy conductors for the TE mode and Equation (3.3-10)
gives that for the TM mode. where kc is the cut-off wave number.
The surface resistance of the conductor, R, is given by:
University of BritTsh Columbia 3-26 BecMcai and Cornputer Engineering
where CJ is the conductivity of the plates, Le., a = 5.813 X lo7 Slm for copper.
The total losses due to the conductor and dielectric are then given by:
3.2.2.3 Limitations
The above methods of Ioading a parallet-plate waveguide to achieve the
desired ratios simultaneously for N o different modes and frequencies have
their limitations. First of all, it is possible that for certain frequencies, no suitabfe
configuration will yield a feasble design. Furthemore, the range limitation of 20°c
go c ?O0 for single frequency arrays still applies. For smaller angles, higher order
modes may be excited. As for larger angles, the loadhg must be of materials with
higher pemitfivities. Cunentty, such matenals have high tosses hence they rnay not
bs suitable for this design.
University of B&h Columbia E!ectricaL and Cornputer Engineering
Chapter 3 - Types of TcavdIing-Slot Anay Antmnaç
References
[t] A. Palumbo and S. Cosentino, 'Circularly Polarized L-band Planar Anay for Aeronautical Satellite Use*, European Microwave Conference, September, 1 969, pp. 22-1 -22-1 5
[2] J. Hirokawa and al., "Waveguide-Fed Parallel Plate Slot Anay Antenna," IEEE Transactions on Antennas and Propagation, Febniary 1 992, pp. 21 8-223.
[3] R. C. Honey, "A Flushed-Mounted Leaky-Wave Antenna with Predictable Patterns", IRE Transactions on Antennas and Propagation, October 1959, pp. 320- 328.
[4] E. D. Sharp, and E. M. T. Jones, "An Antenna Anay of Longludinally-Slotted Dielectric-Loaded Waveguides," IRE Transactions on Antennas and Propagation, March, 1962, pp. 1 79-1 87.
[5] A. F. Stevenson, Theory of Sots in Rectangular Waveguide," Journal of Applied Physics, Vol. 19, 1 948, pp. 24-38.
[6] H. P. Muhs, 'Mm-Wave Antenna". Microwave Journal, Vol. 28, No.7, July 1 985. Cover Stoiy.
m E.C. Jordan and K.G. Balmain, EIectromagnetic Waves and Radiating Systems, 2nd ed., Prentice Hall Inc., New Jersey, 1968.
[8] G.W. Slade et ai., &A Study of Slotline Leaky-Wave Antennas," IEEE Transactions on Antennas and Propagation, March IWO, pp. 41 1-41 4.
(9) F. L. Whetten and C. A. Balanis, 'Meanderhg Long Sot Leaky-Wave Waveguide Antennas", E€€ Transactfons on Antennas and Propagation, November 1991, pp. 1553-1 559.
[IO] M.M.Z. Kharadly, and A. Chan, 'New Antenna Concept for Efficient Ka-band Termina[ Operations in Rain." Fourol Ka-Band Utiliatatio Conference Proceedîngs, November 2 - 4. Venice, ItaIy, IW8, pp. 223-230.
[t II H. Jasik, Radio Engineemg Handbook, McGmw-Hill Book Company, New York. 1961,1& Ediüon,
02) L.O. Goldstone and A. A. Oliner, 'Leaky-Wave Antennas I: Rectangular Waveguidesn, IRE Transact8otw on Anfennas and Propagaüon, Oct. 1959, pp. 307- 31 9-
University of Bcitish Columbia 3-28 Uectn*cat and amputer Engineering
[13] J. N. Hines et al.. Traveiiing-Wave Slot Antenna<, Proceedings of the IRE, Nov. 1953, pp. 1624-1 631.
[ i 4 C.H. Walter, Travelling Wave Antennas, McGraw-Hill Book Company, New York, 1965 pp. 22.
(1 5) D.M. Pozar, Micruwave Engineering, Addison-Wesley Publishing Company, New York, 1990, pp. 144.
[161 J. D. Kraus. Antennas, McGraw-Hill Book Company, New York, 1988.
[ln R.C.Honey, "A Flush-mounted Leaky-wave Antenna with Predictable Patternsn, IRE Transactions on Antennas and Propagation, Oct. 1959, pp.320-329.
[18] D.K. Cheng , Fundamentals in Engineerfng Electromagnetics, Addison-Wesley Publishing Company, New York, 1993.
[19] C.A.Balanis, Antenna Theory, Design and Analysis, John Wiley & Sons, New York, 1982.
[20] R. €.Collin, Foundations for Microwave Engineering, McGraw-Hill Book Company, New York, 1966.
[211 R.F. Hamngton, Field Computatfon by Moment Methods, Macmillan, New York, 1968.
1221 R. E. Collin, M d Theory of Guided Waves, IEEE Press. New York, 1991.
Uniuersity of British Columbia Eiectricai and Cornputer Engineering
Chapter 4
An important part of any antenna array design is the feed structure. There
are several desirable characteristics one should consider in selecting a feed.
First, since the feed delivers power to the array elernents, in order to minimize
losses ni the array, the feed shouId be power efficient. Secondly, the feed
assembly contributes to the amplitude and phase excitation of the anays thus
flexibility to incorporate tapering is afso important Thirdly, the feed network
should have a wide enough bandwiâth such that it does not inhibit the
performance of the amy. FRially, the ease of manufacturing must be taken into
account such that the feed is suited for large-scak production.
A parallei-plate waveguide can be excited by various methods. Examples
of these include a hom-Lens combination, and a resonant dot. The hom-lens
combination is a continuous type of excitation while the resonant dot array
Chapter 4 -Feed Considerations
is a discrete form of excitation. One may be more suitable than the other
depending on the application.
A broadband feed can be Rnplernented using a hom-lens combination. A
typical feed structure consists of a horn, a lens, and a 180' bend for locating the
feed beneath the radiating aperture. This is shown in Figure 4.1-1. The hom-
lens assembly has two features that make it ideal for this application. First, this
combination can accommodate wider bandwidths than a distributed feed.
Second, the hom and lens combine to give a cosine-squared taper to the
amplitude distribution in the parallekplate waveguide. The latter feature serves
to reduce leakage from the sides of the finite parallei-plate waveguide as well as
to improve the sidelobe performance of the array.
In order to reduce the buikiness of the hm-lens combination, a 180' bend
can be used to locate the hom-lens assembly below the radiating structure. The
configuration of the feed anangement for a longitudinaf dot anay equipped with
the bend is show in Figure 4.1-2. The horn, the lens and the 180' bend are
dlscussed in the following sections.
University of British Columbia ElectrM and Cornputer Engineering
C hapter 4 -Feed Conside rations
i to bend
Figure 4.14 Horn-Lens Feed Shycture
University of British Columbia Electricaf and Cornputer Engineering
Chapter 4 -Fe& Considerations
Parailel-Plate Waveguide
1 m
t. u
Flange Horn Lens \ / 180 degree bend
Figure 4.14 Radiating Aperture with Hom-Lens Feed Structure Locatad Beiow
€iectricaI and Cornputer Engineering
Chapter 4 -Feed Considerations
4.1.1 HORN DESIGN
The design of the horn is similar to the design of a horn radiator [Il [21
consisting of the input into the horn, the flare angle. and the length of the hom.
Since this antenna is intended for mm-wave frequencies, a rectangular or square
waveguide feed is used as opposed to a probe feed to reduce loss. The flare
angle and the length can be selected together to give a gradua1 transition. The
larger the fiare angle, the shorter the horn but at the expense of a more abrupt
transition at the throat region which can potentially degrade the return loss. In
the design of the flare, especially if a large angle is required, one must ensure
that the wave emerging from the horn has minimal higher order components.
These higher order modes can result in a non-unifom amplitude distribution with
fluctuations in phase.
The mouth of the hom should be designed to be approximately the size of
the parailel-plate waveguide in order to couple the fields efficiently. The fields at
the horn aperture should be matched to that of the desired mode in the paraIlel-
plate waveguide. An H-sectoral hom can be used €0 obtain a TM mode of the
parallel-plate waveguide and simiîariy an E-sectoral horn is used to obtain a TE
mode. These two types of horn are discussed with respect to their aperture
distributions.
UniVersÏty of Briüsh Columbia EIectrical and Cornputer Engineering
Chapter 4 -Feed Considerations
4.1.1.1 H-sectoral Horn
An H-sectoral horn is shown in Figure 4-19. The field distribution of an H-
sectoral horn aperture propagating the TEM mode has a cosine amplitude taper
and a cylindrical phase front. The E-field is given by the equation below [3]:
The above equation can be analysed in ternis of the amplitude and phase
components. The cosine amplitude taper across the horn aperture has two
effects on the excitation of the parallel-plate waveguide. First, just as a H-
sectoral hom radiator has good sidelobe performance in the E-plane due to this
taper, the excitation to the array will also be tapered in the same way thus
reducing sidelobe levels in the E-plane pattern of the array. Second, since the
parallel-plate waveguide is finite in width, the cosine taper reduces the fields at
the edges of the parallel-plate waveguide hence reducing leakage. The cosine
taper allows metal walls be placed at aie edges of the parallet-plate waveguide if
needed tu fomi a broad rectangular waveguide to provide electrornagnetic
s hielding.
As for the phase component, Equation (4.1-1) shows a cylinddcal phase
front. This is because the wave along the edges of the horn has travelled a
longer distance cornpared to the wave in the middk of the hom. However, in
order to excite the parallel-plate waveguide in the desired TM mode, a unifom
University of British Cottimbk 4 4 Bectriai and Cornputer Engineering
Chapter 4 -Feed Consideratrons
phase front is required. To correct for the cylindncal phase fronts. a dielectric
Iens 1s used at the hom aperture. This is discussed in Section 4.1.2.
Figure 4.1-3 H-Sectoral Horn
University of British Columbia ffectrical and Computer Engineering
Chapter 4 -Feed Considerations
4.1 .i .2 E-sectoral Horn
An E-sectoral hom is show in Figure 4.1-4. Theoretically, the following
equation gives the approximate aperture distribution [3]:
However, this equation is not accurate enough because for an E-sectoral hom,
the E-field is perpendicular to the sides of the hom. As show in Figure 4.1-5,
this results in diffraction and may lead to higher order modes within the parallel-
plate waveguide. In order to reduce this effect, a comigated hom is used [4].
This comigated surface is designed such that a capacitive surface reactance is
presented to the wave, which forces the E-field to zero at the walls. The design
of the comgated surface based on [4] is given in Appendix H. The resulting
aperture has an additional cosine taper which can be given by the following
equation [5]:
Similar to the H-sectoral hom, this cosine taper will reduce the sidelobe level in
the anay as well as reducing leakage at tfw sides of the waveguide. A lens can
correct the cylindricat phase cornpanent of Equation (4. t .3).
University of British Columbia EIectn'cai and Cornputer Engineering
Chapter 4 -Fe& Considerations
Univetsityof British Columbia
Figure 4.14 E-Sectoral Horn
ElectrW and Cornputer Engineering
Chapter 4 -Fe& Consideliftions
Diffraction '/
Figure 4-16 Difhaction
Unlvemity of British Columbia 4-1 0 Bectrical and Cornputer Engineering
Chapter 4 +eed Considerations
AIhough various types of lenses can be used to obtain a uniform phase
front in a horn, for the scope of the thesis. only dielectric lenseç will be covered
since i€ is more suited to mm-wave frequencies. The dielectric lem design is
based on geornetric optics also known as ray theory [6]. The basic function of
the lens is to equalise the paths of the wave such that a unifonn phase front is
achieved at the aperture of the hom-lens structure. The details of detemining
the lens profile are given in Appendix 1. Note that the design of a lens for an H-
sectoral hom is different from that of an E-sectoral horn. This is because in the
first case, h, z h, but it is not so in the latter.
The performance of a lens-corrected hom. in general. is degraded by
reflections from the two air-dielectric boundaiy surfaces. This applies to virtually
al1 lens antennas, resulting in loss of power. Hence a matching section can be
used m [81. One method that is suitabie for high frequencies is to comgate the
lens [9] [IO]. The theory for a comgated lens is given in Appendix J. By
applying this matching, diffraction and higher order modes can be minimized.
The consequence of htroducing the lens is the addition of an amplitude
taper in the wave emerging from the feed assembly. The taper across the
aperture c m be readily calculated and is given by [I II:
Unkeirsity of British Columbia
Chapter 4 -Fe& Considerations
with 0 as shown in Fjgure 4.1-5. It can be shown that the lower the pemittivity,
the steeper the taper. This near-cosine taper, combined with the taper in the
anay is advantageous Ri obtaining lower sidelobe levels.
The theory and computational methods for cafculating waveguide bends
have been studied by many but most solutions are either rigorous and difficult to
solve or inaccurate. Weisshaar introduced a computationally efficient method in
[12]. The modal solutions for the bend are derived by means of the Method of
Moments. The modes found in a straight waveguide section are used to
constnict the modal expansion in the curved region. Mode matching is then used
to obtain the scattering matrix of a single bend discontinuity. The generalized
scatterhg mat& technique [13] is used to obtain the scattering parameters for
cascading two bends with an optional straight section in between. The
computations are given in Appendk K.
University of British Columbia 4-1 2 UectricaI and Cornputer Engineering
Chapter 4 4 e e d Considerations
One advantage of the hom-lens combination is that it is suited for
excitation of the parallei-plate in two polarizations. By using an orthornode
transducer at the input to the hom, two modes of €wo different frequencies can
propagate down the hom. The design of the lens, however, is more cornplex.
Examining Equation (1-2) in Appendix 1, the design of a lens rnay be
unique for a particular frequency and mode. Two different frequency and mode
lenses will not necessary have the same profile. In order to design a lens for
dual frequency and dual polarization, one can either find a dielectrïc the
penittivity of which has the appropriate function with respect to frequency and
polarization, or a loading configuration where the lens only partially fills the height
of the hom. Since the first may not necessary exists, the second, although
bandwidth Iimited, is more suitable.
From Equation (1-2), if the same profile must be used for the two waves,
then the following must be true:
where subscripts TE and TM correspond to that of the TE and TM waves,
respectively. Here, the subscript d iç used to indicate the region of the lens
whether it is fully or partialiy loaded. If WhTE r kdhm for a lens which
completely fiils the height of the hom. then the centre-loading configuration
should be used. Otherwise, side loading should be used. Ushg Equations (3.3-
Univers'w of British Columbia 4-1 3 Etectcical and Cornputer Engineering
Chapter 4 -Fe& Considerations
1) to (3.3-8), by vaiying the thickness, t, and the pemittiity of the dieleetrie, E , it
is possible to satisfy Equation (4.1-6). Note that this partial loading method is
lirnited to frequencies that are not too far apaR The further the frequencies are
separated, the higher the pemittivity, and thus the increased likelihood for
higher-order modes.
Another method of exciting the paralleCplate waveguide in the desireci mode
is to use a resonant dot array in a rectangular waveguide. The resonant dot
array is a more compact structure than the hom-lens combination discussed in
Section 4.1. Its major advantage is aiat it can be used to achieve specific
excitations. Although the resonant slot array feed requires l e s material for
construction, the tolerances may require milling which can increase the cost for
rnanufacturing. Finally, the resonant dot array feed is bandwidth limited due to
its resonant chaiacteristics. The following sections describe the theory for a
resonant dot array.
University of British Columbia Elecîrkai and Cornputer Engineering
Chapter 4 -Fe4 Considerations
In order to excite the desired mode, the slots must be positioned such that the
fields across the slot match up with the fields in the parallet-plate waveguide.
E.g., to excite a TE mode in the parallel-plate waveguide, the resonant slot can
be either offset-transverse broadwall. centied-inclined sidewall, or centred-
inclined broadwall since only these slots generate an E-field which is parallel to
the TE mode in parallel-plate waveguide. Similady to excite a TM mode, only the
longitudinal broadwall slot can be used. These configurations aie shown in
Figure 4.2-1 . Aside from matching the field orientation, it is important to match up the
phasing. For instance, for a TM, or equivalently, a TEM mode. the wave must
assume a unifom phase front along the parallet-plate waveguide. For the
IongitudinaCbroadwaII dot, the dots spaced at A& must be staggered as shown
in Figure 4.2-1. For a TE, mode, in the case of centred-inclined sidewall slots,
u2 spacing can be used if adjacent dots are inclined Ri the opposite direction as
shown in figure 4.2-1. For centred-inclhed broadwall slots, they must be spaced
at A, apart and likewîse offset transverse broadwall dots since &/2 spacing
yields 180' phasing.
University of British Columbia Eiectrical and Cornputer Engineering
Chapter 4 -Feed Considerations
Offset Transverse Broadwa II
Centred Inclined Broadwail
Figure 4.2-1 Resonant Slot Anay Feed Structures
University of British Columbia Elmcal and Cornputer Engineering
Chaptet 4 -Fe& Considerations
For higher order modes, it is possible to use more than one resonant army.
An array of resonant slot arrays can be used to excite the appropriate phasing for
higher order modes in the parallel-plate waveguide. In addition, since each slot
elernent can be RidMdualIy adjusted for excitation, the wave entering the parallel-
plate waveguide can be designed with a specific taper for lowering sidelobe
levels. These are advantages that the resonant dot feed structure has ove? the
hom-lens structure.
Due to the use of resonant stots, this type of feed is not as wide band as the
hom-lens combination. Experimentation shows that typical bandwidth of a
resonant slot is approximately 5%. The bandwidth of the anay decreases with
increasing size. At the edges of the band, the incremental phasing between the
elements will cause the slot array radiation pattern to be skewed to one side.
One method to achieve more bandwidth is to excite the resonant slot anay from
the centre. However, for large anays, the incremental phasing will cause a nuIl
in the centre of the beam at the band edges. Furthemore, this type of feed is not
suited for dual frequency operation hence the leaky-wave dot anay design must
be restricted to the configuration where the paraIlel-plate structure is fed from
opposite ends.
University of British Columbia Bectri& and Cornputer Engineering
Chapter 4 -Feed Considerations
The following summarizes the advantages and disadvantages of the hom-
Lens combination and the resonant dot array feeds:
Advantages of Horn-Lens Combination:
1) Wide bandwidth
2) Less stringent tolerances required for manufacturing
3) Suited for dual-frequency and duai-polarized design
Disadvantages of HomLens Combination:
1) More expensive since more material is required and the dielectric Ri the lens
can be costly
2) Bulky, a bend is required to make a more compact solution
3) Limited to cosine taper excitation
4) More diff~cult to excite non-fundamental modes in the paralle[-plate
waveguide
Advantages of Resonant Slot Array:
1) Compact
2) Suitable for various types of excitations
3) Less expensive due to less matefial required
Unhersity of British Columbia 4-1 8 Eiecûi-cat and Cornputer Engineecing
Chapter 4 -Fe& Considerations
Disadvantages of Resonant Slot Airay:
1) Limited Bandwidth
2) Tighter tolerances required
3) Suited for single frequency and single mode excitation
The hom-lens combination has an inherently wider bandwidth and less
stringent tolerances. It is amenable to the dual-frequency, duai-polarized
operation since it allows for the excitation of two waves from a common feed.
The hom-lem combination has an inherent cosine-squared taper in the
amplitude of excitation. Its size makes it preferable to use a 180' bend such that
it can be located below the parallekplate waveguide.
The resonant slot array is a more compact structure. Its major advantage
is that 1 can be used to achieve specific excitations, Although the resonant slot
array feed requires less material for construction, the tolerances may require
milling which can increase the cost for manufacturing. Finally, the resonant slot
array feed is bandwidth limited due to its resonant charactenstics. The following
sections describe the design details for both types of feeds.
Unkersity of British Columbia Electcïcat and Cornputer Engineering
Chapter 4 -Feed Considerations
References
[Il S. A. Schelkunoff and H.T. Friis, Antennas: Theory and Practice, John Wiley & Sons, Inc., New York, 1952.
[2] L. J. Chu and W. L. Barrow, "Electromagnetic Hom Design", Transactions - Electn'cal Engineetiing, Vol. 58, July 1939, pp. 333-338.
[3] C. A. Balanis, Antenna Theory AnaIysis and Design, John Wiley & Sons Inc., New York, 1982.
[4] A. D. Olver, "Corrugated Homs",tectroniics & Communication Ehgineering Journal, Feburaiy, 1992, pp. 4-1 0.
[5] C. A. Mentzer, and L. Peters, Jr., 'Properties of Cutoff Comigated Surfaces for Comigated Hom Design", IEEE Transactions on Antennas and Propagation, Vol. AP-22, No. 2, March 1974, pp. 191 -1 96.
[6] H. Jasik,, Antenna Engineering Handbook, McGraw-Hill Book Company, New York, 1961.
m E. M. T. Jones et al., "Measure Performance of Matched Dielectric Lenses", IRE Transactions on Antennas and Propagation, January, 1 956, pp. 31 -33.
[8] E. M. T. Jones and S. B. Cohn, "Surface matching of Dielectric Lenses", Journal o f Applied Physii, Volume 26, Number 4. Aprii, 1953, pp. 452-457.
[9] T. Monta and S. B. Cohn, "Microwave Lens Matching by Simulated Quarter- Wave Transfomiers", IRE Transactions on Antennas and Propagation, Januaty, 1956, pp. 33-39.
[l O] R. E. Collin and J. Brown, TThe Design of Quaiter-wave Matching Layers for Dielectric Surfaces, Proceediigs of instifute of Electn*cal Engiieers, vol. 103, part C, September, 1955, pp. 153-158.
[I 11 J. A. Cummins, Side Lobe Reducttion the Radiation Field of Lens Comted H-Plane Homs, Master's Thesis, Laval University, August 1960.
El21 A- Weisshaar et al.. "A Rigomus and Efficient Method of Moments Solution for Cwed Waueguide Bends*, IEEE Transacffins on Microwave Tneoiy and TechnrQues, Vol. 40, No. 12, December, 1992-
[t 3) T. Itoh, eeneraf Scattering Mat& Technique", Nurnerhil Techniques for Microwave and Millrinefre-Wave Passke Stmcfures, New York: Wifey, 1989.
Univetsityof Briüsh Columbia 4-20 Uectncaî and Cornputer Engineering
Chapter 5
In order to perfom an experimental investigation of the theory presented
in Chapters 3 and 4, a design example, based on a set of predefined
specifications, has been selected to be built and test&. The proposed antenna
is a dual frequency dual polarized anay antenna since 1 covers both horizontally
and vertically polarized arrays. The feed configuration is one where the parailel-
plate waveguide is fed from oppasite ends. The feeds selected are hom-fans
combinations. Using the design information presented in Chapters 3 and 4, this
chapter gbes aie design details for the construction of such an antenna using the
machine shop at the university. The availability of materials, machine shop
limitations and test facilities requirements have been considered.
Chapter 5 -A Design Example
The design example is intended for Ka-band satellite communication
terminal applications. Due to the Iimited facilities and materials available, design
specifications have been modified from those in Table 1-1. These modifications
are such that the antenna concepts discussed in the last two chapters can be
tested with minimal deviation from original objectives.
The modified specifications for a typical satellite terminal antenna are
given in Table 5.1-1. Typically, Ka-band satellite systems use 30 and 20 GHz for
the transmit and receive functions, respectively. These bands have been scaled
to allow both frequencies to share a common waveguide system (WR-28) in
order to minhize equipment and resources. Although typical satellite antennas
are larger, a 25-cm X 25-cm aperture was sefected due to the standard size of
copper-cladded dielectric sheets that are readily available. Furthenno re, this size
is more appropriate for the iimited size antenna range at the university. The look
angle of Vancouver for the ACTS satellite is about 30° from the horizon but this
design was changed to 28' due to the thickness and the permittivity of the
dielectrics available.
University of British Columbia ETectricaC and Cornputer Engineering
Chapter 5 -A Design Gcarnpfe
ITEM # 1 PARAMETER SPEClFlCATiON
1, Aperture size (cm) 25 cm
2- Gain (dBi)
Transmit 31
Receive 25
3. Frequency Band (GHz)
Transmit 39.25-39.75 GHz Receive 27.25-27.75 GHz
4. Polarization
Transmit Horizontal
Receive Vertical
5. Sideiobe levels - t7dB
6. C ross-polarkation (dBi) -20 dB
7. Elevation Angle 0, -28'
8. Eficiencv 50%
9. 1 VSWR
fable 5.1 -1 Antenna Example Specifications
Aside from these specifications, the antenna design will attempt to address the
desired attributes discussed in Chapter 1.
University of Brit'sh Columbia EtectrCcaE and Cornpufer Engineering
Chapter 5 -A Design Example
As illustrated in Figure 5.2-1. the proposed antenna is a slotted parallet-
plate waveguide antenna with two arrays of long slots perpendicular to one
another, one for the horizontally polarized function and the other for the vertically
polarized. The horizontally polarized anay is composed of longitudinal leaky-
wave slots which have an inherent tiit of the main beam in the direction of slot to
facilitate the look angle of the satellite. The vertically polarized dots are unifonn
dots with th8 anay excited in a leaky-wave fashion such that the main beam
collocates in space with that of the horïzontally polarized amy. Although the
arrays share a common aperture, each set of slots is excited by a different mode
and hence they have different propagation coefficients in the roaded paraliei-
plate waveguide structure. lt is their respective phase coefficients that determine
the angles of the main beams.
In order to excite the parallet-plate waveguide in the two required modes,
two unique horn-lens feed assernblies are used at opposite ends of the
waveguide. In order to conserve space, these feeds extend below the paraIlel-
plate structure by the use of 18Wegree bends. The antenna is also shown
wittiout the 180-degree bends in Figure 5.2-2.
University of 6-Et Columbia Etectrkd and Cornputer Engineering
Chapter 5 -A Design BarnpIe
@ Transmit
r 1
Transmit Hom
Feed
Figure 5.2-1 DuaCFrequency DuaCPolarization Antenna
University of BRtish Columbia Eiectrid anci Cornputer Engineering
Chapter 5 -A Design Example
Lenî Lens
Tb, mode
X I
Figure 5.2-2 DuaCF requency Dual-Polarization Antenna without 1 80° Bends
University of British CoEumbia 5-6 Etectrical and Cornputet Engineering
Chapter 5 -A Design Example
To illustrate the operating principles of the antenna, first consider the
horizontally polarized path. The horizontally polarized signal is applied as a TElo
wave at the rectangular waveguide section whose walls flare in the E-plane to
fom the E-sectoral hom. As the wave propagates in the hom region, it assumes
a spherÏca1 wave front, which is then collimated by the lens. The resulting wave,
which has an equi-phase front, travels through a 180' bend to the parallei-plate
waveguide. Given the polarization of the wave ernanating from the E-sectoral
horn, the partially loaded parallel-plate waveguide is excited in a quasi-TE1 mode
where the horizontally polarized slots interseet the surface currents of the
waveguide walls perpendicularly. These slots disnipt the surface currents of the
loaded parallei-plate structure resulüng in an €-field across the slot apertures.
Since these slots radiate as the wave propagates and hence the name
longitudinal leaky-wave dot, the resultant radiation pattern is a horizontally
polarized beam in the direction of propagation in the parallel-plate waveguide.
In the vertically polarked path, the rectangular waveguide is also excited
in the TEto mode but the feed is posiüoned at 90' with respect to the horizontally
po larized input waveguide sueh that the hom can be flared in the H-plane Ristead
to fom an €4-sectoral hom. Although the wave travels through the lens and 180°
bend as in the horkontalIy polarked case, the wave emerging into the loaded
parallelglate structure propagates a quasi TEM mode due to the polarization of
the wave from the Ksectorai hom. An Ïmpoctant difference between the
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Chapter 5 -A Design Example
horizontally polarized and vertically polarized waves is that the quasi TEM mode
surface cunents on the walls of the waveguide flow perpendicular tu those of the
quasi TE1 mode. ConsequentIy, the vertically polarized slots are at 90' with
respect to the horizontally polarized slots in onier to disrupt these surface
currents. The verücally polarked anay is ananged in the direction of propagation
such that the progressive phase excitation between radiating elements is chosen
for the verticaliy polarized beam to be collocated in space with the horizontally
polarized beam.
As discussed previously, the proposed antenna has a number of
advantages over existing antennas. To date, this is the only waveguide antenna
type in which separate arrays share a common aperture. A simplified block
diagrarn of the high level design showing the individual cornponents is delineated
in Figure 5.3-1. The antenna structure is decomposed into its key functional
blocks: the horizontally pofarized and vetticalIy polarized dot radiators and
anays, the parallel-plate waveguide, and the feed structure. Each block is further
sub-divided into its individual components. The design of the antenna can be
divided into the radiating slot anays, the parallel-plate waveguide and the feed
structure.
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Chapter 5 -A Design Example
Figure 5.34 High Level Diagram
University of British Cohmbia €iedricai and Ccmputar Eneineering
Chapter 5 -A Design Example
5.4 PARALLEL-PLATE WAVEGUIDE STRUCTURE
In the design of the antenna, first, an investigation is conducted of whether
the appropriate p&, can be achieved given the selected modes, and the look
angle. 0,. The type of dielectric loading, the dielectric properties, and the
separation of the waveguide walls must be setected. Using the design equations
in Chapter 3, and sunreying the available dielectric rnaterials, a list of design
parameters was detemined as shown in Table 5.4-1. The dielectrïc loading
configuration is side- loading, and the dielectric material selected is RT/duroid@
5880 PTFU randorn microfibre-glass. It is available in 300 X 300-mm2 sheets
with the thickness of the dielectric equal to 40 mil or 1.02 mm. This resuks in a
look angle of 0,=28° for the anays given a = 7.0 mm. Two sheets with copper
cladding on only one side fom the top and bottom plates of the paraIfel-plate
waveguide. This is advantageous because it allows the dot arrays to be etched
as opposed to being milled onto a metal plate, Using this process, the cost of
manufacturing approaches that of microstrip patch anays. Furthemiore, typical
etching processes allow for fher tolerances cornpared to mechanical milling.
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Chapter 5 -A Design Example
Table 5.44 Design of ParaIlel-Plate Waveguide Structure
Next in the design, the radiating elements and the anay are investigated. The
slot radiators considered here are of two types: longitudinal dots for the transml
anay and transverse slots for the receive anay. At this stage of the design. the
critical parameten that must be determined are the dot dimensions, and the
tapers of the anay. Given the parallei-plate waveguide size selected, and
leaving the distance of 25 mm on al1 sides of the plates. the aperture size is
determined to be 250 mm.
The dot widths were detemiined by experiment. In the transmit case. the
dimensions of the dots were selected based on the maximum width for high
eficiency and not exciüng higher order modes, narnely the TEi odd mode. In
the receive case, 1 was found that as the slot width was increased, there was a
point where the reflected lobe started to ïncrease rather than decrease as
predicW in Chapter 3. The selected width was the best compromise between
minimizing the reffected lobe and maximizing the efficiency of the anay.
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Chapter 5 -A Design Example
In bath arrays, the taper along the direction of the propagation was chosen to
be trapezoidal to maximize the aperture efficiency of the arrays. In the other
direction, the taper was uniform since the feed contributes a cosine taper. Table
5.5-1 gives the design details for the longitudinal slot array white Table 5.5-2
gives the design details for the transverse dot array.
I l . I
1 Slot lenoth. / 1 250 mm 12 i SIO~ wi& i t centre
I
1 2.0 mm 1 3.
I 1
1 Slot width at end 1 1.5 mm 4. Taper along slot Tfapezoidat 5. S pacing 5.0 mm 6. Taper of anay Uniform 7. Number of f lements 50 8. Phase coefficient 727.8 m-1 9. Elevation Angle 00 -280 1 O. Attenuation Coefficient -8 neperslm 11. Eff icie ncy >60%
Table 5.5-1 Design of Horizontally Polarized Array
PARAMETER DESIGN Slot tength, [ 250 mm Slot width at centre of mav 2.0 mm Slot width at end of array 1.5 mm Taper alona dot Uniforni
I
Spacing 5.4 mm Taper of Anay Trapemidal Number of Elements 50 Phase coefficient 645 nt Elevation Angle O0 -280 Attenuation Coefficient -8 neperim EfficLencv
Tabk 5.5-2 Design of Vertically Polarizeâ Amay
EletMd and Cornputer Ehgineering
Chapter 5 -A Design Exarnpte
In this design, the feeds of the arrays can be considered separately.
Using the design information in Chapter 4, a comgated hom was designed for
the longitudinal slot anay since 1 is an E-plane sectoral hom. A comgated lens
was also designed for this hom. As for the transverse dot anay, it was not
required for the horn to be comgated. Although corrugating the lens would have
improved the matching of the lens, this was not done since machine shop time
was expensive. The detail design of the feed for the longitudinal slot anay is
given h Table 5.6-1 and that of the transverse dot array in Table 5.6-2.
A 180' bend was also constructed. The dimensions are given in Table
56-3 below.
PARAMETER 1 DESIGN
Apemire Height 1 7.0 mm Fiare Angie 30 degrees Comgations Teeth spacing 3+0 mm Teeth wi'dth 2.1 mm Teeth height t .9 mm
Input Waveguide WR28 Lens Pemitüvity 2.55 Lens Comgations Teeth spacing, D 3.2 mm Teeth width. t : 1.9 mm Teetb height, d 1.8 mm
Tabb 5.6-1 Design of Longitudinal Slot Array Hom-bns Feed
University of Britrsh Columbia and Cornputer Engineering
Chapter 5 -A Design Example
ITEM# 1 PARAMETER 1 DESIGN 1. AperRire Width 300 mm 2. Apemire Height 7.0 mm 3. Rare Angle 30 degrees 4. Comgations None 5, Input Waveguide WR28 6. Lens Pemiitavity 2.55
Lens Comaations None a
Table 5.6-2 Design of Transverse Slot Array Horn-lens Feed
# 1. Outer radius 300 mm
Il 5. 1 1
I Insertion L o s I <O.i dB 1 ,
Table 5.6-3 Design of 180° Bend
Based on the design goals set in Section 5.1 and the design information in
Chapters 3 and 4, an exampfe of a dual-frequency duaf-polarized slotted
waveguide anay is presented. The design given in the previous tables is not
necessarily an optimal design, but one based on the availability of materials and
simplicity of construction based on the university resources such that a prototype
can be buik quickry and most cost effectiveîy. In the next Chapter, this design is
built and tested to verify the design objectives.
L L
University of British Columbia BectrÏcal and Cornputer Engineering
7.0 mm 300 mm >30 dB
2. 3, 4.
lnner radius Length of bend Return Loss
Chapter 6
Based on the design information given in Chapter 5, an experimental
prototype was buiit by the university machine shop. The implementation process
was carriad out in steps in conjunction with verification testing. First, the hom-
lens designs were built up and measured. Then each array was built ont0 the
correspondhg hom-Iens structure and tested independently. A band was
designed and built to locate the hom-lens structure below the parallekplate. The
final step was the integration of the two arrays into a shared apeiture fed by the
two hom-Iens structures. Bends were not used in the final structure because the
mechanics were too invoIved for the machine shop at the univenity. This
chapter describes the test setups and g h the tests results for each stage of the .
implementaüon process includhg the fmaE integration.
Chapter 6 -Gcperimental Investigation
Three test setups were used in the experhental investigation: an aperture
measuring setup, a voltage standing wave ratio measurement setup, and an
antenna range to obtain radiation measurements. The setups were designed
based on the equipment available at the university supplemented by fixtures
constnicted by the department machine shop. The following sections describe in
detail each of the setups.
An aperture measurement setup was developed to measure the amplitude and
phase distribution across the hom-lens apertures. This information was used to
verffy the individual performance of the hom-lens structures before adding on the
parallei-plate waveguide.
The setup required connecting the hom-lens structure to a frequency source.
An open-ended waveguide, which was conneded to a detector, sampled the
wave along the aperture. This sarnpler was mounted on a manual dovetail
posiüoner equipped with a scale to indicate position at which the amplitude and
phase nifocmation was collected. The detector output and a reference signa1
were connected to the receiver to give both relathe ampiitude and phase
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Chapter 6 -Experimental investigation
information. The receiver outputs were connected to a computer data acquisition
card ushg a simple analog interface circuitiy designed in-house. The computer
recorded the amplitude and phase information using a program wntten in Pascal.
Various pieces of absorbing material were placed in the setup to minimize
scattering and reflection. The positioner could be manuafly placed within 1 mm
of desired position; the accuracy of the amplitude was usually within 0.5 dB and
the phase was within 4 - 1 5 degrees. This setup is shown in Figure 6.1 -1. Table
6.1 -1 gives the list of equiprnent used for the setup.
t . Positioner Dovetail with scale 2. Open-ended Waveguide Coupler WR28 3. Detector Narda WR28 4. Source HP Sweep Osciilator 86908 5. Receive r Scientific Atlanta 6. Data Acquisition Card PC UO 7, Personai Cornputer 486 66 MHz 8, Absohincr Material Various
Table 6.14 Equipment List for Aperture Measurement
University of British Columbia Ei ecriid and Computer Engineerüg
Chapter 6 -Gcperimental Investigation
Amplitude and Phase
information
Personai Computer with Data Acquisition Card
Reference Signal
Open Ended Waveguide
on Positioner
Figure 6.14 Aperture Measuring Setup
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Chapter 6 -Gcperimental Investigation
The voltage standing wave ratio (VSWR) measurement setup was used to
measure the VSWR perfomance of the antenna as well as the effect of the
bend. The device under test, whether 1 is the hom-lans structure, with or without
a bend or the integrated antenna, was connected to the frequency source with a
stot-line in between. The device under test had to be properly temiinated. In this
case, absorbing material was used to minimize reflections from the surrounding
area. A probe, coupled to a detector, sampled the amplitude of the wave along
the dot Iine. The output of the detector was fed Rit0 the VSWR meter. The ratio
of the maximum and minimum yielded the VSWR. The VSWR measurement
setup is shown in Figure 6.1.2. Table 6.1-2 gives the Iist of equipment used in
the setup.
3. Slot Line Narda 4, Source HP Swee~ Oscilfator 86900
Table 6.1-2 Equipment List for Return Loss Measurement
University of British Columbia 6-5 ETectrlcal and Cornputer Engineering
Square Wave Modulation
Chapter 6 -Experimental Investigation
Source Snu' Detector
VSWR meter
Device Under Test
Slotted Line Absorbing
Figure 6.1-2 Voltage Standing Wave Ratio Measurernent Setup
University of British Columbia Eiectricat and Cornputer Engineering
Chapter 6 -Experimental Investigation
An outdoor range was assembled on the roof of the electrical engineering
building for the radiation pattern measurement. For most far field ranges, a
distance of 2 &A is used, where d is the antenna diameter or equivalent.
However, the distance available is only about 10 metres, or equivalently, 1.2 8~
for the highest frequency and 1.8 &/A for the lowest frequency. The impact is a
slight degradation in the sidelobe levels and widening of the main beam. Given
the accuracy of the equipment for the range, this is negligible.
The setup consists of a rectangular hom, connected to a frequency source,
which was used as a transmit source at one end of the range. At the other end
of the range, the antenna under test was set up as a receive antenna mounted
on an automated tumtable. Two mounting setups were available for mounting
the antenna under test, namely for either the E-Plane or the H-plane
measurernents. These are shown in Appendk L. The signal received by the
antenna under test was connected to a detector, which in tum was connected to
a receiver via a low loss cable. The receiver measured the relative amplitude
reading between a reference signal detected at output of the frequency source
and the detector output of the antenna under test. The amplitude information
was recorded &y the computer using a data VO card. The computer also
controlled the tumtable using this data UO card. A program usïng Labview was
University of British CoCumbïa 6-7 Ektrkat and Cornputer Engineering
written to synchronize the turntable and collect the data from the receiver.
Various pieces of absorbing material were placed in the setup to minimize
scattering and reflection.
The tumtable iç capable of rotathg the antenna in fractions of a degree. The
accuracy of the data VO card controller is within a degree. The amplitude data is
within 4 - 1 dB. This setup is shown in Figure 6.1-3. Table 6.1-3 gives a list of
equipment for the setup.
8. 11 1 Absorbing Matenal 1 Various (
ITEM # 1. 2, 3. 4, 5, 6, 7.
Table 6.1-3 Equiprnent List for Antenna Range
ffecWicai and Cornputer Engineering
--
EQUIPMENT Tumtable Detector Transmit Hom Source Receiver Data VO Card Persona1 Cornouter
MOOEUWPË In house design Narda WR28 Pyramidai HP Sweep Oscillator 86908 Scientific Atlanta 1750 National Instruments Pf OU
Chapter 6 -Experimental Investigation
The hom-lens aperture verkation was tested using the setup described
in Section 6.1.1. This experiment was dMded into two steps: the horn by itself
and the horn together wkh the lens. In each step, both the phase and amplitude
information were recorded at 5-mm increments.
- - - -
Figure 6.24 shows the amplitude across the E-sectoral hom with and
wlhout the l e m The results are retatively close to those predicted by the theory
in Chapter 4 where the E-sectoral hom has a cosine taper and the lens adds an
additional cosine taper. At the edges of the hom, the measured results deviated
frorn the theory due to diffraction. A comgated tans was used tu minimize the
mismatches. The experimental resufts for the phase are compared with the
theoretical results for aperture with and without the lens in Figure 6.2-2. The
measured resufts matched very welI to the theory. Similar ta the amplitude
measurements, the measured phase results deviated from the expenmental
results due to diffraction at the edges of the aperture.
University of British Columbia EledcM and Cornputer Engineechg
Chapter 6 -Experimental Investigation
' * " Measured without Lens - Distance Along Aperture (cm) Theoretical without Lens
* " Measured with Lens - Theoretical with Lens
Figure 6.2-2-1 Amplitude Measurement for the E-Sectoral Hom Aperhire
University of British Columbia ElecWai and Cornputer Engineering
C hapter 6 -Experimental investigation
-i
r
O D 1 1 I l 1
5 10 15 20 25 30
* * * Measured without Lens Distance Along Aperture (cm) - Theoretical without Lens '- Measured with Lens - Theoretical with Lens
Figure 6.2-2 Phase Measurements for the E-Sectoial Hom Aperture
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Chapter 6 -€xperimental Investigation
Figure 6.2-3 shows the amplitude across the H-sectoral hom with and
without the lens. In the case without the lens, the measured results match the
experimental results well. Note that the H-sectoral hom has a unifom E-field
while the lens contributes a cosine taper. Since the E-field is parallel to the
edges of the flared walls, it can be seen that the diffraction in the H-sectoral horn
is less than that of the E-sectoral hom. In the case with the lens, since this lens
is not comigated, there are mismatches at the dielectric-air interfaces. This is
the cause for the amplitude ripple in the measured results with the lens.
Figure 6.2-4 shows the phase distribution across the H-sectoral horn with
and without the lens. Similar to the E-sectoral hom. both phase measurements
match well with the theory* Again, the E-field is paraIIel to the edges of the flared
walls, therefore the effect of diffraction are minimal as cornpareci to the E-sectoral
hom.
6-1 3 ffectrlcat and Cornputet Engineering
Chapter 6 -ExperimentaI Investigation
-
3
-
1°io I I i I 1
5 10 15 20 25 30
Measured without Lens Distance Along Aperture (cm) - Theoretical without Lens
* * Measured with Lens - Theoreticai with Lens
Figure 6.2-3 Amplitude Measurements for the H-Sectoral Hom Aperture
University of British Columbia EiectrkaI and Cornputer Engineering
l8OC
t6OC
l4OC
1200
1 O00 - U)
8 eoo b 8 , 600 u3 Ca 5=
400
200
Chapter 6 -Exparimentai Investigation
Measured without Lens Distance Along Aperture (cm) - Theoretical without Lens + +* Measured wÎth Lens - Theoretical with Lens
Figure 6.2-4 Phase Measurements for the H-Sectoral Hom Aperture
University of Bnüsh Columbia Electcicai and Cornputer Engineering
Chapter 6 -Expertmental Investigation
Due to the mechanical constraints of the antenna, bends could not be
easily constructed to locate both homs below the aperture. Despite that, a bend
was made and tested using the E-sectoral hom. VSWR measurements were
conducted for the transmit hom with and without a bend. In both cases, the
output was teminated by the use of absorbing material. The following table
summarizes the retum Ioss measurement results. lt can be seen that the
addition of the bend caused minimal degradation to the return bss.
1 1 FREQUENCY 1 VSWR II 1 Horn alone
I
I Il I
1 39.00 GHz 1 1.18 I
39.25 GHz 1 .O9 39.50 GHz 1.23
Horn & Bend
Figure 6.3-1 VSWR Measurernents for the E-Sectoral Hom-lens Structura with and without the Bend
University of British Columbia Eiectric& and Cornputer Engineering
C hapter 6 -Experimental investigation
6.4 RADIAT~ON TESTING OF THE INDIVIDUAL SLOT ARRAYS
After the hom-lens structures were verified, two sets of slot arrays were
constnicted: a longitudinal dot array, and a transverse slot anay. The dot
pattern was etched on with a tolerance of +/- 0.5-mm. Each dot array was
connected to the appropriate hom-lens structure. Apart from the alurninum
spacers at the edges of the array, closed ceIl expanded polystyrene foam (er =
1.03) was used as spacers throughout the structure to maintain the separation
between the paraltel-plates. Copper tape was also used to minimize the
radiation leakage at the hom-lens-paraIfel-plate-waveguide junction. Absorbing
material was used to terminate the residual power at the end of the arrays to
minimize radiation in undesired directions.
The radiation tests were done using the experimental setup described in
Section 6.1.2. The tests for each array included CO-polar measurernents for the
E and H planes. These measurements served to verify the performance of the
individual arrays prior to combining the amys into a single aperture. The test
results for the individual anays are given in the following sections.
University of British Columbia ffectricaI and Cornputer Engineering
Chapter 6 -ExperirnentaI Investigation
Several longitudinal slot anays were constnicted with varyhg widths and
tapers. Generally, the wider the slot, the less residual power at the end of the
array, and the lower the beam of the reflected wave. For instance, when the
width was doubled, the beam of the reflected wave was lowered by
approximately 3 dB. Likewise, the steeper the taper, the smaller the effective
aperture of each slot and higher the beam of the reflected wave. However, the
greater the slot aperture, a higher order mode, namely, the TE, odd, became
more prominent. The best compromise of dot width and taper was found to be
3.1 mm wide in the middle tapering d o m lineariy to 2.3 mm at the ends. This
longitudinal dot array was connected to the E-sectoral hom-lens structure. After
careful location of spacers, followed by a laborious process of adjustments of the
screws to prevent buckling of the plates, the following patterns were achieved.
The measured E-plane pattem and the theoretical pattem are shown in Figure
6.4-1. The measured and theoretical patterns for the Hglane are shown in
Figure 6.4-2. The E-plane pattern matched up well with the theoretical with the
exception that the measured beam width was slightly wîder. The H-plane pattern
shows that the attenuation constant a = 8. Note that there is a slight indication of
the higher order mode between 65O to 115'. Overall, both patterns were
comparable to the theoretical.
Uniketsity of British Columbia Electrïcaf andi Cornputer Engineering
Chapter 6 -Expehental Investigation
Angle (degrees)
Figure 6.44 E-pfane Experimental Radiation Pattern for the Longitudinal Slot Amy
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Chapter 6 -Experimental investigation
Angle (degrees)
Figure 6.4-2 H-plane Expetimental Radiation Pattern for the Longitudinal Slot Array
University of British Columbia ElectricaI and Cornputer Engineering
Chapter 6 -kpenmental Investigation
Similar to the previous sets of experiments pecfomied for the longitudinal
dot amys, several sets of transverse dots were constmcted with varying widths
and tapers. Unlike with longitudinal slots, a wider dot did not necessanly result
in a lower beam of the refiected wave as shown in Chapter 5. lt was found that
after the centre slots reached a width of 3.1 mm, widening the slot actually
increased the power in the reflected wave. The best compromise of slot width
and taper for the best sidelobe and reflected wave combination was found to be
2.3 mm wide for slots in the middle tapering down lineariy to 0.5 mm for slots at
the ends. The measured E-plane pattem and the theoreticai pattern are shown in
Figure 6.4-3. The measured and theoretical patterns for the Hglane are shown
in Figure 6.4-4. The measured E-plane pattern had higher sidelobe levels as
well as the beam of the reflected wave was higher than expected. Part of the
reason for the high reffected-wave was the difficult faing of the absorbing
matefial for this particular structure. The rneasured and theoreticai H-plane
patterns matched well with the exception that the measured sidelobe levels were
slightly higher. Overall, both patterns were comparable to the theoretical.
University of British Columbia and amputer Engineering
C hapter 6 -Experimentallnvestigation
-40 L O 20 40 60 -' Expenmental - Theoreticai
80 100 120 140 160 180
Angle (degrees)
Figure 6.4-3 E-plane Exparimental Radiation Pattern for the Transverse Slot Anay
University of British Columbia ElectcM and Cornputer Engineering
Chapter 6 -Experimental Investigation
- - Experimentai Angle (degrees) - Theoreticai
Figure 6.4-4 H-plane Experimental Radiation Pattern for the Transverse Slot Array
University of British Columbia €lectgicd and Cornputer Engineedng
Chapter 6 -Experimentat Investigation
The integration of the antenna consists of combining the selected
structures of the previous experiments to yield a complete antenna system. First,
a parallel-plate waveguide structure etched with the selected dot arrays
combined into a single aperture was constructed. Then the two hom-lens
structures were added to the parallel-plate structure. Again, foam spacers and
copper tape were used. A wide band temination was used at input of one hom
when the other horn was being used. This termination replaced the absorbing
material in the last experiments.
The integration testing consists of three tests: the VSWR measurement.
the gain rneasurernent and the radiation pattern measurements. The following
sections describe the results of these in detail.
6.5.1 VOLTAGE STANDING WAVE RATIO MEASUREHENT
The voltage standing wave ratio measurement as described in Section 6.1
was used to obtain resufts for both the transmit and receive bands. Tabte 6.5-1
summarizes the resuits for high, middle and low fraquencies of each band. The
results indicated a good match at both poits.
Universify of British Columbia Eiect~i and Cornputer Engineering
PORT 1 FREQUENCY 1 VSWR Transmit
39-00 GHz 1-08 39.25 GHz 1 -05 39.50 GHz 1.10
Receive 27.00 GHz 1 .O5 27.25 GHz 1 ,IO 27.50 GHz 1 -08
Table 6.5-1 Voltage Standing Wave Ratio of Antenna
6.5.2 Gain
The gain measurements used the antenna range setup described in
Section 6.1. The range was calibrated using a rectangular hom whose gain was
calculated based on its dimensions. The accuracy of the measurernents is within
2 dB. Table 6.5-2 sumrnarizes the calculated and experimental results. Here,
the directivity is the maximum achievable gain of the array given the site and the
taper. The theoretical gain is based on known losses descn'bed in Table 6.5-3.
These losses included the residual powet at the end of the array, tosses in the
construction of hom, fosses in the dielectric of the lens, tosses in the propagation
of the waveguide both dielectric as well as conductive tosses, leakage from the
side of the aperture shce the hom aperture is made wider fhan the radiation
aperture, and fïnally losses in the junction of the hom-lens structure and the
parallelglates. GNen these losses, and the accuracy of the measurhg system,
the measured gains were approximately those of the expected gains.
University of British Columbk EFmcaC and Cornputer Engineering
Chapter 6 -EXperimentat Investigation
FREQUENCY DIRECTIVITY THEOR~CAL GAiN 1 1 GAiN 1 IYEAÇUREDI Transmit
Table 6.5-2 Gain of Antenna
27.00 GHz 27.25 GHz 27.50 GHz
ANTENNA 1 TYPE OF LOSS 1 11
39.00 GHz 39.25 GHz 39.50 GHz
32.1 dB 32.3 dl3 32.4 dB
35.8 dB
Transmit
35.9 dB 36.0 dB
25.3 dB 25.5 dB 25.6 dB
Hom (constniction)* Dielectric of Lens
1
25.0 dB 25.8 dB 25.2 dB
Residual Power -0.8 dB -0.4 dB
Propagation in waveguide Side of aoerture
31.2 dB 31 -3 dB 31 -4 dB
-1 .O dB
-0.4 dB -0.8 dB
I I 1
30.1 dB 30.5 dB 31 .O dB
Total
Receive
Propagation in waveguide Side of aoetture
Table 6 5 3 toss in the Antenna
Hom-waveguide junction*
Residual Power Hom (construction)* Dielectric of Lens
-0.4 dB -0.8 dB
Totd
University of British ColumbM
-1.2 dB 4.6 dB
-3-0 dB -0.8 dB -0.6 dB
Electrlcat and Cornputer Engineering
* Note that these are rouaih estimates (diff~cult to verifv)
I
Hom-waveg uide junction' -1 -2 dB -6.8 dB
Chapter 6 -Experimental Investigation
The combined aperture was measured using the setup dexribed Section
6.1. These included CO-polar patterns for both arrays. In addition, cross-polar
patterns were taken to compare the performance before and a€ter the combining
of the apertures. The results are divided into that of the longitudinal slot anay
and the transverse slot array.
6.5.3.1 Longitudinal Slot Atray
The theoretical and expenmental longitudinal slot anay €-patterns are
shown in Figure 6.5-1. Note that comparing this pattem to Figure 6.44, the
beam width is similar but the sidelobe levels are slightly higher. This can be
explained by looking at the pattern in the other plane. The measured H-plane
pattern in Figure 6.5-2 shows that there were two additional sidelobes emerging
between the main beam and the beam of the reflected wave. Calculations
showed that these were caused by the TE1 odd mode and its reflected wave.
This mode resulted in less residual power and hence a smaller beam from the
reflected wave but more power in other directions, namely 66' and 123O. This
higher order mode is probably the cause of the higher sidelobe levels in the €0
plane as well. Figure 6.5-3 shows the cross-polar performance of the
Iongitudhal anay before and after the aperture combining. The diierence is
RisignïfÏcant.
University of 6ritlsh Cofumbia 6-27 EfectricaC and Compter Engineering
C hapter 6 -Experimental Investigation
Angle (degrees)
Figure 6.5-1 E-plane Experimental Radiation Pattern for the Longitudinal Slot Array
University of British Cohmbia E k ~ ~ c a l : and Cornputer Engineering
Chapter 6 -Experimental Investigation
-
-
3
-
-
-
I
r I I I
1 l
- O 50 100 150 200 250 300 350 Y
- - Experimentai - Theoreticai
Angle (degrees)
Figure 6.5-2 H-plane Experimental Radiation Pattern for the Longitudinal Slot Ariay
University of Bnüsh Columbia Ekdrical and Cornputer Engineering
Chapter 6 -€xperimental Investigation
l' -40 O 10 20 30 40 50 60 70 80 90
' ' Before combining Angle (degrees) - Mer combining
Figure 6.5-3 Cross-polar Experimental Radiation Pattern for the Longitudinal Slot Array
University of British Columbia 6-30 EIectrïcai and Camputer Engineering
Chapter 6 -Expenrnental investigation
6.5.3.2 Transverse Slot Array - - - -
The theoretical and experimental transverse dot anay E-patterns are
shown in Figure 6.5 -4. Note that comparing this pattern to Figure 6.4-4, the
beam is narrower but the sidelobe levels are similar. The H-plane patterns in
Figure 6.5-5 show that since the antenna is now terminated properly, the
reflected wave is much lowered than before, and likewise the sidelobe tevels.
Figure 6.5-6 shows the cross-polar performance of the transverse slot anay
before and after the aperture combining. Again, the difference is insignificant.
University of BntLsh Columbia UectricaE and Cornputer Engineering
Chapter 6 -Experimental Investigation
- - Experimentd Angle (degrees) - Theoreticai
Figure 6.54 Eplane Experimental Radiation Pattern for the Transverse Slot Amy
University of British Columbia Electrîcat and Cornputer Engineering
Chapter 6 -Experimental Investigation
- - Experimental - Theoreticai
r I I I
f = 27.5 GHz
70 80 90 ioo n o 120 130 Angle (degrees)
Figure 6.5-5 Kplane Experimental Radiatîon Pattern for the Transverse Slot Amy
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Chapter 6 -Experimental Investigation
- - Before combining Angle (degrees) - Afier combining
Figure 6-54 Cross-polar Experimental Radiation Pattern for the Transverse Slot Array
University of British Columbia 6-34 Elemcaf and Cornputer Engineering
Chapter 6 -ExpecimentaC investigation
6.5.3.3 Beam Squint
One of the important parameters in the design of the antenna is the
bandwidth. Depending of the type of array, the look angle of the main beam will
Vary with frequency. The effect, often referred to as beam squint. can be the
Imiting factor in the bandwidth of the design. Table 6.54 shows the theoretical
and experimental results for the look angle of the main bearn, 8,. Note that in the
TE1 mode, the squint is insignificant This is because 8, is a function of the p$po
and the two p's increase approximately the same amount with an increase in
frequency. For the TEM mode, since 8, is not just a function of P$Po but the
spacing of the elements, d, as well, the beam has a greater squint with
frequency. Overall, the theoretical and measured results compared well.
' FREQUENCY THEORETEAL (DEG) MEASURED (DEG) TE1 mode
39.00 GHz 1 29.0 1 29.1
39.25 GHz 1 28.75 1 28.8
39.50 GHz 1 28.4 1 28.3
27.00 GHz 1 23.96 1 22.5
27.25 GHz 1 26.5 1 25.5
27.50 GHz 1 28.8 1 28.7
Table 6.59 Theoretical and Measured Values of Look Angle
Unhersity of British Columbia Efedricaî and hmputer Engineering
Chapter 6 -Experimental Investigation
The following table is a cornparison of the specification in Table 5.1-1 and
the experimental results. In summary, although the experimental model has not
met the gain, sidelobe levels, cross-polar performance and power efficiency
objectives, 1 has achieved its fundamental goal in combining two long-slot anays
of different frequency and polarkation into a single aperture.
ITEM # PARAMETER SPECIFICATION RESULTS
1 . Aperture size (cm) 25 cm 25 cm
2, Gain (dBi)
Transmit 31 30.1 minimum
Receive 25 25.0 minimum
3. Frequency Band (GHz) Transmit 39.25-39.75 GHz 39.25-39.75 G Hz
Race ive 27.25-27.75 GHz 27.25-27.75 GHz
4. Potarization
Transmit Horizontal Horizontal
I Receive I Vertical I Vertical
5. Sidelobe levels - 17d8 - 7 dB worst case
6. Cross-polarkation (dBi) -20 dB -1 6 dB worst case
7, Elevation Angle Bo -28O 22.5 O -29.1' over frequency range
8. Efficiency 50% -25% wosst case
9- VSWR 1.5:f maximum I -1O:i maximum
Table 6.54 Cornpliance Table
University of British Columbia Eiectcical and Cornputer Engineering
Chapter 7
hcreasing attention has been paid in the last few yem to satellite communication in mm-
wave region, in particular 20/30 GHz. With these higher frequencies, designers are faced
w*th p a t e r challenges in both systems and cornponent design. One area in pmicular is the
Iosses associated with the precipitation since power is at a premium. Recent expetimentai
work reveaied that not only is the precipitation in the propagation path important, but
moisture on the antema plays a signifiant mie [LI (21 [3]. Extensive experiments c m e d
out under simulated r a h conditions on the Advanced Communication Technology Satellite
(ACTS) terminai antema at the University of British Columbia reveaied that for a 12 m off-
axis feed parabolic dis& with an devation mgie of about Mo, attenuation due to moisnire on
~e feed hom and dish cm cause as much as 8 cil3 attenuation at 27.5 GHz [4][q[q. The
attenuation due to antenna wening was fomd to be a function of the nin type, rain intensity,
d a c e conditions, and wind velocity and direction.
Chaprer 7 - bwerted Configuration
In one class of antennas, more specificdy in arrays the main beams of which are at an
abirbiuary mgie to the aperture, one cm minimize the antenna-wetting problem by using the
inverted configuration [7J [a]. The inverted configuration aiiows for the antenna structure
itself to protect the radiating aperture from rain and snow. As a result, dwing adverse
weather conditions, the Iosses are reduced to that of the path.
Since the travelling-wave long-dot array can be designed with the main beam at an
arbitrary angle, it is a suitable candidate for the inverted configuration 191. This chapter
descnbes the concept of using the slotted waveguide antenna in the inverted configuration.
An experirnental investigation of the antema's performance in simulated min is also
presented.
The inverted configuration, shown in Ergure 7-14, is accomplished by directing the
radiating aperture downwards at an angle, a, with the horizontal plane, while the main beiun
is in the direction of the look angle, o, such that = a - go. The angies a and go provide two
degrees of freedom in the design of inverted array. When designing for use in the inverted
configuration, Bo should be mllumized to achieve the srndest a such that the radiating
sudace is less exposed to precipitation. This configuration is particularly usehil for Iow
elevation angie appiications where is smd. For Iarger elevation angles, a microwave-
transparent extension, which does not intedere *th the beam, may be added to antema to
furùier shield against min and snow. Figure 7.1-2 shows the sIotted waveguide antema used
University of BrWh Columbia
Chptcr 7 - inverted Configm~on
in the inverted configuration.
/ direction of // radiation
# antenna radiating surface **<
Figorr 21-1 Inverted Configuration
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Microwave Transparent
Figure 7.1-2 Slotted W a v w d e Array in Inverted Configuration
Llnkersity of British Cdumbia Electrid and Cornpiter Engineering
The array in Section 5.1. L was used to hvestigate the performance of the antenna used in
the inverted configuration. The antenna was sealed by heat-shrink plastic to avoid water
seeping into the antenna prototype during simuhted min tests. Two types of experiments
were conducted: the antenna with the aperture exposed io precipitation, and the use of the
antenna in the uiverted configuration. The simulated min shower setup is shown in Figure
7.2-1. A photopph of the setup is shown in Figure 7.22. The antenna used in each
configuration is placed under the shower senip. Fust, the antenna is acting as a receive
antenna under dry conditions for 3 minutes as a reference. Then the simulated min is tumed
on for 3 minutes. Two different experiments were conducted.
This experiment was intended to test the antenna performance at 39.5 GHz The
experimentai set-ttp involved using an elevated inasmitbg (broad-beam) hom, whiie the test
antenna was used for receiving. By using the array with = 32'. the radiation from the
û;uismitting hom in this direction was detected by the receiver. It was ascertained that
negligible couphg existed between the transmit and receive antemas. The results with and
without simuIated rain are depicted in Figure 7.2-3, with the antenna in the conventionai
configuration (a) aod in the inverted codiguration (b), These results are selfexplanatory.
Uokersity of Bntis h Columbia 7-5 Bectricd and amputer Engineering
in this case, the test array antema, used for receiving. was pfaced in the far field of a
directive hom radiating in the horizontai direction. The expenment was perfomed on both
array antemas (00 = 32@ and = 604. The results are shown in Figure 7.2-4, and 7.2-5,
respectively, with the antema in the conventional configuration (a) and in the inverted
configuration (b). Again. the results are self-explanatory.
University of British Columbia Eiectrïcal and Cornputer Engineering
Simulated Rain Source
Figure 7.24 Ekpecimentai Sehip with Simufateci Rain Source
University of B&h Columbia Eleckkai and Cornputer Engineering
Figure 72-2 Photograph oCExperimentai Setup with SimuIated Raùi
University of Briüsh Columbia Elmcal and Cornputer EngineerÏng
1 Simulated Rair
l ncident Radiation
l ncident \ Radiation
* 3Z0f\
Time (s)
Figure 7.2-3 Sample Atteouation Data for Experiment 1 (go = 32")
University of British Columbia ElectrÎcal and Cornputer Engineering
f = 3 9 J GHz Start of 1 Simulated Rain
- Incident Radiation
l ncident Radiation #
Tim (s)
Figure 7.2-4 Sample Attenuation Data for Experiment IL (go = 324
University of British Columbia Bectrical and Cornputer Engineering
Start of Simulated Rain
Incident (b) Radiation 6oo / I\ - - - - - - - * - - *
l ncident Radiation - ,-y
Time (s)
Figure 7.24 Sample Attenuation Data for Experiment C[ (go = 60")
University of Briüsh Columbia Electn'cal and Cornputer Engineering
Chriprer 7 - ütvened Configuration
A viable solution to the problem of performance degradation of mm-wave antemas due
to min has been proposed. It involves the use of planar m y antemas designed such that
their main beam is at an arbitnry angle 0. with the ndiating surface in an inverted
coafiguntion. The application of this scheme at Ka-band and higher frequencies would result
in an improvement o f severai decibels in the fade margin of satellite Links.
It may also be noted that there is flexibility in the design of suitabk m y s for this
scheme: there are Little or no restrictions on the type of m y element to be used, and there is
considenble "leeway" in the choice of beam angle, go, and inclination angle, a. provided that
the condition = a - 8. is satisfied. The inverted arny configuration may also be used to
adviuitage in Local Muitipoint Distribution Systems (LMDS) applications, in which case the
transmission is usually in the horizontai direction.
University of Briüsh Columbia
References
[ I l J.Y.C. Cheah, "Wet Antenna Effiect on VSAT Rain Margin", IEEE Transaction on Commttnications, vol. 4 1, 1993, pp. 12384244.
[2] M. M. 2 Kharadly et ai., "Andysis of the ACTS-Vancouver Path Propagation Data", Proceedings of the N A P M XX und ACTS Propagation Siiidirs Mini Workshop, Fairbanks. Alaska, JPL hbiication 96-20. 1996.
[3] V. N. Bringi and J. Beaver. Presentation of the 9'" ACTS Propagation Studies Worksliop (APSW CX), Hmdon, Virgina, JPL Publication 97-3, 1997.
[4] M. M. Z Kharadly and R. ROSS, "Andysis and Modeling of Comipt Propagation Data due to A n t e ~ a Surface Wetting During Rain Events", COST 255 Workshop, ESTEC Noordwijk, The Netherlands. 28-29 October 1998.
[q M. Kharady and R. ROSS. " EEfect of Wet Antema Attenuation on Propagation Data S tatistics," AP-2000 Millennim Conference on Antmnas & Propagation, Davos, Switzerland, 9-14 Apd. 2000, W375.
[6] M. M. 2. Kbandiy and R. Ross, " Performance of Some Conventionai &-band Antennas in (Simulateci) Rain". AP-2WO Millennium Conference on Antennas & Propagation, Davos, Switzerland, 9- 14 Aprii, 2000, Pû472.
(7J MM. M. 2 Kharacüy and A. Y. Chan, "Mm-wave h t e ~ a Arrays with Minimal Degradation of Performance in Precipitation ". 21" ESTEC Antenna Workhop on Arroy Antenna Technolbgy, ESTEC, Noordwijk The Nethertands, May 6-8, 1998.
[8] US Patent Application, 09/035.879 Patent AUowed.
[9] M. Kharady and k Chan, ' A Mm-Wave Antema with 'Non-DegradableT Performance in Raia " AP-2000 Millemtim Corrfrence on Antennas & Propagation, Davos, Switzerlaad, 9-14 Apd, 2000, P0375.
Unkersity of British Columbia ffectrical and Cornputer Engineering
Chapter 8
The motivation for this work was to develop a small aperture, low profile, dual
frequency, dual polarked antenna suitable for portable and mobile satellite
teminals. The design was to incorporate the elevation angle of the satellite for a
given location such that, when in operation, aie radiation aperture is approximately
in a horizontal position, white the pointing in the azimuth plane is achieved
mechanically. lt is suggested that the work presented in the thesis has futfilled
these requirements, with some limitations to be discussed latet. During the
progress of this work, however, and because of the experience gained at UBC on
the ACTS project, the idea of an antenna used h an invertad configuration was
conceived in order to mînimize the effect of attenuation due to wet antenna surfaces
during rain events. This idea would be applicable in both satellite teminals and
LMDS applications. The findings of this work are summarized as follows.
Chapter 8 -Conclusion
8.1 ANTENNA CONCEPT AND DESIGN
The design concepts can be divided into four categories, the basic array
design, the parallei-plate waveguide design, the feed design and the inverted
configuration.
Basically, the design uses travelling-wave slots arrays because of their wide
bandwidth and low tolerance requirements. Two types of slot anays were
discussed, the hofizontally polarked, and the vertically polarized arrays using
longitudinal slots and transverse slots, respectively. TE modes are used to excite
the fnst type of slots, Mile TM modes are used to excite the latter. In general, it
was found that ushg the fundamental TE, mode had inherently better sidelobe
performance because higher order modes can be easily suppressed, The quasi-
TEM mode is also a fundamental mode. Howevet, 1 is readily shown that the array
is more frequency sensitive and 0. tends tu exhibl more beam squint across the
frequency band, thus it is suited only tu nanow-band applications. However, it is not
inconceivable that in certain applications, higher order modes can be used with the
lower propagation modes sufficiently suppressed.
Unikersïty of British Columbia Uectrical and Cornputer Eiigineeri'ng
In this work, the slot anay is excited by a parallekplate waveguide structure.
It is the p$po ratio of the waveguide that detemines the angle of the main beam. A
horizontaliy polarized anay and a vertically poIarUed array can have their main
beams collocated in space given that the appropriate Pd$, ratios are achieved
simuftaneously for the different modes and frequencies. In addition, it was found
that there is a limitation on the ratio of &/Pa and hence a limitation on the range of
O,. For most designs, the preferred range of $ is between 20' and 70'. This iç
because for smaller angles, the separation of the waveguide, a, requires tighter
tolerances, and consequently, $&JO varies more with frequency, hence the
bandwidth is limited. For Iarger angles, care must be taken to ensure that higher
orders are suppressed.
Two types of feeds were compared: the hom-fans arrangement and the
resonant slot waveguide. For the dual-frequency dual polarized antenna, a hom-
iens structure was selected because it was simple to design and can accommodate
wide bandwidths. In addition, it is suited for dual frequency and dual polarization
anays. However, it is bulky and the material for the Iens can be expensive for
manufacturing. A more attractive alternative would be the resonant slotted
University of British Columbia 8-3 EiedrÎcslr and Cornputer Engineering
Chopter 8 -Conclusion
waveguide, which is more compact and can be more economicaliy manufactured.
Its drawback is that 1 is limited for namw-band operation. This would be the
preferred feed for most applications that do not require wide bandwidths.
Travelling-wave dot anays are suited for use in the inveited configuration
where the radiating antenna surface is shielded from precipitation. It has been
shown that this significantly reduces the performance degradation of the antenna
under rain conditions. Here, the angle of the main beam, go, should be as small as
practically possible in order to achieve the smallest angle. a, for maximum shielding
effect.
8.2 RECOMMENDATIONS FOR FURTHER WORK
With the experience gained from this work, it is suggested that there are
three areas that may be worthwhile pursuing. They are the resonant slot waveguide
feed, alternative dielectric materîals, and circular polarization application.
EIedcicai and Cornputer Engineering
Chapter 8 -Conciusion
In the course of the experiments, the hom-lens arrangement, although
reliable, was curnbersome due to its size and weight. In hindsight, the resonant dot
waveguide feed would be preferable from an aesthetic point of view as well as for
ease of mounting. It is suitable for travelling-wave slot anays because it is relatively
compact and potentially lower in cost. Although the basic theory of resonant slot
anays was covered in the thesis, the actual effect of the slots radiating into a
paralteCpIate waveguide has not been thorough[y investigated. Four types of
resonant dots mentioned in the Chapter 4, narnely the longitudinal broadwall. offset-
transverse broadwall, centred-inclined broadwall, and centred-inclined sidewall, can
be used to couple power into the paraltel-plate waveguide. They should be
compared in t ens of suitability for the various modes, bandwidth, tolerances and
punty of modes in the parallebplate waveguide. The contribution to unwanted higher
order modes and the bandwidth must be examined. The limitations of the various
types of resonant slots and the achievable excitation tapers should be investigated.
An experimental investigation to verity these concepts would be valuable.
During the experimenta! hvestigation phase of the work, the experimental
antenna was difficult to assemble due to the lack d rigidity of parallel-plate
waveguide walls made from copper-cladded dielectric matem. Various expanded
poIystymne spacers were required throughout the parallel-plate structure to support
University of British Columbia 8-5 BectriCa[ and Cornputer Engineering
Chapter 8 -Conclusion
the plates. Furthemore, the tightening of the boks used for attaching the dielectric
rnaterial necessitated careful adjustments in order not to cause buckling. The
material currently used was selected for low loss and quick availability. But during
the course of this work, lower cost microwave materials such as the Rogers R04003
have become available. This and others types of Teflon fibreglass blends may be
suitable for this antenna. Although R04003 rnaterial may have a slightly higher loss
tangent, approximately 25% higher, the resulting additional loss is less than 0.1 dB.
Another additional specification to look for in selecting the dielectric is the change in
permittivity over temperature. For example, for the longitudinal slot anay of this
paiticular design. a mere 5% increase in permittivity equates to a shift of 0.5' in the
look angle. These are al1 factors to consider when evaluating the dielectric material
for a commercial antenna.
In the work on the dual-frequency dual-polarization application, it was shown
that coupling between the longitudinal slot army and the transverse dot anay is
negligible. Hence the possibil'Ry of a circularfy potarized array based on the
concepts discussed in Chapter 3 encourages further investigation. Future work on
circular polarkation application may include he use of electronic phase shifters for
electronic polarkation selection, which is a useful feature especially for direct
broadcastng service reception.
Universityof British Columbia Uectn*caC and Cornputer Engineering
There is a growing dernand for new types of Ka-band satellite terminal
antennas that are capable of meeting a number of challenging requirements and
desirable attributes. The work in this thesis investigates the use of travelling-wave
long-slot anays suled for portable and mobile applications. The main contributions
of this work are the combining of two long-slot anays into a single aperture for
transmitting and receiving at dual frequencies and dual polarkations and the
possibility of using the antenna effectively in rain.
An experimental version of a dual-frequency dual-polarized anay was
constnicted and tested. In general, it met the design requirements with some
limitations. This type of antenna would also be suitable for use in an Riverted
configuration to minimize the effect of moisture on the antenna in precipitation.
ln ternis of cost, the construction of this antenna is likely to be more
involved than some of the commonly used antennas such as the parabolic dish, and
hence it is expected to be more expensive. Whether the cost can be justified for
any particular application remains to be seen. Overall, the proposed design has
achieved its intended goals.
University of Briti'sh Columbia Uectncal and Cornputer Engineeflng
Appendix A THE EFFECT OF TBE SLOT LENGTH, 1, AND TB[E PHASE
COEFFICIENT RATIO ON THE VARIOUS RADIATION CHARACTERISTICS
Two parameters that affect the radiation pattern are the slot lengih I, and the
P$po ratio. Figure A-t compares the radiation pattern for slots of 1= 5 A,, and 10 A,, with
p$po = cos 0, = 1.1 5 (Le., 0, = 29.6" which is the etevation angle of the Advanced
Communications Technofogy Satellite (ACTS) for a Lower Mainland British Columbia
terminal). As the length of the slot increases, the radiation pattern becomes more
directbe and the angle of maximum radiation approaches 0,. Atso of interest is that the
sidelobe levels for the 1 = 10 h, are lower than those of 1 = 5 &. Figures A-2 to A-5
illustrate the effect of the stot length, 1, and the phase coefficient ratio P&, on the
various radiation characteristics The following can be concluded:
1) As the length of the dot, 1, increases:
a) The maximum angle of radiation approaches 8,.
b) The directivity increases.
c) The beamwidth decreases.
ci) The sidelobe levets decrease wiüi respect to the main beam for large
values of
2) As încreases:
a) The maximum angle of radiation increaçes, the relationship is
cos 8, = j3&, for large values of I.
b) The directivity remains fairly constant.
c) The beamwidth decreases slîghtly.
d) The sidelobe levels decrease with respect to the main beam.
It is also found that if attenuation is taken into account, the beamwidth and the
sidelobe [evels are increased, while the directivity is decreased; the angle of maximum
radiation is unaffected.
Figure A-1 Radiation Patterns of a 5 ko and a 10 ho Array
BCTRICAL AND ~ M P U T E R ENGINEERING
Siot Length (&)
Figure A-2 Angle of Maximum Radiation with 0, as Parameter
10 15 20
Slot Length (A,)
Figure A-3 Directlvity with pdpo as Parameter
Figure A94 Beamwidth versus && with Length as Parameter
Slot Length 1 (ho)
Figure A-5 Sîdelobe Level Relative to Main Beam Venus Slot Length with e0 as Parameter for a Uniform Slot
Appendix B
The effect of the dot decreases value of the waveguide phase coefficient, P,,
slightly. From [1 1, the equations for values of 4/h, for very narrow dots with zero wall
thickness are given by (BI) and (82).
The vafiables 8, and denote the phase coeffcient and the wavefength of the
waveguide wïthout a dot. Experimentaf resuks for A& were detemiined using the
following procedures:
1) The positions of the minimums are noted for a predetetmined length of slotted
waveguide with one end shorted.
2) Replacing the slotted waveguide with a unslotted waveguide of equal lengths
or shply, seal off the slot with copper tape, the shift in the minimum, s, is
obtained.
3) it can be shown that regardless of the length of the unslotted portion of the
waveguide. Here n can be any real integer. To determine the value of n, the
above procedures can be applied to a difierent length of waveguide with an
equal dot wîdth replacing n with m. Given Uh, is the same for the two slotted
waveguides, n and m can be detemined ushg trial and enor. This procedure is
most suited for slotted waveguides with short lengths (Le., a few wavelengths).
The experimental and calculated results are show in Figure B-1. The data are
obtained for f =29.6 GHz, a = 7.1 1 mm, b = 3.55 mm, and I = 10 cm.
0.01 5 0.02 0.025 0-03 0.035
Slot Width w (mil)
Theoetical
x Experimentai
Figure B-1 Theoretical and Experimental Phase Coefficient
References
(1) L.O. Gofdstone and A. A. Oliner, "Leaky-Wave Antennas I: Rectangular Waveguidest',-/RE Transucfions on Anfenncrs dnd Propagafiin, Oct. 1 959, pp. 307-3 1 9,
Appendix C
The attenuation coefficient, a. depends on the amount of radiation which is
determined by the width of the dot, must be taken into consideration in order to obtain
a more accurate estimation of the radiation pattern. The attenuation coefficient a is
gïven in (C-1) through (C4) [Il for slot widths wcc b (b is the nanow dimension of the
waveguide) and zero wall thickness.
where y = 1.781 and e = 2.71 8.
Experimental results are shown in Figure Cl. Also shown in this figure, for
comparison purpoçes, are the computed values using equations from [Il, aithough they
are not strictly valid for the range of slot widths used. This range is dictated by machine
shop limitations. The data are obtained for f =29.6 GHz, a = 7.1 1 mm, b = 3.55 mm,
and I = 10 cm.
The discrepancy between the theoretical and measured values is due to the
range of slots widths used. The theoty in [Il is based on a vev narrow slot width
relative to the wavelength. Due to the frequencies of interest and the capabilities of the
machine shop, the range of dot widths used in the experïment were outside the range
where the theoiy was valid. Another source of error was due to the finite thickness of
the waveguide wall. But over all, the measured resuits agreed with the theoretical
results in that as the slot width increased so did the attenuation coefficient.
0.01 5 0.02 0,025
Slot Width w (mil)
Theoretical
Experirnental
Figure Cl Theoretical and Experimental Attenuation Coefficient
(1) L.O. Goldstone and A. A. Oliner, "Leaky-Wave Antennas 1: Rectangular WaveguidesJ:-/RE Tranactlons on Anfennus und Propagcrfion, Oct. 1 959, pp. 307-3 1 9.
Appendix D
For travelling-wave antennas, çince the wave attenuates as it propagates. the
attenuation coefficient must be taken into account when calculating the radiation
pattern, The attenuation adds an exponentially decaying taper along the direction of
propagation. The result is increased sidelobe levels. Figure D-1 shows the effect of
attenuation on the sidelobe levels for a 25 larray with a unifon distribution. Note that
in general, the reflected wave decreases with increased attenuation white the other
sidelobes increase with increased attenuation. One can show that this is the general
affect is the same for other excitation tapers.
ELECTRICAL AND &WUTER ENGINEERING
- 1 neperlm e (Degrees) - - - 3 neperslm - - 5 neperslm -- 10 nepers/m
Figure D 4 Radiation Pattern with Attenuatlon Coefficient as a Parameter
Appendix E THE EFFECT OF AMPUTUDE TAPER ON SIDELOBE LEVEL
It has been shown [35] that it is possible to change the sidelobe level of an array
by applying amplitude taper. Figure E-1 shows the effect of various types of tapers on
sidelobe levels for a 60-element anay. Note that in general, the lower the sidelobe
levels, the wider the main beam and hence the lower the directivity.
- Uniform - - - Triangular - - Cosine - - Square Root
Figure E-1 Radiation Pattern with Attenuation Coefficient as a Parameter
Appendix F
The following are rneasured and calculated radiation patterns for long slots of I=
100 mm at f = 29.6 GHz.
- Theoretical - - Experirnental
Figure F-1 Cornparison of Measured and Predicted Radiation Patterns at 29.6 GHz for 1 = 100 mm and w = 0.46 mm
- Theoretical - - Experimental
Figure F-2 Cornparison of Measured and Predicted Radiation Patterns at 29.6 GHz for f = 100 mm and w = 0.89 mm
Appendix G CALC~LATIONS FOR COUPLING OF SLOTS
The coupling of the dots was investigated using the multi-pipe model. Each slot
cross-junction was model as a junction of pipes as shown in Figure 3.34. The
following is a Mathcad Program used to determine the coupling. First each dot is
modelled as a microstrip line which in tum is modelled by 3 pipes. Each pipe is divided
into 10 segments. The centre of each physical segment is linked to the next via a
cunent segment. The Green's function for each current element is obtained. The pipe is
modelled as a series of current elements with the cunents at both ends equalling to zero.
This is to simplify calculations and given that the ends are suficiently far from the
juncüon, it has negligible effect on the solution. The current segments fom an
intersection at the junction where the sum of the cunents, according to Kirchoffs law must
equal zero. Given these conditions, the method of moments is applied to obtain the
impedance mat& for the structure. The impedance mat& consisüng of sef4mpedoi~es
and mutual ïmpedances can be used to calculate the mutual couplhg between various
segments. Once the impedance matrar is obtanied, an excitation cm be applied to ma&
to simulate the excitation of one siot. The resuiting cunent mat& is used to detemine
the couphg to the other slot.
The following program is the implementation of what is described above. It was
found that for a unifomly excited slot, or for an infinitely thin slot. there is theo retically no
coupling to the cross-slot. Howevet, any asyrnmetry will resuit in a potential difference
across the width of the crosîslot resulting in coupling. The coupling is dependent on two
factors, the width of the stots and the excitation of the excited slot. In conclusion, the
wider the stotç, or the greater the asyrnmetry of the excitation, the greater the coupling.
Hence in the case of a travelling-wave dot, the attenuation along the slot, as well
as any taper applied ta the slot excitation will resuit in coupling. As part of the
investigation, the coupling for various attenuation coefficients and varîous slot widths
were calculated for the frequency of interest. The results are shown in Figure G-1 .
From this graph, one can conclude that the coupling behnreen the stots is negligible.
Filename: mm 1 .mcd
This file calculates the coupling due to a crossed slot using the Momenfs Method.
P = 3 Number of pipes M =10 Number of segments per ami N = 2-M
r =39std' Frequency of lnterest o =2-a.f
Length of Wire
lnterval Length
Defining Segment Intervals
n1 = I..M ml = I.,M n2 =M+ 1.2-M m 2 : = M + 1.. 2-M Defining the spatial CO-ordinates of each segment (beginning, middle and end) Note that the centre of the junction is the origin.
AI nny,, =(nI)-Al- -
2 l
nnxnc = - 2
AI npyn, = (nl )dl t - 2
""Y, = - 2
Current Segments
. - 1 MY^, -- 2
npxl, = (n2 - M)-AL
I VY~, = -
2 Calculating the spacing of pipes based on Green's Function
1
I-5*49910-' Distance of the pipes relative to the centre of the microstrip
Distance From Point m to z IL - I
Distance From Point m to z
Calculatnig Self and Mutual lmpedances
Defining lmpedance Matrix
= lmpedance Matroc
Defhing Voltage Vector with Excitation in on Slot
Defining Cunent Elements
Calculating Coupling
Power in Segments with Excitation P \ 2
c,., ' ('c.n) Power ni Coupled Segments
5 tO 15
AttcnuaUon Coefficient (Neperyrn)
- 5 mi1 (0.127 mm) " ~OmiI(0.254mm) - 20 mil (0.508 mm) - 30 mil (0.762 mm)
Flgure 0-1 Calculated Coupling Vs Attenuatlon Coefficient with Slot Width as Parametet
ELECTRICAL AND COMPUTER ENGINEERING
Appendix H
The comgated horn is the preferred choice of antenna feeds for high-
performance communication systems [I 1. It is superior over than the conventional hom
in sidelobe and cross-polarkation performance because it reduces the fields reflected
off the flared walls. In this application, the horn is used a rectangular waveguide to
parallel-plate translion. Nevertheless, the basic principle of a comgated still applies.
The comigations act as short transmission Iineç where the short circul at the
end is transfened to an open-circuit at the comgation boundaiy. In general, this is
valid for only one frequency. A less stringent approach is to force the tangential
magnetic fields to zero thus preventing surface wave and reduce diffraction. This can
be accomplished by designing the surface of the comigations to simulate a capacitive
surface reactance. A comgation depth, 4 of behveen 0.25b and O.&, where À, is
the free space wavelength, will transfomi the short circuit from the back of the
comgation to saüsfy this reactance requ irement [Z].
There is an implicl assumption in the analysis above: the width of the dots in the
comigations is relatively thin compared to the wavelength. As recomrnended by [21, to
fonn an effective comgated surface, the spacing should be 8 or more comigations per
waveguide wavelength, &. This requirement is onIy necessary at the onset since the
energy is forced away from the flared walls by the comgations. Thicker vanes, f, and
larger spacing such as two €0 four comgations per A, can be used after the first 20
comgations. Wlh the added comgations, the flared walls no longer present a
conducting wall boundary. The resuling wave emitting from this structure assumes a
cosine distribution that has the added benefit of tapering the radiation pattern and
reducing the teakage from the sides of the parallel-plate waveguide.
References
[Il A. D. Olver, "Corrugated Homs", Electronics & Communication Engineering Journal, Febutary, 7992. pp. 4-1 0.
[2] C. A. Mentzer. and L. Peters, Jr., "Properties of Cutoff Comgated Surfaces for Comigated Hom Designn, lEEE Transactions on Antennas and Propagation. Vol. AP- 22, No. 2, March 1974. pp. 191-1 96.
Appendix I
The lens profile can be detemiined using the ray model as illustrated in Figure I-
l . The collimating action of diverging rays by a lens is achieved by wave velocity
retardation as the wave propagates into the dielectrÏc medium of the lem. A plane
wave is obtained if the accumulated phase of any ray along its path from the source to
a plane in front of the Lens is equal to that of any other ray. The curvature of the lens
required to obtain an aqui-phase front can be derived from Equation (1-1).
where 4, is the wavelength in the dielectnc. The collimating action of a lens is shown
in Figure 1-2. This equation can be approximated by Equation (1-2).
Lens
Figure 1-1 Ray Theory of Lens
In general, when a hom is osed to excite a particular mode in a parallel-plate
waveguide, one dimension of the flared waveguide is still relativelysmall, thus 7c, and &
may vaiy differently with frequency. Here, the lens design is bandwidth Iimited.
For a TEM mode excitation, however, which uses an H-sectoral hom, the
aperture dimensions are large enough such that value of h, is approximately that of li,
and & is appmximately h, / elR, where e is the permittivity of the lens,. Here Equation
(1-2) can be fumer reduced:
- f + x n = op,
where n = A,& = dR , is the index of refraction. Note that Equation (1-3) is frequency
independent, hence the lens used for the TEM mode is broadband.
Figure 1-2 Collhating Action of Lens
Appendix J
Two basic rnethods of achieving a matched surface can be used. These are:
1) By appropriate location of metallic obstacles on the surface of the lens, shunt
susceptances can be realised. This reactive wall is placed inside the surface of the
lens, which can be composed of a grid of thin wires, or an anay of thin conducting
disks. Wires have an inductive reactance and tharefore must be embedded 118
wavelength within the dielectric. Disks are capaclive thus they are embedded 318
wavelengths in. It has been shown that this method is very effective [Il [2] but for the
high frequency range required in the present application, it would be vety expensive to
ach ieve the required accuracy.
2) A more suitable method is by use of quarter-wave matchhg [3] [4] [5]. If the
lens is coated with a solid layer of dielectrÏc of thickness, d= h, /4, such that the wave
impedance of the wave in air, &, in the dielectric coating, 21, and in the lem, 4 are
related by:
Often it is difficult to find a dielecttfc coating with the desired permittivity thus a
convenient technique is to simulate this layer by perturbing the shape of the dielectric
boundary. The simulated surface can be composed of comgations of arrays of small
dielectric obstacles such as cylinders or holes. The properadjustment of the depth and
dimensions of the perturbation can obtain a match at the desired frequency and the
angle of incidence. For the ease of construction, the type of perturbation selected are
slots of thickness t and spacing O in the surface of the lens as shown in Figure J-1.
Depending on whetherthe comgations are perpendicular to the E-field as in the case
of an E-sectoral hom or parallei as in an H-sectoral horn, the following equations are
used to obtained the desired penittivity:
To reduce the number of slots, large spacing of D can be used provided that
where 0 is maximum flare angle of the incident wave from normal. This is to avoid
exciting higheporder modes in the lens. By choosing a proper t/D, and using equations
($2) to (J4), the desired effective permMivity can be approximated.
Figure J-1 Matching of Lens by Perturhtion
References
[Il E. M. T, Jones et al., "Measure Performance of Matched Dielectric Lensesn, IRE Transactions on Antennas and Propagation, Jan., 1956, pp. 31 -33.
(21 E. M. T. Jones and S. B. Cohn, 'Surface matching of Dielectric Lenses", Journal of Applied Physics, Volume 26, Number 4, Apr., 1953, pp. 452-457.
[3] T. Monta and S. B. Cohn, "Microwave Lens Matching by Simulated Quarter-Wave Transfomiers", IRE Transactions on Antennas and Propagation, Jan., 1956, pp. 33-39.
[4] R. E. Collin and J. Brown, "The Design of Quarter-wave Matching Layen for Dielectric Suifaces, Proceedinos of lnstitute of Electrical Engheers, vol. 103, part C. Sept., 1955, pp. 153-158.
[5] J. A. Cummins, Side Lobe Reduction in the Radiation Field of Lens Conected K Plane Homs, Master's Thesis, Laval University, August 1960.
Appendix K MATHCAD PROGRAM FOR DESIGN OF BENDS
The design of the bend is based on a method by Weishsshaar [Il, which
combines the method of moments and a mode-matching technique for the analysis of a
bend in a parallel-plate waveguide. This method requires only a few expansion tenns
to achieve accurate solutions. The following program is written for the TM wave; a
similar program is used for the TE wave. Figure K-1 shows the results for the retum
loss of a bend based on the program below.
Microwave Bends - TM mode in parallel waveguide bendstm.mcd
po IO' a = ï - ~ o - ~ Waveguide Height W =a
n =5 Size of Matrix 1 = I o Number of Frequency Points m.= 1-1
fm = ~ t < P + r n d
I . = i&ntity(n-I) Define ldentity Mafa
Radius of Bend
Length of Bend in Radians
z - - lia((fm))*wpi.,,, Defining Diagonal lmpedance Matrix $+(m- l h i r l r n - l h ;
2' k( $,)'
Nomialized Transverse Eigensolutions Will be used as basis and testing functions
C
W J - i Defniing the Following Matrices
L
fQim = (km)' Eigenvalues D P ~ t m - 1)5> = eigenvec (A ( fm) , $q*m) E y D ~ ~ - Q - D I
k(mj2 Y7frn- 1 1 - n . i ~ f m - 1)-II Defining the Admittance Matrix
m(fm) .P O*&.
Let Region i be region before bend. Let Region II be region after bend. Let Region III be region of bend. a = Nomalized coefficient of forward propagating wave of the various modes b = Nomalized coefficient of backward propagating wave of the various modes A =j .vTS-l.~T-~, B -2,-D
Defining a forward propagating wave in the fist mode as exciting wave in region 1. Al1 other modes set to zero. d~+m- Il-* = t
No excitation in Region II. U N ~ S ~ O F B R ~ S H Cocuveu
Forming the Generalized S matrix for bend:
A section of length L is used to connect to the bend.
Length of Section Connecting the bend L =4.@103
Definhg the Matrix of the Section
Combining a bend with fwa sections
F~quency (GHz)
Figure K-l Return Loss for the TM wave Bend as Described in Appendix K
References
[Il A. Weisshaar et al., "A Rigorous and Efficient Method of Moments Solution for Curved Waveguide Bands", EEE Transactions on Microwave Theoiyand Techniques, Vol. 40, No. 12, Decernber 1992.
Appendix L
Figure L-1 shows the mounting configuration used for rneasuring the E-plane of
longitudinal dot array and the H-plane of the transverse dot array. Figure L-2 shows
the mounting configuration used for measuring the H-plane of longitudinal dot anay
and the E-plane of the transverse slot anay.
Antenna
Figure L-1 Mounting Configuration I
Turntable
Figure L-2 Mounting Configuration II
