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2000 IEEE International Solid-State Circuits Conference 07803-5853-8/00 ©2000 IEEE WA 20 .1 A DC Measurement IC with 130nV pp Noise in 10Hz Axel Thomsen, Edwin de Angel, Sherry Xiaohong Wu, Aryesh Amar 1 , Lei Wang 2 , Wai Lee Crystal Industrial and Communications Division, Cirrus Logic, Austin, TX 1 Crystal Industrial and Communications Division, Cirrus Logic, Nashua,NH 2 Texas A&M University, College Station, TX A CMOS DC measurement IC surpasses the noise per formance of prior-art integrated systems. The instrumentation amplifier sur- passes commercially-available stand-alone inamp solutions in 0.1 to 10Hz noise performance. It is targeted for bridge transducer measurements where signal levels are typically on the order of a few mV. Low noise is crucial at these signal levels. In prior-art, inte- grated DC measurement systems the wideband noise of the instru- mentation amplifier is sampled directly without anti-aliasing and thus noise performance is sacrificed for a simpler interface between amplifier and modulator [1,2]. Figure 20.1.1 shows a block diagram of the IC consisting of a p rogrammable gain instrumentation ampli- fier followed by a 4 th -order ∆Σ modulator, a programmable decima- tion filter, and a 3-wire serial interface. The inamp uses the multipath feedforward architecture presented in Reference [3] that offers flexibility in low frequency applications because it separates the low frequency signal path from the high frequency path de- signed for stability. The architecture is modified to reduce the offset introduced by the second stage of the amplifier. It is also modified according to the reduced bandwidth requirements in the DC mea- surement application. Special attention is given the chopper stabi- lization. The opamp architecture used in the inamp is shown in Figure 20.1.2. Multipath feedforward compensation allows the cascading of sev- eral integrators to achieve high gain. It achieves stability by bypassing slower integrator stages at higher frequencies by provid- ing a unity gain path around each integrator. Many arrangements of integrator stages are possible and after gain and stability require- ments are met, more subtle issues determine the optimal arrange- ment of stages. The first stage I1 is chopper-stabilized and must be a low bandwidth integrator to allow filtering of chop artifacts. But since its input transconductance sets the noise performance of the amplifier, it must be large to meet the noise target. To avoid an external capacitor and noise pickup problems that may arise, in Reference [3] this stage is followed by an attenuator that reduces the effective bandwidth of the stage by 1/128x from the 2.56MHz implemented as gm in /C load to 20kHz. But this arrangement leads to a more significant effect of the second-stage offset. A 5mV offset in the second stage must be overcome by –640mV of offset at the output of the first stage. Even with 80dB of gain in the first stage, this leads to an unacceptable 64µ V input offset. This arrangement uses a 1/64x attenuator and places a low-bandwidth second integrator I2 before the attenuator. In this arrangement the integrator provides the drive to cancel the offset of the subsequent stages reflected through the attenuator. The first stage I1 sees only the offset of the new integrator I2. The implementation of the chopper is as follows: Input chop switches are regular voltage switches (Figure 20.1.3). The output chopper is implemented as current steering, rather than as voltage switching, to minimize artifacts at the chop frequency. The voltage swings on the switches p1, p2, n1, n2 are reduced to an effective gate-v on voltage. The phasing of clocks 1 and 2 is simple non-overlap. Clock phases n1 and p1 start after and end before 1, n2 and p2 maintain similar relations to 2. The phases nx and px direct the current to the supply during the crossover switching at the input. Chop frequency is 38.4kHz to maintain input currents below 1nA. The 0.1-10Hz noise results show the effectiveness of the chop circuitry in sup- pressing low-frequency noise (Figure 20.1.4). Noise below 10Hz is kept to within 130nV pp with 20nV measured RMS noise. The offset drift is <70nV/ °C. To maintain gain accuracy of the DC measurement system in the presence of an anti-alias filter, a rough-charging scheme is required (Figure 20.1.5). Otherwise the switched-capacitor current flowing through the anti-alias resistor creates unacceptable temperature coefficients. A signal-dependent rough-charge buffer scheme mini- mizes power consumption and maximizes voltage swing into the ADC. This buffer pre-charges the 16pF of ADC input capacitance so that the path through the anti-alias resistor has to provide for only the remaining charge error. The implementation of the rough-buffer takes advantage of the asymmetric behavior of a two-stage class-A opamp in response to a step on the output. A comparator is used to detect the polarity of the input signal during a preceding clock. Assuming vamp+ is greater than vamp-, all switches labeled p are closed. A switched-capacitor connected to vadcR+ has to be pulled up to vamp+ by the rough-buffer. VadcR+ is pulled down initially by this switched-capacitor, but this downward glitch couples through to node B. With the selected active-pMOS output stage, the pull up is accelerated. Similarly, the active-nMOS output stage efficiently pulls down. The rough-buffer input stages are of a chopper stabi- lized, rail-to-rail folded cascode design. The resulting measured gain drift is 15ppm/ o C. Theswitched-capacitor Σ∆ modulator is chopper stabilized to achieve 70nV/ √Hz in the band of interest. It is dynamically biased as described in Reference [4] to minimize power consumption while providing a wide instantaneous dynamic range. The presence of switched-capacitor circuits and clocked currents due to dynamic biasing do not adversely affect the noise performance of the inamp. Figure 20.1.6 shows a spectrum with a full-scale 20Hz input signal at the 64x gain and 240Hz output word rate setting. The noise rolls off slightly with frequency due to the droop in the Sinc 3 decimation filter response. No appreciable 1/f noise is detectable down to 0.01Hz. The noise density is 6.2nV/ √Hz. The THD is 101dB. The SNR (not counting the harmonic components) is 112dB. At the 64x gain settling, the full-scale input signal is only 28mVrms. Since the inamp is programmable from 2x to 64x gain, this represents a total of 142dB dynamic range over the 120Hz bandwidth. The digital filter uses as its fastest clock the modulator output bit- rate. CIC filters reduce hardware complexity and thus noise[5]. All analog signals are fully-differential to avoid pickup of digital noise. The 16mm 2 chip micrograph is shown in Figure 20.1.7. Power consumption is 41mW from ±2.5V analog and +5V digital supplies. References: [1] Mc Cartney, D. et al, “A Low-noise, Low-drift Transducer ADC”, JSSC, July 1997, pp. 959-967. [2] Kerth, D. A. D. S. Piasecki, “An Oversampling Converter For Strain Gauge Transducers”, JSSC, Dec. 1992, pp. 1689-1696. [3] Thomsen, A. “A 5 Stage Chopper Stabilized Instrumentation Amplifier using Feedforward Compensation”, Proc. VLSI Circ Symp, Honolulu, June 1998, pp. 220-223. [4] Kasha, D. B. et al, “A 16mW, 120dB Linear Switched Capacitor Σ∆ Modulator with Dynamic Biasing”, JSSC, July 1997, pp. 921-926. [5] Hogenauer, E. B. “An Economical Class of Digital Filters for Decimation and Interpolation”, Trans. Acoustics Speech Signal Proc ., April 1981, pp. 155- 162.
