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DECLARATION OF THESIS / UNDERGRADUATE PROJECT PAPER AND COPYRIGHT
Authors full name : NURHAFIZATUL AK MAL BT MD YUSOH
Date of birth : 15TH SEPTEMBER 1985
Title : DESIGN OF 2.4GHZ POWER AMPL IFIER FORTELEMEDICINE TRANSMITTER
Ac ademic Session : 2007/08
I declare that this thesis is classified as :
I acknowledged that Universiti Teknologi Malaysia reserves the right as follows :
1. The thesis is the property of Universiti Teknologi Malaysia.2. The Library of Universiti Teknologi Malaysia has the right to make copies for the purpose
of research only.
3. The Library has the right to make copies of the thesis for academic exchange.
Certified by :
SIGNATURE SIGNATURE OF SUPERVISOR
850915065330 DR. IR ING EKO SUPRIYANTO(NEW IC NO. /PASSPORT NO.) NAME OF SUPERVISOR
Date : 12 MAY 2008 Date : 12 MAY 2008
NOTES : * If the thesis is CONFIDENTIAL or RESTRICTED, please attach with the letter from
the organisation with period and reasons for confidentiality or restriction.
UNIVERSITI TEKNOLOGI MALAYSIA
CONFIDENTIAL (Contains confidential information under the Official SecretAc t 1972)*
RESTRICTED (Contains restricted information as specified by theorganisation where research was done)*
OPEN ACCESS I agree that my thesis to be published as online open access(full text)
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I hereby declare that I have read this thesis and in
my opinion this thesis is sufficient in terms of scope andquality for the award of the degree of Bachelor of Electrical (Electronic) Engineering
Signature :Name of Supervisor : Dr. Eko Supriyanto
Date : 12 May 2008
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DESIGN OF 2.4GHZ POWER AMPLIFIER FOR TELEMEDICINE TRANSMITTER
NURHAFIZATUL AKMAL BT MD YUSOH
A thesis submitted in fulfillment of the
requirements for the award of the degree ofBachelor in Electrical Engineering (Electronic)
Faculty of Electrical Engineering
Universiti Teknologi Malaysia
MAY 2008
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I declare that this thesis entitled Design of 2.4GHz Power Amplifier for Telemedicine
Transmitteris the result of my own research except as cited in the references. The thesis
has not been accepted for any degree and is not concurrently submitted in candidature ofany other degree
Signature :
Name : NURHAFIZATUL AKMAL BT MD YUSOH
Date : 12 MAY 2008
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Specially dedicated to my beloved family who inspires me throughout my journey in
education
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ACKNOWLEDGEMENT
Praised be to Allah for His blessings and giving me the strength along the
challenging journey of completing the project as well as this thesis writing, for without it,
I would not have been able to come this far.
First and foremost, I would like to take this opportunity to express my heartfeltgratitude to my supervisor of this project, Dr Ing Eko Supriyanto who has relentlessly
and tirelessly assisted me in completing this project and has been a good mentor for me,
giving me moral supports and patiently guided me throughout the project. My utmost
thanks also go to my family who has given me endless support and encouragement
throughout my academic years in UTM. My special thanks to these individuals, Mr
Teguh, Al Amin and Lindawati who have brought me into the world of
telecommunication and ADS which is something that really new to me. I may have
stumbled and tripped along the way, however your dedication and patience has made my
learning ladder an easier one to climb.
To all my friends, especially to Haslinah, Asmida, Norafeezah and Zaharah who
have helped and supported me along the way and have willingly offered their help despite
their tight schedule, thank you from the bottom of my heart. Finally yet importantly, to
those who have, in a way or another contributed to the pleasant months of my final year
project. Your presence and your countless effort and support had given me great strength
and confidence.
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ABSTRACT
Telemedicine transmitter is a very high power transmitter generally
designed to take a data, audio and video source like ECG, EEG and EMG,
and transmit them for a very long range from the patient home at the
remote area to the medical center. Thus, in order to make sure that the
transmitter having a very high power and data transmission, power
amplifier comes up as the very important part to be considered. Therefore,the main purpose of this project is to design a 2.4GHz power amplifier
with ouptput power of 32.0828dbm. This output power was obtained
through the propagation equation in ensuring the transmission of data
about 40km of distance. The stability and the matching network of the
circuit were also taken into account in designing it. ADS is the software
that was used to design it and TX-line 2003 calculator was used to
measure the size of the stubs. I hope that this thesis is able to give
sufficient information to anyone who is interested in learning about
designing 2.4GHz power amplifier.
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ABSTRAK
Pemancar telemedik merupakan pemancar berkuasa tinggi yang di
rekabentuk untuk mengambil dan memancar data audio atau video seperti
ECG, EEG dan EMG, lalu memancarkan data-data tersebut pada suatu
jarak yang jauh iaitu dari rumah pesakit di kawasan pedalaman ke hospital
atau pusat perubatan. Oleh itu, bagi memastikan bahawa pemancar itumempunyai kuasa yang cukup tinggi, penguat kuasa muncul sebagai
elemen yang penting dan harus dititikberatkan. Jadi, tujuan utama projek
ini adalah untuk merekabentuk sebuah penguat kuasa berfrekuensi 2.4GHz
dengan kuasa keluaran sebanyak 32.0828dbm. Nilai kuasa keluaran ini
didapati melalui persamaan penyebaran untuk memastikan data dapat
dipancar sejauh 40km. Kestabilan dan penyesuaian masukan dan
keluaran litar turut diambilkira semasa proses merekabentuk penguat
kuasa ini. ADS merupakan perisian yang digunakan untuk merekabentuk
litar penguat kuasa ini manakala kalkulator TX-line 2003 digunakan untuk
mengira saiz stab. Adalah diharapkan agar tesis ini dapat memberi
maklumat dan rujukan kepada sesiapa yang berminat untuk mempelajari
tentang cara-cara untuk merekabentuk penguat kuasa berfrekuensi
2.4GHz.
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TABLE OF CONTENT
CHAPTER TITLE PAGETITL E iDECLARATION iiDEDICATION iii
ACKNOWLEDGEMENT ivABSTRACT vABSTRAK viTABLE OF CONTENT viiLIST OF TABLES viiiLIST OF FIGURES ixLIST OF ABBREVIATIONS xLIST OF APPENDICES xi
1 INTRODUCTION 11.1 Background 11.2 Problem statement 31.3 Project Objective 31.3 Project Scope 4
2 METHODOLOGY 52.1 Flow diagram 52.2 Work breakdown 6
2.2.1 Study 72.2.1.1 Advanced Design System 8
2.2.2 Report writing 92.3.Gantt chart 9
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3 LITERATURE REVIEW 113.1 Introduction 113.2 Stability 213.3 Matching network 253.4 Stubs 27
3.4.1 Single stub 273.4.2 Design procedure 303.4.3 Double stub tuner matching 323.4.4 Stub tuners and E-H tuners 333.4.5 Some comments based on feedback 33
3.5 Microstrip Transmission Line and Design 353.5.1 Microstrip parameters 373.5.2 Some Considerations in the Choice of Microstrip
Substrate Material40
3.5.2.1 Common substrate materials 413.6 Return loss 44
3.7 Voltage Standing Wave Ratio 443.8 Free Space Path Loss 45
4 DESIGN AND SIMULATION OF POWER AMPL IFIERCIRCUIT
46
4.1 Introduction 464.1.2 Select the transistor (noise, gain and price) 484.1.3 Measure the s-parameter of the transistor from
datasheet49
4.1.4 Determining the Stability Factor K 494.1.5 The gain and output value (output power of power
amplifier)51
4.1.6 Computing input and output matching network 534.1.6.1 Computing Input Matching Network 54
4.2 Analyzing the Transistor with Input and Output Matching 554.3 One Stage Power Amplifier 574.4 Two Stages Power Amplifier 584.5 The Amplifier Inclusion of DC Biasing Circuit 59
5 RESULTS AND ANALYSIS 605.1 Stability Factor, K 605.2 One Stage Power Amplifier 625.3 Two Stages Power Amplifier 655.4 The Amplifier Inclusion of DC Biasing Circuit 67
6 CONCL USION 72
7 RECOMMENDATION 73
REFERENCES 74
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APPENDICES 76
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LIST OF TABLES
TABLE NO. TITL E PAGE
4.1.1 A part of S-parameter for transistorFLL 351ME
49
4.2.1 Electrical length 55
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LIST OF FIGURES
FIGURE NO. TITLE PAGE
2.1 Flow Diagram 52.2 Work Breakdown 62.3 Gantt Chart 93.1 Scattering Parameters Analogy 133.2 Input and Output Matching Network 193.3 Two Port Network 21
3.4 A lossless network matching arbitrary load impedance to atransmission
25
3.5 Single-stub tuning circuit shunt stub 293.6 Single-stub tuning circuit series stub 303.7 Data Tranmission 464.1 Simulation of Transistor 504.2 Simulation Result of the Transistor 514.3 TXLINE 2003-Microstrip calculator 574.4 The amplifier With Input and Output Matching Network
(One Stage)58
4.5 The amplifier With Input and Output Matching Network
(Two Stages)
59
4.6 The amplifier With Input and Output Matching Network(With DC biasing)
60
5.1 Simulation of Transistor 615.2 Simulation Result of the Transistor 62
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5.3 The amplifier With Input and Output Matching Network(One Stage)
63
5.4 Stability Factor K 645.5 Maximum available gain 645.6 Output power 655.7 The amplifier With Input and Output Matching Network
(Two Stages)67
5.8 Maximum Gain 67
5.9 Output power 655.10 The amplifier With Input and Output Matching Network
(With DC biasing)
68
5.11 Voltage Standing Wave Ratio 695.12 Simulation Results 705.13 Output Power 71
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LIST OF ABBREVIATION
ADS - Advanced Design SystemdBm - Decibel in miliECG - ElectrocardiographyEEG - ElectrocephalographyEMG - ElectromyographyGHz - Gigahertz
PA - Power AmplifierPwrGain - Power GainRF - Radio FrequencyTEM - TranverselectromagneticVoltGain - Voltage GainVSWR - Voltage Standing Wave Ratio
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CHAPTER 1
INTRODUCTION
1.1 Background
Throughout the decade, there were a number of attempts to
develop medical information systems which are reliable, affordable and
accessible over the entire hospital and beyond. Telecommunication links
are becoming nearly ubiquitous but they do not reach all communities,leaving segment of the population unserved. But, most of the data
transmission is within a restricted distance and restricted frequency so that
the signal cannot be sent at further distance. Most of the available
telemedicine systems are distinctly "low tech" and slow. At present, most
teleconferencing is dependent on communication via fiber optic cable
connection and satellite. High-speed land lines, such as OC3 fiberoptic
cables, are not widely available at this time.Current availability is limited
primarily to large urban areas. Unfortunately, patients receiving care in
rural hospitals who could benefit most from teleconferencing are not likely
to be eligible for these services due to lack of fiber optic infrastructure.
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Satellite-based telemedicine could rapidly fill the gap now present
in the area of high-speed data transmission. Technology Satellite (ACTS),
which is capable of a 622-Mb/sec transmission rate, was recently used in
an experiment that linked physicians at Phoenix Children's Hospital in
Phoenix, Ariz, with consultants at the Mayo Clinic in Rochester, Minn.
But, the major problem in using satellite is that not all of the rural and
urban areas have that facility. So, we are developing a point to point
wireless communication to enable the telemedicine to be applied every
where. The wireless link is utilized to fulfill the need for patient mobility
in a remote area within a specified range of broadcasting and to transmit
real-time medical information and warning within an acceptable time limit
for critical life cases.Solid-state microwave amplifiers play an important role in
communication. Usually, signals provided by the transducers are weak;
typically, it is in the order of microvolt (V) or millivolt (mV). It is not
easy, and sometimes not possible, to have reliable processing for signals
with low levels. For this reason, the need for a signal amplifier arises. In
a transceiver circuit, a signal amplifier has different applications,
including low noise, high gain, and high power amplifiers.
In wireless RF transmitter, RF Power Amplifier (PA) is one of
important device that make many influence to the transmitter performance.
As the first stage of the transmitter, PA required to have certain power for
transmit the signals.
The focus of this research has been the design of power amplifier
using software and propagation equation to determine the appropriate gain
in ensuring the transmission of data.
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1.2 Problem Statement
Telemedicine transmitter is a very high power transmitter generally
designed to take a data, audio and video source like ECG, EEG and EMG,and transmit them for a very long range and in this project, the desired
range of transmission is 40km which considered as a very far of distance.
Nowadays, most of the data transmission is within a restricted
distance and restricted frequency. This is due to the power problem as for
the long range data transmission, the system needs a very high power.
Thus, in order to make sure that the transmitter having a very high power
and data transmission, power amplifier comes up as the very important
part to be considered.
1.3 Objectives of the project
To design a 2.4GHz power amplifier for telemedicine transmitterso that the physiological signals such as EMG, ECG and EEG information
can be transmitted for a 40km of distance which from the patient home at
the remote area towards the medical center.
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1.4 Scope of the project
The main aim of the project is to design RF BJT Class A Power
Amplifier. The transistor chosen for the job is FET FLL351 which comesin SOT-143 package. The maximum IDS sustainable by the transistor is
720mA and VDS = 10 V, with transistion frequency fT = 5GHz, which is
more than sufficient for the job.
The power ampifier using class A operations and suitable for
frequency at 2.4GHz operation.The output power is 33.0282dBm.
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CHAPTER 2
METHODOLOGY
2.1 Flow Diagram
Workflow of this project was divided to some parts like study,design and simulation. The flowchart and Gantt chart of thisproject as follows:
Figure 2.1 Flow Diagram
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2.2 Work Breakdown
The process flows and works have been planned were
separated according to the duration weeks, types of works,difficulties and importance for the project. For this project, all
design and simulation is using Advanced Design System (ADS)
software. Figure 2.2 shows the block diagram for work breakdown
in this project.
Figure 2.2 Work Breakdown
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2.2.1 Study
There are three important parts should be studied before
proceed the project. The parts are telemedicine transmitter, power
amplifier, propagation equation also (Advanced Design System)
ADS software. All those parts took very long duration weeks
because they are ongoing study. Other work like designing and
simulation was done as long as study all those parts.
Study of telemedicine transmitter scope took around seven
weeks. It includes search for telemedicine books, search
information from internet and interview. The telemedicinetransmitter scope were studied are types of data or signal from that
are being transmitted from patient home to the medical center and
output power of access point.
Study of power amplifier scope took around 4 months. It
includes search for characteristics for each type of power amplifier
and how to design to design a high power amplifier. The
characteristics of power amplifier scope were studied are gain,
bandwidth, efficiency, linearity and stability. The design of high
power amplifier scope includes measuring the s-parameter,
determining the stability factor, computing input and output
matching network.
The hardest part in this project is study the software where it
took around 5 months to determine the appropriate software to be
used. I actually had only about 8 weeks to explore the new
software (Microwave Office). However, I found that it was not
friendly user and I was not able to refer to anybody for any inquiry.
I tried out Ansoft Serenade 2000, but I failed to understand it
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better. I had no reference. Finally, I just made up my mind to try
out another software which was ADS (Advanced Design System).
I found that it is better software to design power amplifier. The
examples in the software library helped me a lots and make my
understanding better using stubs in designing power amplifier.
The scope were studied are characteristics of the software, how to
use the software and how to design the circuit using the software.
2.2.2.1 Advanced Design System
Advanced Design System (ADS) is the industry leader in
high-frequency design. It supports system and RF design
engineers developing all types of RF designs, from simple to the
most complex, from RF/microwave modules to integrated MMICs
for communications and aerospace/defense applications.
With a complete set of simulation technologies ranging
from frequency- and time-domain circuit simulation toelectromagnetic field simulation, ADS lets designers fully
characterize and optimize designs. The single, integrated design
environment provides system and circuit simulators, along with
schematic capture, layout, and verification capability - eliminating
the stops and starts associated with changing design tools in mid-
cycle.
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2.2.2 Report Writing
After completing the project, the report should be done. There
were two part of report which is progress report for Final Year
Project I and thesis for the whole project. It took about three weeks
for every report.
2.3 Gantt Chart
Figure 2.3 Gantt Chart
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CHAPTER 3
LITERATURE REVIEW
3.1 Introduction
Amplification is one of the most basic and prevalent microwave
circuit functions in modern RF and microwave systems. Early microwave
amplifiers relied on tubes, such as klystron and traveling-wave tubes, or
solid-state reflection amplifiers based on the negative resistance
characteristics of the tunnel or varactor diodes. But due to the dramatic
improvements and innovations in solid-state technology that have
occurred since the 1970s, most RF and amplifiers today use transistor
device such as Si or SiGe BJTs, GaAs HBTs, GaAs or InP FETs, or GaAs
HEMTs[1]-[4]. Microwave transistor amplifiers are rugged, low cost,
reliable and can easily integrated in both hybrid and monolithic integrated
circuitry. Transistor can be used at frequencies in excess of 100GHz in a
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wide range of applications requiring small size, low noise figure, broad
bandwidth, and low to medium power capacity. Although microwave
tubes are still required for very high power and/or very high frequency
applications, continuing improvement in the performance of microwave
transistors is steadily reducing the need for microwave tubes.
Transistor amplifier design will rely on the terminal characteristics
of transistors, as represented by either S parameter or one of the equivalent
circuit models. Here is discussed about some general definitions of two
port power gains that are useful for amplifier design and subject of
stability. These results will be applied to transistor amplifiers, including
designs for maximum gain and specified gain.
Solid-state microwave amplifiers play an important role incommunication where it has different applications, including low noise,
high gain, and high power amplifiers. The high gain and low noise
amplifiers are small signal low power amplifiers and are mostly used in
the receiver side where the signal level is low. The small signal S
parameter can be used in designing these low power amplifiers. The high
power amplifier is used in the transmitter side where the signal should be
at a high level to cross
The design procedures for a small signal microwave amplifier
consist of selecting the dc bias point for the transistor, measuring the S-
parameters of the transistor, studying the stability, designing the input and
output matching network to achieve the desired goals, building the
amplifier, and performing the measurements. The dc bias point of the
transistor should be determined first. The selection of the dc quiescent for
the transistor amplifier depends on the particular application
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A networks behavior at microwave frequencies can be
characterized using the scattering parameters (S-parameters). These
parameters are defined in terms of travailing waves,relate to the traveling
waves that are scattered or reflected when a network is inserted into a
transmission line.
Figure 3.1: Scattering Parameters Analogy
S-parameters are important in microwave design because they are easier to
measure and to work with at high frequencies than other kinds of
parameters. They are conceptually simple, analytically convenient, and
capable of providing a great insight into a measurement or design
problem.
The relationship between the S-parameters and the incident and
reflected waves can be expressed as follows
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Figure 3.2: Input and Output Matching Network
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A single stage microwave transistor amplifier can be modeled by
the circuit of figure 2 where matching network is used on both sides of the
transistor to transform the input and output impedance to the source
and load impedance and . The most useful gain definition for
amplifier design is the transducer power gain , which accounts for both
source and load mismatch. Thus, from the power gain, we can define
separate effective gain factors for the input (source) matching network, the
transistor itself, and the output (load) matching network as follows:
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3.2 Stability
We now discuss the necessary conditions for the transistor
amplifier to be stable. In the circuit in Figure 3.3, oscillation is possible if
either the input or output impedance has a negative real part: this would
imply that or . Because and depend on the
source and load matching networks, the stability of the amplifier depends
on the and as presented by the matching networks.
Figure 3.3: Two Port NetworkIn terms of reflection coefficients, the necessary conditions for
unconditional stability at a given frequency are
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The necessary and sufficient conditions for a two-port network to be
unconditional stable are [D. Woods, 1976]
In practice, most of the microwave transistor amplifiers are
potentially unstable because of the internal feedback. There are two ways
to overcome the stability problem of the transistor amplifier. The first is
to use some form of feedback to stabilize the amplifier. The second is to
use a graphical analysis to determine the regions where the values ofand (source and load reflection coefficients) are less than one, which
means the real parts of ZIN and ZOUT are positive. Substituting the
values of and in equations (3.7) and (3.8) and solving
for and result in the stability circles. The radii and centers of the
circles are given by [G. Gonzalez, 1984]
Output Stability Circle, input StabilityCircle,
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Then the stability circles need to be plotted in the Smith chart to
determine the stable regions or in other words, the regions where values of
and produce and . Most of the time, microwave
amplifiers used for narrowband or wideband applications face stability
problems at certain frequency ranges. Instability is primarily caused by
three phenomena: internal feedback around the transistor, external
feedback around the transistor caused by an external circuit, or excess of
gain at frequencies outside of the band of operation
These are defined by circles, called stability circles, that delimit
and on the Smith chart. The radius and center of the
output and input stability circles are derived from the S parameters on pg.
614 of Pozar or pg. 97 of Gonzalez. The concept of instability with
varying input or output matching conditions is significant, as we would
desire an amplifier to be unconditionally stable under all expected
conditions of source and load impedances. The example of input stability
circles is shown here.
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If an amplifier is conditionally stable, it can be renderedunconditionally stable by adding resistance to the input and/or output of
the amplifier so that the total loop resistance at the input and output is
positive. The use of resistive loading or feedback can compromise the
noise performance of an amplifier unless accomplished in connection with
an analysis of the amplifier noise figure.
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3.3 Matching network
After the stability of the transistor has been determined, and the
stable regions for and have been located on the Smith Chart, the
input and output matching sections can be designed. Since is fixed for
given transistor, the overall gain of the amplifier will be controlled by the
gains, and , of the matching sections. Maximum gain will be
realized when these sections provide a conjugate match between the
amplifier source or load impedance and the transistor. Because most
transistors appear as a significant impedance mismatch (large and
, the resulting frequency response will be narrowband.
Matching the impedance of a network to the impedance of a
transmission line has two principal advantages. First, all the incident
power is delivered to the network. Second, the generator is usually
designed to work into impedance close to common transmission line
impedances. If it does so it is better behave, the load impedance has no
reactive part which can pull the generator frequency, and the VSWR on
the line is unity or close to unity so the line length is immaterial and the
line connecting the generator to the load is non-resonant.
Figure 3.4: A lossless network matching arbitrary load impedanceto a transmission
Matchingnetwork
Load,
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Impedance matching is often being as a part of the larger design process of
microwave component and system. Impedance is placed between a source
and a transmission line or between a load and a transmission line. The
matching network is ideally lossless, to avoid unnecessary loss of power,
and is usually designed so that the impedance seen looking into the
matching network . Then reflections are eliminated on the transmission
line to the left of the matching network, although there will be multiple
reflections between the matching network and the load. This procedure is
also referred to as tuning. Impedance matching or tuning for the following
reasons:
Maximum power is delivered when the load is matched to the line(assuming the generator is matched), and power loss in the feed lines is
minimized.
Impedance matching sensitive receiver components (antenna, low noiseamplifier, etc) improves the signal-to-ratio of the system.
Impedance matching in a power distribution network (such as antennaarray feed network) will reduce amplitude and phase errors.
As long as the load impedance has some nonzero ea part, a
matching network can always be found. Many choices are available,
however, and will discuss the design and performance of several types
practical matching network. Factors that may be important in the selection
of a particular matching network include the following:
Complexity- As with most engineering solutions, the simplest design thatsatisfies the required specifications generally the most preferable. A
simpler matching network is usually cheaper, more reliable, and less loss
than a more complex design. Bandwidth-Any type of matching network can ideally give a perfect match
(zero reflection) at a single frequency. In many applications, however it is
desirable to match a load over a band of frequencies. There are several
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ways of doing this with, of course, a corresponding increase in
complexity.
Implementation- Depending on type of transmission line or waveguidebeing used, one type of matching network may be preferable compared to
another. For example, tuning stubs are much easier to implement in
waveguide than are multisection quarter wave transformers.
Adjustability- In some applications the matching network may requireadjustment match variable load impedance. Some types of matching
networks are more amenable than others in regard.
3.4 Stubs
Stubs are shorted or open circuit lengths of transmission line
intended to produce a pure reactance at the attachment point, for the line
frequency of interest. Any value of reactance can be made, as the stub
length is varied from zero to half a wavelength.
3.4.1 Single stub
If you look at the smith chart you will find a circle of constant realimpedance r=1 which goes through the open circuit point and the centre of
the chart. If you plot any arbitrary impedance on the SMITH chart and
follow round at constant radius towards the generator, you must cross the
r=1 circle somewhere. This transformation at constant radius represents
motion along the transmission line towards the generator. One complete
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circuit of the smith chart represents a travel of one half wavelengths
towards the generator. At this intersection point your generalized arbitrary
load impedance r + jx has transformed to 1 + jx', so at least the real part of
the impedance equals the characteristic impedance of the line. Note x' is
different from x in general.
At this point you cut the line and add a pure reactance -jx'. The
total impedance looking into the sum of the line impedance and -jx' is
therefore 1 + jx' -jx' = 1 and the line is matched.
Again, look at the SMITH chart and find the outer circle where the
modulus of the reflection coefficient is one. On this circle are the SHORT
and OPEN points, and all values of positive and negative reactance. The
resistance is zero everywhere. To generate a specified reactance, start at a
short circuit (or maybe an open) and follow around towards the generator
until the desired reactance is obtained. Cut the stub this number of
wavelengths long.
It is important to keep the total stub length as short as possible, if
wider bandwidths are required. Every time you add a half wavelength tothe stub length the reactance of the stub comes back to the same value. It
is good design practice to make stubs in the range 0 to 0.5 wavelengths
long. However, this may require an impractically short stub, so then one
can make the stub just a little over 0.5 wavelengths.
If one is allowed to use either short or open stubs at will, one can
always keep the total stub length in the range 0-0.25 wavelengths. A
length of transmission line of 0.25 wavelengths takes us half way roundthe SMITH chart and transforms an open into a short, or vice versa. On
microstrip it is usually easier to leave stubs open circuit, for constructional
reasons. On coax line or parallel wire line, a short circuit stub has less
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radiation from the ends: it is difficult to make a perfect non-radiating open
circuit as there are always some end effects on the line.
Figure 3.5: Single-stub tuning circuit shunt stub
Open orshorted stub
d
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Figure 3.6: Single-stub tuning circuit series stub
.
3.4.2 Design procedure.
It is told, or find out, the load impedance ZL and the transmission
line characteristic impedance Zo. Calculate the normalised impedance z =
(ZL/Zo). Plot it on the SMITH chart. You are told the frequency and the
velocity factor of the line. Calculate the wavelength in meters. (or cm).
d
Open orshorted stub
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Follow the circle of constant radius on the SMITH chart towards the
generator until the locus crosses the r=1 circle. Measure the number of
wavelengths along the perimeter of the SMITH chart between the z point
originally plotted, and the r=1 circle intersection. This tells you how far
from the load to place your stub.
Read off from the r=1 intersection the reactance x' value. Starting
from a short (or open) follow the r=0 circle around the outside of the
SMITH chart until you come to a point of reactance -x'. Measure the
number of wavelengths this represents from short/open end towards the
generator. Cut your stub this long.
The stub is placed in series with one of the transmission line
conductors. In coax this may be difficult to do technically. One therefore
often resorts to shunt stub matching, where the stub and the original
transmission line are connected in parallel. It is easier then to work in
admittances. We notice that the SMITH chart can be used as an
admittance chart merely by rotating it through 180 degrees. Normalised
resistance becomes normalised conductance; normalised reactance
becomes normalised susceptance. Admittances in parallel add; the shortcircuit point has infinite admittance and the open circuits point zero
admittance. The design procedure is the same as for series stubs.
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3.4.3 Double stub tuner matching
Suppose that the load impedance changes. Adjusting a single stub
tuner is very difficult. One has to remove the stub, remake the line where
the break was, and calculate the new stub length and point of attachment.
We can use two stubs permanently attached to the line at fixed
points of attachment, and tune by altering the stub lengths. Two values
have to be matched (r and x) and we have two variables; the length of each
stub.
As before, the generator-end stub has reactance -jx' and is attached
at a point where the line impedance, including the effect of the other stub
at its fixed point of attachment, is 1+jx'. Transforming the unit r=1 circle
towards the load until one reaches the load-end stub attachment, the circle
r=1 transforms to another circle, call it "B", touching the outside of the
SMITH chart, and also passing through its centre.
The load impedance, when transformed towards the generator up
to the load-end stub position, will be a generalised impedance ZL'
different from ZL. The effect of the load-end stub is to add reactance x"
to ZL' so that the impedance value ZL'+jx" lies on the circle "B" above.
We chose the length of the stub to make x" the required value for this to
happen. If we write ZL'=r'+jx' then the effect of adding the stub is to
move the reactance j(x'+x") along the constant r' curve depending on the
size of x".
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3.4.4 Stub tuners and E-H tuners
It is just possible for the r' curve not to intersect the circle "B", in
which case a double stub match is not possible for this value of load
impedance, and stub placements. Generalised adjustable tuners are
therefore designed with three stubs, which are spaced at unequal intervals.
Such a device is called a "Triple Stub Tuner". Sliding shorts are easily
arranged in coax or waveguide.
In waveguide only, there is a special type of tuner called an E-H tuner.This has shunt and series side arms consisting of sliding shorts, attached at
the same point along the guide. There is no equivalent in 2-conductor
transmission line for geometrical reasons. An E-H tuner can always match
any load impedance.
3.4.5 Some comments based on feedback
Stub matching is only desirable for relatively low fractionalbandwidths. For wider bandwidth matching a multi-section quarter wave
transformer can be used, or a tapered line. Impedance matching may be
carried out using the SMITH chart for calculations and design, and lumped
components taking the place of lengths of transmission line. It is possible
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to make undesirable reflections by using a "wrong" stub match, so care
must be taken in applying stub matching in high power (e.g. transmitting)
applications. It is always wise to measure the match before applying
significant input power. In antenna matching situations significant
mismatch can arise from alterations to the near-field environment of the
antenna over time. Thus if a new antenna is added to an existing mast, it
is always wise to check the matching of the pre-existing antennas.
There are practical difficulties at mm wavelengths, eg on
microstrip at above 20GHz. Here, the precision of adjustment of the
lengths of the stubs needs to be +/- 0.01 wavelengths for good quality
matching. At 5mm wavelength this is a precision of +/- 50 microns.There are also practical difficulties at high |gamma| (reflection coefficient
magnitude). Here the purpose of the stubs is to generate an equal and
opposite reflection to cancel out the reflection from the nearly completely
mismatched load. Clearly, to get effective cancellation, the stubs must be
very precisely chosen and constructed, and the fringing-field effects
become important to the point that they can dominate the design. A
standard SMITH chart calculation as in this page is then unlikely to be
very effective.
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3.5 Microstrip Transmission Line and Design
A Microstrip transmission line is a "high grade" printed circuit
construction, consisting of a track of copper or other conductor on an
insulating substrate. There is a "backplane" on the other side of the
insulating substrate, formed from a similar conductor. Looked at on end,
there is a "hot" conductor, which is the track on the top, and a "return"
conductor, which is the backplane on the bottom. A microstrip istherefore a variant of the two wire transmission line.
At frequencies above about 0.5 GHz, sections of microstrip
transmission line can be used to synthesize inductors and capacitors. In
general, open circuited stubs less than wavelength long behave as
capacitors and shorted or series sections less than 1/4 wavelength long are
inductors. The lumped element transmission model below helps explain
why its a good idea to use high impedance line ( 80 - 90 ohms ) for
inductors and low impedance line ( 20 - 30 ohms ) when making
capacitors.
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Given the equation for characteristic impedance of the line, it can be seenthat high im pedance lines contain high L and low C - the desired quality
for an inductor. Similarly, the model for low impedance line has high
values of C and low L.
If one solves the electromagnetic equations to find the field
distributions in the vicinity of a microstrip, one finds very nearly a
completely TEM (transverse electromagnetic) pattern. This means that
there are only a few regions in which there is a component of electric or
magnetic field in the direction of (as opposed to perpendicular to the
direction of) wave propagation. This field pattern is commonly referred to
as a Quasi TEM pattern.
Since some of the electric energy that is stored in this conductor
configuration is in the air, and some is in the dielectric, the effective
dielectric constant for the waves on the transmission line will liesomewhere between that of the air and that of the dielectric. Typically,
the effective dielectric constant will be 50-85% of the substrate dielectric
constant, depending on the geometry of the microstrip.
As an example, in (nominally) an air spaced microstrip, the
velocity of waves would be c = 3 * 10^8 meters per second. We have to
divide this figure by the square root of the effective dielectric constant to
find the actual wave velocity for the real microstrip line. At 10 GHz the
wavelength on that nominally air spaced microstrip is therefore 3 cm;
however, on a substrate with relative dielectric constant of 10, the
effective dielectric constant of the microstrip design may be 7, and the
wavelength is 3/(sqrt{7}) = 1.13 cm. Thus, for example, the maximum
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length for a microstrip "stub" to be used in stub impedance matching,
which is no more than half a wavelength, will be only 5.6 mm when
fabricated using this substrate.
A set of detailed design formulae and algorithms for microstripdesign is presented in T C Edwards, "Foundations for Microstrip Circuit
Design", Wiley-Interscience, New York, 1981. This reference also has a
"rough and ready" monogram for calculating the impedance of a
microstrip line using the dielectric properties and the geometry in the
picture above. The following analysis is developed from that source.
3.5.1 Microstrip parameters
The basic configuration of the microstrip is shown in the picture
above. One of the most challenging problems associated with this
configuration arises from the fact that the small strip is not immersed in a
single dielectric. On one side there is the board dielectric, and on the top
is usually air. The technique that has been developed to handle this
challenge uses, as was mentioned above, the concept of effective relative
dielectric constant, eff This value represents some intermediate value
between the relative dielectric constant of the board material, eff, and that
of air (assumed equal to 1) that can be used to compute microstrip
parameters as though the strip were completely surrounded by material of
that effective relative dielectric constant. One obvious advantage of themicrostrip structure is the "open" line which makes it very easy to connect
components. On the other hand, the configuration doesn't provide the
"shielded" signal line advantage of the stripline. Another advantage is that
microstrips can be packed together with fairly high density (multiple
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channels) with only minimal "crosstalk" interference, and therefore lends
itself well to RF and microwave IC design.
Aside from the difficulty of calculating the value ofeff , there is
another important effect. It is clear that eff will depend on both W and h.
Hence, the phase velocity along the microstrip will depend on these
parameters. Assuming the relative permeability of all materials in the line
design is well approximated by = 1, the phase velocity will be given
by:
Since the characteristic impedance (Zo) of the line will also depend
on these parameters, every time we need to design a microstrip with a new
characteristic impedance, we will be faced with the additional
complication of having to deal with a change in phase velocity (or delay
time) and consequently of the wavelength of waves on that microstrip.
Note that this is not a problem with coaxial cable or stripline design.
To get an idea of the range ofeff, consider the cases of a very wide
W and then a very narrow W. For a wide microstrip, nearly all of the
electric field lines will be concentrated between the metal planes, similar
to the case of a parallel plate capacitor that you studied in physics. Thus:
up = ceff
maximumeff= r
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On the other extreme, for narrow W the electric field lines will be about
equally divided between the air and the board dielectric so that:
minimumeff= 12(r+ 1)
This gives you a range:
Several different equations have been developed for use in calculating
characteristic impedance for microstrip design. Probably the most useful
are the following which are reported to be accurate to within about 1%:
Notice that these are relatively straightforward equations for the
calculation of characteristic impedance, given W, h, and eff . However,
the more useful calculation involves determination of the W/h ratio, given
required characteristic impedance. Here, then, is the design challenge
since the equations are transcendental (don't have a closed form solution)
for the W/h parameter. As you probably guessed, this is a job for
MATLAB and its powerful equation solvers.
Now, just to make things a bit more challenging, we'll introduce a
further "correction" to the above equations which is a consequence ofconsidering the finite thickness (t) of the microstrip. This correction is in
the form of an "effective" microstrip width (We), which is used to replace
W in those equations:
12
(r+ 1) eff r
Zo = 60eff
ln 8 hW
+ W4h
where eff=r+ 1
2
+ r- 1
2
1 + 12 h
W
-1/2+ 0.04 1 - W
h
2for W
h
1
or
Zo = 120eff
Wh
+ 1.393 + 0.667ln Wh
+ 1.444
where eff=r+ 1
2+ r- 1
21 + 12 h
W-1/2
for Wh
1
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These corrections are further subject to the restrictions that both of theserestrictions are usually easily satisfied in practice
3.5.2 Some Considerations in the Choice of Microstrip Substrate
Materials
Important qualities of the dielectric substrate for microstrip designinclude:
1. The microwave dielectric constant2. The frequency dependence of this dielectric constant which gives rise to
"material dispersion" in which the wave velocity is frequency-dependent
3. The surface finish and flatness4. The dielectric loss tangent, or imaginary part of the dielectric constant,
which sets the dielectric loss
5. The cost6. The thermal expansion and conductivity7. The dimensional stability with time8. The surface adhesion properties for the conductor coatings9. The manufacturability (ease of cutting, shaping, and drilling)10.The porosity (for high vacuum applications we don't want a substrate
which continually "outgasses" when pumped)
We = W + t
1 + ln 2ht
for Wh
12
orWe = W + t
1 + ln 4Wt
for Wh
12
t h and t < W2
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Types of substrate include plastics, sintered ceramics, glasses, and single
crystal substrates (single crystals may have anisotropic dielectric
constants; "anisotropic" means they are different along the different crystal
directions with respect to the crystalline axes.)
3.5.2.1 Common substrate materials
Plastics are cheap, easily manufacturability, have good surface adhesion,
but have poor microwave dielectric properties when compared with other
choices. They have poor dimensional stability, large thermal expansion
coefficients, and poor thermal conductivity.
1. Dielectric constant: 2.2 (fast substrate) or 10.4 (slow substrate)2. Loss tangent 1/1000 (fast substrate) 3/1000 (slow substrate)3. Surface roughness about 6 microns (electroplated)4. Low thermal conductivity, 3/1000 watts per cm2 per degree
Ceramics are rigid and hard. They are difficult to shape, cut, and drill.
They come in various purity grades and prices, each having domains of
application. They have low microwave loss and are reasonably non-
dispersive. They have excellent thermal properties, including good
dimensional stability and high thermal conductivity. They also have very
high dielectric strength. They cost more than plastics. In principle, the
size is not limited.
1. Dielectric constant 8-10 (depending on purity) so slow substrate2. Loss tangent 1/10,000 to 1/1,000 depending on purity3. Surface roughness at best 1/20 micron4. High thermal conductivity, 0.3 watts per cm2 per degree K
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Single crystal sapphire is used for demanding applications. It is very hard,
needs orientation for the desired dielectric properties which are
anisotropic, and is very expensive and can only be made in small sheets.
It has a high dielectric constant and so is used for very compact circuits at
high frequencies. It has low dielectric loss with excellent thermal
properties and surface polish.
1. Dielectric constant 9.4 to 11.6 depending on crystal orientation (slowsubstrate)
2. Loss tangent 5/100,0003. Surface roughness 1/100 micron4. High thermal conductivity 0.4 watts per cm2 per degree K
Single crystal Gallium Arsenide (GaAs) and Silicon (Si) are both used for
monolithic microwave integrated circuits (MMICs).
Dealing with GaAs first we have:
1. Dielectric constant 13 (slow substrate)2. Loss tangent 6/10,000 (high resistivity GaAs)3. Surface roughness 1/40 micron4. Thermal conductivity 0.3 watts per cm2 per degree K (high)
GaAs is expensive and piezoelectric; acoustic modes can propagate in the
substrate and can couple to the electromagnetic waves on the conductors.
Now dealing with Silicon we have:
1. Dielectric constant 12 (slow substrate)2. Loss tangent 5/1000 (high resistivity)3. Surface roughness 1/40 micron4. Thermal conductivity 0.9 watts per cm2 per degree K (high)
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The dielectric strength of ceramics and of single crystals far exceeds the
strength of plastics, and so the power handling abilities are
correspondingly higher, and the breakdown of high Q filter structures
correspondingly less of a problem.
It is also a good idea to have a high dielectric constant substrate and a
slow wave propagation velocity. This reduces the radiation loss from the
circuits. However, at the higher frequencies the circuits get impossibly
small, which restricts their power handling capability. For these
applications, one often chooses fused quartz (dielectric constant 3.8).
3.6 Return LossWhen the load is mismatched, not all of the available power from
the generator is delivered to the load. This loss is called return loss (RL)
and is defined (in dB) as
RL = 20log dBSo that a matched load line, ( ) has a return loss of dB (no reflected
power), whereas the total reflection ( ) has a return loss of 0dB (all
incident power reflected).
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3.7 Voltage Standing Wave Ratio
When the load is mismatched, however, the presence of a reflected
wave leads to standing waves where the amplitude of the voltage on theline is not constant.
It is seen that SWR is a real number such that , where
SWR=1 implies a matched load.
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3.8 Free Space Path Loss
Figure 3.7 : Data Tranmission
Path loss or path attenuation is the reduction in power density
(attenuation) of an electromagnetic wave as it propagates through space.
Path loss is a major component in the analysis and design of the link
budget of a telecommunication system.
This term commonly used in wireless communication and signal
propagation. Path loss may be due to many effects, such as free space
path loss, refraction, diffraction, reflection, aperture-medium coupling lossand absorption. Path loss is also influenced by terrain contours,
environment, propagation medium, the distance between the transmitter
and receiver, and the height and location of antennas.
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CHAPTER 4
DESIGN AND SIMULATION OF POWER AMPL IFIER CIRCUIT
4.1 Introduction
A design usually starts with a set of specification and selection of
the proper transistor. Then a systematic solution, aided by graphical
method, is developed to determine the transistor loading the source and
load coefficients for particular stability and gain criteria. A complete of
designing RF power amplifier is shown below:
1. Determine the gain and output value (output power of poweramplifier)
2. Select the transistor (noise, gain and price)3. Measure the s-parameter of the transistor from datasheet4. Study the stability
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5. Design the input and output matching network usingmicrostriplines
6. Build the power amplifier
4.1.1 Determine the gain and output value (output power of poweramplifier)
Calculation of amplifier gain and output power:
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4.1.2 Select the transistor (noise, gain and price)
FET FLL 351ME was selected as the power amplifier transistor
because of its features. Its output power is 35.5dBm.This output powershould be high enough to ensure the transmission of data in real time. The
gain this transistor is 11.5dB. (For others features can be referred to the
FLL 351ME datasheet in Appendix C). Besides that it is available in the
market and the transistor is already unconditionally stable. Thus, it is
much easier to design the power amplifier. The price is also reasonable as
it cost only $7.85.
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4.1.3 Measure the s-parameter of the transistor from datasheet
Table 4.1: A part of S-parameter for transistor FLL 351ME
4.1.4 Determining the Stability Factor K
From the datasheet transistor FET FLL 351ME (Appendix A),
which contained the scattering parameter hat was imported into the
working directory of Advanced Design System (ADS) software. An initial
simulation of the transistor amplifier at 2.4GHz frequency operation
shown in Figure 1 and the result in Figure 2.From calculation used
mathematical equation of the stability factor K.
From the calculation, the transistor is unconditionally stable as K>1.
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Figure 4.1: Simulation of Transistor
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Figure 4.2: Simulation Result of the Transistor
4.1.5 The gain and output value (output power of power amplifier)
Calculation of amplifier gain and output power:
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4.1.6 Computing input and output matching network
With the aid of smith chart (Appendix B and Appendix C) and
restricted of the series and open circuit lossless element. Approximate
transmission length of the element was obtained.
The design of the input matching network using simultaneous
conjugate match technique. Simultaneously gives the value and
required for a simultaneous conjugate match.
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4.1.6.1 Computing Input Matching Network
The design of input matching network using microstrip lines is
illustrated in Appendix B, where the admittance value, associated with
is
From move towards the generator until reach an admittance unit
circles, . A length of series microstrip transmission line
is is obtained by moving from to (a point on the unit
circle impedance circle) in direction towards the generator.The length of the open circuited stub is obtained by moving
from point towards load until the is reached hence
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4.1.6.1 Computing Output Matching Network
The design of output matching network using microstrip lines is
illustrated in Appendix C, where the admittance value, associated with
is
From move towards the generator until reach an admittance unit
circles, . A length of series microstrip transmission line
is is obtained by moving from to (a point on the unit
circle impedance circle) in direction towards the generator.
The length of the open circuited stub is obtained by moving
from point towards load until the is reached hence
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4.2 Analyzing the Transistor with Input and Output Matching
After the length in wavelength unit of series microstrip
transmission line and open circuited stub for input and output are obtained,
the electrical length can be calculated as in the table below:
Table4
.2.Table 4.2: Electrical length
Then TXLINE 2003-Microstrip was used to calculate the
conversion from the length to the physical length in mm needed by real
life microstrip components or MLIN elements in ADS software. And for
the open circuited stub, MLOC elements of ADS software were used.
With the substrate properties included, transmission lines characterizes
impedance of and at 2.4GHz, the physical lengths are as in the table
above.
Electrical length Length() Length(deg) Length(mm) Width(mm)
MLIN(input) 0.231 83.16 15.5995 2.97038MLOC(input) 0.498 179.28 33.63 2.97038
MLIN(output) 0.212 76.32 14.3164 2.97038
MLOC(output) 0.443 159.48 29.9159 2.97038
TERMINATE 0.125
45 8.44127 2.97038
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Figure 4.3: TXLINE 2003-Microstrip calculator
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4.3 One Stage Power Amplifier
Figure 4.4: The amplifier With Input and Output Matching Network (OneStage)
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4.4 Two Stages Power Amplifier
Figure 4.5: The amplifier With Input and Output Matching Network (TwoStages)
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4.5 The Amplifier Inclusion of DC Biasing Circuit
Figure 4.6: The amplifier With Input and Output Matching Network (With
DC biasing)
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CHAPTER 5
RESULTS AND ANALYSIS
5.1 Stability Factor, K
Figure 5.1: Simulation of Transistor
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Figure 5.2: Simulation Result of the Transistor
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5.2 One Stage Power Amplifier
Figure 5.3: The amplifier With Input and Output Matching Network (One
Stage)
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Figure 5.4: Stability Factor K
Figure 5.5: Maximum available gain
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Figure 5.6: Output power
Clearly the design goals do not be met for the gain criterion with a
single stage. The output power is not adequate the transmitter to transmit
the data for 40km of distance. Therefore, a two-stage design was designed
and implemented.
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5.3 Two Stages Power Amplifier
Figure 5.7: The amplifier With Input and Output Matching Network (Two
Stages)
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Figure 5.8: Maximum Gain
Figure 5.9: Output power
Figure 5.9 indicates that the desired output power is achieved to meet the
goal.
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5.4 The Amplifier Inclusion of DC Biasing Circuit
Figure 5.10: The amplifier With Input and Output Matching Network
(With DC biasing)
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Figure 5.11: Voltage Standing Wave Ratio
The voltage standing wave ratio (VSWR) is considered to be as a good
value because the best value of VSWR for microwave power amplifier
should be less than 2.5.
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Figure 5.12: Simulation Results
The simulation results above indicate that a very good return loss
(m2) is achieved, as the best value for return loss (RL) should be less than
-20dB.The highest stability is at 2.4GHz shows that the power amplifier is
really stable at this frequency.
However, gain and stability are traditionally the tradeoffs in any
microwave amplifier design. Also, the noise figure may be increased, as it
depends on the DC resistance configured at the input for DC biasing.
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Figure 5.13: Output Power
Figure 5.13 indicates that the desired output power is achieved to
meet the goal and the value is higher than the power amplifier without DC
biasing circuit.
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CHAPTER 6
CONCLUSION
A two-stage RF PA was designed using FET FLL351 power stage
at 2.4 GHz of frequency for wireless transmitter in telemedicine
application was successfully designed since it almost meets all the
specifications. A large-signal model has been developed and verified for
the GaAs device. The desired output power which is 32.0282 dB is
successfully achieved. The time spent in studying the design process of a
microwave power amplifier served as a great experience and preparation
for the future designing endeavors.
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CHAPTER 7
RECOMMENDATION
The desired power amplifier was designed successfully. However
it has only been proven theoretically and by simulation. Thus, it should be
implemented into hardware in the future works to determine its efficiency.
It is recommended to use AutoCAD software to convert the layout plot of
the power amplifier circuit into layout. After that, the circuit should be
etched on the RF4 board.
It is hope that this system will be able to provide point-to-point and
point-to-multi-point signal transmission through the air over a terrestrial
microwave platform, rather than through copper or fiber cables. As a
result, telemedicine systems do not require satellite feeds or local phone
service and can provide a wireless broadband alternative to cable modem
or DSL connections.
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8. Christian Bourde, Jeff & FullerSteve Long. RF Prototyping Techniques. Journal
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