A Bidirectional High-Power-Quality Grid Interface With a Novel ... · For the bidirectional...

2018 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 5, JUNE 2012

A Bidirectional High-Power-Quality Grid InterfaceWith a Novel Bidirectional Noninverted

Buck–Boost Converter for PHEVsOmer C. Onar, Member, IEEE, Jonathan Kobayashi, Student Member, IEEE,

Dylan C. Erb, Student Member, IEEE, and Alireza Khaligh, Senior Member, IEEE

Abstract—Plug-in hybrid electric vehicles (PHEVs) will play avital role in future sustainable transportation systems due to theirpotential in terms of energy security, decreased environmental im-pact, improved fuel economy, and better performance. Moreover,new regulations have been established to improve the collective gasmileage, cut greenhouse gas emissions, and reduce dependence onforeign oil. This paper primarily focuses on two major thrust areasof PHEVs. First, it introduces a grid-friendly bidirectional al-ternating current/direct current ac/dc–dc/ac rectifier/inverter forfacilitating vehicle-to-grid (V2G) integration of PHEVs. Second,it presents an integrated bidirectional noninverted buck–boostconverter that interfaces the energy storage device of the PHEVto the dc link in both grid-connected and driving modes. Theproposed bidirectional converter has minimal grid-level disrup-tions in terms of power factor and total harmonic distortion, withless switching noise. The integrated bidirectional dc/dc converterassists the grid interface converter to track the charge/dischargepower of the PHEV battery. In addition, while driving, the dc/dcconverter provides a regulated dc link voltage to the motor driveand captures the braking energy during regenerative braking.

Index Terms—Alternating current/direct current (ac/dc)–dc/acgrid interface converter, bidirectional converters, noninvertedbuck–boost dc/dc converter, plug-in hybrid electric vehicles(PHEVs), vehicle-to-grid (V2G).

NOMENCLATURE

Vs Alternating current (AC) grid voltage.L1 Alternating current/direct current

(AC/DC)–DC/AC converter inductor.T16 AC/DC–DC/AC converter insulated gate

bipolar transistors (IGBTs).

Manuscript received June 18, 2011; revised October 17, 2011, January 11,2012, and March 3, 2012; accepted March 19, 2012. Date of publicationMarch 31, 2012; date of current version June 12, 2012. This work was sup-ported in part by the U.S. National Science Foundation under Grant 0801860,Grant 1157633, and Grant 0852013. The review of this paper was coordinatedby Prof. F. Assadian.

O. C. Onar is with the Energy and Transportation Division, Oak RidgeNational Laboratory, Oak Ridge, TN 37831 USA.

J. Kobayashi is with the Department of Mechanical Engineering, Universityof California, Berkeley, CA 94720 USA.

D. C. Erb is with the Department of Mechanical Engineering and alsowith the MIT Energy Initiative, Massachusetts Institute of Technology (MIT),Cambridge, MA 02139 USA.

A. Khaligh is with the Power Electronics, Energy Harvesting, and RenewableEnergies Laboratory, Department of Electrical and Computer Engineering,University of Maryland, College Park, MD 20742 USA.

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TVT.2012.2192459

D12 AC/DC–DC/AC converter diodes.T711 Bidirectional DC/DC converter IGBTs.D37 Bidirectional DC/DC converter diodes.L2 Bidirectional DC/DC converter inductor.Vab Voltage across the grid and L1.C1−2 DC link capacitors.C3 Capacitor in parallel with the battery.Cdc DC link equivalent capacitance.Vdc DC link voltage for both plug-in and driving

modes.Vbatt Battery voltage.Vpeak Peak value of the grid voltage.VC1, VC2 C1 and C2 capacitor voltages.I∗ref Reference grid current (peak).IP Reference grid current (instantaneous).x, y, z Combinational logic Boolean variables.PBatt Measured charge/discharge power.P ∗

ref Reference charge/discharge power.D Duty cycle for DC/DC converter

switching.R Triangular carrier waveform.G Gate signals.V ∗

dc,ref Reference DC link voltage.T (s) DC/DC converter voltage loop controller

transfer function.Ibatt Battery current.I∗batt,ref Reference battery current.fs Switching oscillator frequency.THD Total harmonic distortion.SoC State of charge (of the battery).vce(θ), VCE(sat) On-saturation voltage for IGBT.vF (θ),VF Forward voltage drop for the diode.Eon(θ) Turn-ON commutation loss.Eoff (θ) Turn-OFF commutation loss.ED_rev(θ) Diode reverse recovery loss.η Efficiency.η′ Change in efficiency.Pin Input power.Po Output power.Ploss Power loss.GHG Greenhouse gas emission.V2G Vehicle to grid.ISO Independent system operator.Tn nth switch.

0018-9545/$31.00 © 2012 IEEE

ONAR et al.: GRID INTERFACE WITH BIDIRECTIONAL NONINVERTED BUCK–BOOST CONVERTER FOR PHEVs 2019

I. INTRODUCTION

P LUG-IN electric vehicle (PHEV) technology is the mostpromising candidate for meeting the future’s ever-growing

transportation needs. PHEVs introduce reduced fuel consump-tion, higher performance, and lower emission [1]–[3]. By thetime the next generations of PHEVs are brought to market,certain issues will need to be addressed. One imperative issueis the method by which these vehicles will be recharged and ifthe current grid can sustain the increased demand due to morePHEVs. The additional charging load of the PHEVs will bringin new load levels, patterns, and load characteristics, particu-larly the PHEV battery charger’s load characteristics [4]. Thereare some studies with regard to the load level increasing impactsof PHEV penetration in power systems [5]–[8]. However, theeffects of PHEV chargers are not only limited to the increasedload level but may also affect the power quality in terms of re-active power and harmonic distortions [9]–[14] if conventionalchargers are employed. The negative effects of the harmonicsinclude transmission, distribution, and transformer overloading;additional losses on power system; decreased power systemstability; increased skin effect losses; equipment, appliance, ordevice failures or damages; protection system malfunctioningor failure; insufficient reactive power compensation due toincrease in frequency; resonant effects; interfering with com-munication and phone lines; and control and communicationsystem malfunctioning. According to the U.S. Department ofEnergy Alternative Fuels and Advanced Vehicles Data Center,1 888 971 hybrid electric vehicles (HEVs) had been sold inthe U.S. as of March 2011. The Department of Energy’s targetfor electric vehicles (EVs) and PHEVs is 1 million vehiclesby 2015. When PHEVs achieve a large market share, the gridcould suffer if the PHEVs’ charging is always unidirectional,is uncoordinated with poor power factor, and draws distortedcurrents [15], [16].

Note that PHEVs have the capability of representing a largeenergy source for the grid. Tapping this resource could elim-inate grid issues such as the constant need to provide loadand generation balancing, frequency regulation, transmissioncongestion, time-of-use demand charges, and the need forvoltage regulation, power quality, and renewables integration,which are simply a result of the demand variations that occurevery day in addition to the constant need for voltage andfrequency regulation [17]. To perform grid-connected vehiclebattery applications, an advanced power electronic grid in-terface that can provide V2G bidirectional power flow withhigh power quality is required. An advanced interface sys-tem can respond to the charge/discharge commands that werereceived from an ISO, an aggregator, or a utility to providedemand-side management based on the needs of the powersystems. This advanced interface should accomplish this taskwithout causing any reactive power or power quality issuesto the grid. If this V2G-enabled interface tracks the refer-ence charge/discharge power, coordinated charging would beachieved, which could significantly reduce the potential grid-related problems of PHEVs [18], [19]. In the near-future smartgrid environment, ISOs, aggregator agencies, or utilities may

determine the reference charge/discharge power by consideringthe drivers’ requirements, the SOC of the batteries, and the stateof the grid. Smart grid communication infrastructure is alsoexpected to command the residential appliances or commer-cial/industrial loads to provide regulation or spinning reservesthrough the demand response programs.

A comprehensive review of the bidirectional convertershave been presented in [20] with respect to their advantages,drawbacks, and other aspects. A comprehensive analysis ofpower converters employed in offboard chargers have been de-tailed in [21], including a basic voltage–source inverter bridge,a VIENNA rectifier, a phase-shifted full-bridge converter, ahalf-bridge inductor-inductor-capacitor (LLC) converter, athree-phase interleaved converter, and the phase-shifted ZVSfull-bridge converter. An onboard charger for PHEV appli-cations that consists of a diode-bridge rectifier, followed byan interleaved boost converter with power factor correction(PFC), a high-frequency inverter and a transformer, and adiode-bridge rectifier that is cascaded to a battery filter cir-cuit is proposed in [22]. The proposed topology achievesefficiency up to 94%, a power factor that ranges from 0.97to 0.995, and a THD that ranges from 4% to 24%. Refer-ences [23] and [24] present the energy efficiency in PHEVchargers along with the evaluation and comparison of front-end ac/dc topologies. Evaluated converter topologies includethe conventional PFC boost converter (diode-bridge rectifier,followed by a boost converter), interleaved PFC boost con-verter, bridgeless PFC boost converter, phase-shifted semi-bridgeless PFC boost converter [25], bridgeless interleavedboost converter [26], and bridgeless interleaved resonant PFCboost converter [27]. Among these converters, the semibridge-less boost converter produces 5%–43% at 240 V input at differ-ent load levels, and the bridgeless interleaved boost converterproduces about 4%–42% THD at 240 V input at differentload levels. At 120 V input, the THD of these two convertersvaries from 5% to 24% and from 4% to 14%, respectively. Thethree-phase bidirectional battery charger that was presented in[28] is a three-phase bridge inverter with a bidirectional dc/dcconverter and results in 20.475% of THD. As identified in[9], a report by the California Energy Commission reportedbattery charger input current THD variation up to 28% overthe charging cycle. Reference [9] also states that a Ford Escortonboard battery charger measured input current THD of 59.6%at an output current of 15.7 A. It is clear that high penetrationof PHEV chargers in a distribution network could cause asignificant increase in the system THD.

For the bidirectional noninverted buck–boost part of theproposed system, the comparison may include a conventionalbuck–boost converter. The converters can step up or down theinput voltage but cannot provide bidirectional power flow, andthey require an inverting transformer, because their output volt-age is negative with respect to the input voltage [29]. Althoughsome topologies can be noninverted [30]–[41], [30]–[32] and[34]–[41] employ more than one switch in the pulsewidthmodulation (PWM) mode, resulting in higher switching losses.Among these topologies, although they provide buck or boostoperations, bidirectional power flow cannot be achieved inthe topologies in [30], [33], and [37]. The conventional


two-quadrant bidirectional converters can operate the buckmode in one direction and the boost mode in the other direc-tion; however, they cannot operate vice versa [33], [36]. Bidi-rectional power flow with bucking and boosting capabilitiescan be achieved in two cascaded two-quadrant bidirectionalconverters; however, more than one high current inductor isrequired in these topologies [31], [35]. Although two switchesand two inductors are used in [36], only unidirectional buckingor boosting can be achieved. In the case of a dual-active bridgedc/dc converter, all switches are operated in the PWM mode;therefore, switching losses are four times higher in the half-bridge case or eight times higher in the full-bridge case than theproposed converter. Dual-active bridge dc/dc converters alsorequire a transformer at the middle stage, which increases theoverall losses, size, and cost [38]–[41]. In [38], two inductorsare required in addition to the transformer, and in [39], thenumber of inductors is three. In [40], bidirectional power flow ispossible with ten switches and two inductors. Finally, in [42],the proposed dc/dc converter requires two transformers, withone being multiwinded, which may complicate the structureand add up to the cost, and it does not have the bidirectionaloperating capability.

This paper presents a bidirectional grid interfacing powerelectronic converter that enables the beneficial V2G interac-tions while ensuring that all power that is delivered from orinjected to the grid has a high power factor and negligiblecurrent harmonics. This combination of the multilevel (three-level) bidirectional ac/dc converter with the proposed bidirec-tional dc/dc converter can accomplish these requirements whiletracking the reference charge/discharge power. The dc/dc partis also employed to provide a regulated dc link voltage tothe motor drive in typical driving conditions. The multilevelac/dc converter has low device stress with a relatively smallerinput inductor compared to conventional H-bridge counterpartsand inherently higher waveform quality (low THD) over theother diode bridge or PWM rectifier/inverter-based applica-tions [43], [44]. The noninverted operation capability of theproposed converter totally eliminates the need for an invertingtransformer, which reduces the overall size and cost. The dc/dcconverter part of the proposed system uses only one switchin the PWM mode; therefore, the controls are as simple asthe conventional buck or boost dc/dc converters despite all thecompetences; furthermore, switching losses are not more thanthat of a conventional buck or boost dc/dc converters. The pro-posed grid interface topology can also be scaled to three-phaseapplications, although it is originally a single-phase converter.To achieve a three-phase multilevel converter, one more leg (orbridge arm) should be added in parallel to the existing two-phase legs. In a three-phase application, each phase leg shouldhave two clamping diodes, whereas the number of capacitors atthe dc link and their connections stay the same.

The proposed system, as shown in Fig. 1, has four differentgeneralized operational modes. Two of these modes occur inthe grid-connected mode to supply power to/from the batteryfrom/to the grid. The other two modes occur while driving bysupplying power from/to the battery to/from the dc link whileaccelerating or regenerative braking. Due to these features, thiscomplete topology is an ideal candidate for meeting the inclu-

Fig. 1. Proposed integrated converter with ac/dc (grid interface) and dc/dc(battery interface) converters for EV and PHEV applications.

Fig. 2. AC/DC converter flow path in mode 1.

sive needs of the PHEV industry and HEV to V2G-equippedPHEV conversions.

This paper is organized as follows. In Section II, the topo-logical overview and the operation modes are presented, alongwith the control systems. Simulation results for the proposedconverter are given in Section III. Section IV focuses on theexperimental results to evaluate and validate the capabilities ofthe proposed converter. Finally, the conclusions are drawn inSection V.

II. SYSTEM DESCRIPTION AND OPERATING MODES

The multilevel ac/dc converter and the bidirectional inte-grated dc/dc converter constitute the proposed topology shownon the respective left and right sides of Fig. 1. The bidirec-tional multilevel ac/dc converter is made up of components L1,D1D2, T1T6 with their internal diodes, as well as C1C2.

A. Operational Modes of the Grid Interface Converter

When the vehicle is plugged in, the ac/dc converter switchesthrough six different modes using T1T6. In mode 1, Vdc isapplied across the grid and input inductor by turning T1, T3,and T6 ON, as shown in Fig. 2. During this interval, the voltagesacross both C1 and C2 simultaneously increase or decrease,and the current that was delivered to/from the grid decreases,because a negative voltage is applied on L1. In mode 2, whichis given in Fig. 3, T2, T4, and T5 are turned ON to apply−Vdc across the grid and inductor to simultaneously charge ordischarge C1 and C2. In this operation state, the grid currentincreases due to the positive voltage across L1. That is, in thisstate, both the grid voltage and the applied voltage are negative.




Throughout mode 3, half of the dc link voltage or the C2 volt-age (Vdc/2 = VC2) is applied across the grid and the inductorby switching ON T3, T4, and T6, as shown in Fig. 4. Based onthe instantaneous voltage of the grid and VC2, the current thatis delivered to/from grid either increases or decreases. In thismode, if the capacitor C2 is charged (rectifier mode), positivegrid current flows through the grid, L1, T4, D2, C2, and T4.If the capacitor C2 is discharged (inverter mode), negative gridcurrent flows through C2, D1, T3, L1, the grid, and T6. Thesetwo different current flow paths are shown in Fig. 4, where thesolid lines show the capacitor discharge (inverter mode), andthe dashed lines show the capacitor recharge (rectifier mode).In mode 4, T2, T3, and T4 are turned ON to apply negative halfof the dc link voltage or the negative C1 voltage (−Vdc/2 =−VC1) across the grid and the inductor, as shown in Fig. 5.According to the relationship between Vs and VC1, the currentthat is delivered to/from grid increases or decreases. In thismode, if the capacitor C1 is charged (rectifier mode), positivegrid current flows through the grid, L1, T4, D2, C1, and T2.If the capacitor C1 is discharged (inverter mode), negative gridcurrent flows through C1, D1, T3, L1, the grid, and T2. Thesetwo different current flow paths are shown in Fig. 5, where thesolid lines show the capacitor discharge (inverter mode), andthe dashed lines show the capacitor recharge (rectifier mode).

During mode 5, as shown in Fig. 6, the negative half-linecycle of Vs is shorted across the inductor by keeping T1, T2, andT3 ON. Therefore, the inductor current magnitude increases. Inmode 6, T4, T5, and T6 are turned ON, as shown in Fig. 7, sothat the positive half-line cycle of Vs can be shorted across theinductor; therefore, the inductor current increases once again.




The converter operation is provided by applying three dif-ferent voltage levels Vab to the right side of the inductor,depending on the voltage that the grid applies to the inductorVs. Applying three-level voltages provides waveform shapingfor the grid current; by switching between two different voltagelevels with different pulsewidths, the current that is drawn fromor injected to the grid can be controlled. The grid voltagevariation and the applied Vab voltage levels over a period areprovided in Fig. 8.

Table I presents the gate-switching patterns for each modeand the corresponding Vab voltage applied across the inductorand the grid.

The control strategy for both the ac/dc rectifying and dc/acinverting is provided in Fig. 9, which generates the Booleanx, y, and z, which will determine the mode of operation andthe switching signals for T1 − T6. The controller requires fourfeedback interface inputs: grid voltage Vs, grid current Is, and


Fig. 8. Switching between modes with respect to Vs.

TABLE ISWITCHING PATTERNS FOR DIFFERENT MODES OF OPERATION

dc link capacitor voltages VC1 and VC2. The peak of the Vs

voltage is selected to be the reference dc link voltage. BecauseVdc = VC1 + VC2, the peak of Vs is compared with VC1 + VC2,and the error is fed through the voltage controller, which outputsthe rough current reference. The grid voltage is normalized bydividing Vs by Vpeak so that a unity sine wave in phase with Vs

is obtained. This unity sine wave is multiplied with the outputof the voltage controller. To compensate for the imbalancebetween the dc link capacitors, their voltage difference is addedup to the rough current reference to obtain the actual I∗ref . Theactual grid current is then compared with I∗ref , and error is fedto the current proportional–integral–derivative controller.

Consequently, three Boolean variables x, y, and z are gener-ated based on the instantaneous state of the circuit. Variable xis determined by the current direction, variable y is determinedby the ac voltage region detection, and the sign of the acvoltage determines the variable z. Finally, these variables areinterpreted according to the combination given in Table II thatrepresents the converter’s mode of operation.

B. Operational Modes for the Battery Interface Converter

The proposed dc/dc converter can provide bidirectionalpower flow by bucking/boosting the dc link voltage or buck-ing/boosting the battery voltage. Leading automakers have in-troduced PHEVs that utilize battery packs with rated voltageslower than the dc link (motor drive) voltage but higher thanthe rectified ac voltage. For example, the Toyota Prius plug-incomes with a 345.6 V battery pack, whereas the motor drivevoltage is 650 V [45], and the vehicle can be charged with a120 V standard North American wall outlet. Similarly, theChevy Volt battery pack is 300 V, the motor drive voltageis 320–350 V, and the battery can be recharged from 120 Vsupplies [46]. An add-on battery pack with lower voltage canalso always be used in a retrofitted PHEV conversion [47].Therefore, it is evident that a multifunctional dc/dc battery

interface is required for PHEVs. The different conditions forplugged-in or driving modes are summarized in Table III.

When the vehicle is plugged in, the rectified ac voltage (here,the dc link voltage) should be stepped up for grid charging, andthe battery voltage should be stepped down for grid discharging.While the vehicle is driven, the battery voltage should bestepped up during acceleration, and the dc link voltage shouldbe stepped down during regenerative braking or down hilling.The states of the switches with respect to these operationalmodes are mapped in Table IV.

Modes 1 and 2: Plug-In Charging and Discharging: Whencharging the battery pack, the rectified ac voltage (dc linkvoltage) is usually maintained at the peak grid voltage. Whilestepping up the dc link voltage, D3, T7, L2, T9, D5, and T11

are used to form a boost converter, as shown in Fig. 10. In thismode, T9 is the only switch that is operated in PWM. When T9

is turned ON, as shown by the red dashed line, L2 is energizedby the dc link voltage through D3, T7, L2, and T9. When T9 isturned OFF, both the inductor and the dc link supplies power tothe battery, as shown in the solid and dashed red lines of Fig. 10.

The battery voltage is stepped down with the buck converterthat is made up of T10, D7, L2, D6, D4, and T8. In this mode,T10 is operated in the PWM mode. When T10 is ON, the batterycurrent is delivered to the dc link while energizing the inductorthrough T10, D7 (as shown by the solid blue line), L2, T8, andD4. When T10 is turned OFF, the output current is recoveredby the freewheeling diode D6, decreasing the average currenttransferred from the battery to the dc link (as shown by thedashed blue lines). The current flow paths of this mode arepresented in the blue lines in Fig. 10.

Modes 3 and 4: Acceleration and Braking Modes forDriving: During driving, the battery voltage should be steppedup in acceleration or cruising conditions. A boost converter isformed by T10, D7, L2, T9, T8, and D4, as shown in Fig. 11. Inthis case, PWM switching signals are applied to T9. When T9

is turned ON (as shown by the dashed red lines), battery currentflows through T10, D7, and T9, and the inductor is energized.When T9 is turned OFF, D4 is forward biased, and the inductorcurrent flows to the output, as shown in the solid red lines inFig. 11.

During regenerative braking, the motor drive inverter recov-ers braking energy; therefore, the dc link voltage increases.When a rise is detected in the dc link voltage, the convertershould be operated in the buck mode from the dc link to thebattery. A buck converter can be formed by D3, T7, D6, L2,D5, and T11, as shown in Fig. 11. In this condition, T7 is inthe PWM mode. When T7 is turned ON, current from the dclink passes through D3, T7 (as shown by solid blue lines),L2, D5, and T11 while energizing the inductor. When T7 isOFF, the output current is freewheeled through D6, D5, andT11 (as shown by solid blue lines), decreasing the averagecurrent transferred to the battery. The current flow path for thisoperation is shown the blue lines of Fig. 11.

As shown in Figs. 12 and 13, all operations of the batteryinterface converter are combinations of buck-and-boost oper-ations. For the proposed system, two different controllers areincorporated: one controller is for modes 1 and 2 (plug-incharging/discharging), and the other controller is for modes 3


Fig. 9. Block diagram of the ac/dc–dc/ac converter controller.

TABLE IISEQUENCE OF OPERATIONS AND THE COMBINATIONAL LOGIC

TABLE IIISEQUENCE OF OPERATIONS AND THE COMBINATIONAL LOGIC

TABLE IVSEQUENCE OF OPERATIONS AND THE COMBINATIONAL LOGIC

Fig. 10. Boost and buck power flows for plug-in charging/discharging.

and 4 (acceleration/deceleration during driving). In the drivingmode, it is important to provide a regulated dc link voltageto the motor drive, whereas in the plug-in mode, it is desiredto control the charging or discharging power of the battery.Therefore, a power controller is used for plug-in modes, anda double-loop voltage and a current controller is employed foracceleration/braking modes.

Fig. 11. Boost and buck power flows for acceleration/braking.

Fig. 12. DC/DC converter’s charging/discharging power controller.

Fig. 13. DC/DC converter’s cascaded controller for the driving mode.

The battery charge/discharge power controller, as shown inFig. 12, allows for tracking the reference charge or dischargepower of the battery. Based on the SOC of the battery, userrequirements, and the state of the grid, this reference power canbe determined. If a fast charge is desired, the reference powerP ∗

ref can simply be increased to the desired value, because theproposed converter can dynamically adapt to any changes inreference charge/discharge power. In the case of an applicationwhere “constant current” charge is followed by a “constantvoltage” charge, the power interpretation of such charging canbe implemented by accordingly varying the power. In constantcurrent charging, the battery voltage would gradually increase,resulting in a ramping-up power rate, whereas in constant volt-age, charging the current would gradually decrease, resulting in


a ramping-down power rate. By applying these power rates asthe reference power, both the constant current and the constantvoltage charging can be achieved.

During driving, the high-voltage bus is maintained at the ref-erence motor drive inverter voltage by the cascaded voltage andcurrent controller, as shown in Fig. 13. This enables dischargingof the battery to the dc link during acceleration and regenerativerecharging of the battery from the dc link while maintaining theproper voltage level for the hybrid vehicle.

C. Efficiency Analysis Approach for the Grid Interface andBattery Interface Converter

For the grid interface converter part of the proposed system,semiconductor power losses can be estimated by implementingvsat(θ) × Il(θ) and E(θ) × Il(θ) presented in the manufac-turer datasheet of each device. Here, vsat(θ) is the ON-statesaturation voltage, which is vce(θ) for the insulated gate bipolartransistor (IGBT) and vF (θ) for the diode, whereas E(θ) is theenergy losses in one commutation, where Eon(θ) is the turn-ON commutation loss, Eoff (θ) is the turn-OFF commutation,and ED_rev(θ) is the diode reverse recovery commutation loss.For the loss calculations, simple models of these losses asfunctions of the carried current can be created as vce(Il(θ)),vF (Il(θ)), Eon(Il(θ)), Eoff (Il(θ)), and ED_rev(Il(θ)) for thedevices. The conduction and the switching power losses can becalculated based on the models for each semiconductor deviceof the grid interface converter. The sum of the conduction andswitching losses would give the total losses of the converter.Because the switching losses should be obtained by implement-ing every turn-ON and turn-OFF instants during one referenceperiod, turn turn-ON, turn-OFF, and reverse recovery losses canbe given by

Pon =1T

∑Eon (Il(θ)) (1)

Poff =1T

∑Eoff (Il(θ)) (2)

PD_rev =1T

∑ED_rev (Il(θ)) . (3)

Hence, the total switching losses are equal to the sum ofthe turn-ON, turn-OFF, and reverse recovery losses, which isexpressed by

PSwitching = Pon + Poff + PD_rev. (4)

Here, note that there is not a certain switching frequencyfor the grid-side converter (the switching frequency is totallyrandom, i.e., the converter does the switching whenever nec-essary and transits between modes, depending on the gridvoltage, grid current, dc link voltage, and the dc link capacitorsvoltage imbalance). Although there is not a certain switchingfrequency for the overall grid-side converter, we can obtainthat the switching frequency of the power switches T2 andT6 is equal to the line frequency, which is 60 Hz (see theoperational modes). On the other hand, when the converterswitches between different levels (0 to Vdc/2, Vdc/2 to Vdc, 0to −Vdc/2, and −Vdc/2 to −Vdc), the switching frequency isalso relatively low, i.e., around only a few kilohertz. Therefore,

(1)–(3) should be computed with different reference periods fordifferent devices. For example, switches T2 and T6 would havemuch less switching losses compared with other switches forthe same amount of time.

The conduction losses in the proposed grid interface con-verter occur when a semiconductor device is ON and there isconducting current. The conduction losses for an IGBT is givenby (5), as shown below, whereas (6) expresses the conductionlosses for a diode, as shown below. Therefore, the total lossescan be calculated by (7), as shown below.

Pcond,SW =1

2π

2π∫

0

vsat(θ).Il(θ) dθ (5)

Pcond,D =1

2π

2π∫

0

vF (θ).Il(θ) dθ (6)

Pcond,Total =Pcond,SW + Pcond,D. (7)

Finally, the total losses will be equal to the sum of theswitching and the conduction losses as

PTotal_loss = PSwitching + Pcond,Total. (8)

Because the switching losses of the grid-side converter arerelatively lower, the conduction losses of the proposed con-verter are slightly higher than the H-bridge conventional in-verters. However, H-bridge conventional converters switch atmuch faster switching frequency, resulting in their switchinglosses being higher than the proposed converter. Therefore, wecan obtain that the overall losses of both the proposed gridinterface and the conventional H-bridge inverters can be in closeproximity. Equations (1)–(8) are implemented in a MATLABscript program along with the current variation for a givenperiod time while taking the switching periods of individualdevices into account. Because the efficiency can be describedas a function of the input power and the losses, the efficiency ofthe converter can be calculated by

η = 1 − PTotal_loss

Pin(9)

Therefore, the efficiency of the proposed grid interface con-verter is obtained as a function of the percentage of the inputpower, as given in Fig. 14. The peak efficiency of the proposedgrid interface is 95.25%. The rated power is 18 kW based onthe switch ratings of 600 V and 30 A.

The switching losses of the proposed bidirectional non-inverted buck–boost converter are very similar to a regularnoninverting buck–boost converter. To make the criteria ofcomparison clear, the compared converter should have nonin-verted output and a relatively wider voltage for both the batteryand the dc link. Even if the proposed converter is comparedwith the simplest fundamental buck or boost dc/dc convertersfor each of its modes, the switching losses are identical, be-cause the proposed converter has only one switch in the PWMmode in all of the modes. Because the number of switchesthat operate in PWM is the same for the proposed and the


Fig. 14. Grid interface converter’s efficiency.

conventional converters, we can obtain that there is no increasein switching losses. However, it can be stated that the proposedconverter has relatively slightly more conduction losses in allthe operating modes. The additional conduction loss is mainlydue to the additional switches or diodes in the current flowpaths of the proposed converter. Because the proposed dc/dcconverter has four different operating modes, losses should sep-arately be examined, because all modes introduce different losscomponents.

For example, in the plug-in charging mode, the dc/dc con-verter is operated in the boost mode from dc link to the battery.If the bottom line is a regular boost dc/dc converter, it can befound that the difference in loss is the conduction losses ofa pair of an IGBT and a diode whereas the switching lossesare the same. In dc analysis, the diode conduction loss wouldbe PD = vF .IF , whereas the IGBT conduction loss would bePT = vCE(SAT ).ICE . Therefore, in this mode, the change inlosses can be expressed as

ΔPloss,1 = PD3 + PT7 + PT11. (10)

In the plug-in discharge mode, where the dc/dc converteris operated in the buck mode from the battery to the dc link,it is shown that the additional conduction losses compared toa conventional buck converter are due to a pair of additionaldiode (D7 and D4) and the switch (T8). Therefore, in this mode,additional conduction losses can be calculated by

ΔPloss,2 = PD7 + PD4 + PT8. (11)

For the acceleration in the driving mode, the dc/dc converteris operated in the boost mode from the battery to the dclink; therefore, the additional conduction losses compared toa conventional boost converter are due to a pair of additionalconduction switches (T10 and T8) and the diode (D7). Thisresults in additional losses to be

ΔPloss,3 = PT10 + PT8 + PD7. (12)

Similarly, for the regenerative braking in the driving mode,the dc/dc converter is operated in the buck mode from the dclink to the battery; therefore, additional conduction losses occurdue to the forward biased diodes D3 and D5 and the conductingswitch T11, as given by

ΔPloss,4 = PT11 + PD3 + PD5. (13)

TABLE VCOMPARATIVE CHANGE IN EFFICIENCY OF THE DC/DC CONVERTER FOR

DIFFERENT MODES

To estimate the comparative change in efficiency, η is iden-tified as the efficiency of the conventional buck or boost mode,and η′ is defined as the efficiency of the proposed converter withadditional losses. If Po is the output power and the Pin is theinput power, the change in efficiency can be obtained as

Δη = η − η′ =Po

Pin− Po

Pin + ΔPloss. (14)

The comparative change in efficiency for all of the fourmodes is formulated as a function of Pin, Pout, Ploss, andΔPloss; ΔPloss for different modes is given by (10)–(13). Forthese analyses, as used in the experiments, HGTG30N60A4DIGBT modules and FFPF30U60STTU power diodes fromFairchild Semiconductor are used, where the IGBT’s VCE(SAT )

is 1.6 V, whereas the diode’s VF is 2.1 V, as given in thedatasheets. Changes in efficiency values are summarized inTable V with respect to different operating modes.

D. Cost Comparative Analysis for the Proposed Converter

For the grid interface converter part of the proposed system,cost comparison analysis is based on the kilovolt-ampere rat-ings of the total power electronic semiconductors needed, andthis total kilovolt-ampere ratings of the total semiconductor de-vices needed is compared with a conventional H-bridge invertertopology. In an H-bridge inverter topology, when one phase is inoperation, there are two devices in series, and these two-seriesswitches are subject to the entire Vdc voltage. Assuming thatVdc is 1 p.u., the voltage on per device would be 0.5 p.u. How-ever, in the proposed multilevel grid interface converter, thereare three semiconductor devices in series during the operationof one phase. Moreover, these three devices in series are subjectto only half of the Vdc voltage, because the converter switchesbetween 0 and Vdc/2, Vdc/2 and Vdc, 0 and −Vdc/2, and −Vdc/2and −Vdc. Therefore, each device is subject to 1/3 the half dclink voltage. Two topologies are compared in Table VI in termsof their number of devices, voltage stress on the device, andthe total kilovolt-ampere ratings in per unit. In these analyses,current is assumed to be constant and the same for two of thetopologies, because it depends on the load, and the convertersare analyzed under the same loading conditions. Table VI showsthat the total kilovolt-ampere rating of the proposed converteris 1.32 p.u., whereas a conventional H-bridge inverter’s totalkilovolt-ampere rating is 2.0. p.u. Therefore, although a largernumber of semiconductor switches are utilized in the proposedconverter, their voltage rating can be less, and the proposedconverter would be less expensive due to the reduced deviceratings.


TABLE VICOMPARATIVE KILOVOLT-AMPERE ANALYSES OF THE PROPOSED GRID

INTERFACE CONVERTER AND A CONVENTIONAL H-BRIDGE INVERTER

TABLE VIICOMPARATIVE KILOVOLT-AMPERE ANALYSES OF THE PROPOSED DC/DC

CONVERTERS WITH CONVENTIONAL APPROACHES

For the dc/dc converter, to provide the same functionality,four dc/dc converters would be needed with conventional con-verters: two of these converters would be boost dc/dc converters(one converter for plug-in and one converter for the drivingmode), and the other two converters would be buck dc/dcconverters (one converter for plug-in and one converter forthe driving mode). In this case, instead of one inductor, fourinductors would be needed for each of the converters. However,commercially available EVs and PHEVs do not currently havethe capability to inject power back to the grid. In addition,for the driving mode, they utilize a two-quadrant converterto provide both the boost and buck functions either for ac-celeration or regenerative braking modes. In Table VII, theproposed converter is compared with the conventional approachwith four conventional dc/dc converters and the commerciallyavailable vehicle power electronics, although they do not havethe plug-in discharge capability. As shown in Table VII, theproposed converter adds only two more semiconductor devices;however, it reduces the number of inductors from four toone compared with the two buck–boost converter approach.Because the inductor core and winding materials are extremelymore expensive than the semiconductor devices, it is alwaysdesirable to add two more semiconductor devices to reducethe number of inductors by three. Moreover, inductors wouldrequire much more space compared to the space requirement oftwo switches. Therefore, we can state that the proposed dc/dcconverter would reduce both the cost and the size of the con-ventional approach for the same functionality basis. Comparedwith the commercial approach without the grid discharge mode,the proposed converter has six more semiconductor switches.

TABLE VIIISIMULATION CONDITIONS AND CIRCUIT PARAMETERS

However, it reduces the size, cost, and space requirements of theinductors by 50% while adding the grid discharge functionality.If a unidirectional charging scenario is required, the boostfunctionality of the dc/dc converter (from the dc link to thebattery) would not be needed. In this case, the following twosemiconductor devices can be eliminated: 1) D3 and 2) T7.However, there would not be any significant modification on thegrid interface converter, because it is inherently bidirectional.However, the control strategy can be less complicated in thecase of a unidirectional approach.

III. SIMULATION ANALYSIS AND RESULTS

Simulations for the grid interface converter provide the majorwaveforms and results for a typical application along withthe comparisons in terms of noise emissions and harmonicsrelated to the conventional grid interface converter and theproposed converter. Simulations for the dc/dc converter partprovide how the dc/dc converter would work in a real drivecycle and how it successfully regulates the motor drive inputvoltage while managing the input and output power of thebattery. Simulations are performed using MATLAB, Simulink,SimPowerSystems, Signal Processing Toolbox, and ControlSystem Toolbox products.

For the simulations, the charging conditions of a Toyota PriusPHEV are considered in the proposed system. The specifica-tions of the simulation model are summarized in Table VIII.

The variations of the ac voltage, ac current, and Vab voltageacross the grid and inductor are given in Fig. 15 for 1800-Wreference charging power.

As shown in Fig. 15, the current drawn from the grid is inphase with the grid voltage, resulting in a unity power factor.Moreover, the grid current has near-zero harmonic distortionand has an almost-perfect sinusoidal shape. To test the dynamicperformance of the reference charge power tracking feature ofthe proposed system, a step change in reference power has beenapplied at t = 5 from 1000 W to 1800 W. The voltage thatis applied to the battery, the dc link power that is transferredto the battery, the battery current, the battery SoC, and thegrid current variations are provided in Fig. 16. In this mode,note that the rectified ac voltage (dc link voltage) is boostedmore than the battery’s rated voltage of 345.6 V to chargethe battery. To increase the charging power from 1000 W to


Fig. 15. Simulation results for the grid-charging mode.

Fig. 16. Simulation results for the grid-charging mode. (a) Battery chargingvoltage. (b) Battery charging current. (c) DC link power. (d) SoC of the battery.(e) Current drawn from the grid.

1800 W, the converter applies higher voltage to the batteryterminals, as shown in Fig. 16(a); therefore, the battery chargingcurrent increases, as shown in Fig. 16(b). Fig. 16(c) shows thedc link power that is used to charge the battery through thedc/dc converter. The SoC increment of the battery is displayedin Fig. 16(d) to show how its rate of change differs with twodifferent charging power levels. Because the charging powerincreases, the ac/dc converter draws more current from thegrid [see Fig. 16(e)] to maintain the dc link at the grid’s peakvoltage.

In the grid-connected discharging mode, the ac voltage, accurrent, and Vab voltage are recorded, as shown in Fig. 17for a 1000-W reference discharge rate. Similar to the grid-charging mode, the current that is injected to the grid is almosttotally free of distortions, but it is in reverse phase (180◦),with the grid voltage indicating that the power is suppliedback to the grid. Because the zero crossings of the ac voltageand current are identical, no reactive power is injected to the

Fig. 17. Simulation results for the grid-discharging mode.

Fig. 18. Simulation results for the grid-discharging mode. (a) Battery dis-charge voltage. (b) Battery discharge current. (c) DC link power. (d) SoC ofthe battery. (e) Current injected to the grid.

grid. A step change in the reference discharge power has beenapplied at t = 5 from 800 W to 1500 W to test the dynamicperformance of the reference discharge power tracking featureof the proposed system. The battery voltage during discharge,the dc link power that is injected to the grid, the battery current,the battery SoC, and the current, which is injected to thegrid, are shown in Fig. 18. In the plug-in discharge mode, thebattery voltage is stepped down to slightly more than the peakof the grid’s voltage, and this voltage is inverted to the ac.Once the controller senses that the reference discharge poweris changed, more current is drawn from the battery, as shown inFig. 18(b), to meet this demand. Therefore, the battery terminalvoltage drops, as shown in Fig. 18(a), with respect to the drawncurrent. Fig. 18(c) shows the dc link power that is transferredfrom the battery through the dc/dc converter. Fig. 18(d) showsthe SoC decrease of the battery. When the reference discharge


Fig. 19. Simulation results for the driving buck and boost modes. (a) Loadcurrent. (b) Switches in the PWM mode. (c) DC link voltage. (c) Battery’s SoC.

power increases, the SoC drops faster. The multilevel converterinjects more current to the grid, as shown in Fig. 18(e), to keepthe dc link regulated.

For the second phase of the simulations, the dynamic perfor-mance of the battery interface converter is tested over a portionof a well-known Urban Dynamometer Drive Schedule (UDDS)drive cycle for t = [1200, 1369]. A Toyota Prius PHEV with345.6 V nominal battery voltage and 650 V nominal dc link(motor drive inverter input) voltage is considered. The loaddemand in this portion includes acceleration, regenerative brak-ing, and idling conditions that may occur in a typical drivingscheme. While the vehicle is driven, a regulated dc link voltageshould be provided to the motor drive while supplying the loaddemands and recapturing the braking energy. In Fig. 19, theload current profile that corresponds to the drive cycle [seeFig. 9(a)], map of switches operated in PWM [see Fig. 9(b)],regulated dc link voltage variation [see Fig. 9(c)], and theSoC of the battery [see Fig. 9(d)] are provided. When theload demand is positive (mode 3), the vehicle is accelerating,cruising, or idling; therefore, the power flow direction is from

TABLE IXEXPERIMENTAL CONDITIONS AND CIRCUIT PARAMETERS

the battery to the dc link, and the battery voltage is stepped up.When the load demand is negative (mode 4), the motor drivecaptures the braking energy, and the dc/dc converter is operatedin the buck mode to recover the braking energy. As shown inFig. 19(b), T9 is in the PWM mode, whereas the load demandis positive, and T7 is operated in the PWM mode, whereas theload demand is negative. The regulated dc link voltage at 650 Vis provided in Fig. 19(c) while the vehicle is being driven.Fig. 19(d) presents the SoC of the battery, which decreaseswhen the load demand is positive and slightly increases duringregenerative braking.

IV. EXPERIMENTAL SETUP AND RESULTS

The details of the experimental setup of the proposed con-verter are provided in Table IX.

For the experimental validation, a smaller scale setup hasbeen built in the Energy Harvesting and Renewable EnergiesLaboratory with 30-Vrms ac voltage, 42 V dc link voltage,and 24 V battery voltage both in the grid-connected and high-voltage bus loading modes. Because the proposed topologiesare new and have not been built or tested, it would be moreappropriate to build the small-scale prototypes rather than thefull-scale high-power converters. Moreover, due to safety pur-poses and to protect the students and the laboratory equipment,a smaller scale prototype with a lower voltage rating is preferredto serve as a proof of principle.

Because the dc link voltage is higher than the battery voltage,the battery voltage is stepped up in grid discharging and loadingconditions, whereas the dc link voltage is stepped down in grid-charging and regenerative tests. A picture of the experimentalsetup of the proposed topology is depicted in Fig. 20. Asa feedback and control systems interface, the TMS320F2812digital signal processing (DSP) module from Texas Instrumentshas been employed. For programming the DSP, processing thefeedback signals, and control realizations, the Target SupportPackage 4.1 for the TMS320C2000 and Embedded IDE Linkfor Code Composer Studio from MathWorks Inc. have beenused. These tools allow for deploying generated code onto thereal-time embedded microcontrollers and DSPs.

The experimental results of the grid connection mode arepresented in Fig. 21. In Fig. 21(a), CH-1 is the ac grid


Fig. 20. Experimental setup of the proposed converter.

Fig. 21. Experimental results for the grid-connected mode. (a) Grid-chargingac quantities and the dc link voltage. (b) Grid-charging battery voltage andcurrent. (c) Grid-discharging quantities and the dc link voltage.

Fig. 22. Experimental results for the Vbatt to Vdc boost mode.

Fig. 23. Experimental results for the Vdc to Vbatt buck mode.

voltage reduced with a transformer for small-scale testing,CH-2 is the Vab voltage across the input and the trans-former, CH-3 is the current drawn from the grid, and CH-4is the dc link voltage in the grid-charging mode, all in 20 Vand 20 A per division scales. The dc link voltage (CH-1,20 V/div), battery voltage (CH-2, 20 V/div), and battery current(2 A/div) are presented in Fig. 21(b) for grid charging. Finally,the ac voltage, Vab voltage, current that is injected to thegrid, and dc link voltage are provided in Fig. 21(c) inCH-1–Ch-4, respectively. Again, the grid current is 180◦ outof phase with the ac input voltage, indicating that the poweris supplied back to the grid. Because the battery voltage andcurrent are similar in the grid-discharging mode, they are notincluded; the major difference is the direction of the battery


Fig. 24. Harmonic content and THD comparisons of a (a) multilevel converter, (b) diode-bridge rectifier, and (c) full-bridge PWM converter.

Fig. 25. Noise emissions of the multilevel (CH-1) and full-bridge PWM (CH-2) converters. (a) [0, Fs/2] frequency band. (b) [0, Fs] frequency band.

current. The experimental results verify that the ac power qual-ity is in compliance with the IEEE-1547 and 1519 standards,which deal with the harmonic levels and other power systemquality aspects of the equipment.

To verify the capabilities of the proposed system in thedriving mode, the battery to the dc link boost mode and thedc link to the battery buck mode are tested. The experimentalresults of the Vbatt to Vdc boost mode of operation (accelerationor cruising) are presented in Fig. 22, where CH-1 is Vdc, CH-2is Vbatt, CH-3 is the input current, and CH-4 is the gatesignal of the switch T9 operated in PWM. It can be ob-served in Fig. 22 that the proposed converter can provideregulated dc link voltage at its reference value by boosting24 V Vbatt voltage to 42 V Vdc output voltage.

The experimental results for the Vdc to Vbatt buck mode ofoperation (regenerative braking) is presented in Fig. 23, whereCH-1 is Vbatt, CH-2 is Vdc, CH-3 is input current, and CH-4 isthe PWM switching signals of the switch T7. In Fig. 23, it isshown that the input voltage of Vdc is stepped down to about24 V of the battery terminal voltage.

The THD percentages of the conventional diode bridgerectifier, a conventional PWM converter, and the presentedmultilevel converter are shown in Fig. 24, including all theharmonic components. In addition, the switching noise that isemitted by the PWM converter and the multilevel converteris analyzed and compared in Fig. 25. This figure presents theresults of the spectrum analyzer. According to Figs. 24 and 25,the multilevel topology exhibits higher performance in terms of

power quality and noise emissions compared with the other gridinterface converters.

These results show that the proposed converter resulted inhigher power quality by improving the waveforms and reducingthe THDs from 61.02% to 2.35% compared with the diodebridge rectifier and from 4.71% to the 2.35% compared to thefull-bridge PWM rectifier. At 100 kHz, the proposed topologyreduces the electromagnetic interference (EMI) noise from−41 dB to −71 dB compared to the full-bridge PWM-basedpower electronic interface at the [0, Fs/2] frequency band dueto the improved power quality. At the [0, Fs] frequency band,the proposed converter further reduces the EMI emissions from−83 dB to −119 dB due to the improved power quality.

V. CONCLUSION

Because more PHEVs will be on the roads in the near future,it is important to consider the effects that large numbers of plug-in vehicles might have on the grid. To avoid these issues, a high-power-quality grid interface must handle the energy exchangebetween the vehicle and the grid with minimal current harmonicdistortions, high power factor, and less noise. The presentedgrid interface enables V2G interactions, which could improvethe efficiency of the grid. The proposed dc/dc converter, similarto the grid interface converter, exhibits excellent performanceboth in the grid-connected and driving modes of operation. Thisconverter can step down and step up the dc link and battery volt-age, providing bidirectional power flow. Efficiency analysis and


the kilovolt-ampere-ratings-based cost comparative analysis areincluded to represent the feasibility of the proposed converter.The proposed grid interface converter reduces the THD ofan H-bridge converter from 4.71% to 2.35% and reduces thediode bridge rectifier–boost converter combination’s THD from61.02% to 2.35%. The noise emission level of the proposedmultilevel converter is −77 dB, whereas the noise emissionlevel of the H-bridge inverter was −41 dB, which shows thatthe proposed multilevel converter causes less EMI effects. Thekilovolt-ampere analysis of the grid interface converter showsthat the proposed grid interface converter reduces the totalkilovolt-ampere rating of a conventional H-bridge inverter by66%. In addition, the kilovolt-ampere rating analysis for thedc/dc converter part shows that the number of inductors can bereduced from four to one compared to a conventional approachor it can be reduced from two to one in a commercial approachthat does not have the grid discharging functionality.

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Omer C. Onar (S’05–M’10) received the B.Sc. andM.Sc. degrees in electrical engineering from YildizTechnical University, Istanbul, Turkey, in 2004 and2006, respectively, and the Ph.D. degree in electricalengineering from Illinois Institute of Technology,Chicago, in July 2010.

He is currently with the Energy and TransportationScience Division, Oak Ridge National Laboratory,U.S. Department of Energy, Oak Ridge, TN. He isthe author or a coauthor of more than 40 publi-cations including journals, conference proceedings,

and books. His research interests include power electronics; energy harvest-ing/scavenging; renewable energies, particularly solar, wind, and ocean energyconversion systems; grid interconnection of renewable energy sources; powermanagement for sustainable energy systems; and electric/hybrid electric/plug-in hybrid electric vehicles.

Dr. Onar is a member of the IEEE Power and Energy Society, the IEEEVehicular Technology Society, the IEEE Industrial Electronics Society, andthe IEEE Power Electronics Society. He is the recipient of the 2008–2009Transportation Electronics Fellowship from the IEEE Vehicular TechnologySociety, the Joseph J. Suozzi INTELEC® Fellowship in Power Electronics fromthe IEEE Power Electronics Society in 2009, and the distinguished Alvin M.Weinberg Fellowship from the Oak Ridge National Laboratory in July 2010.

Jonathan Kobayashi (S’08) received the B.Sc. de-gree in electrical engineering from Illinois Instituteof Technology (IIT), Chicago, in 2011. He is cur-rently working toward the M.Sc. degree with theDepartment of Mechanical Engineering, Universityof California, Berkeley.

He was a Power Electronics Researcher with theEnergy Harvesting and Renewable Energies Labora-tory, Electric Power and Power Electronics Center,IIT, where he worked on experimental bidirectionalac/dc and dc/dc power electronic converters for plug-

in hybrid electric vehicles. From August 2009 to December 2009, he was aTeaching Assistant for the robotics lab course. From August 2008 to May 2009,he was with the Formula Hybrid Team, IIT, where he worked on control systemsand data acquisition. Earlier, he was with the Hawaiian Electric Company,Honolulu, HI, where he worked on various projects related to transmissionand distribution substations, system protection, and communications. He is anexpert in computer, web, and electrical power technologies.

Mr. Kobayashi received the First Hawaiian Bank Scholarship in 2007–2008,the Heald Scholarship, and the First Robotics Scholarship in 2007. He wasincluded on the IIT Dean’s List in Fall 2007 and Fall 2009.

Dylan C. Erb (S’07) received the B.S. degreein general engineering from the University ofIllinois, Urbana-Champaign, in 2011. He is cur-rently working toward the M.S. degree in mechani-cal engineering with the Department of MechanicalEngineering, Massachusetts Intitute of Technology(MIT), Cambridge.

He is currently with the Field Intelligence Labora-tory, MIT, under the supervision of Prof. S. Sarma.His research interests include the optimization ofenergy storage packs for hybrid and electric vehicles.

Alireza Khaligh (S’04–M’06–SM’09) received theB.S. and M.S. degrees in electrical engineeringfrom Sharif University of Technology, Tehran, Iran,and the Ph.D. degree in electrical engineering fromIllinois Institute of Technology (IIT), Chicago.

He was a Postdoctoral Research Associate withthe Department of Electrical and Computer Engi-neering, University of Illinois, Urbana-Champaign.He was an Assistant Professor with IIT. He is cur-rently an Assistant Professor and the Director of thePower Electronics, Energy Harvesting, and Renew-

able Energies Laboratory (PEHREL), Department of Electrical and ComputerEngineering, University of Maryland, College Park (UMCP). He is the authoror a coauthor of more than 100 journal articles and conference proceedings. Hisresearch interests include the modeling, analysis, design, and control of powerelectronic converters, electric and plug-in hybrid electric vehicles, biomechani-cal energy harvesting, renewable energies, grid integration of distributed energysystems, and smart grid.

Dr. Khaligh is the recipient of the Ralph R. Teetor Educational Awardfrom the Society of Automotive Engineers in 2010, the Armour Collegeof Engineering Excellence in Teaching Award from IIT in 2009, and theExceptional Talents Fellowship and the Distinguished Undergraduate StudentAward from Sharif University of Technology. He was the Program Chair of the2011 IEEE Vehicle Power and Propulsion Conference (VPPC) and the Grantsand Awards Chair of the 2012 IEEE Applied Power Electronics Conference andExposition (APEC). He is a Program Cochair of the 2012 IEEE TransportationElectrification Conference and Expo (ITEC). He is an Associate Editor forthe IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY (TVT) and aGuest Editor for the IEEE TVT Special Section on Sustainable TransportationSystems and the IEEE TRANSACTIONS ON POWER ELECTRONICS SpecialIssue on Transportation Electrification and Vehicle Systems.

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