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A Cost Effective Method of Reducing Total
Harmonic Distortion (THD) in Single-Phase
Boost Rectifier
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CHAPTER 1
INTRODUCTION
1.1.Introduction:
Because of a large number of high harmonics are injected into power with more and more
of the nonlinear semiconductor devices were widely used Harmonic can cause the distortion in
current in a power system and worsen the power quality. Rectifier circuit is the biggest harmonic
sources. At present,research on the PFC of rectifier devices has an important practical
significance, especially based on the average current control single-phase Boost type APFC
circuit.
PFC is used as a positive method for improving the power quality. Essentially PFC can
eliminate harmonic source of rectifier devices, through input current waveform automatically
with input voltage waveform of the grid, and get the former waveform as sine waveform and
have the same waveform with voltage waveform on phase. PFC is mainly included reactive
power factor correction and active power factor correction (short for APFC).Because there is no
source power factor correction technique in be used actually has certain limitation, and power
factor is not high, only can make the power factor improvement to 0.7 ~ 0.8, and APFC can
make the power factor, so close to one currently used more APFC technology. APFC can be
converted to the input current power input and utility in phase with the sine wave, so as to
improve the power factor of electrical equipment, and to reduce the grid harmonic pollution.
Converters operated in discontinuous-conduction- mode (DCM) and in continuous-conduction-
mode (CCM) are suitable for lighter and higher loads, respectively. A new, constant switching
frequency based single-phase rectifier system is proposed, which operates in DCM and in CCM
for outputs less than and greater than 50% rated load, respectively, covering a wide range of load
variation. The power circuit and the control circuit of the proposed rectifier are easily
configurable for DCM and CCM operations. The measured load current is used to select the
desired operating mode. The peak device current under DCM is limited to rated device current
under CCM without using a device of higher current rating. The input current shaping under
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CCM and DCM are based on the comparison of measured input current with linear and nonlinear
carriers, respectively. A load current feed forward scheme is presented to improve the system
dynamic performance and also to ensure a smooth transition between the two operating modes.
Single-Phase diode bridge rectifiers are gradually being replaced by pulsewidth modulation
(PWM) rectifiers tomaintain a sinusoidal input current at near unity power factorand to satisfy
the necessary harmonic standards . Asingle-phase, single-switch boost rectifier (Fig. 1) is a well-
establishedtopology in the field of acdc power conversion to comply with the above harmonic
standards. The rectifier systemmay be operated in the continuous conduction mode (CCM), orin
the discontinuous conduction mode (DCM) . TheCCMis preferred overDCMbecause of
continuous inputcurrent and low conducted electromagnetic interference (EMI). However, it is
reported to have high input current distortionat light load . For a particular switching frequency
andboost inductance, the amount of current distortion increases as the load decreases . A high
valued boost inductor is necessary at light load to limit the input current distortion . This
increases the size, weight, and cost of the converter and resultsin poor system dynamic response.
Hence, CCM is preferred athigher loads.The above issues are not seen, when the converter is
operated in DCM. However, DCM is always associated with high device current stress and
conducted EMI. Therefore, a high currentrated device and a costly EMI filter are necessary at
higher loads.Thus, DCM is preferred for light loads. The present work deals with a constant
output voltage application,where the load current varies over a wide range (10%
to 110% of rated load current) and the converter is required to comply with the necessary
harmonic standards. It can be seen from the above discussion that neither of the operating modes
(CCM and DCM) alone is suitable and economical for the above application. Therefore, the
optimum choice is to operate the converter in DCM during light loads and in CCM for higher
loads . The load boundary between DCM and CCM operations can be set at a suitable level (say
50%) to limit the peak device current stress under DCM up to the rated device current under
CCM without using a higher current rated device. Similarly, the minimum load under CCM, for
which the converter is required to comply is 50% rated load. This permits us to use a low valued
boost inductor compared to the entire CCM case without any degradation in the performance of
the converter. The main challenge associated with such a mixed-mode operation is to realize the
two distinct operating modes (DCM and CCM) into a single converter system without
introducing any appreciable dynamics during transition between the two operating modes. There
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are two possible ways to achieve this. The first method suggests a single-valued boost inductor
with constant but two different switching frequencies (a low switching frequency for DCM and a
high switching frequency for CCM) for the above operation. The second method requires two
different boost inductors (a high valued inductor for CCM and a low valued inductor for DCM)
with a constant switching frequency for above application. The first method is simpler than
the second method, as it only requires the switching frequency of the converter to be changed.
However, the second method requires the physical inductors to be changed. A converter system,
using two different switching frequencies (2.56 kHz for DCM and 25.6 kHz for CCM) and a
single valued boost inductor has been reported . The use of two different switching frequencies
introduces difficulties in designing the EMI filter. The controller works in the principle of
voltage mode control without using any input current sensor. A current sensor is however
required for over-current protection of the converter. The input current distortion under DCM is
high as there is no lowpass filter connected at the input to the converter. The implementation of
the above control scheme involves complex mathematic operations, such as multiplications,
divisions and square root operations. It also requires the peak value and the zero crossing instants
of the input voltage to compute the unit vectors ( and ). These increase the
complexity and cost of the controller. Addressing the above-mentioned issues and using two
different boost inductors (a high valued inductor for CCM and a low valued inductor for DCM) a
new, constant-switching-frequency based rectifier system is proposed in this paper. The power
circuit of the proposed converter system can be configured either for CCM or for DCM by
performing a simple on-off control of an auxiliary switch. A DCM power topology, with an input
side lowpass filter is obtained, when the auxiliary switch is on. Again, a CCM power topology
(without any input filter) is realized, when the auxiliary switch is off. A simple, input voltage
sensorless, current-mode controller is proposed for the above rectifier system. The controller
works in the principle of one-cycle control or the nonlinear carrier control , without using any of
the above-mentioned complex mathematical operations. The required gating pulses for the
converter switch are generated by comparing the measured input current with one of the two
periodic carriers in a modulator. A linear carrier is used under CCM, while a nonlinear carrier is
selected under DCM. The measured load current is used to select the desired operating mode
(CCM or DCM). A simple load current feedforward scheme is used to improve the dynamic
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response of the converter system, which also ensures a smooth transition from one operating
mode to the other. The proposed concept has been simulated onMATLAB/SIMULINK platform.
A single-phase, single-switch boost rectifier is shown in Fig. 1, where Fig. 1(a) and (b) represent
the DCM and the CCM boost rectifier topologies, respectively. A lowpass filter is used in the
DCM topology for filtering the switching current harmonics, which is absent in the CCM
topology. Further, it can be shown that for the same switching frequency, the value of boost
inductor is much lower than . Thus, for the same switching frequency, the DCM topology is not
suitable for CCM operation and vice versa. Similarly, it can be shown that the control scheme,
suitable for CCM application may not be useful in DCM operation and vice versa. Therefore, a
common rectifier system (power circuit topology and control scheme) is required to be
developed, which is suitable for both CCM and DCM. Such a rectifier system is developed
in this section.
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that no isolated gate drive is required to drive the switch . Now, it is required to develop a
suitable controller for controlling the proposed power topology, which is suitable for both
CCM and DCM operations.
1.1 An Efficient Quasi-Active Power Factor Correction Scheme:
The ac/dc rectifier consisting of a diode bridge rectifier followed by a filter capacitor is
simple, economic and robust in construction, but produces a harmonic-rich ac line current with
poor power factor. To overcome the problem, passive and active power factor correction (PFC)
circuits have been proposed. In general, active circuit methods are more efficient, lighter in
weight, and less expensive than passive circuit methods . Recent international regulations
governing the harmonic content of the input current drawn by electrical equipment have inspired
the development of many new active PFC circuits employing switched mode DC-DC converters.
Active PFC, classified by the system configuration, can be categorized into two-stage and single-
stage (S2) schemes. A two-stage scheme results in high power factor and fast response output
voltage regulation by using two independent controllers and optimized power stages, as shown in
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Fig. 1 (a). The main drawbacks of this scheme are its relatively higher cost and larger size
resulted from its complicated power stage topology and control circuits,
particularly in low power applications. An S2 scheme combines the PFC cell and the dc/dc
power conversion cell into one stage, and typically uses only one controller and shares power
switches, as shown in Fig.1 (b). Usually, in the S2 scheme the high power factor is guaranteed by
operating the PFC boost cell in discontinuous current mode (DCM), while the fast response
output regulation is achieved by the dc/dc cell. Many S2 converters have been reported in the
literature in recent years . Although the single-stage scheme is quite attractive in low cost and
low power applications due to its simplified power stage and control circuit, major issues, such
as low efficiency and difficulty to move to higher power level, and high as well as widerange
intermediate dc bus voltage stress.
To overcome the disadvantages of the single-stage PFC converter, a new concept of quasi-
active PFC has been proposed . In most off-line dc/dc converters, the input current is made up of
a series of discontinuous current pluses. It is this discontinuous current that allows the quasi-
active PFC to function. The concept is based on using standard dc/dc topologies with minor
changes to obtain fast response at a low cost. In this case the main objective is to obtain an input
current with a permitted amount of harmonic content as per the regulation at a low cost and high
efficiency. The quasiactive PFC circuit shown in Fig.1.2 improves the efficiency of a single-
stage converter by preventing the input current from the ac mains from being added to the active
switch . As the circuit uses resonance of circuit parameters to achieve PFC, the control of the
power factor will be very sensitive to the variation of components values.
In this paper a new quasi-active PFC circuit is presented. This PFC circuit is based on
adding two auxiliary windings coupled to the transformer of a cascade dc/dc DCM flyback
converter. The PFC circuit is placed between the input rectifier and the low-frequency filter
capacitor used in conventional power converter. Since the dc/dc converter is operated at high
frequency, the auxiliary windings produce high frequency pulsating source such that the input
current conduction angle is significantly lengthened and the input current harmonics is reduced.
The input inductor can be designed to operate in DCM or CCM. The single-stage PFC circuit has
a slightly higher efficiency when it operates in CCM. However, DCM operation gives a
lowharmonic content compared to that of the CCM operation. Hence, by properly designing the
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converter components, a good trade-off between efficiency and harmonic content can be
achieved.
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Fig. 1 Functional block diagram of PFC converters: (a) two-stages, (b) single stage.
Fig. 1.2 The quasi-active PFC circuit
CHAPTER 2
LITERATURE REVIEW
PROPOSED QUASI-ACTIVE PFC CIRCUIT:
The proposed quasi-active PFC circuit is discussed in this section. As shown in Fig.2.1, the
circuit consists of a bridge rectifier, a boost inductorL4, a capacitorC3, an intermediate dc bus
capacitor C1, and a discontinuous input current power load, such as DCM flyback converter.
Two auxiliary windings L3 & L5 are added to the primary side of the transformer in order to
shape the input current and improve the efficiency. In the proposed quasi-active PFC scheme, the
dc/dc converter section is not directly involved in the PFC; it simply offers a driving power by
applying a series of discontinuous current pulses. The quasi-active PFC network can be
considered as one power stage without an active switch.
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Fig.2.1. The proposed quasi-active PFC circuit diagram
To simplify the analysis, the following assumptions are made:
All semiconductors components are ideal. According to this assumption, the primary
switch and the rectifiers do not have parasitic capacitances and represent ideal short and
open circuits in their on and off states, respectively
The power transformer does not have the leakage inductances because of the ideal
coupling
The values of all capacitors are sufficiently high so that the voltages across them are
considered constant.
Finally, the input voltage of the converter is considered constant during a switching
cycle because the switching frequency is much higher than the line frequency
A. Operation stages of the proposed method:
In the proposed circuit, properly designing the inductorL4 and the turns ratio n1, n3, and
n5 the input inductor current iL4 can be made to operate either in DCM or CCM for a line
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current period. Operating in DCM gives low THD and high power factor while in CCM the
circuit gives better efficiency and lower power factor. Therefore the design objective is to obtain
an input current with a harmonic content which meets the regulations. Cost and efficiency are of
primary concern. A typical CCM input inductor current produced by the proposed circuit is
shown in Fig. 2.2. It can be seen, that during one half of period of the supply the waveform of the
input inductor current iL4 waveform always has two parts, the DCM part and the CCM part.
When the instantaneous input voltage vin is less than the capacitor voltage vc3 the inductor
current is also low and therefore the input inductor current is in DCM. With the increase of
instantaneous input voltage vin vc3 the inductor current increases and becomes continuous.
In steady state operation the topology can be divided into three operating stages, as shown in
Fig. 2.3. The converter waveforms for both CCM and DCM parts are shown in Fig. 2.4 and Fig.
2.5 respectively.
Stage 1(to-t1): At t= to the switch (SW) is off and diode D1 is also off due to VC1 (which is
always higher than the rectified input voltage). The inductorL4 absorbs the input energy, and the
inductor current iL4 is increased linearly. The current i3 through the windingL3 is equal to (iL4)
at this stage. The capacitorC3 gets charged during this stage. At this time, the output diodeD2 is
forward biased and the stored energy on the transformer primary winding is transferred through
the transformer secondary winding to the output load, so the transformer secondary current i2 is
decreasing. AsL1,L2,L3 andL5 are windings of an ideal transformer, based on Amperes law,
it has
n1i1+n3i3-iD1n5-i2n2 = 0 (1)
Where, n1, n2, n3 and n5 are the no. of turns of the transformers primary, secondary and the two
auxiliary windings respectively. Without loss of generality, n2 is assumed to be equal to one.
Since i1= im, iL4= i3 and iD1= 0 in this stage
i2=n1im+n3iL4 (2)
As it can be seen from (2), the load absorbs energy not only from the magnetizing inductor but
also from the input source through the auxiliary winding n3. Therefore,there is a direct power
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transfer from the input source to the load during this period. As a result, the efficiency is
improved. This stage ends when the secondary current i2 becomes zero at (t= t1).
Fig. 2.2 Input inductor current IL4 illustrating its conduction modes during half line current
period
Stage 2(t1-t2) (current source stage): During this stage the switch SWand both diodesD1,D2are turned-off. The input current continues to flow through the resonant circuit formed by Vin,
L4,L3, C3. The capacitorC3 continues to get charged by Vin at this stage. The current iL4(=i3)
slightly decreases when Vc3>Vin (for DCM) and is constant when Vc3=Vin (for CCM part). This
stage ends when the switch (SW) is turned-on at t= t2.
Stage 3(At t= t2): This stage can be explained into two partsFor the DCM part (where vin < vc3): when the switch is turned on, the bulk capacitor C1
discharges through the transformer primary and hence the primary winding current i1 increases.
Once the voltage across C1 drops, diode D1 starts also conducting, then inductors L4, L3 and
capacitorC3 release their energy to the primary windings L1 andL5, so more energy is stored in
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the primary side windings of the transformer. This means that the transformer magnetizing stored
energy is not only from C1 but also from the input. Therefore, C1 is in the discharging mode.
Note that because vin is low the current iL4 will decrease to zero (DCM) before the switch is
turned off. Once iL4=0 the inductor current i3 continues to supply the energy to the primary
winding of the transformer. At t= t4, the switch is off again, and i1 decreases to zero, therefore
D1-off and D2 is on. The current i3 reverse its direction again where it equals to iL4 and the
switching cycle repeats again.
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Fig. 2.3 Equivalent circuits for the three stages of operation of the proposed PFC circuit
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Fig. 2.4 Key switching waveforms of the proposed PFC with CCM operation
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Fig. 2.5 Key switching waveforms of the proposed PFC with DCM operation
Based on Amperes Law
n1i1+n3i3-n5iD1-n2i2 = 0 (3)
At this stage i2 = 0 and substituting foriD1=i3+iL4 therefore:
Since the magnetizing inductance Lm >> L1, therefore the switch current through the main
switch (SW) is approximately isw -i1. From Fig. 5 (stage 3) the switch current is given by:
isw =-i1 = iD1+ic = iL4+ i3 + ic (5)
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2.1 A New Hybrid Active Power Filter for Harmonic Suppression and Reactive Power
Compensation:
Harmonic currents generated by power electronicsrelated appliances in industry are
causing more and more attention. Harmonics can not only influence the normal work state of
electric equipments, but also interfere communication equipments. To improve the power
quality, some methods for harmonic suppression have been used in power systems. Active power
filters (APF) have been an effective way since they can compensate harmonics dynamically .
However, due to the high cost of high-speed and high-power switching devices such as power
IGBTs, IGCTs etc, APF can not be widely applied in industry.
In order to reduce the cost and capacity of APF, some hybrid active power filters (HAPF),
combining passive power filters and active power filters connected in series or parallel, have
attracted widespread concernt. In 1990, Fujit.H proposed a typical topology of the shunt hybrid
active power filter, combined of passive filters and a small-rated active filter connected in series
with each other. The passive filters remove load harmonics, and the active filter plays a role in
improving the filtering characteristics of the passive filter. It results in a great reduction of the
required rating of the active However, in order to further reduce the capacity of active power
filter leading to a practical and economical system. power filter, passive power filters are
designed as a high fundamental impendence, there is little fundamental current flowing through
active filter, so the structure is only used to suppress harmonic currents of the grid usually. In
case that harmonic suppression and high capacity reactive power compensation are both needed,
the filter can not satisfy the requirement.
The harmonic current detection process, current control for active power filter is another
important aspect. Because the reference current of the system is periodic, traditional
PIcalgorithm is not suitable Control methods based on neural network or fuzzy controllers are
commonly used. However, when they are applied to eliminate the harmonic currents, the method
based on neural network is too complex to realize and the method based on fuzzy controllers is
difficult to achieve the ideal precision.A method having both high precision and fast
dynamicresponsecharacteristic is desired. In this paper, a new type hybrid active power filter,
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containing a high capacity passive power filter and an active power filter, is proposed. Prediction
method is proposed to detect the harmonics to decrease the delay caused by sampling and
calculation, and a recursive integral PI algorithm is proposed to control the harmonic currents in
order to improve the control precision and dynamic response characteristic. Simulations and
industrial application results have shown that the proposed hybrid active power filter has good
performance in both reactive power compensation and harmonic suppression.
2.2 PRINCIPLES OF OPERATION:
The system configuration of the proposed hybrid power filter is shown in Fig.2.6. It consists of
an active power filter and passive power filters. In this system, passive power filters are tuned at
the 100Hzand 250Hzto suppress part of harmonic currents and improve the power factor. Theactive power filter generates harmonic currents opposite to the harmonic currents caused by non-
linear loads and improves the performance of passive power filters. The compensating current is
injected to the power grid through coupling transformer and capacitor CF. The active power
filter shown in the fig.2.6 focuses on suppressing harmonic currents and improving the
performance of passive power filters, it is not necessary to compensate reactive power.
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Fig.2.6 Topology ofthe proposed hybrid active filter.
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2.3 Design and Analysis of Passive Power Filter:
The single-phase equivalent circuit of the proposed passive power filter is shown in fig. 2.7
(
Fig.2 .7 The single-phase equivalent circuit of passive power filters with coupling transformer
L] and C] are tuned at fundamental frequency. V] will be a low voltage, it makes the active
power filter realize easily by semiconductor switching device. CH C] and L] are tuned at 100Hz
and CFis mainly used to compensate reactive power. The impendence ofLo is so big that the
influence ofLo to the passive filters can be neglected. C2 and L2 are tuned at 250Hzto suppress
the fifth order harmonic and compensate reactive power. So the values of CH Cj, Lj, C2 andL2
should satisfy the following equations:
W0L1=1/w0C1 (1)
2w0L1 = cF+c1/2w0(c1. cF) (2)
5w0L2 = 1/5w0C2 (3)
wo is fundamental frequency.
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The equivalent circuit of active power filter is shown in Fig.2.8.
Fig.2.8 The single-phase equivalent circuits of injecting branch.
When the system works, part of the compensating currents generated by APF will flow through
fundamental series resonance circuits. If APF produces the fifth order harmonic current, it will
mostly flow into the fifth resonance branch, so APF is used to produce the seventh order and
higher order harmonics. The equivalent impedance ofLj, C] at 350Hzcan be obtained from the
following equation (4), and the equivalent impedance ofCFat 350Hzcan be obtained through
the equation (5). Thus, the proportion of the current between CFand the auxiliary winding of the
transformer can be calculated by equation (6)
The ratio will increase as the harmonic frequency increases; therefore, mostly harmonic currents
produced by active power filter can be injected into power grid. As a result, the proposed active
power filter in fig.2.6 can inject plentiful harmonic current and compensate high-capacity
reactive power at the same time.
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2.4 RECURSIVE INTEGRAL PI CONTROL ALGORITHM APF:
The proposed recursive integral PI controller is composed of recursive integral PI and
fuzzy control as shown in Fig.2.9, where e stands for the input of the controller, De is for
variational rate of e; (x represents the x phase's compensating current; vde and v;e are the de side
voltage of PWM inverter and its reference value, respectively, while isx and is! respectively
denote the source current and source fundamental current obtained from ip ' iq theory; iREF
represents the reference signal.
Fig.2.9.structure of proposed controller
The input of the controller is periodic and its period is 20ms. It is composed of a series of
sinusoid whose periods are multiple of 20ms. Adopted is traditional PI algorithm, it will integrate
every sampling point. The output ofthe controller is
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u and e respectively denote the output and input of the controller; , andK; respectively stand
for process gain and integral constant, andKis sample time.
Recursive integral PI algorithm is proposed on the basis of the traditional PI algorithm, it
integrates e in every period, which is equivalent to n PI controllers work together, thus a fast zero
steady-state error adjusting is realized(n is an integer).
2.5.Study of a single-phase series active power filter with hysteresis control:
Active power filters (APF) are promising tools to improve the quality of the electrical
energy . Series active power filters are applied meanly to remove distortions from the good
quality form of the AC grid voltage . The most widely used power circuit topology of the single-
phase filter is the fullbridge shown in Fig.2.10. The operation of APF is based on adding the
produced instantaneous value of the voltage UFto the voltage of the grid US(adding means also
subtraction of those voltages when required) in order to produce a good sinusoidal waveform of
the voltage UL that supplies the load. The filterLF, CFis added to eliminate the high frequency
harmonics due to the switching. The transformer ratio of Tr depends on the value of the DC
supply (storage) voltage. The control implementations may be of different nature: pulsewidth
modulation, predictive control, sliding mode control, etc. . Some implementations maintain a
constant switching frequency , and some apply varying switching frequency, e.g. the hysteresis
control (which may be limiting the maximum switching frequency). This last method allows
minimizing the power losses in the switches, simplifying the design of the filterLF , CF , and
thus improving the electromagnetic compatibility of the APF.
The aim of this study is to investigate the operation of the series APF shown in Fig.2.10 when its
control is based on hysteresis, both with and without limitation of the maximum switching
frequency.
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Fig.2.10 Power stage of a single-phase full-bridge series active power filter
2.5 OPERATION:
The hysteresis operation of this APF is based on the comparing the instantaneous value of a
reference sine wave UREF(synthesized in phase with the grid voltage US) to the load voltage
UL . The comparison is performed within the defined hysteresis voltage interval H . If the
instantaneous value of the load voltage is lower than that of the reference ( UREF H ), the
transistors VS1,VS3 turn on. In case it is higher than the value UREF+ H , then the transistors
VS2,VS4 turn on. The reference sinusoid is generated by measuring the instantaneous voltage atthe point of common coupling (PCC) US. If the reference sinusoid value would be kept equal to
that of the first harmonic of the USthen only a filtering will be performed. When the reference
sinusoid has
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Fig 2.11. Building blocks and time diagrams illustrating the tracking down of the reference curve
at a not limited maximum switching frequency
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2.6 Hysteresis control at limited maximum switching frequency:
Fig.2.12 illustrates the concept of the implementation of the hysteresis control with an imposed
limitation of the maximum switching frequency. The turning on of the power switches is not
allowed immediately at reaching the value UREFHforVS1,VS3 and the value UREF+Hfor
VS2,VS4 . The turning on is performed after arriving at the corresponding values and delayed
synchronously by the length of a clock signal. The frequency MAX Fof that signal will limit the
maximum switching frequency of the power switches.
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Figure 2.12. Building blocks and time diagrams illustrating the tracking down of the reference
curve at a not limited maximum switching frequenc
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CHAPTER 3
PROPOSED CONVERTER
3.1The fundamental principle of APFC:
In fact,APFC is meaning that the rectifier voltage which the input alter-current (short
for AC) signal is converted direct-current (short for DC) voltage through the bridge diode is
changed into the current signal by DC/DC converter and the proper control methods. The current
wave which can auto track the DC voltage wave is changed with a sine wave, and get a steady dc
output voltage.
The fundamental principle frame of APFC is shown in Figure3. 1
.
Fig 3.1. The fundamental principle frame of APFC
Figure 3.1 input by rectifier after rectifying, alternating current will get sinusoidal voltage waveform signal as
the input current IC simplifies PFC reference waveform and then by simulation on time-multiplier operations, will
get as the result of current waveform reference, and the value of the current value and the actual sampling
comparison, then after driving circuit to control signal generated driver circuit DC/DC current output and output
voltage.
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3.2The main circuit topology of APFC:
The main circuit topology is usually brought out with DC/DC converter. The main circuit
topology is consisted of buck, boost/buck, flyback and boost circuit.
Buck circuit is rarely used as the big noise and the bad filtering. Boost buck circuit has a
complexity circuit. Flyback circuit is usually used in low power application. The last one is a
simple current control circuit because of the high PF value, the low total harmonic distortion and
the high efficiency. The peak current of boost APFC is nearly equal to the input current. The
amplitude of the peak voltage of boost APFC is higher than the grid-side voltage.
The advantages of boost APFC are as follows.
the continuous input current and easy filtering.
Can contain a higher PF in the range of all input voltage
output DC voltage than input DC voltage peak.
3.3.The control methods of input current: At present, the strong practicality control method is CCM which is brought out with
multiplier. CCM mode is included directly and indirectly control. Directly control methods is
consisted of Average Current Mode Control (short for ACMC), Peak Current Mode Control
(short for PCMC), lag current control and current tracking control. The advantages of ACMC
mode are frequency stability, CCM mode. Low switching current and so on. The four control
methods, CCM is most frequently used, especially in the boost APFC.
The frame of CCM of boost APFC is shown in Figure 3.2.
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Figure 3. 2. The frame of CCM of boost APFC
Figure 3.2 APFC circuit average current pressor type control circuit principle diagram the
average current control type pressor type of active power factor correction simulation In order to
validate the circuit topology average current control type pressor type of active power factor
correction effect, based on MATLAB simulation software was founded on the circuit topology
simulation, and the simulation results circuit to carry on the detailed analysis and research.
3.4 APFC BASED ON THE CIRCUIT TOPOLOGY AVERAGE CURRENT CONTROL:
In order to validate the circuit topology average current control type of active power factor
correction effect, based on MATLAB simulation software was founded on the circuit topology
simulation, and the simulation results circuit to carry on the detailed analysis and research. In
this paper the design of index parameters: the AC voltage source for standard sinusoidal RMS is
220V voltage, frequency f = 50Hz ac inductance, L = 0.5 mH, C = 960, R = 2. Control circuit of
the main average current control. Standard sine signal from power signals, modulation signal is
for the triangle 5kHz frequency wave, output voltage given value for 400V.
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Proposed Circuit:
An ac to dc converter consisting of a line frequencydiode bridge rectifier with a
large output filter capacitor is cheap and robust, but demands a harmonic rich ac
line current. As a result, the input power factor is poor. Due to problems associated
with low power factor and harmonics, harmonic standards and guidelines, which
will limit the amount of current distortion allowed into the utility, is introduced.
Thus the simple diode rectifiers may not in use. To correct the poor power factor
and reduce high harmonic current contents, passive and active circuits can be used.
In general, active methods are more efficient, lighter in weight, and less expensive
than passive circuit methods. In active power factor correction techniquesapproach, switched mode power supply (SMPS) technique is used to shape the
input current in phase with the input voltage. Basically in this technique power
factor correcting cell makes the load behave like a resistor leading to near unity
power factor. Fig. 1 shows the circuit diagram of basic active power correction
technique. There are different topologies for implementing active power factor
correction techniques including the boost converter and the buck converter. For
reasons of simplicity and its popularity, the boost converter is used to improve the
power factor. In boost circuit, the switching device handles only a portion of the
output power and this property can be used to increase the efficiency of the
converter. The boost converter may be designed to operate either in the continuous
conduction mode (CCM) or in the discontinuous conduction mode (DCM).
Compared with the CCM approach, a converter operating in DCM provides a
simpler control scheme, which requires only one (voltage) control loop to
modulate the on-time, Fig 2. Furthermore, operating a boost converter in
discontinuous mode avoids the output diode reverse recovery problem and
alleviates the high switching loss in continuous mode operation. One drawback of
DCM PFC approach is that its input current waveform is not always purely
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sinusoidal. The input current will contain certain distortion due to the modulation
of inductor current discharging time. This waveform distortion is found to be a
function of the ratio of the peak line voltage to the output voltage of the PFC
circuit.
Generally, to reduce the harmonic current contents of DCM boost converter, the
duty cycle of the rectifier switch needs to be properly modulated during a rectified
line period instead of being kept constant. Recently, a number of duty cycle
modulation techniques for the DCM boost rectifier have been introduced to reduce
the total harmonic distortion (THD) of the input current for singlephase
and three-phase systems. Specifically, the approach based on variable switching
frequency control was presented and analyzed. However, since the switching
frequency directly depends on the input voltage and output power variations, the
variable switching frequency method suffers from very wide frequency range
which decreases the efficiency and makes the rectifier design and control circuit
more complex. To improve the performance of the DCM boost converter at
constant switching frequency, harmonic injection methods have been introduced,
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which gives high quality input current at the cost of complicating the control
circuitry. The purpose of this paper is to propose a new simple,
low cost harmonic reduction method, which has the simplicity of voltage follower
technique. In the proposed method a periodic signal proportional to the rectified ac
line voltage is injected into the control circuit to modulate the duty cycle of the
power switch, S, such that the amplitude of third-order harmonic of the line current
is reduced and THD is improved. The generation of the injected signal with unity
amplitude is simplified by sensing the output voltage of the bridge rectifier Vg
(Fig.1). As a result, the additional circuit required to generate and synchronize the
second-order harmonic signal, Fig. 2, proposed in is eliminated. Experimental and
simulated results show the effectiveness of the proposed method.
ANALYSIS OF THiE CURRENT WAVEFORM DISTORTION BOOST
RECTIFIER:
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The voltage follower (DCM) boost converter is shown in Fig. 3, assuming that the
converter operates in DCM with a constant switching frequency, fs. The diode
bridge rectifier is used to rectify the ac line voltage, and the semiconductor power
switch S, the inductor L, and the output diode Do operate as a boost chopper. The
Power switch, S, is operated at high switching frequency and the output voltage is
regulated by varying the duty cycle of the switch, S. A capacitor C0 is used to
reduce the ripple in the output voltage. The EMI input filter is used to filter out the
high frequency components in the input current.
In voltage-follower PFC circuit, the on-time, Ton, is designed to change slowly
and is almost constant over an ac line cycle. In general, the input current in
constant switching frequency boost converter is composed of a charging
component and discharging component, as shown in Fig. 4. Assuming sinusoidal
input voltage (Vs=V1sinwt), the peak inductor current, Ipk, is determined as
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Where T is the period of a switching cycle, D is the duty cycle. The peak inductorcurrent follows an envelope of the input voltage. At the end of on-time, the
inductor current is discharged to the output and is reset by a voltage of VO - Vs,
where VO is the output voltage of the boost converter. The discharging time Td, is:
The ac line, in effect, sees an average inductor current waveform due to the
presence of the input filter capacitor and the stray line inductance.
Where I1(avg) is the line current, I,, (avg) is the average of inductor current during
the on-time, and Id(avg) is the average of inductor current during discharge
time.
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Using Fourier analysis, Iin can be shown to contain several low-order harmonics.
The significant lower-order harmonics are the third, fifth, and the seventh. Among
the three lower-order harmonics, the third harmonic component has the largest
amplitude and has to be attenuated using PWM.
NEWLOW COST INJECTION SIGNAL IMPLEMENTAION:
In the proposed circuit as shown in Fig. 6, a signal d(t) proportional to the rectifiedac line voltage is injected into the control circuit to modulate the duty cycle of the
power switch, S, such that the amplitude of third-order harmonic of the line current
is reduced and THD is improved. In this paper, the generation of the injected signal
with unity amplitude is simplified by eliminating the additional circuit required to
generate the second-order harmonic signal proposed , Fig 2. It also eliminates the
need to employ phase detecting and phase-locking circuits to properly synchronize
the injected signal with rectifier input current. The proposed harmonic injection
technique uses a voltage signal which is proportional to the rectified ac input
voltage. As a result, the injected signal is naturally synchronized with the input
voltage. Therefore, to modulate the duty cycle of the power switch, we can employ
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the output of the diode bridge rectifier of the power stage, which contains a
second-order harmonic and higher-order components such as 4th, 6th 8 ,...etc.
Therefore the injected signal can be expressed as:
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Where 0 < m < 1 is the modulation index, and M is the input frequency. A filter
capacitor, Cin, is placed between the bridge rectifier and the boost inductor L, in
order to reduce the high frequency ripple of the injected signal. Cin can also reduce
the peak inductor current as well as the power switch peak current due the DCM
operation. Therefore, using the modulated duty cycle, which optimizes the time-on
of the power switch, and the capacitor Cin, the switching loss of the power switch,
can be reduced. It is Possible to get the optimal values of modulation index m and
the capacitor Cin , which make the total harmonic distortion THD as small as
possible for a given values of input and output voltages. The optimal values of m
and Ci, that result in a THD of less than 400 are found out through the simulation
and later implemented in the experimental converter. The modulation of the duty
ratio during a line cycle can be expressed as:
Dmod (t) = D[1 + d(t)]
Where Dmod is the modulated duty cycle, and D is duty cycle in the absence of the
modulation,with the modified duty cycle Dmod defined in the average input
current in the presence of the signal injection can be described as
Where [d(t)]2 term is neglected, since it is much smaller than the unity. Since the
third-order harmonic is the dominant harmonic with constant switching frequency
PWM control, the input current can be approximately expresses as
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Where I, and I3 are constant values. Substituting for D with the modified duty
cycle Dmod , and by setting n=2 for the second-order harmonic, since it has the
highest magnitude, the input current can be rewritten as
Furthermore, if we consider the fifth-order harmonic in input current, the term [-
mI3 sin(5ay)] can reduce the amplitude of this harmonic component by small
amount of (-m13). Therefore, the total harmonic distortion (THD) of the input
current with the modulated duty cycle is given by
Obviously THD with injection is much smaller than THD I3 without injection.
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CHAPTER 4
SIMULATION RESULTS
MATLAB is a high-performance language for technical computing. It integrates computation,
visualization, and programming in an easy-to-use environment where problems and solutions are
expressed in familiar mathematical notation. Typical uses include-
Math and computation
Algorithm development
Data acquisition
Modeling, simulation, and prototyping
Data analysis, exploration, and visualization
Scientific and engineering graphics
MATLAB is an interactive system whose basic data element is an array that does not require
dimensioning. This allows solving many technical computing problems, especially those with
matrix and vector formulations, in a fraction of the time it would take to write a program in a
scalar non-interactive language such as C or FORTRAN.
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The MATLAB system consists of six main parts:
(a) Development Environment
This is the set of tools and facilities that help to use MATLAB functions and files. Many of these
tools are graphical user interfaces. It includes the MATLAB desktop and Command Window, a
command history, an editor and debugger, and browsers for viewing help, the workspace, files,
and the search path.
(b) The MATLAB Mathematical Function Library
This is a vast collection of computational algorithms ranging from elementary functions, like
sum, sine, cosine, and complex arithmetic, to more sophisticated functions like matrix inverse,
matrix Eigen values, Bessel functions, and fast Fourier transforms.
(c)The MATLAB Language
This is a high-level matrix/array language with control flow statements, functions, data
structures, input/output, and object-oriented programming features. It allows both "programming
in the small" to rapidly create quick and dirty throw-away programs, and "programming in the
large" to create large and complex application programs.
(d) Graphics
MATLAB has extensive facilities for displaying vectors and matrices as graphs, as well as
annotating and printing these graphs. It includes high-level functions for two-dimensional and
three-dimensional data visualization, image processing, animation, and presentation graphics. It
also includes low-level functions that allow to fully customize the appearance of graphics as well
as to build complete graphical user interfaces on MATLAB applications.
(e)The MATLAB Application Program Interface (API)
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This is a library that allows writing in C and FORTRAN programs that interact with MATLAB.
It includes facilities for calling routines from MATLAB (dynamic linking), calling MATLAB as
a computational engine, and for reading and writing MAT-files.
(f) MATLAB Documentation
MATLAB provides extensive documentation, in both printed and online format, to help to learn
about and use all of its features. It covers all the primary MATLAB features at a high level,
including many examples. The MATLAB online help provides task-oriented and reference
information about MATLAB features. MATLAB documentation is also available in printed form
and in PDF format.
(g) Mat lab tools
(i) Three phase source block
The Three-Phase Source block implements a balanced three-phase voltage source with internal
R-L impedance. The three voltage sources are connected in Y with a neutral connection that can
be internally ground.
(ii) VI measurement block
The Three-Phase V-I Measurement block is used to measure three-phase voltages and currents in
a circuit. When connected in series with three-phase elements, it returns the three phase-to-
ground or phase-to-phase voltages and the three line currents
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(iii) Scope
Display signals generated during a simulation. The Scope block displays its input with respect to
simulation time. The Scope block can have multiple axes (one per port); all axes have a common
time range with independent y-axes. The Scope allows you to adjust the amount of time and the
range of input values displayed. You can move and resize the Scope window and you can modify
the Scope's parameter values during the simulation
(iv). Three-Phase Series RLC Load
The Three-Phase Series RLC Load block implements a three-phase balanced load as a series
combination of RLC elements. At the specified frequency, the load exhibits constant impedance.
The active and reactive powers absorbed by the load are proportional to the square of the applied
voltage.
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Three-Phase Series RLC Load
(v) Three-Phase Breaker block
The Three-Phase Breaker block implements a three-phase circuit breaker where the opening and
closing times can be controlled either from an external Simulink signal or from an internal
control signal.
Three-Phase Breaker block
(vi) Gain block
Gain block
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The Gain block multiplies the input by a constant value (gain). The input and the gain can each
be a scalar, vector, or matrix.
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Discrete,
s = 5e-007 s.
powergui
V o l t a g A n d C u r r e n t P f
Vac
v+-
V3
v+-
V2
v+-
V1
S c o p e 5
S c o p e 4
S c o p e 3
S c o p e 2
P r o d u c t 2
P r o d u c t 1
P r o d u c t
O u t p u t V o l t a g e
g
m
D
S
Mosfet
L2L1
L
1s
I n t e g r a t o r 1
1
s
I n t e g r a t o r
I n p u t C u r r e n t
i+
-
Imes1
i+
-
Imes
1 / 4
G a i n 3
1 / 2
G a i n 2
- K -
G a i n 1
1 / 2
G a i n
ma
k
D5
m
a
k
D4m
a
k
D3
m
a
k
D2
m
a
k
D1
2
C o n s t a n t 2
1 / 1 0 0
C o n s t a n t 17 0 0
C o n s t a n t
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Fig.4.1 voltage and current power factor output
Fig.4.2 input current wave form
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Fig 4.3 output voltage wave form
Fig 4.5 total harmonic distortion wave form
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CHAPTER 5
CONCLUSION
Active power factor correction circuit is analyzed and studied in this paper. The simulation
model is built in MATLAB. The results show that the grids-side input current wave is changed
with a sine wave, has the same phase with the grid-side input voltage wave. In some way , the
power factor is improved. In addition, the structure of this circuit is very simply, easy to bring
out and widely used in the big power Occasion.
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REFERENCES
[1] Wang Zhaoan,Yang Jun,Liu Jinjun,Wang Yue,harmonics suppression and reactive power
compensation , the Second Edition , [M] , China Machine Press (CMP),2009.
[2] Athab , H.S. , Lu, D.D.-C. , An efficient quasi-active power factor correction scheme Power
ElectronicsIEEE Transactions on Volume:25 ,pp 1103-1109,2010.
[3] Chen Zhe, Boost APFC device design[J], Electrical Technology, 1:pp 45-50,2010.
[4] Qi Lei , Xi ZiqiangXin Zhanqiang, Huang Wencong, Research on single phase APFC with
method of hysteresis control[J] , Editorial Department of Hubei University of Technology, 25(1):
pp 52-54, 2010.
[5] Jovanovic,M.M.,Jang, Y..State-of-the-art, single-phase, active powerfactor- correction
techniques for high-power applications - an overview. Industrial Electronics.IEEE Transactions
on Volume: 52 , pp :701 - 708 ,2005.
[6] Lin Fei , Du Xin. The MATLAB simulation of power electronics application
technology[M].China Electric Power Press,2009.