A LOW-DISTORTION CLASS-AB AUDIO AMPLIFIER WITH HIGH
POWER EFFICIENCY
BY
CHAITANYA MOHAN, B.Tech
A thesis submitted to the Graduate School
in partial fulfillment of the requirements
for the degree
Master of Sciences, Engineering
Specialization in: Electrical Engineering
New Mexico State University
Las Cruces, New Mexico
March 2011
“A Low-Distortion Class-AB Audio Amplifier with High Power Efficiency,” a the-
sis prepared by Chaitanya Mohan in partial fulfillment of the requirements for the
degree, Master of Sciences has been approved and accepted by the following:
Linda LaceyDean of the Graduate School
Dr. Paul M. FurthChair of the Examining Committee
Date
Committee in charge:
Dr. Paul M. Furth
Dr. Jaime Ramirez-Angulo
Dr. Jeffrey Beasley
ii
DEDICATION
Dedicated to my father Chandolu Rama Mohan Rao, mother Chandolu
Hemalatha, sister Srujana Mohan Rao.
iii
ACKNOWLEDGMENTS
First I would like to thank my parents Chandolu Rama Mohan Rao and
Chandolu Hemalatha, sister Srujana Mohan Rao and brother-in-law DeepakNadh
Tammana for supporting me at every level of my life. They are the reason behind
my success at every corner in the journey of life. Srujana has been more of a
friend, guide and advisor than a sister.
Dr. Paul M. Furth, the coach of VLSI V6 team is man behind the success
of this thesis. I can proudly say, the knowledge I acquired from him in Electronics
is more than what I have earned in my entire bachelors. The approach towards
every problem and level of analyzing things before hand is what I would like to
get from him.
I would also like to thank Dr. Jaime Ramirez-Angulo for imparting knowl-
edge on analog concepts.
A special note of thanks to my childhood friend Hareesh Gottipati (Nani),
Vidhul Dev and Arka who are more than just friends. I still remember the fights
we had on every other day on almost every topic. The topics included more of
politics, movies, places and almost every current situation, but the discussion
never involved studies.
Swetha Peri is one other person in my life who is more than a friend. She
had the patience to hear everything and take any situation casually with a calm
iv
mind. I would like to thank Swetha for being such a great friend and who always
supported my every decision.
A thanks is just not enough for Harish Valapala. I cannot forget the help
that I got from him, every time when I was supposed to meet the deadline.
I would also like to thank Alex from math department for giving an op-
portunity to work as a math tutor in the final semester. He has been very humble
during my defense and allowed me to work based on my availability, which was
very helpful
Finally I would like to thank all my friends and roommates: Sravan (Buggi),
Varun (Jaffa), Lalith (Makku bro), Venu, Madhusudhan Nagireddy (Madhu),
Suresh (Debri), Nikhilesh (Hadavidi) and especially the V6 group Punith (the
buss), Rajesh, Ramesh, Venkat and Harish.
v
VITA
December 17, 1986 Born in Hyderabad, India.
Education
2004 - 2008 B.Tech. Electronics and Communication Engineering,Jawaharlal Nehru Technological University, India
2009 - 2010 Teaching Assistant,New Mexico State University,USA
Since 2008 M.S in Electrical Engineering,New Mexico State University, USA
Awards and Achievments
2008 - 2011 In-State Tuition, NMSU,USA.
March - 2011 Third place in Graduate Research and Arts Symposium,NMSU, USA.
Field of Study
Major Field: Electrical Engineering (Analog Microelectronics/VLSI Design)
vi
ABSTRACT
A LOW-DISTORTION CLASS-AB AUDIO AMPLIFIER WITH HIGH
POWER EFFICIENCY
BY
CHAITANYA MOHAN, B.Tech
Master of Sciences, Engineering
Specialization in Electrical Engineering
New Mexico State University
Las Cruces, New Mexico, 2011
Dr. Paul M. Furth, Chair
Place: Thomas & Brown Room-207
Date: 03/17/2011 Time: 2:00 PM
A low-distortion three-stage Class-AB audio amplifier is designed to drive
a 16-Ω headphone speaker. High power efficiency in the design was achieved by
using fully-differential internal stages with local common-mode feedback networks
and replica biasing of the output stage. The threshold voltage of NMOS transistors
were made comparable to PMOS transistors by biasing the p-substrate in order
to achieve high linearity. The stability of the amplifier is achieved using multiple
compensation techniques. The audio amplifier is designed to drive widely varying
capacitive loads from 10 pF to 5 nF. The peak power delivered to the load is
vii
93.8mW. The quiescent power of the amplifier is 1.43mW. The output signal
swing is 2.45Vpp for ±1.5V supply. The THD of the amplifier is measured as -
79dB. The design has been implemented in a 0.5µm CMOS process and occupies
0.35 mm2 of area.
viii
TABLE OF CONTENTS
LIST OF TABLES xii
LIST OF FIGURES xiii
1 INTRODUCTION 1
2 BASE FOR AUDIO AMPLIFIERS 4
2.1 Audio Amplifier Specifications . . . . . . . . . . . . . . . . . . . . 4
2.1.1 Headphone Speaker Load . . . . . . . . . . . . . . . . . . . 5
2.1.2 Total Harmonic Distortion in an Amplifier . . . . . . . . . 5
2.1.3 Power Efficiency of an Amplifier . . . . . . . . . . . . . . . 6
2.2 Output Stage Classification . . . . . . . . . . . . . . . . . . . . . 7
2.2.1 Class-A Amplifiers . . . . . . . . . . . . . . . . . . . . . . 7
2.2.2 Class-D Amplifier . . . . . . . . . . . . . . . . . . . . . . . 9
2.2.3 Class-AB Amplifier . . . . . . . . . . . . . . . . . . . . . . 9
2.3 Multi-Stage Amplifiers . . . . . . . . . . . . . . . . . . . . . . . . 11
2.3.1 Pseudo Class-AB Amplifier . . . . . . . . . . . . . . . . . 11
2.3.2 True Class-AB Amplifier . . . . . . . . . . . . . . . . . . . 13
2.4 Common-Mode Feedback Network . . . . . . . . . . . . . . . . . . 14
2.5 Compensation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
2.5.1 Miller Compensation . . . . . . . . . . . . . . . . . . . . . 16
ix
2.5.2 Reverse-Nested Miller Compensation . . . . . . . . . . . . 17
2.6 Three-Stage Class-AB Amplifier from [1] . . . . . . . . . . . . . . 18
2.6.1 Design from [1] . . . . . . . . . . . . . . . . . . . . . . . . 18
2.6.2 Experimental Results from [1] . . . . . . . . . . . . . . . . 19
2.7 Replica Biasing . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
3 DESIGN OF THE THREE-STAGE CLASS-AB AUDIO AMPLI-FIER 23
3.1 Architecture and Key Aspects of the Audio Amplifier . . . . . . . 23
3.2 Transistor Level Three-Stage Design . . . . . . . . . . . . . . . . 25
3.3 Bias circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
3.4 Input-Stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
3.5 Second-Stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
3.5.1 PMOS differential amplifier . . . . . . . . . . . . . . . . . 31
3.5.2 NMOS differential amplifier . . . . . . . . . . . . . . . . . 32
3.6 Output-Stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
3.7 Compensation used in the Design . . . . . . . . . . . . . . . . . . 35
3.8 Small-Signal Models . . . . . . . . . . . . . . . . . . . . . . . . . 36
3.9 Pole-Zero Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . 40
4 SIMULATION RESULTS 43
4.1 DC analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
4.2 AC analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
4.3 Transient analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
4.4 THD analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
5 HARDWARE TESTING 54
x
5.1 Layout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54
5.2 Experimental Setup . . . . . . . . . . . . . . . . . . . . . . . . . . 54
5.3 DC Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . 56
5.4 Transient Measurements . . . . . . . . . . . . . . . . . . . . . . . 57
5.5 THD Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . 58
6 DISCUSSION AND CONCLUSION 69
APPENDICES 74
A. HARDWARE TEST PROCEDURE 75
B. POLE/ZERO ANALYSIS USING MAPLE 85
C. MATLAB CODE TO PLOT WAVEFORMS 88
REFERENCES 95
xi
LIST OF TABLES
2.1 Comparison of measured results . . . . . . . . . . . . . . . . . . . 20
3.1 Transistor Dimensions . . . . . . . . . . . . . . . . . . . . . . . . 26
3.2 Poles and Zeros . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
4.1 Design Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . 43
4.2 AC Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . 46
4.3 Transient Simulation Results . . . . . . . . . . . . . . . . . . . . . 52
5.1 Hardware Measurements . . . . . . . . . . . . . . . . . . . . . . . 68
6.1 Summary of Hardware Test Results . . . . . . . . . . . . . . . . . 69
6.2 Comparison of results with state-of-the-art ([1]) . . . . . . . . . . 70
6.3 Simulation vs Hardware (LIQ) . . . . . . . . . . . . . . . . . . . . 71
6.4 Simulation vs Hardware (LTHD) . . . . . . . . . . . . . . . . . . 72
6.5 Simulation vs Hardware (MIQ) . . . . . . . . . . . . . . . . . . . 72
6.6 Simulation vs Hardware (HCL) . . . . . . . . . . . . . . . . . . . 73
xii
LIST OF FIGURES
2.1 Schematic of a Class-A amplifier . . . . . . . . . . . . . . . . . . . 7
2.2 Basic design of a Class-D amplifier . . . . . . . . . . . . . . . . . 9
2.3 Schematic of a Class-AB amplifier . . . . . . . . . . . . . . . . . . 10
2.4 Schematic of a three-stage pseudo class-AB amplifier . . . . . . . 12
2.5 Schematic of a three-stage true class-AB amplifier . . . . . . . . . 13
2.6 Schematic of a fully-differential amplifier with common-mode feed-back network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
2.7 Architecture of Miller compensation for two-stage amplifier . . . . 16
2.8 Architecture of Reverse-Nested Miller compensation for three-stageamplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
2.9 Architecture of the three-stage class-AB amplifier of [1] . . . . . . 19
2.10 (a) Schematic of two-stage pseudo class-AB amplifier (b) Replicabias circuit to control quiescent at the output stage . . . . . . . . 21
3.1 Architecture of the proposed three-stage class-AB amplifier . . . . 24
3.2 Schematic of the three-stage class-AB audio amplifier . . . . . . . 27
3.3 Schematic of the bias circuit . . . . . . . . . . . . . . . . . . . . . 28
3.4 Schematic of the first-stage . . . . . . . . . . . . . . . . . . . . . . 30
3.5 Schematic of the second-stage PMOS differential amplifier . . . . 31
3.6 Schematic of the second-stage NMOS differential amplifier . . . . 32
3.7 Schematic of the output-stage . . . . . . . . . . . . . . . . . . . . 34
xiii
3.8 (a) Left-half of the input-stage (b) small-signal model for left half. 36
3.9 (a) Right-half of the input-stage (b) small-signal model for right-half. 37
3.10 (a) PMOS differential amplifier (b) small-signal model PMOS dif-ferential amplifier. . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
3.11 (a) NMOS differential amplifier (b) small-signal model NMOS dif-ferential amplifier. . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
3.12 (a) Schematic of output-stage (b) small-signal model for output-stage 40
3.13 Small-signal model of the designed three-stage class-AB audio am-plifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
4.1 Schematic of the DC test-bench . . . . . . . . . . . . . . . . . . . 44
4.2 DC-analysis output . . . . . . . . . . . . . . . . . . . . . . . . . . 45
4.3 Schematic of the AC test-bench . . . . . . . . . . . . . . . . . . . 46
4.4 AC output of LIQ circuit . . . . . . . . . . . . . . . . . . . . . . . 47
4.5 AC output of LTHD circuit . . . . . . . . . . . . . . . . . . . . . 47
4.6 AC output of MIQ design . . . . . . . . . . . . . . . . . . . . . . 48
4.7 AC output of HCL circuit . . . . . . . . . . . . . . . . . . . . . . 49
4.8 Schematic of the Transient test-bench . . . . . . . . . . . . . . . . 49
4.9 Transient output of LIQ circuit . . . . . . . . . . . . . . . . . . . 50
4.10 Transient output of LTHD circuit . . . . . . . . . . . . . . . . . . 50
4.11 Transient output of MIQ circuit . . . . . . . . . . . . . . . . . . . 51
4.12 Transient output of HCL circuit . . . . . . . . . . . . . . . . . . . 52
4.13 Schematic for THD measurement . . . . . . . . . . . . . . . . . . 53
4.14 Transient output for measuring THD . . . . . . . . . . . . . . . . 53
5.1 Layout of LIQ amplifier . . . . . . . . . . . . . . . . . . . . . . . 55
5.2 Layout of LTHD amplifier . . . . . . . . . . . . . . . . . . . . . . 56
5.3 Layout of MIQ amplifier . . . . . . . . . . . . . . . . . . . . . . . 57
xiv
5.4 Layout of HCL amplifier . . . . . . . . . . . . . . . . . . . . . . . 58
5.5 Layout of the frame with two LIQ and MIQ. . . . . . . . . . . . . 59
5.6 Layout of the frame with two LTHD and HCL. . . . . . . . . . . . 60
5.7 Micrograph of the chip. . . . . . . . . . . . . . . . . . . . . . . . . 61
5.8 Transient response of LIQ design (an offset of 900mV is addedintentionally for visibility) . . . . . . . . . . . . . . . . . . . . . . 62
5.9 Transient response of LTHD design (an offset of 900mV is addedintentionally for visibility) . . . . . . . . . . . . . . . . . . . . . . 62
5.10 Transient response of MIQ design (an offset of 900mV is addedintentionally for visibility) . . . . . . . . . . . . . . . . . . . . . . 63
5.11 Transient response of HCL design (an offset of 900mV is addedintentionally for visibility) . . . . . . . . . . . . . . . . . . . . . . 63
5.12 THD measurement for LIQ design . . . . . . . . . . . . . . . . . . 64
5.13 THD measurement for LTHD design . . . . . . . . . . . . . . . . 65
5.14 THD measurement for MIQ design . . . . . . . . . . . . . . . . . 66
5.15 THD measurement for HCL design . . . . . . . . . . . . . . . . . 67
xv
Chapter 1
INTRODUCTION
The size of portable devices are decreasing with advances in technology; simi-
larly battery size is also decreasing [2],[3],[4]. The portable devices available in
the present day market such as laptops, cellphone, iPods and other music play-
ers require audio amplifiers that are capable of driving small resistive loads and
wide range of capacitive loads (headphone speakers). Audio amplifiers require
high current at the output stage to drive low resistive loads [5]. The main fea-
tures of audio amplifiers are low power dissipation, high output power and low
distortion[1],[2],[5],[6],[7]. The ideal choice for audio amplifiers are class-AB and
class-D amplifiers [6]. Though class-D amplifiers have high efficiency, low power
dissipation and low distortion [5], class-AB amplifiers are preferred for designing
audio amplifiers because they have better power supply rejection ratio (PSRR)
than class-D amplifiers [2],[6]. Moreover, class-D amplifiers are subject to electro-
magnetic interference [1],[5],[7].
A three-stage pseudo class-AB amplifier from [3] experiences a large qui-
escent current when the output stage current increases. An adaptive biasing
technique is used to transform the pseudo class-AB amplifier to a true class-AB
amplifier [3],[4]. However, the gain experienced by the load through PMOS output
transistor is different from the gain experienced by the NMOS output transistor.
This results in asymmetry at the output, which in turn causes severe distortion.
Although, the bias current of the amplifier is low, the distortion is large.
1
In order to obtain symmetry at the output, the load must experience the
same gain through both the NMOS and PMOS output transistors. Biasing the
output transistors at a low quiescent current is achieved using replica bias. The
replica biasing circuit is used to generate the required bias voltages at the gates
of the output transistor [8]. A local common-mode feedback network is used for
symmetrical gain and to generate a desired common-mode output voltage. Thus
it simultaneously improves the power efficiency and reduces distortion.
As the number of stages in an amplifier increases, the stability starts to
degrade [3],[4],[9],[10],[11]. Thus compensation networks are used to improve the
stability of a multi-stage amplifier. Some of the commonly used compensations
are Miller compensation with nulling resistor, nested Miller compensation and
reverse-nested Miller compensation for multi-stage amplifiers. Miller compensa-
tion with nulling resistor proposed in [10] is used to create a RHP zero to split
poles. Reverse-nested Miller compensation is more desirable for multi-stage am-
plifiers than nested Miller compensation as it improves the bandwidth [3],[9],[10].
Based on the class-AB amplifier in [3],[4], this thesis reports on the design of
a new three-stage class-AB amplifier. The class-AB amplifier has fully-differential
internal stages. A common-mode feedback network is used to provide the symmet-
rical gain and to generate a common-mode voltage at the output. Low quiescent
current at the output-stage is obtained using the replica bias circuit. Substrate
biasing technique is used to attain linearity at the output.
Chapter 2 describes specifications for designing an audio amplifier, the
types of output stages that can be used in the design of an audio amplifier, the
purpose of using using multi-stage amplifiers, the stability issues of multi-stage
amplifiers, and the compensation networks that are used to improve the stability,
bandwidth and transient response of the multi-stage amplifiers. A summary of
2
architecture and experimental results of three-stage class-AB amplifier from [1]
is described. The results of [1] are used as basis for designing a new three-stage
class-AB amplifier with improved figure of merit (FOM).
Chapter 3 explains the architecture of each stage of a three-stage class-AB
amplifier designed in this thesis. The working of replica bias circuit to generate low
quiescent currents at the output stage is discussed. The compensation networks
used for stabilizing the amplifier and the small signal models that explain how the
variation in compensation capacitor values improve the stability is also explained.
Chapter 4 discusses the results that determine the functionality of the
amplifier. The test-benches for DC, AC and transient analysis are explained. A
comparison of results for four designs with variation in compensation capacitor
values is summarized. A test-bench for measuring the total harmonic distortion
(THD) is explained.
Chapter 5 explains the hardware implementation of the three-stage class-
AB audio amplifier. The layout of the design is discussed and the test-setup of
the design for determining the quiescent current is explained in the DC testing.
The hardware testing results obtained are compared with simulation results.
Chapter 6 discusses about the figure of merit (FOM) of the designed am-
plifier. A summary of results obtained in [1] are compared with the hardware
testing results obtained in chapter 5.
APPENDICES contains the test procedure for testing the circuit in real-
time, the Maple work that determines the poles and zeros in an amplifier based
on the small-signal model and the code used for plotting the waveforms.
3
Chapter 2
BASE FOR AUDIO AMPLIFIERS
This chapter give an introduction to specifications of audio amplifiers, types of
amplifier output-stage, multi-stage amplifiers and compensation networks.
2.1 Audio Amplifier Specifications
Amplifiers are used in every electronic device. Though general purpose
op-amps can be used to drive a variety of loads but driving small resistive loads
is a tough task. The modern portable devices such as laptops, cellphones, Ipods
and other music players require audio amplifiers for driving headphone speakers.
The resistance of headphones is small. It can vary from 32-Ω to a much smaller
value depending on the supply voltage.
The device dimensions and supply voltages are decreasing with advances
in technology. In order to provide nearly constant output power, corresponding to
the human perception of loudness, reduced voltages necessitate reduced headphone
speakers. The output power POUT is given as
POUT =V 2RMSRL
(2.1)
Thus for small device dimensions and supply voltages, the load resistance must
be made small to maintain a constant output power. The important aspects in
an audio amplifier are load, total harmonic distortion and power efficiency.
4
2.1.1 Headphone Speaker Load
The output resistive load for an audio amplifier in portable devices is very
small. As the supply voltage in these devices is very small, the resistance of the
headphone must be made small to provide constant output power as shown in
(2.1). If the resistance of the headphone speaker is made large, then for small
supply voltages the output power of the amplifier decreases. Thus the loudness
is reduced. Modern day headphone speakers have resistance as small as 8-Ω.
The quiescent current is small for small supply voltages. The distortion in an
amplifier is inversely proportional to the quiescent current. As the quiescent
current decreases, the distortion in an amplifier increases.
2.1.2 Total Harmonic Distortion in an Amplifier
Total harmonic distortion is very important for amplifiers that drive large
capacitive loads and low resistances such as audio amplifiers. Harmonic distortion
occurs in amplifiers when the AC component of the drain current id is comparable
to DC component of the drain current ID [12]. Every amplifier produces harmonics
for a given fundamental frequency. The level, or amplitude, of these harmonics
is smaller than the amplitude, or level, of its fundamental frequency. The total
harmonic distortion is an important amplifier specification. It is given by
THD =
√V 22RMS + V
23RMS
+ V 24RMS ........+ V2nRMS
V1RMS(2.2)
where V1RMS is the rms voltage of the fundamental frequency and VnRMS is the
rms voltage of the nth harmonic.
The total harmonic distortion is measured in percentage (%). The lower
the value better the sound quality. Audio amplifiers with a THD of less than
0.5% produce an audio signal with noise which is hardly perceived by the human
5
ear. Thus audio amplifiers must be designed with a THD less than 0.5% for high
sound quality. In order to reduce distortion, feedback networks are used.
Apart from distortion, another important feature of an audio amplifier
is power efficiency. The power efficiency of an audio amplifier is made high by
lowering the quiescent current in the amplifier. This is discussed in the next
section.
2.1.3 Power Efficiency of an Amplifier
The battery life of a portable system is very important [1]. There will
always be demand in the market for systems that run for longer time. Thus in
order to have an efficient battery life, the quiescent power of the system should
be minimized. The power efficiency of the amplifier is defined as the ratio of
power delivered to the load to power supplied by the battery [1],[12]. The power
delivered to the load is given in (2.1). The power efficiency (η) of the amplifier is
given by
η% =POUTPS
× 100 (2.3)
where PS is the power supplied by the battery given by
PS = IQ(VDD − VSS) (2.4)
Thus, from (2.3) and (2.4), if PS is reduced, the efficiency of the amplifier will be
improved. In order to reduce PS, the quiescent current (IQ) can be reduced.
As supply voltages are smaller for new technologies, multi-stage amplifiers
are used to drive the load. The quiescent current increases with increase in number
of stages. The quiescent current also depends on the type of amplifier. The
linearity in the operational region and low power dissipation are important aspects
6
for audio amplifiers. Thus two types of amplifiers that provide either of the two
properties or both are discussed in the following sections- Class-A and Class-AB
amplifiers
2.2 Output Stage Classification
The output stage of amplifiers are classified based on the circuit config-
uration and the type of operation [8]. Three output stage classifications of an
amplifier are discussed in the following sections.
2.2.1 Class-A Amplifiers
The high linearity in the operational range of class-A amplifiers make them
ideal for audio applications. Owing to their high power dissipation these ampli-
fiers are replaced with class-AB amplifiers for audio applications. Thus class-A
amplifiers are limited to applications that require only small changes in the output
voltage, as the power consumption can be very low for small output signals. The
schematic of the class-A amplifier is shown in Fig. 2.1.
VB
M2
M1VIN
VOUT
RL
IB
Figure 2.1: Schematic of a Class-A amplifier
7
The gate of the transistor M2 is connected to a DC bias voltage (VB).
This turns the transistor M2 ON and a constant current (IB) is sourced from VDD
to bias transistor M1. Thus it acts as a current source. The input for a class-A
amplifier is at the gate of transistor M1. The gain of the amplifier is Gm1 ·Ro, where
Gm1 is the transconductance of the transistor M1 and Ro is the resistance at the
output node given by Ro = ro1‖ ro2‖RL. The current from transistor M1 increases
when the gate experiences a large signal at the input. The sinking current in
the amplifier is not limited but the sourcing current is limited to IB as shown in
Fig. 2.1. The upper-half cycle at the output observes distortion [8]. Moreover the
efficiency of the class-A amplifier is low. The efficiency of an amplifier is given in
(2.3). For class-A amplifiers the load power is given as
PL =V 2OUT2RL
(2.5)
and the power from battery PS is given as
PS = 2ID(VDD − VSS) (2.6)
The efficiency is maximum when VOUT = ID·RL = (VDD - VSS). The efficiency of
class-A amplifier accounts to 25% [13] using equations (2.3), (2.5), (2.6). Hence
makes it hard to be used in present day market of audio amplifiers.
To overcome the problem of low efficiency, class-AB amplifiers are preferred
over class-A amplifiers. The class-AB amplifiers have low power dissipation and
low distortion. The other classification of output stage amplifier known as class-D
amplifier, also provides low power dissipation and low distortion. This is explained
in the following section.
8
2.2.2 Class-D Amplifier
Class-D amplifiers find applications in audio amplifiers and pulse genera-
tors [8]. The basic design of a class-D amplifier is shown in Fig. 2.2. The output
of a Class-D amplifier is a sequence of pulses. The frequency of the output signal
is higher than input signal frequency. Passive filters are used at the output-stage
to eliminate undesired harmonics.
+
–
+Precision triangular
Wave generator
RL
VoutA1
Figure 2.2: Basic design of a Class-D amplifier
The amplitude of the output pulses is fixed. The conduction of switching
devices occurs during the transition of states. Hence, the power dissipation is
reduced as the transition is only for a short duration. Though class-D amplifiers
have an efficiency up to 95%, the power supply rejection ratio is higher when
compared to class-AB amplifiers, as one of the supply voltage is the output voltage
[8]. Hence, variations in supply voltage create variations at output.
Thus class-AB amplifiers are preferred for audio applications, as they have
higher power supply rejection ratio over class-D amplifiers.
2.2.3 Class-AB Amplifier
The basic schematic of the class-AB amplifier is shown in Fig. 2.3. The
transistor MN is ON for the negative half cycle of the output signal and the
transistor MP is ON for the positive half cycle. Thus, only one transistor sources
9
current for each half cycle while other is turned OFF. This kind of operation
provides a push-pull action. This minimizes the quiescent current of the amplifier.
The Vbat is used to turn ON both MN and MP for the short time when the output
signal is near zero. Thus it acts as a class-A amplifier for very small output signals.
The result is minimized crossover (crossing through zero) distortion.
MN
MP
VIN VOUT
RL
Vbat
Vbat
Figure 2.3: Schematic of a Class-AB amplifier
The efficiency of the class-AB amplifier is determined using (2.3). The load
power is given in (2.5). The power from the battery is the total average power
drawn through the power supplies given by
PS =2 · VOUT · (VDD − VSS)
(π + θ) ·RL+ 2IQ(VDD − VSS) (2.7)
where IQ(VDD - VSS) is the quiescent power (VOUT = 0) per transistor and (π
+ θ) is the conduction angle in radians, (π + θ) ≤ 2π. Thus the maximum
efficiency accounts to 50% ≤ η ≤ 78.5% [13]. The push-pull action, low distortion
and high power efficiency of class-AB amplifiers make them ideal choice for audio
applications.
10
A single stage class-AB amplifier does not provide the enough gain to
drive the load. Thus multi-stage amplifiers are used for driving small loads. As
the number of stages increase, the stability starts to degrade. To stabilize these
multi-stage amplifiers compensation networks are used. The following sections
focus on multi-stage amplifiers and compensation networks.
2.3 Multi-Stage Amplifiers
In the past, a single-stage amplifier was sufficient for providing large gain.
As the device dimensions were large, voltage levels were large to drive resistive
loads. A single-stage amplifier has only one pole, thus single-stage amplifiers are
highly stable. Advances in technology has made the device dimensions smaller.
Low voltages are required to operate these transistors. Hence a single-stage gain
is not sufficient to drive resistive loads.
In order to overcome this problem, two or more stages are cascaded together
to provide a gain that is the product of each gain stage [14],[15],[16].As the number
of stages increase, the stability starts to degrade. A compensation network is
required to provide stability. The compensation network becomes more complex
when the number of stages increases beyond four. A large number of multi-stage
amplifiers are proposed in the literature [9],[10],[11],[13]. A multi-stage pseudo
class-AB amplifier proposed in [17] is used as the basis for audio amplifier designed
in this work.
2.3.1 Pseudo Class-AB Amplifier
The schematic of a multi-stage amplifier is shown in Fig. 2.4. The first-
stage is a folded cascode differential amplifier formed with transistors N1−5 and
P1−4. The gain of the first-stage at node V1 is given by GM3 ·R1, where R1 is the
output resistance at V1. The second-stage is realized using a NMOS common-
source amplifier N6.
11
Vin+
Vb1
Vb3
Vb2
Vin-
Vb2
Vb3
N1 N4 N5 N6
N2 N3
P3
P1
P4
P2
P6
P5
N7
P8
N8
P7
RL CL
Rm3 Rm1Cm3 Cm1Cm2
Ccb
V1 V2
V3
Vout
Figure 2.4: Schematic of a three-stage pseudo class-AB amplifier
The output stage for sinking the current is designed using a NMOS common-
source amplifier N8. The third-stage or the output stage for sourcing the current
is realized using N7 common-source amplifier and P7 and P8 current mirror. Thus
the push-pull action is provided by P8 and N8 transistors.
The gain in the first-stage is a inverting gain. The common-source ampli-
fiers N6 and N8 of second-stage have inverting gain configuration. The gain of the
third stage realized with transistors N7 and P7−8 is a non-inverting gain. As the
two-stages of inverting gain from P8 and N7 common-source amplifiers are cas-
caded to obtain a non-inverting gain. Thus the overall gain from Vin+ to Vout is
non-inverting. This multi-stage pseudo class-AB amplifier is a low voltage design
for driving small loads.
Though it can drive small resistive loads, it cannot be used as audio ampli-
fier. As stated earlier, the low power dissipation and low distortion are the main
features of audio amplifiers. The quiescent current in this amplifier increases with
increase in current at the output stage. Thus the power dissipation of the ampli-
12
fier is large. In order to reduce the high quiescent current in this amplifier a new
technique called adaptive biasing has been proposed in [3],[4].
2.3.2 True Class-AB Amplifier
To minimize the current through the transistor P7 in Fig. 2.4, two resistors
Rad1 and Rad2 are used, as shown in Fig. 2.5. The diode connected transistor P7
and the resistors Rad1 and Rad2 form the adaptive biasing network [3],[4]. The
value of the resistors Rad1 and Rad2 are made large to minimize the current through
transistor P7 .
Vin+
Vb1
Vb3
Vb2
Vin-
Vb2
Vb3
N1 N4 N5 N6
N2 N3
P3
P1
P4
P2
P6
P5
N7
P8
N8
P7
RL CL
Rm3 Rm1Cm3 Cm1
Cm2
Ccb
V1 V2
V3
Vout
Rad1
Rad2
Adaptive biasing
Figure 2.5: Schematic of a three-stage true class-AB amplifier
The transistor P7 is in cutoff region when the output current is very low.
The load at node V3 is simply Rad2. The phase margin of the amplifier is improved
with the presence of resistor Rad2 and it moves the non-dominant pole to high
frequencies [3],[4].
When the load experiences a large sourcing current through the transistor
P8, the transistor P7 moves to the saturation region. Now, the load experienced
by node V3 is (1
GM7+ Rad1) ‖ Rad2. For large values of resistors, the gain at the
node V3 in Fig. 2.5 is large compared to the gain at node V3 in Fig. 2.4 [3],[4].
13
Thus the power efficiency and gain of the amplifier are improved. Though
the amplifier has good power efficiency but it cannot be used for audio amplifica-
tion, as the distortion in the amplifier is high. The gain experienced by the load
through P8 common-source is a three-stage gain, whereas the gain experienced by
the load through N8 common-source amplifier is a two-stage gain. This creates
non-linearity at the output and leads to distortion. To attain low distortion, load
must experience same gain through the push-pull transistors.
One way to achieve this is by using fully differential internal stages. The
fully differential amplifiers require additional circuitry to attain balanced outputs.
This additional circuitry is a common-mode feedback network. The working of
the common-mode feedback network is explained in the following section.
2.4 Common-Mode Feedback Network
A common-mode feedback network is used for generating a known voltage
at the output of a fully-differential amplifier [12] as shown in Fig. 2.6. The differ-
ence between the inverting terminal and the average of non-inverting terminals of
the common-mode feedback amplifier are amplified with a gain at the output.
If the difference between Vb2 and the average of Vout+ and Vout- is large,
then the voltage VCMFB increases. This increases the VGS of transistors N1 and
N4 and the current through them. Thus the voltage at the output nodes reach
close to Vb2. Hence a known voltage is obtained at the output. The circuit also
works for differential inputs. If one of the input voltage moves above the other
then one of the output goes above Vb2 by a small amount and the other output
moves down the Vb2 by the same amount.
There also exist some concerns while using these feedback networks. The
stability of the amplifier changes and it requires compensation networks to sta-
14
Vb2P1
Vin+Vin-N2 N3
N1 N4
CMFB++ -
P2
Vout+ Vout-
VCMFB
Figure 2.6: Schematic of a fully-differential amplifier with common-mode feedbacknetwork
bilize the differential amplifier, as well as common-mode feedback network. The
following section gives an introduction to compensation.
2.5 Compensation
Stability is an important aspect for amplifiers. The amplifiers with single
stage are highly stable as they have single dominant pole. In multi-stage amplifiers
the number of dominant poles are not limited to one. Thus the stability goes
15
down. Compensation networks are used, to improve the stability of multi-stage
amplifiers. Two types of compensation networks are reported in the following
sections.
2.5.1 Miller Compensation
A Miller compensation uses a resistor and a capacitor in series between
the input and output of an inverting stage [9],[11]. The architecture of the Miller
compensation is shown in Fig. 2.7.
-Av1 -Av2
CM
VIN VOUT
RM
R1
-gm1 -gm2
C1RL
Figure 2.7: Architecture of Miller compensation for two-stage amplifier
If the two poles of the two-stage amplifier are near each other, then a
Miller compensation can be used to split the poles. The dominant pole is moved
towards the low frequencies and the non-dominant pole is moved towards the
high frequencies. The series resistor RM and capacitor CM create a zero. The
dominant pole is given by P−3dB =1
CM (RM+GM2R1RL), the non-dominant pole is
P2 =RM+GM2R1RLCL(R1+RM )RL
, and the zero is 1(RM− 1GM2
)CM. Thus the RHP zero can be
eliminated by selecting RM =1
GM2[9],[11], [13].
16
2.5.2 Reverse-Nested Miller Compensation
A two-stage amplifier is designed by cascading two inverting stages. Sim-
ilarly, a three-stage amplifier is designed by cascading two inverting stages and
a non-inverting stage. In a three-stage amplifier design the first stage is invert-
ing stage that is implemented using a differential amplifier and the next stages
are implemented with common-source amplifiers. If the second stage is made non-
inverting, then nested Miller compensation is used. Likewise, if the second-stage is
implemented with inverting gain configuration and third-stage with non-inverting
gain configuration, then reverse-nested Miller compensation can be used to stabi-
lize the amplifier. The architecture of the reverse-nested Miller compensation is
shown in Fig. 2.8.
-Av2 Av3
CM2
VIN VOUT
RM
-Av1
CM1
R1 R2C1 C2
RL
-gm1 -gm2 gm3
Figure 2.8: Architecture of Reverse-Nested Miller compensation for three-stageamplifier
The dominant pole for this compensation is given by
P−3dB =1
CM1GM2GM3R1R2RL(2.8)
The transfer function adapted from [9] is given by
17
Arnmc =GM1GM2GM3R1R2RL(1− s(
CM2GM2
+CM1
GM2GM3R2)− s2 CM1CM2
GM2GM3)
(1 + s 1CM1GM2GM3R1R2RL
)[1 + s(CM2CLGM3CM1
− CM2GM2
+CM2GM3
) + s2CM1CLGM2GM3
]
(2.9)
The equation 2.9 shows that the amplifier has three poles and two RHP
zeros. The dominant pole is given in equation 2.8. The other two poles are high
frequency poles.
Thus in any amplifier after the required compensation scheme is applied,
the values of compensation capacitors and resistors are determined to move the
non-dominant poles and zeros to high frequencies. This stabilizes the circuit
and improves the bandwidth. A large number of compensation techniques are
proposed to stabilize the multi-stage amplifiers [9],[10],[11],[15],[16],[17]. These
multi-stage amplifiers are used in a wide range of applications.
A three-stage class-AB amplifier is proposed by [1] to drive 16-Ω headphone
speakers. The design and results are summarized in the following section. This
design is taken as a reference to compare the results with our work.
2.6 Three-Stage Class-AB Amplifier from [1]
The architecture of the proposed design is shown in Fig. 2.9. It is a three-
stage class-AB amplifier that drives 16-Ω headphone speakers and a wide range
of capacitive loads.
2.6.1 Design from [1]
The first stage is implemented using a folded-cascode amplifier with an
inverting gain configuration. The second-stage is implemented using common-
source amplifiers with positive gain configuration. A damping factor control stage
is used in the amplifiers that drive large capacitive loads [1],[11]. The damp-
ing factor control stage is used in amplifiers that have large swing to improve
18
Gm2 -Gm3
CC
VIN VOUT
RB
CC2
RL
-Gm1
GmD
RC
CD2CD
VB
CL
Figure 2.9: Architecture of the three-stage class-AB amplifier of [1]
the bandwidth and transient response [11]. The output stage is designed with
common-source amplifiers to provide the required push-pull action. The output
stage of the amplifier is biased at ±1V, and the rest of the amplifier is biased at
±0.6V.
2.6.2 Experimental Results from [1]
The total quiescent current of the amplifier is 730 µA. The THD of the
design is -84.8dB for 1.4VPP , 1 kHz sine-wave output. A figure of merit (FOM)
was defined to compare with other designs. The figure of merit defined by [1] is
the ratio of peak output power to the supply power. A comparison of measured
results is shown in Table 2.1.
Taking the above design as a reference, a new three-stage amplifier is de-
signed. The concept of replica biasing is used in the design to generate the bias
19
Table 2.1: Comparison of measured results
Parameter [18] [19] [20] [1]
Technology -0.35 µmCMOS
65 nmCMOS
130nmCMOS
Capacitance load 0-300 pF 0-300pF 0-12 nF 1 pF - 22 nF
Supply 3.0V 0.8V 2.5V 1.2V/2.0V
THD+N @ max.output
-90dB -69dB -68dB -84dB
Total compensationcapacitance
- - 35pF 14pF
Quiescent power 12.0mW 2.5mW 12.5mW 1.2mW
FOM 8.1 1.3 4.3 33.3
voltages for the output stage. The following section gives a brief introduction to
replica biasing.
2.7 Replica Biasing
A replica bias circuit is used for generating bias voltages for the output
stage [21]. A replica bias circuit can be used in a class-AB amplifier to bias the
common-source amplifiers of the output stage. The bias voltages generated by
the replica bias circuit acts as Vbat as shown in Fig. 2.3. When no input signal
is present, transistors of the output stage are in ON state but in a non-linear
region. The quiescent current in the output stage is set by the current through the
replica bias circuit [8]. The output stage acts as a class-A amplifier, as the output
transistors are either sourcing or sinking current all the time. This eliminates the
dead band region and minimizes the distortion in class-AB amplifiers.
20
The schematic of a two-stage pseudo class-AB amplifier and the replica
bias circuit to control quiescent current adapted from [8] are shown in Fig. 2.10.
VCTRL
M4 M5
2IB
VI+VI– M3M2
M1
M6
M7
M8
M9
M11
M10
VB CC
RCCC
RLargeVout
yx
(a)
VCTRL
M5C
IB
VrefM3C
M1C
M6C
MB
CC
RC
MP
MN
IBVBVB
(b)
Figure 2.10: (a) Schematic of two-stage pseudo class-AB amplifier (b) Replicabias circuit to control quiescent at the output stage
The two-stage pseudo class-AB amplifier has a fully-differential first-stage.
The transistors M6 and M7 are common-source amplifiers that provide the push-
pull action. The replica bias circuit is used for controlling the quiescent current
through output-stage transistors M6 and M7. The transistors M5C , M3C , M1C
and M6C are replicas of M5, M3, M1 and M6 respectively. The transistor MB
21
is biased at a voltage VB such that the current through MB is IB. The current
through transistors MP , M5C and M5 are similar, as they have same VSG. Thus,
the voltage at the drain of M5C is similar to the voltage at node ’y’. This causes
the current through M6 to be same as the current through M6C (IB). Hence, a
quiescent current of known value is obtained at the output of a two-stage pseudo
class-AB amplifier.
A feedback network is used along with the replica bias circuit to generate
bias voltages for the output-stage. Three-stage class-AB amplifier designed in our
work use the concept of replica bias to generate bias voltages and control quiescent
current at the output stage. The design of the amplifier is discussed in Chapter. 3.
22
Chapter 3
DESIGN OF THE THREE-STAGE CLASS-AB AUDIO AMPLIFIER
A three-stage class-AB audio amplifier is designed to drive 16-Ω headphone speak-
ers. The audio amplifier has high power efficiency and low distortion, and it is
also capable of driving a wide range of capacitive loads.
This chapter deals with the design of the three-stage class-AB amplifier.
It is organized as follows: The key aspects and architecture of the audio amplifier
is discussed in the first section. It is followed by the design of each stage with
small signal models. The last section of the chapter deals with the stability of the
amplifier that is analyzed with poles and zeros.
3.1 Architecture and Key Aspects of the Audio Amplifier
The architecture of the designed three-stage class-AB audio amplifier is
shown in Fig. 3.1. The design is implemented using fully-differential internal
stages. The first stage is a fully-differential folded cascode amplifier with an
inverting gain configuration. The second-stage is implemented with two two dif-
ferential amplifiers. A non-inverting gain configuration is used in this stage. The
third stage is implemented with PMOS and NMOS common-source amplifiers for
the push-pull action.
The gain experienced by the load at the output through NMOS and PMOS
common-source amplifiers is same. The symmetry in gain is achieved using two
differential amplifiers in the second-stage. In the absence of input signal, a dead
band region is created at the output, as the NMOS and PMOS common-source
23
+
-
-
+
+
-
+
-
Vout
Rc1Cc1
Cc2
Cc2
Rc2
2·Cc3
Cc3
Vin+
Vin–
A1
A2
A2
A3
A3
-
+
+
-
2·Cc3
-
-
Vo1–
Vo1+
Vo2P+
Vo2N+
Cc3
Figure 3.1: Architecture of the proposed three-stage class-AB amplifier
amplifiers are turned OFF. This leads to crossover distortion. To minimize the
distortion, a common-mode feedback network is used in combination with replica
bias in the second-stage to generate bias voltages for the third-stage . This turns
ON the transistors of the third-stage and the dead band region is eliminated.
The linearity in the design is achieved using a technique called substrate
biasing. The threshold voltage of all NMOS transistors are made comparable to
threshold voltage of the PMOS transistor. This is attained by connecting the bulk
of the NMOS transistor to a voltage lower than source voltage. This is explained
in detail in the following sections.
24
3.2 Transistor Level Three-Stage Design
The transistor level schematic of the three-stage class-AB audio amplifier
is shown in Fig. 3.2. The first stage is a fully-differential folded cascode amplifier
realized with transistors M1-M12. The transistors M13-M20 in combination with
resistors R1 and capacitors CS form the common-mode feedback network. The
second-stage is realized using two differential amplifiers. A NMOS differential
amplifier is formed with transistors M40-M47 and the PMOS differential amplifier
is formed with transistors M21-M28. The transistors M37-M39 and M56-M58 are
replica bias circuits. The output-stage is realized with transistors MP and MN .
A Miller compensation is used from the output of third-stage to negative output
terminal of the first-stage. A reverse-nested Miller compensation is used between
input and output of the second-stage. The transistor dimensions are given in
Table 3.2.
In this work, four designs are implemented. The difference in each design
is the dimensions of the transistors M39 and M58, the value of compensation
capacitors and resistors, and the input bias current. A trade-off has been observed
between the total harmonic distortion and quiescent current. This is discussed in
Chapter 4. The design of each stage is explained in the following sections.
3.3 Bias circuit
The schematic of the bias circuit is shown in Fig. 3.3. The bias circuit
internally generates four bias voltages Vb1, Vb2, Vb3 and Vb4. The voltage Vb2
is one VSG below VDD. This is generated by diode connecting the transistor M1A
as shown in Fig. 3.3. Similarly the voltage Vb1 is generated.
A long L (length) diode connected transistor is created by connecting the
gates of transistors M4P1-M4P5 as shown in Fig. 3.3. The voltage Vb3 at the gate
of transistor M3A is 2VDSSAT + VTHP . The current through this transistor is given
25
Table 3.1: Transistor Dimensions
Device Dimensions
M1A, M1B, M1C , M19, M20, M35, M36, M50, M5120µm1.2µm
M11, M12, M27, M28, M48, M5620µm1.2µm , m = 2
M120µm1.2µm , m = 4
M4720µm1.2µm , m = 8
M2C , M17, M18, M33, M3420µm0.9µm
M9, M10, M31, M32, M49, M5720µm0.9µm , m = 2
M220µm0.9µm , m = 4
M4620µm0.9µm , m = 8
M4A, M4B, M4C , M54, M55, M15, M16, M31, M3260µm1.2µm
M4, M5, M13, M29, M37,60µm1.2µm , m = 2
M5, M6, M21, M40, M41, M44, M4560µm1.2µm , m = 4
M23, M2660µm1.2µm , m = 6
M3A, M3B, M3C , M52, M5360µm0.9µm
M7, M8, M14, M30, M3860µm0.9µm , m = 2
M22, M42, M4360µm0.9µm , m = 4
M4P1, M4P2, M4P3 , M4P4, M4P530µm1.2µm
M1N1, M1N2, M1N3 , M1N4, M1N510µm1.2µm
M39150µm0.6µm , m = 4,6
MN150µm0.6µm , m = 40
M58300µm0.6µm , m = 4,6
MP300µm0.6µm , m = 40
26
Vo2
N+
Vo2
P+
Vo1+ R
c2
Cc2C
c2
Vb3
Vbia
s
Vb4
Vb2
Vb3
Vb2
Vb
2
Vb
3
Vb
4
Vb1
M1
CM
1A
M1B
M2
C
M3
C
M4
C
M3B
M4B
M3A
M4A
M4P
1
M4P
2
M4P
3
M4P
4
M4P
5
M1
N5
M1
N1
M1
N2
M1
N3
M1
N4
Vin
+
Vb
4
Vb1
Vb4
Vb3
Vb2
R1
R1
Vin
-
Vb4
Vb2
Vb3
Vr
Vo1+
Vo1-
M1
M2
M11
M9
M12
M10
M19
M17
M20
M18
M3
M4
M7
M5
M8
M6
M14
M13
M15
M16
R2
R2
Vo
1!
Vo1+
Cc3
Cc3
Vb
4
Vb3
Vb4
Vb3
Vb2
Vb
2V
b2
Vb
3
Vo2N!
Vo2N+
Vrn
M35
M33
M36
M34
M28
M26
M27
M25
M24
M23
M22
M21
M30
M29
M38
M37
M32
M31
M39
Vb3
R2/2
R2/2
Vo
1!
Vo1+
2·C
c32·C
c3
Vb
4
Vb1
Vb3
Vb
4
Vb1
Vb
4
Vb1
Vo2P+
Vo2P!
Vrp
M50
M51
M49
M48
M57
M56
M52
M54
M53
M55
M42
M40
M43
M41
M46
M47
M58
M44
M45
Vo2
N+
Vo2
P+
Vo
1!
Cc1
Rc1
Vo
ut
MP
MN
CS
CS
CS
CS
CS
CS
Bia
s ci
rcu
itF
irst
-sta
ge
Sec
on
d-s
tag
eT
hir
d-s
tag
e
Rep
lica
-bia
s
Rep
lica
-bia
s
Vc
Vrc
Vrs
Fig
ure
3.2:
Sch
emat
icof
the
thre
e-st
age
clas
s-A
Bau
dio
amplifier
27
Vb3
Vbias
Vb4
Vb2
Vb3
Vb2 Vb2
Vb3
Vb4
Vb1M1CM1A M1B
M2C
M3C
M4C
M3B
M4B
M3A
M4A
M4P1
M4P2
M4P3
M4P4
M4P5
M1N5
M1N1
M1N2
M1N3
M1N4
Vb3
IB2IB1
Figure 3.3: Schematic of the bias circuit
by
ID =µN · COX
2· W3AL3A
· (VGS − VTHP )2 (3.1)
As the current
IB1 = IB2 (3.2)
The current through the Long ’L’ transistor formed by the transistors M4P1-M4P5
is given by
ID =µN · COX
2· W4PL4P
((2VDSSAT + VTHP )− VTHP )2 (3.3)
This equation can be rewritten in terms of VGS as
ID =µN · COX
2· W4PL4P
· 4(VGS − VTHP )2 (3.4)
28
Thus from (3.1), (3.2) and (3.4) we obtain
W4PL4P
=1
4
W3AL3A
(3.5)
The transistor M4A is biased at the edge of the saturation region, thus pulling
more current from transistor M4A, moves it from saturation to triode region [12].
Hence, the length of the long L transistor is assumed five times the length of M3A
rather than four times. This generates a voltage Vb3 that is VSDSAT away from
Vb2.
Similarly, the voltage Vb4 is generated that is VDSSAT away from Vb1. The
bias voltages are replicated to others stages of the amplifier through transistors
M1C-M4C . The current through the branch M1C-M4C is mirrored to the next
stages based on the aspect ratio of the current mirror.
3.4 Input-Stage
The first stage of the amplifier is realized using a fully-differenatial folded
cascode amplifier, as they have wide swing and high gain. The schematic of the
first stage is shown in Fig.3.4. Transistors M1-M12 represent the folded cascode
amplifier. The transistors M13-M20 in combination with resistors R1 and capac-
itors CS form the common-mode feedback network. A common-mode feedback
network is used for generating a known voltage Vr at the output of the folded
cascode amplifier.
The operation of the first-stage is as follows. The voltage Vr is set to 0V,
this allows the current to pass through the transistor M16. The voltage at the
node VX increases. The transistors M20, M11 and M12 form the current mirror.
The current through the transistors M11 and M12 depends on the aspect ratio
29
Vin+
Vb4
Vb1
Vb4
Vb3
Vb2
R1R1
Vin-
Vb4
Vb2
Vb3
Vr
Vo1+
Vo
1-
M1
M2
M11
M9
M12
M10
M19
M17
M20
M18
M3 M4
M7
M5
M8
M6
M14
M13
M15 M16CS CS Vc
VX
Common-mode
Feedback network
Figure 3.4: Schematic of the first-stage
of the mirror. Hence the voltages Vo1+ and Vo1- are pulled towards VSS. The
voltage Vc acts as virtual ground, as it is the center voltage of Vo1+ and Vo1-.
The gain of the first-stage is determined by the gm4,3 resistors R1, as R1�
ro of the transistors. The tail of the folded cascode amplifier and common-mode
feedback network are cascoded to provide better matching between the transistors
and to obtain good precision in matching the currents.
Under equilibrium condition, the voltage Vo1+ and Vo1- are approximately
at 0V. This voltage is used for biasing the second stage. The implementation of
second-stage is explained in the following section.
3.5 Second-Stage
The second stage of the amplifier is implemented using a NMOS differential
amplifier and a PMOS differential amplifier. The NMOS differential amplifier is
used for biasing PMOS common-source amplifier of the third-stage and the PMOS
30
differential amplifier is used for biasing NMOS common-source amplifier of the
third-stage.
3.5.1 PMOS differential amplifier
The schematic of a PMOS differential amplifier is shown in Fig. 3.5
R2 R2
Vo1– Vo1+
Cc3 Cc3
Vb4
Vb3
Vb4
Vb3
Vb2Vb2Vb2
Vb3
Vo
2N–
Vo2
N+
VrnM35
M33
M36
M34
M28
M26
M27
M25
M24M23
M22
M21
M30
M29
M38
M37
M32M31
M39
CS CS Vrc
CMFB
Replica bias
Figure 3.5: Schematic of the second-stage PMOS differential amplifier
Transistors M21-M28 form the PMOS differential amplifier. The common-
mode feedback network (CMFB) is realized with transistors M29-M36 and resistors
R2 and capacitors CS. Transistors M37-M39 form the replica bias circuit. The cur-
rent through this circuit is determined by the current source formed by transistors
M37 and M38. The voltage Vrn is at VGS above VSS. The common-moded feedback
network generates the known voltage Vrn at the output of the PMOS differen-
tial amplifier. The gain of the amplifier is determined by resistors R2 and the
transconductance gm23,24 .
31
Under equilibrium, the common-mode voltage Vrc is equal to V o2N+. This
voltage is used as the input for the NMOS common-source amplifier of the third-
stage. Since, V o2N+ is equal to Vrn, transistors M39 and MN form a virtual
current mirror. Thus, the quiescent current of the output stage is determined by
the current through M39. The ratio of currents depends on the ratio of multiplicity
of transistor dimensions.
3.5.2 NMOS differential amplifier
The NMOS differential amplifier is similarly designed. The schematic of
the NMOS differential amplifier is shown in Fig. 3.6.
Vb3
R2/2 R2/2
Vo1– Vo1+
2·Cc3
Vb4
Vb1
Vb3
Vb4
Vb1
Vb4
Vb1
Vo2
P+
Vo2
P–
Vrp
M50 M51
M49
M48
M57
M56
M52
M54
M53
M55
M42
M40
M43
M41
M46
M47
M58
M44 M45
CS CS
2·Cc3
Vrs
CMFB
Replica bias
Figure 3.6: Schematic of the second-stage NMOS differential amplifier
Transistors M40-M47 form the NMOS differential amplifier. The common-
mode feedback network (CMFB) is realized with transistors M48-M55 and resis-
tors R22
and capacitors CS. Transistors M56-M58 form the replica bias circuit. The
32
current through this circuit is determined by the current-source formed by tran-
sistors M56 and M57. The voltage Vrn is at VSG below VDD. The common-moded
feedback network generates the known voltage Vrp at the output of the NMOS
differential amplifier. The gain of the amplifier is determined by resistors R22
and
the transconductance gm44,45 .
Under equilibrium, the common-mode voltage Vrs is equal to V o2P+. This
voltage is used as the input for the PMOS common-source amplifier of the third-
stage. Since V o2P+ is equal to Vrp, transistors M58 and MP form a virtual
current mirror. Thus, the quiescent current of the output stage is determined by
the current through M58. The ratio of currents depends on the ratio of multiplicity
of transistor dimensions.
The dimensions of the transistors M46 and M47 are doubled to obtain twice
the bias current than the current through M21 and M22 of Fig. 3.5. Thus, the
gain of the NMOS differential pair is 2·gm44,45·R22 , which is equivalent to the PMOS
differential amplifier gain gm23,24·R2. The gain of the PMOS differential amplifier
is made equal to the NMOS differential amplifier to attain symmetry at the output
of the amplifier.
Unlike the first-stage, the second-stage is a single-ended differential ampli-
fier. The output of the NMOS and PMOS differential amplifiers are the inputs
for PMOS and NMOS common-source amplifiers of the third-stage, respectively.
The design of the third-stage is discussed in the next section.
3.6 Output-Stage
The output-stage of the amplifier is implemented with huge PMOS and
NMOS common-source amplifiers. In order to have same current ID through
transistors MP and MN , the dimensions of the PMOS transistor MP is made twice
the size of NMOS transistor MN , as the mobility of the electrons is approximately
33
about 2.5 times the mobility of holes. The schematic of the output-stage is shown
in Fig. 3.7
Vo2N+
Vo2P+
Vo1–
Cc1Rc1
Vout
MP
MN
Figure 3.7: Schematic of the output-stage
The input capacitance CGS associated with transistor MP is twice the
capacitance CGS of the transistor MN , as dimensions of the transistor MP is
twice the size of transistor MN . The pole of the second-stage PMOS differential
amplifier is approximately at 1R2·CGSN
. In order to have the same pole at the output
of NMOS differential amplifier, the resistance R2 is halved and the compensation
capacitance is doubled. The pole of NMOS differential amplifier is approximated
as 1R22·(2CGSN )
.
34
Output swing at the second-stage NMOS differential amplifier must be
equal to the output swing of PMOS differential amplifier to attain linearity at the
output. The swing at the output of second-stage is determined by voltages Vrn
and Vrp shown in Fig. 3.5 and Fig. 3.6. The voltage Vrn is made equal to Vrp
to obtain same swing at the input of third-stage and this is achieved by biasing
the bulks of NMOS transistors at a voltage lower than VSS. This increases the
threshold voltage of NMOS transistors. Hence, more voltage is required at the
gate of transistor M39 to allow the current from transistors M37 and M38 to pass
through. Thus, VGS of transistor M39 increases and this is comparable to Vrp.
The designed amplifier has three stages. Hence, the stability of the am-
plifier cannot be achieved without a compensation network. Miller compensation
and reverse-nested Miller compensation are used to stabilize the three-stage class-
AB amplifier. A brief description of the compensation network used in this design
is given in the following section.
3.7 Compensation used in the Design
The first-stage is an inverting gain configuration and second stage is a non-
inverting gain configuration. The output-stage is implemented with an inverting
gain configuration. A Miller compensation network is applied between the output
of first-stage and the output of the third-stage. Reverse-nested Miller compensa-
tion is used across the second-stage for NMOS and PMOS differential amplifiers
as shown in Fig. 3.1.
A symmetry is maintained while using the compensation networks at the
second-stage. This simplifies the small-signal model of the amplifier. The small-
signal model of each stage is discussed in the next section.
35
3.8 Small-Signal Models
The small-signal model of a symmetric folded-cascode amplifier is repre-
sented with a current source and a resistor parallel to it. The input-stage in this
design is not symmetric, as the compensation capacitor CC1 is connected to Vo1-
and there is no compensation capacitor to Vo1+. Hence, the circuit is divided
into two equivalent circuits. The left-half of the folded-cascode amplifier and its
corresponding small signal model is shown in Fig. 3.8, where gm1 is the transcon-
Vb4
Vb3
Vb2
Vin+
Vb1
Vb4
R1
Vo1
-
CS
M1
M2
M11
M9
M3
M7
M5
(a)
R1
gm1·Vin/2
Vo1-
(b)
Figure 3.8: (a) Left-half of the input-stage (b) small-signal model for left half.
ductance of the transistor M3. The input signal is assumed asV in2
, since Vin =
(Vin+ - Vin-) and Vin+ is only half of the signal Vin. The resistor R1�ro. Thus
R1 is the output resistance at node Vo1-. One end of the resistor R1 is connected
to node Vo1- and the other end is at virtual ground.
Similarly, the small-signal model for right-half of the folded cascode am-
plifier is as shown in Fig. 3.9, where gm1 is the transconductance of M4 and Rc2
is the compensation resistor. Under quiescent conditions the current through M3
36
Vb4
Vb3
Vb2
Vin-
Vb1
Vb4
R1Vo1+
Rc2
Vx
CS
M1
M2
M12
M10
M4
M8
M6
(a)
R1gm1·(-Vin/2)
Vo1+Rc2
Vx
(b)
Figure 3.9: (a) Right-half of the input-stage (b) small-signal model for right-half.
is equal to current through M4. Thus the gm’s are equal. The signal Vin- is
represented as −V in2
. The resistor R1 is approximately equivalent to resistor R1
shown in Fig. 3.8(a).
The second-stage is implemented with NMOS differential amplifier and
PMOS differential amplifier. Thus for each amplifier, the small-signal models are
drawn separately. The PMOS differential amplifier has symmetry at compen-
sation, hence only the output side of the differential amplifier is considered for
drawing the small-signal model as shown in Fig. 3.10.
The transconductance of the transistor M24 is gm2. The resistor R2 is
the output resistance at node V o2n+. Cc2 and Cc3 are compensation capacitors.
Capacitor C2 in Fig. 3.10(b) is the input capacitance CGS of the transistor MN
of the output-stage. The voltage Va is (Vo1+ - Vo1-) and the voltage Vo1+ is
represented as V a2
.
37
R2
Vo1+
Cc3
Vb4
Vb3
Vb2
Vo2n+
Cc2
Vx
CS
M28
M26
M24
M22
M21
(a)
Cc3
R2
Vo2n+ Vo1+Cc2
C2gm2·Va/2
Vx
(b)
Figure 3.10: (a) PMOS differential amplifier (b) small-signal model PMOS differ-ential amplifier.
The part of NMOS differential amplifier and it corresponding small-signal
model are shown in Fig. 3.11. Similar to the PMOS differential amplifier, the
compensation in NMOS differential amplifier is symmetric.
The input capacitance C2 of the transistor MP of output-stage is twice
the input capacitance of NMOS transistor MN of the output-stage, as the size of
PMOS transistor MP is twice the NMOS transistor MN . Thus to provide the same
pole frequency at the output of second-stage the common-mode feedback resistor
is made R22
and the compensation capacitor Cc3 is doubled. The transconductance
of the transistor M45 is 2·gm2, as the current through the transistor M45 is twice
the current through transistor M24 in Fig. 3.10. The signal Vo1+ isV a2
. The
current through current source in Fig. 3.11(b) is given by 2·gm2 · V a2
38
R2/2
Vo1+
2Cc3
Vo2p+
Vb3
Vb1
Vb4
Vx
Cc2
M43
M41
M46
M47
M45
(a)
2Cc3
R2/2
gm2·Va
Vo2p+Vo1+Cc2
2·C2
Vx
(b)
Figure 3.11: (a) NMOS differential amplifier (b) small-signal model NMOS differ-ential amplifier.
The third stage is implemented with NMOS and PMOS common-source
amplifiers. These transistors provide the required push-pull action for the class-AB
output-stage. The schematic of output-stage and its corresponding small-signal
model are shown in Fig. 3.12.
The transconductance of transistors MN and MP is gm3, as the quiescent
current through transistors MN and MP is same. Cc1 and Rc1 are the Miller
compensation capacitor and nulling resistor across the outputs of input-stage and
output-stage. Cout is the load capacitance. Rout is the output resistance of the
amplifier and is given by
Rout = RL‖roN‖roP (3.6)
39
Vo2N+
Vo2P+
Vo1–
Cc1Rc1
Vout
MP
MN
(a)
Rout
Vout
Cout
gm3·Von+
gm3·Vop+
Vout-
Rc1Cc1
(b)
Figure 3.12: (a) Schematic of output-stage (b) small-signal model for output-stage
Rout ≈ RL as RL � roN,P
The complete small-signal model of the designed three-stage class-AB au-
dio amplifier is shown in Fig. 3.13. The equations for current at each node are put
in a software Maple to determine the poles and zeros in the amplifier. The equa-
tion of the currents at each node and equations used to determine the pole/zero
frequencies is shown in APPENDIX B.
3.9 Pole-Zero Analysis
The gain of the amplifier is determined from the equations in APPENDIX B
as
Gain = gm1 ·R1 · gm2 ·R2 · gm3 ·Rout (3.7)
The designed audio amplifier has six poles and five zeros. The poles and zeros
obtained are shown in Table 3.2. The fourth pole is at high frequency, hence the
last two high frequency poles are neglected. The third zero is a RHP zero. The
40
R1
gm
1·V
in/2
Vo1-
R1
gm
1·(
-Vin
/2)
Vo1+
Rc2
Vx
Cc3
R2
Vo2
N+
Cc2
C2
gm
2·V
a/2
2·C
c3
R2/2
gm
2·V
a
Vo2
P+
Cc2
2·C
2
Rout
Vout
Cout
gm
3·V
o2
N+
gm
3·V
o2
P+
Rc1
Cc1
Fig
ure
3.13
:Sm
all-
sign
alm
odel
ofth
edes
igned
thre
e-st
age
clas
s-A
Bau
dio
amplifier
41
other two zeros are high frequency zeros, thus neglected. The first zero (ωZ1) is
used, to cancel the second pole (ωP2). Similarly the second zero ωZ2 cancels third
pole ωP3. Thus the system acts as a two-pole system, where the fourth-pole is
considered as second-pole.
Table 3.2: Poles and Zeros
Poles / Zeros
ωP12
gm2·R1·R2(2·Cc2+2·gm3·Rout·Cc1+3·Cc3)
ωP22·Cc2+2·gm3·Rout·Cc1+3·Cc3
Cc1·(2·gm3·Rout·R2·C2+2·R1·Cc2+3·R1·Cc3)
ωP32·gm3·Rout·R2·C2+2·R1·Cc2+3·R1·Cc3
R1·R2·C2·(6·gm3·Rout·Cc3+4·gm3·Rout·Cc2+2·Cc2+3·Cc3)
ωP4gm2·(2·gm3·Rout+1)Cout·Rout·gm2+2·C2
ωZ12
Cc1·R1+2·R2·C2
ωZ21
R2·C2 +2
Cc1·R1
ωZ3 -gm22·Cc3
42
Chapter 4
SIMULATION RESULTS
To test the functionality of the designed three-stage class-AB audio amplifier, the
amplifier is subjected to DC, AC and transient analysis tests. The following sec-
tions discuss the response of the of the amplifier. The obtained waveforms are
plotted using the Matlab code given in APPENDIX C. Four designs have been im-
plemented by varying the input bias current, dimensions of M39 and M58 shown in
Fig. 3.2, compensation capacitor and resistor values and its corresponding results
are plotted. The designs are named after their performance as LIQ (Low quies-
cent current), LTHD (Low THD), MIQ (Moderate quiescent current) and HCL
(High load capacitance). The design parameters of the four designs are given in
Table 4.1.
Table 4.1: Design Parameters
Design IB Rc1 Rc2 Cc2 Cc3 M39 & M58
HCL 8 µA 1 kΩ 1 kΩ 500 fF 500 fF m = ×4
LTHD 9 µA 1 kΩ 2 kΩ 200 fF 300 fF m = ×4
MIQ 9 µA 1 kΩ 2 kΩ 200 fF 300 fF m = ×6
LIQ 8 µA 1 kΩ 2 kΩ 300 fF 300 fF m = ×6
43
The bulk terminal of all the NMOS transistors are connected to -3V for all
the tests performed.
4.1 DC analysis
The DC analysis determines the symmetry and linearity of the amplifier.
The test-bench for DC analysis of the amplifier is shown in Fig 4.1. The input is
a DC voltage varied from -2mV to 2mV. To obtain the differential input voltage,
two voltage controlled voltage sources (VCVS) are used. One VCVS is with 0.5
gain and the other is with -0.5 gain. The input bias current of the amplifier is
8µA.
vdd
vdd
vss
vss
Vin-
Vin+
Vr
Vout
vss_sub
vss
_su
b
biasI
biasV
Vb2
CL
+
+-
+-
Vin
egain= -0.5
egain= 0.5 RL=16 Ω+ +
+
Figure 4.1: Schematic of the DC test-bench
The simulation results obtained for a resistive load of 16-Ω and capacitive
load of 500 pF are shown in Fig 4.2
The maximum current through the NMOS and PMOS transistors of the
output stage is 83.5mA. The maximum swing at the output is observed as ±1.25V.
4.2 AC analysis
The AC analysis determines the stability of the amplifier. The test-bench
for the AC analysis is shown in FIg. 4.3. Input is a 1V AC signal given at positive
input terminal of the amplifier. A large resistor and capacitor are used in the
feedback network to provide open-loop operation.
44
−2 −1.5 −1 −0.5 0 0.5 1 1.5 2−1.5
−1
−0.5
0
0.5
1
1.5
Out
put V
olta
ge (V
)
−2 −1.5 −1 −0.5 0 0.5 1 1.5 2
0
20
40
60
80
100
DC Input (mV)
Out
put−
Stag
e Cu
rrent
(mA)
MN = OFFMP = ON
MN = ONMP = ON
MN = ONMP = OFF
ID, MN ID, MP
Figure 4.2: DC-analysis output
The simulation results of the LIQ design is given in Fig. 4.4. The circuit
has input bias current of 8µA. The open loop gain of the amplifier is 48.4 dB
The input bias current of the LTHD design is made 9µA. The magnitude
and phase plots of the design is given in Fig. 4.5. The open-loop gain of the
amplifier is 53.7 dB.
The MIQ design is tested with a input bias current of 9µA. The simulation
results of the design is given in Fig. 4.6 and open-loop gain of the amplifier is
50.6 dB
45
+
vdd
vdd
vss
vss
1F
1Vac
Vin-
Vin+
Vr
Vout
vss_sub
vss
_su
b
biasI
biasV
Vb2
1GΩ
CLRL=16 Ω
+
++
Figure 4.3: Schematic of the AC test-bench
Table 4.2: AC Simulation Results
Design GainPhaseMargin
GainMargin
Gain Bandwidth
HCL 51.5 dB 72.04 ◦ 15.52 dB 1.23 MHz
LTHD 53.7 dB 70.8 ◦ 11.5 dB 2.03 MHz
MIQ 50.6 dB 70.3 ◦ 14.5 dB 1.56 MHz
LIQ 48.4 dB 70.7 ◦ 16.4 dB 1.24 MHz
The simulation results of the HCL design is given in Fig. 4.7. The circuit
has input bias current of 8µA.The open loop gain of the amplifier is 51.5 dB
The open loop gain, gain margin, phase margin and unity gain frequency of
four designs are tabulated and given in Table 4.2. The gain-bandwidth product of
the LTHD design is 2.03 MHz. MIQ design has the highest phase-margin among
the four designs.
46
100 101 102 103 104 105 106 107 108−100
−50
0
50
Gai
n (d
B)
100 101 102 103 104 105 106 107 108−400
−300
−200
−100
0
100
Frequency (Hz)
Phas
e (d
eg)
Figure 4.4: AC output of LIQ circuit
100 101 102 103 104 105 106 107 108−80
−60
−40
−20
0
20
40
60
Gai
n (d
B)
100 101 102 103 104 105 106 107 108−400
−300
−200
−100
0
100
Frequency (Hz)
Phas
e (d
eg)
Figure 4.5: AC output of LTHD circuit
47
100 101 102 103 104 105 106 107 108−80
−60
−40
−20
0
20
40
60
Gai
n (d
B)
100 101 102 103 104 105 106 107 108−400
−300
−200
−100
0
100
Frequency (Hz)
Phas
e (d
eg)
Figure 4.6: AC output of MIQ design
4.3 Transient analysis
The transient analysis determines the time domain response of a amplifier.
A square wave of ±100mV and 50kHz frequency is given at the negative input
terminal. The rise time and fall time of the input is 10 nS. The amplifier has a
inverting gain of ’4’. The test-bench for transient analysis is shown in Fig. 4.8.
The simulation result for LIQ is shown in Fig. 4.9. The load capacitance
of the amplifier is varied from 10 pF to 1.5 nF.
LTHD design has no ringing at the output up-to 1 nF. The output of the
amplifier for different load capacitances is shown in Fig. 4.10.
48
100 101 102 103 104 105 106 107 108−80
−60
−40
−20
0
20
40
60
Gai
n (d
B)
100 101 102 103 104 105 106 107 108−400
−300
−200
−100
0
100
Frequency (Hz)
Phas
e (d
eg)
Figure 4.7: AC output of HCL circuit
vdd
vdd
vss
vss
Vin-
Vin+
Vr
Vout
vss_sub
vss
_su
b
biasI
biasV
Vb2
40KΩ
CL
10KΩ
Va=100mV
Freq=50KHz
RL=16 Ω
+
+ +
+
Figure 4.8: Schematic of the Transient test-bench
49
5 10 15 20 25
−0.1
0
0.1
Inpu
t (V)
5 10 15 20 25
0
0.5
1
1.5
2
2.5
Time (µS)
Out
put (
V)
CL = 1.5 nF
CL = 500 pF
CL = 10 pF
Figure 4.9: Transient output of LIQ circuit
5 10 15 20 25
−0.1
0
0.1
Inpu
t (V)
5 10 15 20 25
0
0.5
1
1.5
2
2.5
Time (µS)
Out
put (
V)
CL = 1 nF
CL = 500 pF
CL = 10 pF
Figure 4.10: Transient output of LTHD circuit
50
The simulation result for MIQ is shown in Fig. 4.11. The load capacitance
of the amplifier is varied from 10 pF to 1 nF.
5 10 15 20 25
−0.1
0
0.1
Inpu
t (V)
5 10 15 20 25
0
0.5
1
1.5
2
2.5
Time (µS)
Out
put (
V)
CL = 1 nF
CL = 500 pF
CL = 10 pF
Figure 4.11: Transient output of MIQ circuit
The load capacitance for HCL design is varied from 10 pF to 5 nF, and
there was no ringing at the output. The slew-rate of the design is 1.25 V/µS. The
simulation result for HCL design is shown in Fig. 4.12.
4.4 THD analysis
A transient analysis has been performed on the amplifier to determine the
total harmonic distortion. The amplifier is connected in inverting gain configura-
tion with gain 1. The input is a sine wave of 2.45VPP and frequency of 1kHz. The
test-bench for the measuring THD is shown in Fig. 4.13.
The input has a dead time of 10 µS. The quiescent power of the amplifier is
determined at 5 µS, when there is no input signal. The THD of designs is measured
by using the THD-option of the simulator for a sample of the output signal. To
51
5 10 15 20 25
−0.1
0
0.1
Inpu
t (V)
5 10 15 20 25
0
0.5
1
1.5
2
2.5
Time (µS)
Out
put (
V)
CL = 500 pF
CL = 10 pF
CL = 5 nF
Figure 4.12: Transient output of HCL circuit
Table 4.3: Transient Simulation Results
Design CLOAD THDQuiescentPower
PeakPower
FOM
HCL 10 pF - 5 nF -77 dB 1.77 mW 97.6 mW 55.23
LTHD 10 pF - 1 nF -80.89 dB 1.98 mW 97.6 mW 49.27
MIQ 10 pF - 1 nF -78.56 dB 1.66 mW 97.6 mW 59.40
LIQ 10 pF - 1.5 nF -77.29 dB 1.47 mW 97.6 mW 66.25
measure the THD precisely, the time step parameters have been modified. The
step is given as 1µ and Max. step as 1µ.
The output waveform for measuring the THD is shown in Fig. 4.14. The
THD and quiescent power measurements for four designs are shown in Table 4.3.
52
vdd
vdd
vss
vss
Vin-
Vin+
Vr
Vout
vss_sub
vss
_su
b
biasI
biasV
Vb2
40KΩ
CL
40KΩ
Va=1.25V
Freq=1KHz
RL=16 Ω
+
+ +
+
Figure 4.13: Schematic for THD measurement
0 0.5 1 1.5 2 2.5−1.5
−1
−0.5
0
0.5
1
1.5
Inpu
t (V)
0 0.5 1 1.5 2 2.5−1.5
−1
−0.5
0
0.5
1
1.5
Time (mS)
Out
put (
V)
Figure 4.14: Transient output for measuring THD
53
Chapter 5
HARDWARE TESTING
The three-stage class-AB audio amplifier is fabricated in ON-SEMI 0.5µm pro-
cess through MOSIS. The hardware test is the real-time test for determining the
functionality of the amplifier. This chapter describes the layout techniques used
in the design, the experimental setup used for testing and the tests performed to
determine the operation of the audio amplifier.
5.1 Layout
The layout of four designs is implemented using the virtuso environment
of Cadence. A common-centroid technique is used to attain matching between
devices. The current at the output-stage is huge, hence strapping technique is
used for the huge current to flow through the transistor of the output-stage. In
order to avoid noise that can be coupled with bias voltage, the bias voltage Vb1
and Vb4 are shielded with substrate biasing voltage and Vb2 and Vb3 are shielded
with VDD. The layout of the four designs are shown in Fig. 5.1, Fig. 5.2, Fig. 5.3
and Fig. 5.4.
The designs are arranged in the frame of 40 pins. The layout of the frame
with the designs is shown in Fig. 5.5 and Fig. 5.6. The micrograph of the chip is
shown in Fig. 5.7.
5.2 Experimental Setup
The following apparatus are used for measuring the outputs:
54
Figure 5.1: Layout of LIQ amplifier
1. Oscilloscope: Hewlett Packard: 54600B: 100MHz, Digital Storage Oscillo-
scope.
The oscilloscope is used for plotting the input and output waveforms for a
transient response.
2. Function Generator: Agilent: 33120A: 15 MHz Function/Arbitary Wave-
form Generator.
The function generator is used for generating the input signals/waveforms.
3. Digital Multimeter: Agilent: 34401A: 612
Digit Multimeter.
55
Figure 5.2: Layout of LTHD amplifier
The Digital multimeter is used in this work for determining the DC voltages
AC voltages and DC currents.
4. Stanford Research System: SR770 FFT Network Analyzer.
The Stanford research system is used to determine the THD in the design.
The DC, Transient and THD measurements are obtained using the above
apparatus. The test procedure for these measurements is given in APPENDIX A.
5.3 DC Measurements
The amplifier is connected as a voltage follower and both input terminals
of the amplifier are connected to ground. The output voltage is measured using
56
Figure 5.3: Layout of MIQ amplifier
the digital multimeter. Under quiescent conditions the voltage obtained at the
output is offset voltage. The offset voltages of all the designs are discussed in
Chapter 6. The quiescent current is measured with the digital multimeter by
creating an open circuit between the VDD and the VDD pin of the chip.
5.4 Transient Measurements
The test-setup for transient measurements is similar to the test-bench used
in Fig. 4.8. The results obtained for each design for different capacitive loads are
plotted using the Matlab code given in APPENDIX C.
57
Figure 5.4: Layout of HCL amplifier
5.5 THD Measurements
The amplifier is used in inverting gain configuration with a gain of 1. A in-
put signal of 2.45VPP and 1kHz frequency is applied at the negative input terminal
of the amplifier. The output amplitude is measured using a digital multimeter.
The output is connected to the Stanford research system (SRC) to measure the
THD at the output. The SRC gives the amplitude of fundamental frequency and
its harmonics. The amplitude of first 10 harmonics for each design are shown in
Fig. 5.12, Fig. 5.13, Fig. 5.14 and Fig. 5.15.
58
LIQ
MIQ
Figure 5.5: Layout of the frame with two LIQ and MIQ.
The noise floor is at -106dB. The hardware measurements for all the designs
are shown in