106 IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 32, NO. 2, MAY 1990

A Novel RF-Insensitive EED Utilizing an Integrated Metal-Oxide-Semiconductor

Structure

Absfmcf-The description and characterization of a novel electro- explosive device (EED) is presented here. The structure is designed, using microelectronic fabrication techniques, to be inherently immune to radio frequency (RF) radiation and also offers protection against stray signals associated with RF-induced arcing. A detailed discussion of the structure, which includes the fundamental mechanisms of operation, fabrication techniques, the device’s frequency response and sensitivity to RF-induced arcing, and its compatibility with present fire control systems, is provided.

Preliminary test results of the prototype device are discussed and show a significant improvement in the system’s overall EM1 immunity. These results include bench and field measurements of the structure’s RF response for frequencies of 10-225 MHz and field measurements of the device’s sensitivity to RF-induced arcing. The measurements indicate a significant reduction in real power dissipated bq an EED employing the novel structure over an EED employing a conventional bridgewire (20 dB at 90 MHz) .

I. INTRODUCTION

VER THE PAST four decades, the electromagnetic 0 environment of electro-explosive devices (EED’s) has changed dramatically, and as an obvious consequence, the as- sociated EM1 problems have changed as well. The necessary operation of high-power radar and communication equipment in the proximity of EED’s (e.g., an aircraft carrier flight deck) has resulted in a typical operating environment that includes high-intensity electromagnetic fields and the possibility of in- termittent RF-induced arcing.

High-intensity RF fields associated with the EED’s environ- ment present a serious EM1 compatibility problem [ 11-[ 111. These fields can couple electromagnetic energy either through a direct or indirect path to an EED and cause accidental firing. Typical examples of these two types of EM1 problems are when RF radiation is incident on the device’s chassis (i.e., the EED acts as the load of a receiving antenna) or when RF-induced arcing occurs in the vicinity of the ordnance and couples energy to the EED (e.g.., via a conducting umbilical cable) [ 121.

An RF-induced arc over (discharge) results whenever suffi- cient electrical energy (charge accumulation) is present across

Manuscript received February 1. 1989; revised September 15, 1989. This work was supported by the Hazards of Electromagnetic Radiation to Ordnance (HERO) group at the Naval Surface Warfare Center, Dahlgren, VA.

The authors are with the Electrical Engineering Department, Auburn Uni- versity, Auburn, AL.

IEEE Log Number 9034 185.

‘L’ L

‘Pi‘

“T’

-+I++ b

Fig. 1. L , Pi, and T filter equivalent circuits

an air gap to initially ionize the gas and sustain an ionized channel. For the purpose of EM1 filtering, the arc-over phe- nomena is usually considered to occur whenever a given peak voltage of sufficient magnitude (geometry dependent) is de- veloped between electrified conducting surfaces.

In addition to the two sources of EM1 problems discussed above, the electrical environments of EED’s located on sur- face ships may contain signal components due to rectifica- tion of RF radiation. Rectification of RF radiation on ships is due to simple metal contact diode action. This is generally caused by poor electrical connections, such as those caused by corrosion of contacts or incorrectly connected fasteners [ 131. The rectified signal may have signal components that are at much lower frequencies than the source RF radiation and also contain a dc component, any of which (rectified dc and low- frequency components and source RF) may couple to the EED and cause ignition.

The first method of solving a given EM1 problem usually involves the installation of one or more passive filters. Sev- eral standard types of passive filters exist that can be utilized to attenuate stray RF signals. These filters can usually be classified as either L , Pi, or T types, or as combinations of each (see Fig. 1) [14], [15], and have historically been used as a first measure of eliminating electromagnetic interference (EMI) problems. In many instances, they provide sufficient attenuation of the interfering signals. However, there are an ever-increasing number of cases when conventional passive fil-

0018-9375/90/0500-0106$01 .OO @ 1990 IEEE

BAGINSKI AND BAGINSKI. NOVEL RF-INSENSITIVE EED 107

ELECTROMS - / DIELECTRIC

PCK =OVER THREADED

/-TERMINAL

I I I I I I

Fig. 3 . Structural diagram of typical EED.

layer construction, and many other techniques that cause the circuitry to be inherently immune to EMI.

Fig. 2 . Typical L-type RF filter.

ters provide inadequate EM1 protection or do not meet size, durability, cost, or other requirements. This condition is espe- cially acute on Naval surface ships, as was alluded to earlier.

There are a host of problems related to the implantation of the standard filters cited previously that do not involve the filter’s EM1 characteristics. It is beyond the scope of this study to describe in detail the totality of relevant nonelectromagnetic problems. However, because a general understanding of these nonelectromagnetic filter considerations is invaluable to those working in this area, several nonelectromagnetic realistic filter constraints will be briefly discussed.

Conventional filters are usually constructed from standard passive components assembled on printed circuit boards or hard wired within a metal chassis with size minimization be- ing of secondary importance. A typical example of this type of RF filter is shown in Fig. 2. The filter consists of a circular multilayer ceramic capacitor and a wound toroidal inductor. A physical requirement that is often placed on a iilter and difficult to achieve is the allowable size it may occupy [16]. Size limitations are a major consideration for filters used in conjunction with EED-related weapons systems [ 171. There- fore, a conventional filter may simply be too large for some applications.

Another factor that enters into filter selection, especially if large-scale installation of the device is involved, is the cost of the device. Even though this and similar filters have relatively few components, the cost of components and assembly may result in per-unit prices that are relatively high in comparison with the cost of an EED [ 181.

All of the problems associated with standard commercially available filters discussed above may be eliminated or signif- icantly reduced if the filter system is based on an integrated microelectronic design; this is the focus of the research pre- sented here. The fabrication of a microelectronic filter system reduces the device’s size greatly just as it does to the cost per device. Integrated circuit fabrication allows passivation, multi-

11. DESIGN CONSIDERATIONS

Prior to the discussion of the filter and EED fabrication, an overview of the important relative EM1 effects that must be considered in the specific EED filter design will be given (a typical EED is shown in Fig. 3). The possible electrical paths the interfering electromagnetic energy can take to cou- ple the interfering signal to the EED must be identified. In conventional filter systems, where the EED and filter are in separate stages, the interfering signal may couple through the filter to the EED or penetrate through small apertures (seams) and circumvent the filter entirely. If nonpropagating magnetic fields (near fields) are present, an induced EMF may also exist across the EED via closed-loop induction (Fig. 4).

Both of these EM1 effects can be mitigated by wafer scale integration of the EED and filter system together. The problem of interfering signals coupling directly through the filter with insufficient attenuation can be eliminated by simply increasing either the number of stages or the complexity of the integrated circuit filter without increasing the device’s cost or spatial requirement. EM1 effects due to closed-loop induction are directly proportional to the area associated with the closed conducting loop. Since microelectronic fabrication techniques allow the device’s maximum feature size and hence surface area to be reduced to hundreds of micrometers, the possibility of inductive EM1 is almost nonexistent.

111. FABRICATION

Starting material for the structure was a (100) oriented, 3-in diameter, 18-mil-thick p-type silicon wafer. The wafer is thermally oxidized to form a 100-nm-thick layer of silicon dioxide (Si02). The electrical properties of Si02 make it an ideal dielectric material for the device design. Its permittivity remains constant well into the gigahertz region [2 11.

A layer of nichrome was next sputtered onto the Si02. Nichrome was arbitrarily chosen for the initial study as a matter of convenience (other metal alloys could easily be uti- lized in future efforts). A low resistivity layer of copper was sputtered onto the nichrome to allow lead attachment by sim- ple soldering procedures. Aluminum was evaporated onto the

108 IEEE TRANSAC

r7Tk / MAGNETiC FIELD

CROSS SECTIONAL 7 + INDUCED EMF 1 B = u.H

EMF = VIA uH

Fig. 4. Schematic equivalent of magnetic field coupling

I I I 1

+Silicon Waler (18 mills Ihickl 4000i Aluminum-

I I I I

ills

Copper Conlacl Pad-

I I

I Nlchrome Healing IElemenl A 0 I 1 I

I W D i s f r i b u f e d Copocilance

Fig. 5 . Schematic representation of MOS structure.

backside of the wafer to provide an ohmic contact to the sili- con substrate. The copper and nichrome were then selectively etched to produce the pattern shown in Fig. 5. Dimensions were chosen such that the nichrome resistor had a value of 4.5 R. A photograph of the device is shown in Fig. 6, and a simplified lumped parameter representation is shown in Fig. 7. The structure described above has been tested (discussed in a latter section) and found to provide a significant amount of immunity to EMI. A qualitative discussion of the specific device features and its operation is given next.

The metal areas over the Si02 form a distributive capacitive structure that is in parallel with the resistance of the nichrome wire (Fig. 5) . Any stray RF signals that couple onto the leads essentially see a capacitor ( 1250 pF) in parallel with a resistor (4.5 0). The values of the capacitive reactance and resistance are selected to ensure that interfering RF signals will see an effective short circuit across the oxide layer and hence no appreciable real power dissipated by the nichrome resistor (no heating).

At high frequencies, the self-inductance of the nichrome wire and copper coating further enhances the structure’s im-

‘TIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 32 , NO 2, MAY 1990

Fig. 6 . Photograph of MOS structure.

I_

Fig. 7. Lumped parameter model of MOS structure.

munity to high-frequency signals. The series-inductive reac- tance of the nichrome resistor is proportional to frequency, making it an effective block to high-frequency wire currents.

EM1 problems related to possible inductive effects are prevented by the device’s geometry. The problem has been discussed previously and the device designed to minimize the cross-sectional area (-250 square mils associated with copper-oxide closed path) of any closed conducting path in the proximity of the EED [19].

The failure of an EED to ignite due to structural damage (e.g., vibration induced) is a serious problem that must be considered here since the EED is included in the wafer scale integration. In a conventional EED, this type of failure can usually be attributed to either the bridgewire breaking, a sev- ered connection of the bridgewire to the binding post, or the bridgewire melting at a hot spot caused by a material defect in the wire where the pyrotechnics have not ignited [22]. In the integrated circuit design presented here, these problems are reduced dramatically. The nichrome bridgewire is integrated onto a silicon wafer that serves as an effective chassis. In or- der for the series bridgewire circuit to be electrically opened due to structural damage, the integrated circuit itself would have to be fractured. This is extremely unlikely considering integrated circuit designs have the ability to withstand mechan- ical shocks associated with proximity fuses on artillery rounds

Nonuniformities in the wire that cause localized heating are associated with the standard processing used and can be sig- nificantly reduced by microelectronic fabrication processes. Standard wires are usually drawn from a die and have the problems of contamination, thickness variations, and a variety

1201 ’

BAGINSKI AND BAGINSKI: NOVEL RF-INSENSITIVE EED 109

of material defects such as disloca1.ions [23]. When a signal current (possibly a very low-level EMI) is passed through a flawed wire, the wire may melt apart at a point without ignit- ing the EED. By integrating the heating element onto the filter using microelectronic fabrication techniques, a much greater level of metal uniformity is achieved [24]. The level of metal purity and a host of other material properties of deposited films are also superior to that of conventional wires.

High K dielectric ceramic capacitors are included in the fil- ter system for the purpose of attenuating EM1 resulting from RF-induced arcing in the vicinity of the EED. RF-induced arcing has a wide spectral range that includes a dc component of relatively short duration [20] that may be especially trou- blesome. The ceramic capacitor is mounted externally from the fabricated device in parallel with the heating element. When arcing occurs, the ceramic capacitor limits the magni- tude of voltage developed due to the short-duration dc current by increasing the net capacitance of the bridgewire system ( V - Q / C , Q - s i D c d t ) and hence the power dissipated by the EED. Values of 0.1 p F for the ceramic capacitors are used here and found to effectively prevent detonation of the EED in the proximity of moderate RF-induced arcing.

A typical requirement of this and most filter systems is that they must operate reliably once installed for long periods of time. The device designed here may be used on naval surface ships and must be resistant to the associated environmental ef- fects. All materials used in the structure remain inherently sta- ble over an indefinite period of time. The metal systems have been chosen to eliminate any electrolytic action that could lead to corrosion. To further protect the device, the package could be hermetically sealed with either a low-temperature nitride deposition or a potting compound. Both these techniques are standardly used in the microelectronic industry and are rou- tinely employed to meet mil-spec requirements.

IV. THEORETICAL ANALYSIS

A lumped parameter model of the structure will serve as the basis for the theoretical and numerical analysis for frequencies ranging from 10-100 MHz. Use of lumped parameter model- ing in this range of frequencies is justified since the maximum electrical feature size of the prototype considered is much less than the minimum source wavelength (i.e., maximum struc- ture size - 0.6 cm, A,,, - 3.3 m).

The planar resistive film of the EED is modeled as a 4.5- R resistor R F . The self-inductance of the film LF and all subsequent impedances were measured using an HP 48 15A vector impedance meter. The value of L F was measured by examining an identical planar resistor deposited on a very thick (3-pm) dielectric layer. The value measured was 8 nH.

The electrical circuit that models current flow through the silicon wafer can be approximated as a series LRC configu- ration. The value of CS was calculated as

where C S is the shunt capacitance, eo represents the permit- tivity of free space = 8.84 x lo-'' F/m, E R represents the

TABLE I TABII I .ATED VALUES OF ,!,s AS A FLXCTION OF F R E Q U E N C ~

Frequency (MHz) Ls (nH)

1. 10 2. 20 3. 30 4. 40

5. 50 6. 60 7. 70 8. 80 9. 90

1.6 1.14

.93

.80

.72

.65

.61

.57

.54

relative permittivity of Si02 = 3.8, A represents the cross- sectional area of the capacitor = 6.35 x 1.27 mm2, and tox is the oxide thickness = 100 nm.

The factor of 0.5 results from two equivalent capacitors being in series. The calculated value of Cs is 1250 pF. The measured value was approximately 1200 pF.

The resistance of the silicon substrate was calculated as follows

where p is the resistivity of the silicon = 3 R-cm, L is the effective length of the resistance = 1.27 mm, w is the width of the substrate = 6.35 mm, and t is the substrate thickness = .457 mm. These values yield a resistance of 0.25 Q.

The inductance of the silicon substrate Ls was calculated by modeling the substrate as a rectangular conductor. Since the calculation can be found elsewhere, only the resultant ex- pression is shown [21]:

Ls = 1 / ( ~ . x ( w + L ) o ~ f) (3)

where U is the conductivity of the silicon, 6 represents skin depth = 1 /(~fpu)'/~, where p represents the permeability of Si = 47r x H/m, and f is the frequency.

The coupled power P c , which reaches the resistor R,G , is given as

P c = [ V / ( Z L , +Rs)I2Rs (4)

where V is an EMI-induced voltage at the input of the EED

ZLF = jWLF ( 5 )

where Z L ~ represents the impedance of LF . The total power coupled to the device PT is given as the

power coupled to the LRC branch consisting of Ls , R s , and CS (PRc) plus the power coupled to the R L branch consisting of LF and RF ( P L R ) :

PT = PLRC = P L R . (6)

The ratio of (PR,)/(PLRc + P L R ) gives the effective power reduction of the device. A numerical routine (Microcap 11) has been utilized to calculate the reduction in power. The numerical value of Ls was calculated (see Table I), and a

110 IEEE TRANSAC

- Theoretical Ileducllon

U - IO 20 30 40 50 60 70 80 90 100

FREOUENCY (MHZ)

-40 d6

Fig. 8. Theoretical and measured power reduction of MOS structure.

TABLE 11. TABULATED VALUES OF Cs AS A FUUCTION OF FREQUENCY

Frequency (MHz)

1. 10 2. 25 3. 50

4. 15 5 . 100

cs (Pf)

100,000 10,Ooo

6,000 3,000 1,500

"1 0--4 Theoretical fleduclion

-Measured Reduction [L W 3 -10 dB 0 a

- I O 20 30 40 50 60 70 80 90 100

4 -30 dB 0 U

-40 d6

FREOUENCY (MHz)

Theoretical and measured power reduction of MOS structure with Fig. 9 . monolithic ceramic capacitor.

TIONS ON ELECTROMAGNETIC COMPATIBILITY. VOL. 12 . NO 2 . MAY 1990

monitor waveforms at various points on the structure (specif- ically, points A and B in Fig. 5 ) . The voltage across each end of the resistor was measured with respect to ground. No appreciable phase difference in the waveforms existed, and therefore, the element appears resistive. This allows the real power dissipated by the element to be easily calculated.

Using a 4.5-12 resistive element, a series of measurements were performed. The voltage at point A was held at 2-V peak- to-peak, whereas the frequency of the signal was swept be- tween 10-100 MHz. Voltage readings were taken at A and B, and the differences between them were used to calculate the power being dissipated in the resistor. With no dielectric layer under the nichrome, all of the voltage at A dropped across the resistor. Thus, in all following figures, 0 dB was chosen as a reference point to make a ratio between the real power coupled to the resistor without a dielectric layer and the real power coupled with a dielectric layer and with a dielectric layer plus a ceramic capacitor in place.

Fig. 8 shows the response of the structure with a silicon dioxide dielectric layer. Note that at low frequencies, the structure looks purely resistive, and the power absorbed is nearly as much as in the case with no dielectric layer. As the frequency increases, the coupled power begins to roll off. The curve drops off sharply after 70 MHz.

This abrupt cut-off frequency may be lowered considerably by using a thinner dielectric layer to increase the capacitance of the structure. The next-generation prototype being con- structed includes this alteration. In addition, in this test struc- ture, no backside contact was employed in an effort to examine the response of the simplest structure with the least amount of processing involved. It is noted even in this case that coupled power is markedly reduced (-20 dB) at 90 MHz. It is also noted that the theoretical and measured power attenuation are in excellent agreement.

To extend the bandwidth of the EED's response and pro- vide protection against arcing, a 0.1-pF ceramic capacitor was added as an additional shunting element. The ceramic capac-

series of computations were performed with this information. The theoretical power reduction is shown in Fig. 8.

To further improve the low-frequency performance of the structure, a 0.1-pF monolithic ceramic capacitor was soldered in parallel with the planar resistor. The permittivity of the dielectric used in the capacitor decreases as a function of fre- quency. Therefore, an average capacitance value as a function of frequency was determined by measuring the impedance of numerous capacitors and averaging the results. The values were used in the simulation to describe the system and are shown in Table 11. The simulation results are shown in Fig. 9.

V. TEST AND EVALUATION

A . Benchtop Measurements To verify the response of the structure, several series of

measurements have been performed. The first series of mea- surements was performed utilizing a benchtop setup. An RF signal generator was used to drive a RF power amplifier. The EED was attached as the load. An oscilloscope was used to

itor utilized is a BaTi03 composite multilayer ceramic and represents the state of the art in this area because it combines the characteristics of highest capacity per unit volume, ex- treme stability, and a very high self-resonant frequency [25]. The volume of this element (12.6 x lop9 m3) is small enough to be easily incorporated into the smallest of EED's. The re- sponse of the system with the additional element in place is shown in Fig. 9. Below 100 MHz, the high K ceramic di- electric provides shunting. At the lower end of the frequency range (10 MHz), it is noted that the reduction is 30 dB. This configuration provides better than a 25-dB reduction in real power coupled to the resistor over the entire frequency range of 10-100 MHz. Beyond 100 MHz, the dielectric layer on the silicon provides shunting. It is also noted that excellent agree- ment exists between the theoretical and experimental values.

To further substantiate the effectiveness of the structure, a series of standard tests were performed by placing wax next to the resistive element. The wax has a melting temperature of - 113°F and was used here to clearly indicate if this temper- ature was exceeded. First, a calibration curve for the structure was determined. This curve establishes the amount of dc cur-

BAGlNSKI AND BAGINSKI: NOVEL RF-INSENSITIVE EED 111

1 400'F

1 1 l I I I I I I I I I I I I I I 2 3 4 5 6 7 8 9 IO II 1213141516

POWER DISSIPATED (x IO-^ WATTS)

Fig. 10. Calibration curve of MOS structure.

z 0 I- U -

W

W > + 4

W

CALIBRATION PULSES

Fig. 12. Typical calibration of instrumented EED

FIRING BAND IGNITER

Fig. 13. Schematic equivalent of 2.75-in folding fin aircraft rocket (FFAR). -*--I =;2; ' Fig. 11. Instrumentation system to measure bridgewire current

rent necessary to melt a wax element placed on the resistive element (Fig. 10). Note that 200 mw of real power is required to melt the wax.

The element with wax in place was next reconnected to the RF power supply, and as large a signal as the power source could produce was fed into the structure. The corresponding voltage levels were at least 2.5 V p-p and at most 10 V p- p between frequencies of 10-100 MHz. In no case was the wax (1 13°F melting point) noted to have melted and verifies that coupled real power has been significantly reduced via the additional dielectric elements.

B. Field Measurements A preliminary series of field tests of the structure were

performed at Naval Surface Weapons Center in Dahlgren, VA. Due to several factors, including facility time constraints, the structure was tested using only a single frequency of 17 MHz. The structure had its resistive element broken so that it appeared to be a purely reactive element. The broken re- sistive element was soldered in parallel with a 4 . 5 4 resistive match and a 0.1-pF ceramic capacitor (same capacitor used for benchtop measurements).

The resistive match was previously instrumented with a small thermocouple connected to a high-gain preamplifier. This element was utilized for reasons of compatibility with the instrumentation system. The preamplifier's output was cou- pled, via a fiber-optic cable, to an external high-gain ampli- fier used to drive a strip chart recorder. The arrangement is shown in Fig. 11.

To calibrate the resistive match, a varying dc current was injected through the bridgewire and the corresponding deflec-

tion on the strip chart recorded (Fig. 12). The rms value of current flowing through the bridgewire during field tests on the ground plane can be determined by proportional scaling of the output to the calibrated deflection.

The response time of the entire system was measured to be approximately 20 ms. In all fields tests, the coupling was maintained for tens of seconds in order to avoid any prob- lems with instrument response time. The minimum detectable bridgewire current was measured to be 5 mA (the instru- mented resistive match was placed in a 2.75-in rocket shown schematically in Fig. 13). The exposed firing band was con- nected directly to the input of the resistive match for the pur- poses of simulating a worst-case electrical scenario. This ar- rangement is particularly susceptible to RF because a direct path for a stray signal exists from firing band to rocket case (rocket case was used as the electrical ground reference).

The rocket with the instrumented resistive match was placed on the ground plane in the proximity of an RF communica- tion antenna and subject to peak electric field strengths of 25, 50, 100, and 125 V/m. An unaltered resistive match was also irradiated and used as a control. It was observed that an ungrounded technician touching the firing band of the rocket with a bare index finger induced a current of 29 mA in the resistive match for an electric field strength of 25 Vlm. This event represents the case of direct RF coupling, and when the reactive network was installed, the current was measured to be less than 3 mA. This corresponds to a reduction of 20 dB in real power coupled to the resistive match.

Increasing the field of 50 Vlm for the conditions described above produced a current sufficient to burn out the unaltered resistive match. At this field strength, measurements were taken of the induced currents in the resistive match with the network installed. Electrical contact to the firing band was made by touching the firing band with a bare index finger as before as well as by touching the firing band with a metal lead held by the technician. The measured currents in the resistive match (50 V/m field strength) were 6.5 mA when contact was made by hand and 38 mA when the metal lead was used. The field strength was increased to 100 and 125 Vlm and contact made by touching a bare index finger to the firing

112 IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 32, NO. 2 , MAY 1990

TABLE 111 TABULATED VALUES OF 1NI)UCED CURRENT

Unaltered Match

Field RF Intensity Coupling Arcing

25 V/m 29 ma 50 Vlm

100 Vlm 25 V/m

Capacitor Network With Match

RF Coupling Arcing

3 ma 14.5 ma 6.5 ma 38 ma

22 ma -- 26.5 ma --

band (metal lead was not used for 100 and 125 V/m field strengths). The currents were measured to be 22 and 26.5 mA for field strengths of 1 0 0 and 125 V/m, respectively. All induced currents are shown in Table 111.

Considering the tabulated values of the induced currents in the resistive match, it is evident that a significant reduction in dissipated power was achieved via the shunting capacitors. The use of shunting capacitors in this manner will probably have applications to other similar bridgewire structures.

IV. SUMMARY

The research conducted here has shown that an EED with significant EM1 immunity can be successfully fabricated us- ing integrated circuit design procedures. Standard benchtop and field tests of the structure indicate that the structure is inherently immune to the probable types of interference com- mon to its environment. These include wide-band RF radi- ation (10-100 MHz) and RF-induced arcing. Measurements made on an resistive match show a significant reduction in real power coupled to the bridgewire and the structure in place (20 dB at 90 MHz). In addition to the structures electronic and electromagnetic characteristics, it is relatively inexpensive to manufacture, exhibits long-term stability and reliability, and is compatible with present fire control systems.

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u51

t161

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H. W. Ott, Noise Reduction Techniques in Electronic Systems. New York: Wiley, 1988. H. A. Schwab, “Electrical properties of radio frequency glow dis- charges in air at atmospheric pressure,” NWL Tech. Rep. TR-2124, Nov. 1967. W. H. Hayt, Engineering Electrornagnetics. New York: McGraw- Hill, 1981. R. H. Thompson and J. G. Stuart, “Determination of electroexplosive device (EED) sensitivity,” Franklin Res. Center Tech. Rep. 606-25 16- 029, 1987. K. M. Ralls, T . H. Courtney, and J. Wulff, Introduction to Material Science and Engineering. A. B. Glaser and G. E. Subak-Sharpe, Integrated Circuit Engineering. Reading, MA: Addison-Wesley, 1979. M. Kahn, “Multilayer ceramic capacitors: Materials and manufac- ture,’’ Tech. Note 683, AVX Corp.

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