ANTENNA THEORY by Constantine A. Balanis Chapter 2.2.5 – 2

Hanyang University

1/29 Antennas & RF Devices Lab.

WEEKLY SEMINAR

Sunryul Kim2019.01.23

Hanyang University


Paper Review I Paper Review II

Single-Layer Circularly Polarized Antenna With Fan-Beam Endfire Radiation

IEEE ANTENNAS AND WIRELESS PROPAGATION LETTERS, VOL. 16, 2017

FREQUENCY

5.8 GHz Band

STUCTURE

M-dipole

Double-side slot-coupled line

(DSSCL)

Double-side parallel strip line

(DSPSL)

E-dipole

COMPARISON

Bandwidth / Beamwidth /

Dimensions

SUMMARY

Fig.1 Antenna geometry

FIGURES

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SUBSTRATE

F4B material (εr = 2.65, tanδ = 0.0013)

PARAMETERS

Fig.2 Antenna configuration :

(a) geometry ; (b) exploded perspective view.


CONFIGURATION

(b)

(a)

FIGURE

d = 1 mm s = 1.5 mm w =31 mm

l = 16.58 mm h = 1.5 mm lm = 7.78 mm

we = 1.5 mm le = 21.64 mm lec = 5.2 mm

lp = 1.35 mm gp = 0.2 mm wp =0.2 mm

dd = 0.4 mm ld = 7.96 mm wd = 0.8 mm

gd = 0.6 mm d0 = 0.6 mm df = 2.7 mm

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE

PARAMETERSVERIFICATION

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TWO IDENTICAL CONDUCTORS

VIA HOLES

By terminating one of its transverse

edges with via holes, a substrate

integrated shorted parallel-plate cavity

is formed

Its open aperture plays the role of

equivalent magnetic dipole

Fig.3 Part 1 : Magnetic dipole


EXPLANATIONM-DIPOLE

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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Since the electric field near the edge of the

cavity is sufficiently large while the

electric current is quite small, this

structure can efficiently realize the energy

coupling without generating obvious

spurious radiation.

Fig.4 Part 2 : T-shaped double-side slot-coupled line

(DSSCL)


DSSCL EXPLANATION

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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A meandered double-side parallel strip

line (DSPSL) with chamfered cutting

edges is directly connected to the

outstretched end of the DSSCL


Fig.5 Part 3 : Meandered double-side parallel strip line

(DSPSL)

DSPSL EXPLANATION

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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Two conductor arms are printed along the

opposite directions with a total length of le,

while lec marks their overlapped length.


Fig.6 Part 4 : Electric dipole

E-DIPOLE EXPLANATION

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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At a point along the endfire direction, the

electric field can be expressed as


Fig.7 Equivalent model from the view of field

synthesis

FIGURE EXPLANATION

If the design goal is realize LHCP at the

exactly endfire direction,k = 1

β = 270˚

E=θ Eθ+φ k ∙ Eθ ∙ e jβ

k = |Eφ| / |Eθ|

β =∠Eφ−∠Eθ

According to the definition of axial retio

(AR)

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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Relative magnitude and phase delay of

power flowing from the magnetic dipole to

the electric dipole are marked as

parameters m and α

And they are respectively relevant to

parameters k and β

M-dipole has omnidirectional radiation,

while E-dipole radiate at the positive half-

space beside xz-plane for the existence of

backward conductors, the value of k is

about twice of m


CLIPBOARD EXPLANATION

E=θ Eθ+φ k ∙ Eθ ∙ e jβ

k = |Eφ| / |Eθ|

β =∠Eφ−∠Eθ

Fig.7 Equivalent model from the view of field

synthesis

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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Overall phase difference at the observation point comes

from two aspects

The feeding phase difference caused by the traveling

of electromagnetic wave from the shorted parallel-

plate cavity to the printed arms

The spatial phase difference caused by the

noncoincidence of the positions of two

complementary dipoles along the direction of wave

propagation


PHASE

M-dipole has omnidirectional

radiation, while E-dipole

radiate at the positive half-

space beside xz-plane for the

existence of backward

conductors, the value of k is

about twice of m

MAGNITUDE

k ≈ 2 × m Required feeding phase delay for generating ideal CP

wave

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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The widths and lengths of DSSCL, DSPSL, and the electric dipole all have influences on the

relative power and phase delay between the load and the source.

To simplify the design procedure, only θp and θd are chosen as the variables to realize the

tuning of m and α here.

Shortening the distance between two dipoles is beneficial for achieving a wider AR beamwidth

and higher gain

DSPSL of a narrow linewidth is adopted, which makes its characteristic impedance higher

than the input impedance of a normal half-wavelength E-dipole.

By increasing the length of the E-dipole, real and imaginary parts of its input impedance rise,

which makes it inductive.

Two printed arms are designed with an overlapped, which can be modeled as a capacitor.


Fig.8 Equivalent model from the view of microwave matching circuit

PROCEDURE

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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The width of DSPSL is chosen to be

0.8mm, which correspond to a

characteristic impedance of

approximately 170.1 Ω

le = 21.64 mm and lec = 5.2 mm

Thanks to stable directivity of the dipole

antenna over a sufficient large length

variation, the final radiation pattern is

almost unaffected after the modification


IMPEDANCE EXPLANATION

Fig.9 Effects of lengths le and lec on the input

impedance of the electric dipole at 5.8 GHz.

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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Impedance of Port 2 is defined as

169.7889 + j 0.1793 Ω

m can be effectively controlled by lp

m is almost independent of ld


MAGNITUDE EXPLANATION

Fig.10 Effects of lengths ld and lp on relative

magnitude between two dipoles.

Port 1 : power feed

Port 2 : electric dipole

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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PHASE EXPLANATION

Fig.11 Effects of lengths ld and lp on relative phase

delay between two dipoles.

Impedance of Port 2 is defined as

169.7889 + j 0.1793 Ω

α can be effectively controlled by either lp

or ld

Final design point marked by the

pentagram shows | S21 | = − 4.4 dB , which

corresponds to m = 0.56

α = 312.1°, which is also close to the

required 308.6° Port 1 : power feed

Port 2 : electric dipole

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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PROTOTYPE

Fig.12 Photogtaph of the planar endfire CP anternna

prototype.

Using a standard PCB fabrication process

Agilent’s N5230A vector network analyzer

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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VERSUS FREQUENCY

Fig.13 (a) S11 and efficiency versus frequency and (b)

AR and gain versus frequency

Both simulated and measured results are

in good agreement

Measured −10 dB bandwidth is from 5.7

to 5.9 GHz (3.5%)

Radiation efficiency at the desired band is

more than 90%

Measured 3 dB AR bandwidth is found to

be 5.65 – 5.9 GHz (4.3%)

The discrepancies are mainly attributed to

the fabrication tolerance and test

environment

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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RADIATION PATTERN

Fig.14 Radiation patterns at 5.8 GHz: (a) azimuth

plane and (b) elevation plane.

The measured gain at the exactly endfire

direction is 3.99 dBi

Wide coverage is obtained as can be

clearly seen

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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PERFORMANCE COMPARISONS

CONFIGURATIONDESIGN

FIELD

DESIGN

MATCHING

DETERMINE


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Wearable Dual-Band Magneto-Electric Dipole Antenna for WBAN/WLAN Applications

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 63, NO. 9, SEPTEMBER 2015

APPLICATION

WBAN/WLAN

FREQUENCY

2.45 GHz Band

5 GHz Band

STUCTURE

Magneto-Electric Dipole

ANALYSIS

S11 / Radiation Pattern / Gain /

Efficiency / Bending / SAR

SUMMARY

Fig.15 Antenna geometry

FIGURES

Fig.16 Bending insensitive

antenna

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SUBSTRATE

3-mm-thick Felt textile

(εr = 1.3, tanδ = 0.044)

CONDUCTOR

0.17-mm-thick ShieldIt Super conductive

textile (conductivity = 1.18 × 105 S/m)

Based on the conventional magneto-

electric dipole topology, which combines a

planar magnetic dipole with an electric

dipole.

Two U-shaped slots are cut in the planar

electric dipole.

Fig.18 Felt textile, ShieldIt Super


GEOMETRY FIGURE

Fig.17 Antenna geometry.

Physical dimensions in mm.

GEOMETRY THEORYRETURN

LOSS

RADIATION

PATTERNEFFIC.&GAIN BENDING SAR

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Wideband

Low back radiation

By positioning an electric and a magnetic

dipole orthogonally with respect to each

other, the antenna forward radiation can

be enhanced while simultaneously

reducing the back radiation.Fig.19 Operating theory of magneto-electric dipole

antenna.


THEORYFIGURE


LOSS

RADIATION


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Without slots 2.45 GHz : Two resonances can be combined

to form a wide band

5 GHz : The electric and magnetic

resonances are sill separated

Bringing the two resonances closer

together by the addition of the slots Two resonances are combined to form a very

wide upper band from 4.57 to 6.28 GHz

This however results in a bandwidth

degradation in the first band – from 670

MHz for the antenna without slots to 490

MHz for the slotted antenna

Nonetheless this 490 MHz band width is sill

more than sufficient to meet the

WBAN/WLAN requirements

Fig.20 Simulated and measured reflection coefficients.


SIMULATIONS-PARAMETER


LOSS

RADIATION


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Lower band Two measured resonances are slightly more

separated from each other compared to the

simulations

This result in a maximum S11 of -7.5 dB

within the required band, which is still

acceptable in many real applications

Theses slight disagreements are mainly

caused by the fabrication inaccuracies

and the inhomogeneous material

properties

Upper band The same tendency is observed, but less

severe

S11 never goes above -10 dB

The measured bandwidth is wider than the

simulated one

Fig.20 Simulated and measured reflection coefficients.


MEASURES-PARAMETER


LOSS

RADIATION


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Stable radiation characteristic is

observed throughout the

operating frequency band

Broadside pattern with high

front-to-back ratio (FBR)

The higher cross polarization in

the upper band is caused mainly

by the feeding pin, which has a

considerable length compared to

the wavelength

Fig.21 Simulated and measured radiation patterns : in the xz plane at (a) 2.3

GHz ; (b) 2.6 GHz ; (c) 5.2 GHz ; (d) 5.8 GHz ; in the yz plane at (e) 2.3

GHz ; (f) 2.6 GHz ; (g) 5.2 GHz ; and (h) 5.8 GHz.


Red : simulated copolar

Magenta : measured copolar

Black : simulated cross-polar

Blue : measured cross-polar

RADIATION PATTERN


LOSS

RADIATION


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The measured forward realized gain is at

least 4.7 and 3 dB in the lower and upper

frequency band, respectively

The radiation efficiency and the total

efficiency are both above 50% and 60% in

the lower and upper band, respectively

Fig.22 Simulated efficiency and gain of the antenna.


MEASUREEFFICIENCY & GAIN


LOSS

RADIATION


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The antenna mounted on a vacuum

cylinder with a varying radius r.

When bending radius is reduced

Lower band increases

Upper band narrows

But the changes are insignificant

Directivity & Gain are lower

Since the bending decreases the

electrical size of the ground, especially

in the lower band

Nevertheless, these value are still

acceptable for on-body communication.

From this evaluation, it is clear that

once an accurate fabrication can be

established, the antenna is observed to

be very robust against bending.

Fig.24 Simulated S11 under bending conditions.

Bending along a cylinder.


CYLINDERBENDING CONDITIONS


LOSS

RADIATION


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This mimics the normal wrinkling of cloth

on the human body

As the wrinkling becomes denser, the

bandwidth of the antenna decreases in

both bands

However, even wrinkling angle is the most

extreme, i.e. 60˚, the bandwidth are

maintained at 115 and 655 MHz in the

lower and upper band, respectively.

Also, no significant degradations occur in

terms of gain and FBR


NOTCHED CURVEBENDING CONDITIONS

Fig.25 Simulated S11 under bending conditions.

Bending along a notched curve, p = 10 mm.


LOSS

RADIATION


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Human tissue model

3-mm-thick skin layer

7-mm-thick fat layer

60-mm-thick muscle layer

300 mm × 300 mm sized

6 mm from the antenna ground

The SAR value is calculated based on the

IEEE C95.1 standard and averaged over

10 g of biological tissue

It is clear that the estimated maximum

SAR value is 0.045, which is far below the

European threshold of 2 W/kg


SAR

Fig.26 SAR distributions at different frequencies. (a)

2.45 GHz. (b) 5.2 GHz. (c) 5.8 GHz.


LOSS

RADIATION


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THANK YOU

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ANTENNA THEORY by Constantine A. Balanis Chapter 2.2.5 – 2

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