Auxiliary Power Supply for Medium-Voltage ModularMultilevel Converters

Dimosthenis Peftitsis, Michael Antivachis and Jurgen BielaLAB FOR HIGH POWER ELECTRONICS SYSTEMS, ETH Zurich

Email: peftitsis@hpe.ee.ethz.ch, URL: http://www.hpe.ee.ethz.ch/

Keywords<<MV isolation>>, <<resonant converter>>, <<auxiliary power supply>>, <<modular multilevelconverter>>, <<LLC converter>>

AbstractDesign considerations for a modularized auxiliary power supply suitable for variable voltage modularmultilevel converters, in terms of electrical performance and isolation against high voltages, are pre-sented. In the presented auxiliary supply, an LLC resonant converter along with several individual floatedvoltage sources containing high-frequency and high-voltage-isolation transformers are employed. Thedesign process of the individual floating voltage sources is based on circuit and FEM simulations. Partialdischarge measurements are also shown for the proposed high-voltage transformer design. Last but notleast, electrical measurements performed on a down-scaled laboratory prototype reveal the stable andproper operation of the system at various operating points.

1 IntroductionHigh-voltage direct current (HVDC) transmission for long distances, and from/to multiple energy sourcesand loads, has been a research topic of high importance during several decades. In addition to this,medium-voltage direct current (MVDC) transmission systems seem to be a promising technology inorder to overcome problems associated with the alternative current (ac) transmission of electricity indecentralized power generating systems (e.g. bulky passive components etc.) [1]. Moreover, MVDCsystems provide flexibility in terms of local energy distribution, while they are also preferable when itrefers to fault current management compared to HVDC (i.e. lower fault currents etc.) [2, 3].MVDC can serve as an interface between decentralized energy generation and various locally-distributedloads. In particular, collector grids for photovoltaic and off-shore wind generation systems count astwo application examples where MVDC seems to be advantageous compared to the traditional ac grids(flexible control of power etc.). Moreover, the electrification of the various ac and dc load in electricships can be efficiently done by using a MVDC distribution system on ships [3]. Thus, bulky and lossyac transformers can be eliminated.Power electronics interfaces are vital components in order to operate such systems reliably and efficiently.Medium direct voltage test facilities are necessary in order to evaluate the performance of various com-ponents and converters. However, as the voltage level increases, there is only a narrow area of powerelectronics converters that are suitable for such test facilities.For testing the MV dc components, the modular multilevel converter (M2C) seems to be a very promisingtopology when it refers to variable medium-voltage and high-power direct voltage source used as labo-ratory experimental facility. In such cases, the M2C operates as bidirectional rectifier having a variableac input voltage and supplies a variable direct output voltage (dc) to test dc loads. Such a MV 250 kVAvariable dc voltage source, based on the M2C concept shown in Fig. 1 is developed at ETH for testingpurposes.Such a variable high-voltage lab power supply must have two features. The first one deals with currentlimitation under fault conditions. This can be solved by using full-bridge SMs which allow to limit thecurrent or to turned-off completely if this is necessary. The second design feature is associated withsupplying the auxiliary power to the system. The term “auxiliary power supply” refers to power supplieswhich energize the gate-drive circuits, controllers etc in each SM. Considering the modularized structureof the M2C and the floating modules, the major constraint of such a supply is to ensure that a certainisolation voltage-level exists among the individual power supplies for each SM. In addition to this, theoperation of the auxiliary power supply must be independent on the operation of the M2C as such. This

Figure 1: Circuit diagram of a phase-leg of the M2C with the resonant auxiliary power supply.

means, that even if the direct voltage, Vd on the output of the M2C is zero (Fig. 1), the auxiliary powermust be present in order to assure that all features of the SMs are enabled (e.g. communication amongthe SMs, fault-detection circuits etc.). The same applies when the voltage is ramping up slowly.A concept used as auxiliary power supply, which employs a high-voltage tapped-inductor dc/dc buckconverter with a high step-down ratio was proposed in [4]. In particular, the input stage of this converteris directly connected to the capacitor-tank of the module and a low voltage is supplied on the output.However, it is not a suitable solution for M2Cs operating with a wide range of input and output voltagesif the auxiliary power must be present before the main power is supplied.Therefore, in this paper, an auxiliary power supply for modularized power electronics converters withinput and output voltages having large variations is presented. A circuit diagram of this auxiliary powersupply connected to a phase-leg of an M2C is shown in Fig. 1, while Tables I and II summarize thedesign parameters of the M2C and the auxiliary power supply, respectively. The main concept is basedon a resonant converter which supplies power to several isolated rectifier units via transformers based onring ferrite cores. A low-voltage design of such a converter has already been described in [5], while ahigh-voltage inductive power supply is shown in [6]. In the later work, however, the high-voltage designis not discussed.

Table I: Design parameters of the M2C.

Line-line input voltage, Vin 0...9 kV/50 HzOutput voltage, Vd 0...35 kVSubmodules/phase N=30

Rated power, P 250 kVA

In this paper, a simple medium-voltage isolation concept for such an auxiliary power supply is presented.An overview of this systems is presented in Section 2. The isolation requirements and design procedureof the rectifier units from electrical and mechanical points-of-view are treated in details. Furthermore,design considerations of the resonant tank with respect to the isolation voltage-level (Sections 3) andleakage inductance (Sections 4) are explained. More specifically, it is shown that a trade-off between the

Table II: Design parameters of the auxiliary supply consisting of 30 rectifier units.

Input voltage, Vdc 400 VOutput voltage, Vout 14 V

Rated power of each rectifier unit 30 WClamped output voltage, Vout,cl 28 V

Range of the supplied power, ∆P 27.5 W ±10%Switching frequency 45-55 kHz

Resonant capacitor, Cr 22 nF (Epcos B32672L8223, 9 series connected and 3 parallel branches)Turns-ratio of the transformer 2:2Leakage inductance, Lσ,1...N 660 nH

Magnetizing inductance, LM,1...N 20 µHRing core 2 x Kasche R102/65.8/15, K2006

value of the leakage inductance caused in the ring-core transformer and the isolation voltage level mustbe taken into account for a safe operation. It is the leakage inductance that governs the performance ofthe resonant tank, on the one hand, while on the other hand, the distance of the primary winding fromthe core must also be properly adjusted so that a non-destructive electric field is present. The detaileddescription of the resonant converter is shown in Section 5, while experimental results are presented inSection 6. Last but not least, conclusions are given in Section 7.

2 Auxiliary Power Supply System

Figure 2: Visual illustration of a part of the rectifier units with their housing.

A simplified circuit diagram of the modularized resonant power supply connected to a phase-leg of anM2C is shown in Fig. 1. It consists of a resonant converter (full-bridge converter with a resonant tank) andthe rectifier units which are supplied via high-frequency transformers (T/Fs). The series resonant tankconsists of a capacitor Cr and the series connected primary windings of the T/Fs. In fact, each primarywinding is represented by the leakage and magnetizing inductances, Lσ and LM, respectively, while theT/F is assumed to be ideal. As shown in Fig. 1, the T/F T-models are series-connected constituting theinductor of the resonant tank.Fig. 2 depicts an illustration of the rectifier units. Considering that a three-phase M2C contains a numberof N SMs per phase-leg, 3*N individual rectifier units are required in order to supply the auxiliarypower to the converter. Each of the rectifier units is mounted on side of a SM as depicted in Fig. 2. Acertain distance between the neighboring rectifier units must be kept for isolation and space-arrangementreasons.A single rectifier unit contains two stacked ring cores which are potted with a material having highpermittivity, εr = 4.5. Therefore, the electric field is pushed in the air between the primary winding

and the cores. Moreover, a plastic housing is used in order to mechanically fasten the potted cores onthe rack with the SMs. The permittivity of the plastic housing is also higher than unity. The primarywinding, on the other hand, is a two-turns winding which passes through an aluminum tube which isgrounded (field-shaping tube) and through the core as well. Design considerations of the field-shapingtube in terms of shielding against electric fields and eddy current losses are presented in Section 3 and 4,respectively. In order to achieve a more uniform distribution of the magnetic field generated in the T/Fs,the returning path of the primary winding is split in two symmetrical paths as also shown in Fig. 2. Field-shaping aluminum tubes also also used for the returning paths of the primary winding. These tubes, aswell as, the tube which passes through the cores are mechanically fastened on the top and bottom of thewhole construction. Thus, the primary winding can be properly adjusted so that the distance between theprimary winding and the cores will be kept constant. Therefore, no mechanical support of the primarywinding is required on each rectifier unit, where there is only air between the field-shielding tube and thecores. Moreover, it must be noted that taking into account the range of the switching frequencies wherethe resonant converter is aimed to operate, the primary winding consists of a litz wire.The creepage distance between the field-shaping tube and the secondary winding on each rectifier unitis approximately equal to 75 mm. According to DIN EN 60 664-1/VDE 0110 standard, this distance isadequately long to ensure a safe operation with a peak voltage of 22.5 kV.In Fig. 2, a cross-section of a single rectifier unit is also depicted, where the arrangement of the twostacked ring cores is shown. From this figure, it can be seen that a distance of approximately 5 mm existsbetween the cores and the inner walls of the plastic housing. This has intentionally been done in order toleave space for the potting of the cores. Moreover, notches are also designed on the plastic housing, sothat there is adequate space for the secondary winding to enfold the ring cores. The reason of using twostacked ring cores in each rectifier unit is the reduction of the core losses. A more detailed analysis ofthe electrical design of the converter, resonant tank and rectifier units is given in Section 5.

3 High-Voltage Design ConsiderationsIn order to evaluate the requirements for isolation between the primary and secondary windings, a 3Dfinite-element-method (FEM) simulation of the electric field distribution was performed. From the designinputs of the M2C, the maximum direct voltage that the isolation must withstand is equal to 22.5 kV.This is derived as follows. The corresponding voltage level equals 22.5 kV, which is defined as 35/2kV (maximum output voltage Vd = 35 kV divided by two arms per phase-leg) plus 5 kV direct voltageoffset. It must be noted that the worst case voltage of the primary winding equals 0 V, so that thevoltage difference is the highest possible. All in all, the high-voltage design must be done considering avoltage difference between the SM and primary winding which equals 22.5 kV. Thus, the simulation wasperformed by assuming the worst case in terms of voltage difference between the primary winding andrectifier units.For simplicity, in the 3D FEM simulation model, only 3 rectifier units were considered. In particular,these are considered to be connected to the SMs placed closer to the mid-point of the M2C. The permit-tivity of the potting material is equal to εr=4.5, while the aluminum field-shaping tube has an outer andinner diameters of 20 mm and 18 mm, respectively. A top view of the electric field distribution betweenthe primary and secondary windings in the rectifier unit connected to the mid-point of the M2C is shownin Fig. 3. Taking into account that there is only air between the aluminum grounded tube and the outerwall of the potting, the electric field must not exceed 2.4 MV/m [7]. From Fig. 3, it is clear that theelectric field in the area between the potting and the field-shaping tube is kept lower than the electricbreakdown field of the air.The effect of displacing the field-shaping metallic tube far from the center of the rectifier unit has alsobeen investigated. In particular, the field-shaping tube was moved further from the secondary windings,as indicated with the red arrow in Figs. 3 and 4a, at steps of 1 mm. Based on the results of the 3DFEM simulations, the peak electric fields between the primary winding and the wall of the housing andbetween the primary and secondary winding were extracted. Fig. 4b shows the variation of the twoaforementioned peak electric fields as a function of the tube displacement for various diameters of thetube. It is observed that the minimum peak electric field (2.05 MV/m) is obtained when a field-shapingtube having a diameter of 20 mm is placed in the center of the ring core. Other positions and diametersof the tube result in higher electric field as shown in Fig. 4b.In addition to these constraints, a crucial characteristic of the potting is associated with being free fromair-bubbles.

4 Parasitic components of the auxiliary power supplyIf the concept of the presented modularized supply is carefully examined two major challenges must beaddressed. The first one is associated with the leakage inductance. In particular, the leakage inductancedepends on the geometry of the core and the distance of the primary winding from the core. The closerthe winding is mounted on the core, the lower the leakage inductance that is obtained. A close placement

Figure 3: Top-view of the 3D FEM simulation results of the electric field distribution in the rectifier unit usingfield-shaping tubes having 20 mm of diameter.

(a)

(b)

Figure 4: (a) Schematic drawing shown the displacement of the field-shaping tube showing the paths for the twoelectric fields examined and (b) peak electric field between the primary winding and the core and between theprimary and secondary windings for various positions of the field-shaping metallic tube.

of the primary winding, however, will reduce the isolation distance between the primary winding andthe rectifier units. This counts as the second design challenge of the presented concept. It must be notedthat, under normal operation, maximum isolation voltage for the rectifier units equals 22.5 kV.The expected leakage inductance caused in each rectifier unit, was estimated using 3D FEM simula-tions. In particular, the same geometry of 3 rectifier units, as in the case of electric field simulations,was considered. A current excitation was provided in the primary winding which passes through thering cores, while half of this current was assumed to flow through each of the primary winding returningpaths. Moreover, it is assumed that the total magnetic flux in the ferrite cores equals zero (equal Ampere-turns in both primary and secondary windings). Taking into account these simulation inputs, the leakageinductance, Lσ, in each rectifier unit was estimated by integrating the magnetic energy. Fig. 5a showsthe leakage magnetic field in the area between the primary winding and the ferrite cores for the afore-mentioned geometry consisting of 3 rectifier units. Based on the simulation results, it is found that theleakage inductance on the primary side for the aforementioned geometry is Lσ,FEM = 3.7 µH. Hence, anaverage Lσ,FEM,av per rectifier unit is assumed to be equal to 3.7/3=1.23 µH.

In order to verify the 3D FEM simulation results, a measurement of Lσ was also performed. The stack of3 rectifier units which is shown in Fig. 2 was built. Fig. 5b shows the equivalent circuit of the 3 rectifierunits that is used for measuring the leakage inductances. In particular, the secondary windings of therectifier units were short-circuited, while the leakage inductance was measured on the primary winding(between points A and B shown in Fig. 5b). From these measurements, the total leakage inductance of 3rectifier units was found to be Lσ,m = 3.96 µH, which gives an average Lσ,m,av = 1.32 µH per rectifier unit.A deviation of approximately 7% is observed between the value for Lσ obtained from FEM simulations

(a)

N1:N2Lσ1

LM1

Ip

Ip

N1N2

Lσ1

N1:N2Lσ2

LM2

Lσ2

N1:N2Lσ3

LM3

Lσ3

A

B

Ip

N1N2

Ip

N1N2

(b)

Figure 5: (a) 3D FEM simulation results of the leakage field caused in the primary winding and (b) Test circuit formeasuring the leakage field caused in the primary winding.

and lab measurements.Taking into account the T/F T-model circuits shown in Fig. 5b, the values for Lσ1,Lσ2 and Lσ1 canbe obtained. It should be noted that in the circuit shown, the leakage inductances are reflected in theprimary side of the T/Fs, whereas the T/Fs are considered to be ideal. The corresponding values for theleakage inductances are Lσ1 = Lσ2 = Lσ3 = 1.32/2 = 0.66 µH. These values are taken into account forthe calculations presented in Section 5 regarding the electrical performance of the converter. Moreover,the total magnetizing inductance of the 3 rectifier units was also measured by keeping the secondarywindings open-circuited. It was found that the total magnetizing inductance equals approximately 60µH, which, on average, corresponds to average values of LM1 = LM2 = LM3 =20µH.A further design constraint is related to the induced eddy currents in the metallic field-shaping tube.An additional design requirement is, therefore, associated with suppressing the induced eddy currentson the metallic tube. The effect of the metallic tube thickness on the induced eddy currents was alsoinvestigated. Using the geometry shown above, the effect of the induced eddy currents on the field-shaping metallic tubes was studies by means of 3D FEM simulations. The outer and inner diameters ofthe examined field-shaping tube are equal to 20 mm and 18 mm, respectively, while the height of thetube was set to 270 mm. It must also be noted that for the current excitation in the 3D FEM simulationmodel, a current source of 6 A peak-peak current at a frequency of 60 kHz was used. Under these valuesthe worst-case operation of the converter with 30 rectifier units at rated conditions is expected.Fig. 6 illustrates the 3D FEM simulation results for the induced eddy currents. Based on the FEMsimulation results, the actual eddy-current losses, in the case of 30 rectifier units, were found to be 0.045W.

5 Design of the Resonant Converter

Figure 6: 3D FEM simulation results of theeddy current induced on the field-shapingmetallic tubes.

Apart from the HV isolation and parasitic components is-sues, the electric performance of the resonant converter mustcarefully be studied. In particular, the operating range of theresonant converter must be such that it is able to supply anoutput power equal to the rated power. An additional designconstraint of such a converter deals with supplying a con-stant output voltage also for changing output power. Thiscan be achieved for a constant switching frequency, fsw if theresonant tank is properly designed. As already shown in Sec-tion 4, the leakage inductance is high so that to be consideredin the design process of the resonant converter. This is dueto the fact that both Lσ and LM are basically determined bythe HV isolation requirements of the rectifier units (distanceof the primary winding from the cores, space requirements,etc.). Therefore, the examined resonant converter is of theLLC type [8], as it is also shown in the schematic diagram inFig. 1. In the LLC resonant converter, three passive elementsare contained: the leakage and magnetizing inductances, Lσ

and LM, respectively and the resonant capacitor, Cr. Thus,two resonant frequencies exist. A unity gain, which is in-dependent on the load variations, is expected at the series-

resonant frequency: f0 = 1/(2 ·π ·√

(N ·Lσ ·Cr)). On the contrary, the pole-resonant frequency is givenby: fp = 1/(2 ·π ·

√(N · (Lσ +LM) ·Cr)). Under no-load conditions, the peak resonance appear at fp,

while as the load increases the peak resonance moves to frequencies closer to f0.The design process of the resonant converter is based on the flowchart shown in Fig. 7. The most crucialdesign constraint deals with assuring the HV isolation between the primary winding of the T/Fs and thecores. Along with this, a proper range for the switching frequencies and a suitable value for the capacitorCr must also be chosen. In particular, these two are the only independent parameters which can be tunedso that the system will operate within the design requirements Taking into account the flowchart in Fig. 7,the HV isolation requirements count as the starting design criterion which dictates the core selection (i.e.core size) and the value of the leakage and magnetizing inductances (Lσ and LM). Based on these designinputs and also considering the input voltage of the resonant converter, Vdc and the constraints for theoutput voltage, Vout and the supplied power to each rectifier unit, Psm, the resonant capacitor Cr and theswitching frequency, fsw can be properly chosen. In addition to this, the turns-ratio, n of the T/Fs mightalso be adjusted if it is necessary. All in all, the three independent design parameters that can be adjustedare the resonant capacitor, Cr, the switching frequency, fsw and the turns-ratio, n.A further crucial design requirement is associated with the power losses generated in the cores. Theselosses depend on the core size and material, the operating switching frequency and the magnetic fluxdensity, B. In this specific design case, it is only the two latter parameters that can be properly adjusted.However, the choice of fsw is mainly governed by the desired performance of the resonant tank. It is,therefore, the reduction of B which can result in lower core losses. Considering the space limitationin terms of height in the rectifier units, B that corresponds to a specific operating point can be reducedby stacking two ring cores in each rectifier unit. In addition to this, the voltage across the leakage andmagnetizing inductances are governed by the operating point of the resonant converter, so that ILM canonly be reduced by means of increasing LM. This is basically the reason that the primary winding consistsof two turns. Along with this, lower magnetizing current is also associated with lower reactive powerdrawn from the power source. The values for LM and Lσ, as those have been measured on the examinedrectifier units geometry, are shown in Table II.Considering the values for LM and Lσ, several iterations using different values of Cr, fsw and n have beenperformed in order to reach an output voltage, Vout , having the lowest possible variations as the outputload is changed. This counts as the last design constraint shown in Fig. 7. The design parameters of theresonant tank are summarized in Table II. It must be noted that for the calculations a separate resonanttank is assumed to be connected on each phase-leg of the M2C. In other words, only N=30 rectifier unitsare supplied by each separate resonant tank. Furthermore, in order to meet the operating voltage andcapacitance requirements of the resonant capacitor, Cr consists of 3 parallel-connecting branches of 9series-connected single capacitors.

Figure 7: Flowchart shows the design procedure of the auxiliarypower supply.

The possibility to employ a commonresonant tank for the whole three-phasesystem was also investigated. As-suming that the primary windings areseries-connected, the leakage induc-tance becomes three times higher com-pared to the single-phase configuration.Thus, the corresponding value of Crmust be lower (a factor of 3) in orderto obtain the same resonant frequencyand the range of fsw must also prop-erly be adjusted in order to achieve thedesired operation. Moreover, the turns-ratio might also need to be properly ad-justed in order to obtain the requiredoutput voltage and power.An additional design requirement is as-sociated with supplying a constant out-put voltage, Vout regardless of the deliv-ered power, Psm. As shown in Table II,the expected operating range of the sys-tem in terms of supplied power on eachrectifier unit is Psm = 27.5 ± 10% W.Considering a fixed value for fsw, a variation in the output voltage, as Psm varies, is expected. In particu-lar, as Psm is reduced, an slightly higher Vout is expected. As will be examined in the following paragraph,the worst-case over-voltage across the output of a rectifier unit is expected when an open-circuit appears.When a SM of the M2C has a reduced auxiliary power consumption due to a failure, the output ofthe rectifier unit supplying this SM will be subjected to an over-voltage. Considering N rectifier units,an equivalent circuit diagram under a worst-case open-circuit (OC) condition of a single rectifier unit isshown in Fig. 8. Under the assumption of an OC on the output of one rectifier unit, the design parametersshown in Table II and N=30 the simulated output voltages, Vout , of a healthy rectifier unit and the OC

(N-1)LM

(N-1)Lσ

Lσ

LM

Lσ

A

B

(N-1)Lσ

(N-1)Rac

DZ Tcl

RZ

Figure 8: Schematic diagram of the resonant tank under a worst-case open-circuit of a single rectifier unit. Theclamping circuit for over-voltages is also shown.

(a) (b)

Figure 9: (a) Simulation results of the full system containing 30 rectifier units under an open-circuit condition ofone rectifier unit without any clamping circuit on the outputs and (b) simulation results showing the operation ofthe clamping circuit against over-voltages.

unit are shown in Fig. 9a. As expected, an over-voltage appears across the OC rectifier unit. The lowersubplot in Fig. 9a illustrates the voltages across the primary windings of the T/Fs which correspond tothe waveforms shown in the upper subplot. It is clear that the primary voltage across the OC rectifierincreases, while the primary voltages of the healthy rectifier units are slightly reduced. The reason isthat the sum of the voltages across the primary windings of the T/Fs is constant for a specific switchingfrequency.Therefore, an over-voltage protection circuit is required on the output of each rectifier unit in order toensure a safe operation of the loads. Such a circuit might consists of a thyristor which is controlledby a zener diode having a reverse breakdown voltage slightly lower than the clamped output voltage,Vout,cl = 28 V (Table II). The circuit diagram of the over-voltage protection scheme is shown in Fig. 8.When the thyristor is triggered, the output voltage of the OC rectifier unit in that circuit, which is detectedby the main controller, resulting in a system shut down. In the upper subplot of Fig. 9b, the simulatedclamped output voltage is shown.Potential tolerances in the parameters of the T/Fs (i.e. Lm and Lr) should also be taken into accountduring the design process of the system. Considering a tolerance of 10% in Lm and Lσ of the T/Fs,simulations of the full system at the rated operating conditions were performed in order to evaluate anychanges in the output voltage. In particular, Lm of one T/F is reduced by 10% compared to the valuegiven in Table II, while Lm of the rest 29 T/Fs is increased by 10%. Thus, the extreme case in terms oftolerances in Lm is investigated. A voltage difference of approximately 8.3% was obtained between theoutput voltages of the 29 units (Vout = 14.26 V) and the single rectifier unit (Vout = 13.1 V).

6 Experimental ResultsIn order to evaluate the performance of the presented auxiliary power supply, a down-scaled laboratoryprototype was constructed. It basically consists of the full-bridge resonant converter shown in Fig. 10awhich supplies three rectifier units as they are illustrated in Fig. 10b. In Fig. 10b the field-shaping tubesof the primary winding are also shown. The parameters of the full-scale system shown in Table II havebeen adapted to those shown in Table III. Thus, the operation of the down-scaled prototype in terms ofoutput voltage, switching frequency and delivered power will be identical to the expected operation of the

(a) (b)

Figure 10: Picture of the (a) converter laboratory prototype and (b) the down-scaled rectifier units consisting ofthree T/Fs and field-shaping tubes with a diameter of 10 mm, which are replaced by tubes with 20 mm in the finalsetup.

full system containing 30 rectifier units. Apart from the input voltage, Vdc, which was reduced to 10% ofthe rated value, also the capacitor of the resonant tank, Cr, was scaled to match the full system operation.In particular, Cr increased by ten times in order to meet the desired performance of the down-scaledexperimental system.

Table III: Parameters of the down-scaled auxiliary supply consisting of 3 rectifier units.

Input voltage, Vdc 40 VOutput voltage, Vout 14 V

Rated power of each rectifier unit 30 WClamped output voltage, Vout,cl 28 V

Range of the supplied power, ∆P 27.5 W ±10%Switching frequency 45-55 kHz

Resonant capacitor, Cr 220 nFTurns-ratio of the transformer 2:2

Total leakage inductance per rectifier unit, 2∗Lσ,1,2,3 1.32 µHMagnetizing inductance per rectifier unit, LM,1,2,3 20 µH

Ring cores per rectifier unit 2 x Kasche R102/65.8/15, K2006

Measurements of the down-scaled experimental system employing three rectifier units were performed atvarious loads on the output of the rectifiers and a wide range of switching frequencies, fsw. Thus, severaloutput voltage, Vout and supplied power characteristics were obtained. Based on this data, the variation ofVout at various supplied power levels for given frequencies are shown in Fig. 11. As already mentionedabove, a desired operation of the system is associated with supplying a flat output voltage while thedelivered power is varying at Psm = 27.5± 10% W and the switching frequency is kept constant. Withthe presented design and at a switching frequency of fsw = 53.52 kHz, Vout is varying between 13-16 Vas shown in Fig. 11.

Partial discharge measurementsFor validating the isolation design, partial discharge (PD) measurements were performed on a rectifierunit with potting. Fig. 12 shows the experimental setup for the PD measurements. The field-shaping tubehaving an outer diameter of 20 mm was fastened in the center of the ring cores, which will also be thecase in the real setup employing 30 rectifier units. By applying a peak voltage of 22.5 kV and using theOmicron MPD600 measurement equipment, PDs were measured to be lower than 2 pC. It must also benoted that the background partial discharge is lower than 1 pC.

Figure 11: Measured output voltage of a single rectifier unit as a function of the supplied power for variousswitching frequencies.

Figure 12: Picture of the partial-discharges measurement setup.

7 ConclusionsA modularized auxiliary power supply suitable for energizing the drive and control circuits of modu-larized converters or high-voltage series connected switches is studied in terms of electrical and high-voltage isolation designs. A resonant converter and rectifier units on the output consisting of ring-core-based transformers, count as vital parts for supplying floated voltages to the SMs. It is shown that aspecial design effort must be made on the rectifier units in order to comply, on the one hand, with therequired electrical performance, while on the other hand with the high-voltage isolation requirements.From 3D FEM simulations, as well as measurements using an impedance analyzer, the leakage and mag-netizing inductances in the transformers of the rectifier units were estimated and measured, respectively,and based on these, the resonant tank was designed. Additionally, from the distribution of the electricfield in the rectifier units, and by using a metallic grounded shielding of the primary winding, it is shownthat for certain core and field-shaping tube geometries, high-voltage isolation with only air can be as-sured. A worst-case peak electric field of approximately 2.05 MV/m was expected in the air between thefield-shaping tube and the ring cores.Experiments performed on a down-scaled prototype consisting of three rectifier units show that usinga constant switching frequency of fsw = 53.52 kHz, the rectifier units can supply power in the rangeof Psm = 27.5± 10% W with a variation on the output voltage in the range of 13-16 V. The sensitivityof the supplied loads at over-voltages must be carefully studied before choosing the specific operatingconditions of the auxiliary power supply. However, a clamp circuit can be employed in order to clamppotential over-voltages on the rectifiers outputs. Last but not least, from partial-discharge measurementsat the maximum peak voltage of 22.5 kV, partial discharges were found to be lower than 2 pC, which arenot destructive for the transformers.

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