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L\- s.- Brunel Technical College VOL: 1 .;3; "

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Department of Aeronautical Studies SECT 2 - . CHAP 7

CHAPTER 7

m o ' Frequency Amplifiers

INTRODUCTION

d t~ q l i f y s p e e ~ h frequencies, the upper audible l i m i t being about 18 kHz. I n airborne communication systems, bandwidths a re generally more r e s t r i c t ed than t h i s i n order t o reduce noise, Audio frequency amplifiers are used i n both t ransmit ters and receivers. In the first case the purpose is t o supply audio power su f f i c i en t t o modulate the radio frequency carp ier wave, I n the second case the purpose is t o supply su f f i c i en t power t o operate some form of transducer e.g. headphones o r loud- speakers.

2. I n most cases more than one stage of amplification is required so t h a t one o r more small signal pre-amplifier o r d r ive r stages precede the f i n a l stage. This f i n a l s tage i s always a power amplifier operating under la rge signal conditions which means tha t a cer tain amount of d is tor t ion i s inevi table , Distortion can be minimised by good design and special c i r c u i t techniques, par t icu lar ly the use of negative feedback, described i n Chapter 8.

Temperature Stabi l i sa t ion .

(a) Unstabilised Circuit . (b) S tabi l

FIG.1. Temperature Stabi l i sa t ion

T- S r b N c ) l

o Gt-rPu t I=, 1 + +

72 -0 =c

-

u

ised Circui t .

VOL 1 a. - - P SECT 2 '1 CHAP 7

3 . ~ i g . l ( a ) shows a simple small signal audio frequency amplifier 1 - -I m* -

s ULWLLSS~U 111 m a p e r 3 . lvote tna t two separate d. c. b ias supplies a re required, VCc and V~B,

4. The c i r c u i t of FYg.l(a) is prone t o variat ions i n performance due t o temperature change. Consider, f o r example, a r i s e i n ambient temperature which causes an increase i n the t r ans i s to r leakage current. A s the leakage current forms a p a r t (although a very small p a r t ) of the t o t a l co l lec tor current, t h i s a l so increases. The r i s e i n col lector current r e su l t s i n an I S n L, he power dissipated i n the t r a n s i s t o r and hence there w i l l be a fur ther r i s e i n temperature a t the collector- base Junction. Thus t h i s is an accumulating e f f e c t and a small increase i n ambient temperature can r e s u l t i n a s igni f icant change i n col lec tor current. ~t best t h i s means a change i n the working point, possibly resul t ing i n reouced gain and/or increased d is tor t ion . A t worst, the increased power d iss ipa t ion i n the 7 t r ans i s to r can destroy the device.

5 The c i r c u i t of Fig. l (b) is a f a r more prac t ica l c i r c u i t . It only requires one doc . supply, Vcc , the base b ias being obtained by means of the potent ial divider rl and r The 2. design of the c i r c u i t a l so reduces the e f fec t s of temperature fluctuations, par t icu lar ly by the inclusion of re.

Action of re and ce The doc . b ias between the base and the emitter i s equal t o the vol t s drop across r minus the vol t s drop across r :- 2

e

Base - h i t t e r d o c . voltage = I r 2 2 - 'ere

i n practice I r w i l l need t o be s l i g h t l y greater than Iere. If 2 2 ' . t

now temperature increases, t o t a l t r ans i s to r current increases and 1 I increases. But the increase i n Ie resu l t s i n an increase i n e

I r and a reduction i n the base-emitter forward b ias - thus e e

* causing a reduction i n t r ans i s to r current. The c i r c u i t therefope has a s t ab i l i s ing action, with the d.c. currents being reasonably independent of temperature. The capacitor ce performs a s imilar function t o the decoupling capacitor i n a valve cathode bias c i r cu i t . Its reactance i s such tha t it is ef fec t ive ly a short c i r c u i t f o r a l l signal frequencies.

7. Typical values f o r the components of F ig . l (b) a re :- r L 5 kilohms r e

1 kilohm

'e loo p F

r 1 80 kilohms r 2 10 kilohms

---- - -- --- . - --

. - ,, p- - "3.*&,

VOL 1 . ".bn " ";& ' SECT 2 7.5 1'6

7 8. I n pract ice addi t ional measures may be taken t o achieve

- a -gigher aegree o r iemperature s t ab i i i s a t ion , using ex t r a components such as diodes o r thermistors.

Coupling Between Stages.

R-C Coupling. I f more than one stage of amplification is required, the output of one s tage must be applied t o the input of t he next stage. Two stages of the amplif ier of ~ i g . l ( b ) may be coupled a s shown i n Fig.2,

If the co l l ec to r of T1 was d i r e c t l y coupled t o the base of T 2 the l a t t e r would be incor rec t ly biased. The d o c . blocking capacitor C is therefore inser ted t o i s o l a t e T base from T 2 1 col lec tor , The value of C i s chosen so t h a t i ts reactance is negl igible a t al l s ign i f i can t s ignal frequencies. For audio frequency applications, C needs t o be a t leas* 1 0 p F .

10. Transformer Coupling. Transformers, although bulky and r e l a t ive ly expensive, automatically give d o c . i s o l a t i o n between primary and secondary. They a r e used i n a . f , s tages where it is necessary t o match the output impedance of the first s tage t o the input impedance of the second stage, o r where it is necessary t o obtain a pa r t i cu la r shape of response curve. Transformer coupling is i l l u s t r a t e d i n Fig.3.

VOL 1 SECT 2 CHAP 7 ,' \

FIG.3. Transformer Coupling.

Direct ( d o c ) Coupling. This method of coupling saves on the number of components and can r e s u l t i n higher gains and b e t t e r thermal s t a b i l i t y than other coupling methods. ~ i g . 4 shows some typica l methods of achieving d i r e c t coupling. ( a ) In Fig.4(a), the f i r s t s tage is i n common col lec tor (emit ter follower) configuration, the output being taken

from the emitter. This enables the cor rec t d o c . bias t o be applied t o the base of the second stage.

(b) Fige4(b) i l l u s t r a t e s the use of a PNP-NPN pai r , the co l lec tor current of the first t r a n s i s t o r becoming the base current of the second t r ans i s to r .

( c ) ~ i g . 4 ( c ) i l l u s t r a t e s a very common cal led a " ~ a r l i n g t o n pair".

form of configuration

- "'-4 - - - ---.

- - - - --;* - +.- .

VOL 1 i * Z .y?f f v *-r,-, . I)

SECT 2 CHAP 7

Frequency Response

12. The normal frequency response of an audio-frequency amplif ier is shown i n Fig.5. The useful l i m i t s of operation are generally regarded as fl and f2, the frequencies a t which the gain has dropped by 3 dBs compared with mid-band.

FIG. 5. Gain - Freauencs Res~onse.

13. I n a high qua l i ty audio amplifier, f may be a s low a s a few Hz 1 and f is the upper l i m i t of the audlble range i.e. about 18 kHz. 2 I n a communications receiver, where f i d e l i t y ( i . e . freedom from d i s to r t ion ) i s not so important, and where bandwidth requires t o be s m a l l t o reduce the e f f ec t s of noise, f l may be about 300 Hz and f2 about 3 kHz.

14. Causes of Fall-Off i n Gain. Both the ac t ive device and its associated c i r c u i t components cause the reduction i n gain a t high and low frequencies, and t h i s is i l l u s t r a t e d i n Fig.6.

F I G . ~ . Effects of Coupling and Input Capacitances.

I n t h i s diagram the s ignal source (which may be the output of a preceding amplif ier) is applied t o the amplif ier input v i a a coupling capacitor Cc. The amplifier input impedance cons is t s of a res is tance r i n pa ra l l e l with a Capacitor Cin. Cin i s in' bas ica l ly the emitter-base junction capacitance but possibly increased due t o Miller Effect.

VOL 1 SECT 2 CHAP 7 \

15. High Frequency Gain Reduction. A s frequency increases X,. -&A*

decreases and a smaller proportion of v is developed across the amplif ier input terminals. In the l i m i t XCin is zero and the input is short c i rcu i ted . It is simple t o design t r a n s i s t o r s t o operate up t o the a.f. l i m i t but t r ans i s to r s required t o operate a t higher frequencies must have su f f i c i en t ly low values of C i n .

16. Low Frequency Gain Reduction. A t the lower audio frequencies, in

1- uec i n relevant. However, as frequency decreases, X increases. A g rea te r proportion of V s is developed across Eg and a smaller proportion across r . In the l i m i t , a t zero frequency, a l l of Vs w i l l be develope?inacross Cc, none across the amplif ier input terminals. Note t h a t i n the case of d i r e c t coupling, operation is theore t ica l ly possible a t frequencies approaching zero.

Class A Operation.

I,

FIG. 7. Class A Bias.

17. Class A Operation i s when the b ias i s such t h a t the input signal swing does not take the ac t ive device e i t h e r i n t o sa tura t ion or beyond cut-off. Severe d i s to r t ion w i l l r e s u l t i f these l i m i t s a re exceeded i n the type of amplifier so f a r discussed. If very la rge s ignal swings a r e t o be accommodated, the b ias point should be approximately half way along the load l ine , a s shown i n Fig.7.

18. Power Dissipation. I n the case of Fig.7, the co l l ec to r d.c. current and voltage a r e 2mA and 5 v o l t s respectively. Thus the power supplied t o the t rans is tor , i r respec t ive of the magnitude of the s ignal is :

. " . ~ *

VOL 1 SECT 2 CHAP 7

Note t h a t under no-signal conditions, t h i s l O m W must be dissipated, i n the form of heat, i n the t r ans i s to r .

19. Maximum Power Dissipation. Every t r a n s i s t o r has a maximum safe 9 l i m i t f o r power diss ipat ion. This l i m i t is determined by the design

of the t r a n s i s t o r and the ease by which the excess heat can be diss ipated. Most power t r a n s i s t o r s a re mounted on a "heat sink" which conducts heat away from the collector-base Junction. Every device has a "maximum col lec tor dissipation", which, i n the example of Fig.7, i s 10mW. The PC curve shown i s the locus of points

1 where the product IcVc = l h W . It is therefore e s sen t i a l t h a t the / t r a n s i s t o r is not biased above the curve, i n the shaded area. 20. Conversion Efficiency. In the case of Fig.7, i f the s ignal

amplitude is zero, there is no s ignal power output and a l l the power from the supply is dissipated i n the device. The m a x i m u m

r power output occurs when the s ignal swings over the whole length .. .. . ..

. ..-

- : :.: . ; of the load l ine , then :-

-, . . -.

Conversion eff ic iency = Power out = - 5mW = 0.5 or 50% dc power i n l O m W

and t h i s is the maximum possible value i n a Class A stage.

Simple Class A Power Amplifier.

FIG.^. Single h d e d Class A Power Amplifier.

21. A simple power amplifier output stage, transformer coupled t o a loudspeaker is shown i n ~ i g . 8 . The transformer is commonly used t o convert a low loudspeaker impedance t o a higher value t o match the t rans is tor .

~ ..

VOL 1 SECT 2 cwip 7 /-?

Example: Suppose the loudspeaker impedance Zs is r e s i s t i v e and e q u a l t o 25 ohms. O p t i m u m load impedance f o r the t r a n s i s t o r Z 2 . 5 kilohrns (As shown i n Fig.7). The turns r a t i o required i8:-

22. Although, i n t h i s case, the ac (s ignal) co l l ec to r load is 2.5 kilohms, the dc co l l ec to r load is zero (assuming the dc resis tance

voltage is therefore equal t o Vcc (= 5 v o l t s i n t h i s case) and when a s ignal i s applied, the instantaneous co l l ec to r voltage swings above and below t h i s value (see Fig.7 and 8 ) .

Class A Push Pull (Double-Ihded)

F I G . 9. Push Pull Amplifier.

23. The c i r c u i t shown i n Fig.9 employs two t r ans i s to r s therefore giving twice the power output of a s ingle ended s tage but with ce r t a in other advantages i n addition. The important fea ture of the push-pull c i r c u i t is i t s symmetry, the two t rans is tors , TI and T2, have ident ica l cha rac t e r i s t i c s and the transformers a r e accurately centre- tapped.

24. The inputs t o T1 and T2 a r e equal i n magnitude but i n antiphase (achieved i n Fig.9 by the centre-tapped input transformer although there a r e other methods). Their outputs w i l l a l s o be i n an t i - phase, so i f i is increasing, say, ic2 w i l l be decreasing.

C1 However a s i and i a r e flowing i n opposite d i rec t ions i n the

C1 C2 transformer primary, they produce inphase voltages i n the output transformer secondary.

, y-%- VOL 1 .:%. ; X SECT 2 CHAP 7

25. Reduction i n Second Harmonic Distortion. Because of the uneven +-..+ +LC. m-4-nn .1 r r - I - 4 nrr 1 e n m n c+ - Q ~ C be cryur, c , r l r ; ~ & b u b ~ ~ * > u ~ b u , ureAru W

d'istoried (see Chap. 5 para.48). A s i n e wave output w i l l produce outputs a t the fundamental frequency ( the undis tor ted portion) plus unwanted harmonics o r overtones, pa r t i cu la r ly second and th i rd . Fig.10 shows outputs from T1 and T2, d i s t o r t e d due t o the presence of the second harmonic. When "added" i n the output transformer secondary, the second harmonics cancel.

FIG.10. Cancellation of Second Harmonics.

A Class A push-pull amplifier can therefore produce twice the power output of a s ing le ended stage with a lower l eve l of d i s tor t ion ; o r more than twice the output power a t the same l e v e l of d i s tor t ion .

26. Other Advantages of Push-Pull Operation. ( a ) A s icl is equal and i n antiphase t o iC2, no s ignal current

flows through the power supply c i r c u i t . This s impl i f ies power supply design and reduces the l eve l of supply r ipple . Also no decoupling capacitor is required f o r R,.

VOL 1 .-. - SECT 2 - CHAP 7

(b) The dc co l l ec to r current flows i n opposite d i r ec t ions t-gh+hp-+.ransfr\rmpy ~ r i = r ~ s +,!xc t h e magnetisation is zero. This enables a smaller output transformer t o be used as there is, i n theory, no poss ib i l i t y of magnetic saturat ion.

27. Integrated Ci rcu i t s f p r Push-Pull Operation. Many "operational amplifier" type integrated c i r c u i t s a re designed t o produce two outputs, equal i n magnitude but i n antiphase. A s ing le integrated c i r c u i t is therefore commonly used instead of the two separate

Class B Push-Pull.

FIG. 11. Class B Push-Pull Operation

28. I n c lass B operation, the act ive device i s biased approximately a t cut-off, and so conducts f o r only half the input cycle. The r e s u l t i n a s ingle ended audio frequency s tage would be unacceptable d is tor t ion . In a push-pull configuration howevqr, each device -\ operates on a l t e rna te half-cycles, the one " f i l l i n g i n the gaps" missed by the other. In pract ice the devioes a re of ten biased s l i g h t l y forward of cut-off t o minimise cross-over d i s to r t ion . e

29. Conversion Efficiency. c u n o a L a s - I

(~oJLR & UV(tCL(3e $uPf&Y CUM

-- -- p u w t

/ *c

FIG.12. Power Supply Current i n a Class B Push-Pull Amplifier.

VOL 1 I . " SECT 2 j I CHAP 7

From F'ig.12 it may be seen that the average current drain from the power supply-i-sA=s 2.c - vaLue+iM.s fn a Class A amplifier working at optimum efficiency, the current drain is equal to the peak a.c. This means that the conversion efficiency of a Class B amplifier is higher than that of a Class A and it can be shown that its maximum theoretical value is 78.5s. Furthermore, under no signal conditions the power supply current is zero.

Push Pull Amplification with Complementary Pairs.

The requirement to provide a phase splitter input i.e. equal magnitude but opposite phase is avoided by the use of a complementary pair of transistors. This is a matched pair, having identical characteristics, but one of the transistors is PNP, the other NPN. A simple circuit employing this principle is shown in Fig.13. Note that if the input signal is "going , negative" say, ic will be increasing and ic2 will be decreasing - that is the outpu%s are in push-pull.

FIG.13. Push Pull Amplification using a Complementary Pair. F - 3 e 4-44 s;OCLd 4 ,4i t3L & -y+* w-

BRUNEL TECHNICAL COLLEGE, BRISTOL . DATE: JUNE-' 1983 .