CACOPY - NASA...demodulation, and analog computation tech-niques. Linear product modulation is...

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TECHNOLOGY UTILIZATION

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COPY

ELECTRONIC CIRCUITSFOR

COMMUNICATIONS SYSTEMS

A COMPILATION

NATIONAL AERONAUTICS AND SPACE ADMINISTRATION

https://ntrs.nasa.gov/search.jsp?R=19720011636 2020-03-23T14:51:44+00:00Z

ForewordThe National Aeronautics and Space Administrat ion and Atomic Energy

Commission have established a Technology Util ization Program for the rapiddissemination of information on technological developments which havepotential util i ty outside the aerospace and nuclear communities. By encouragingmultiple application of the results of their research and development, NASA andAEC earn for the public an increased return on the investment in aerospaceresearch and development programs.

The compilation of electronic circuits for communications systems has beendivided into thirteen basic categories, each representing a unique .area of circuitdesign and application. The circuits and modular subassemblies are onlysamples of many similar items that are available through the TU program. Theenormous volume of information available in this fertile area is indicative ofthe important role that communications technology has played in the aerospaceprogram.

In general, the compilation items are moderately complex and as such,would appeal to the applications engineer. However, the rationale for the selec-tion criteria was tailored so that the circuits would reflect fundamenta l designprinciples and applications, with an additional requirement for simplicity when-ever possible.

Additional technical informat ion on indiv idual devices and techniques canbe requested by circling the appropriate number on the Reader Service Cardincluded in this compilation.

Unless otherwise'stated, NASA and AEC contemplate no patent action onthe. technology described.

We appreciate comment by readers and welcome hearing about the rele-vance and u t i l i ty of the informat ion in this compilation.

Jeffrey T. Hamil ton, DirectorTechnology Utili:alion OfficeNational Aeronautics and Space Administration

NOTICE • This document was prepared under the sponsorship of the National Aeronautics and SpaceAdministration. Neither the United States Government nor any person acting on behalf of the UnitedStates Government assumes any liability resulting from the use of the information contained in thisdocument, or warrants that such use will be free from privately owned rights.

For sale by the National Technical Information Service, Springfield, Virginia 22151. $1.00

C ' o n l c i , ; s

SECTION!. Modu la t ion TechniquesDouble-Emit ter , Suppressed-Carrier Modula tor 1Remodula tor Filter 1Resistance-Controlled Linear Product Modula tor 2Ferrite Antenna Modulator 2

SECTION 2. Frequency ControlFrequency Converter with High Signal-to-Noise Ratio 3Frequency Mul t ip l i e r for Remote Operation 4Varactor Frequency Divider 5Frequency Doubler with Negligible Dis tor t ion 5

SECTION 3. FM Design ConsiderationsCircui t Reduces Distortion of FM Modula tor 6Frequency Offset in Linear FM/CW Transponder

El iminates Clut ter 7Opt imum FM Preemphasis: A Concept 7

SECTION 4. Fil tersA Self-Tuning Filter 8Composite Fi l ter Steepens Rejection Slopes

in Microwave Appl ica t ions 9Tunable Bandpass Filter wi th Variable Selectivity s)

SECTION 5. Phase DetectorsCont inuous ly Variable, Voltage-Controlled Phase Shif ter 10Phase Detector Synthesizes Own Reference Signal 11Phase Detector Has High Eff ic iency in Presence of Noise 12

SECTION 6. Gain and Frequency ControlWide-Range Automat ic Gain Control Circui t 12Simple, Accurate Automat ic Frequency Control Circuit 13Gain Control Limiter Circui t 14

SECTION?. RF OscillatorsLow-Power, Compact, Voltage-Controlled Oscillator 14Automat i c Frequency Control of Voltage-Controlled

Oscillators 15A 225 MHz FM Oscillator wi th Response to 10 MHz . 16Voltage-Controlled Oscillator 16

SECTIONS. RF Ampl i f i e r sVariable Gain A m p l i f i e r 17Low-Noise, Wide-Bandwidth l.F. Preampl i f ie r 18Complementary Pair Broadband Transistor A m p l i f i e r 19Gain and Phase-Tracking A m p l i f i e r 20

SECTION9. RF Measurement TechniquesDynamic Lineari ty Measurement Technique 20Simpl i f ied Method for Measur ing the Impedance

of RF Power Sources 21

Linear Signal-Noise Summer Accurately Determinesand Controls S/N Ratio 21

SECTION 10.. Personal Communications SystemsSelf-Contained Minia ture Electronics Transceiver

Provides Voice Communication in Hazardous Env i ronmen t . . . 22Personal Communications Assembly - 23Audio Signal Processor 24

SECTION 11. Phase-Locked Loop TechniquesHigh Noise Immuni ty , Wideband Bi-Phase Modulat ion

Scheme 25Phase-Lock Loop Phase Modulator with High Modulat ion

Index, Low Distortion 25RE Receiving System with Improved Phase-Lock

Characteristics -. : . 26

SECTION 12. Video Circuits for TV ApplicationsVariable Word-Length Encoder Reduces TV Bandwidth

Requirements , 27Mul t ip lex Television Transmission System . 27TV Synchronization System Features Stability

and Noise I m m u n i t y 28

SECTION 13. MixersAdded Diodes Increases Output of Balanced Mixer Circuit . . . . 29Compact Microwave Mixer,Has,High Conversion Efficiency . . . . 29Signal Mixer Provides Outputs with Matched Impedance

Characteristics 30

Section 1. Modulat ion Techniques

DOUBLE-EMITTER, SUPPRESSED-CARRIER MODULATOR

This suppressed carrier modulator develops asignal-to-carrier output ratio of 40 dB or greaterwith a signal input of 2.5 V peak-to-peak. Notun ing potentiometers are required wi th in the

frequency range from 450 to 520 kHz, and allsideband harmonics are less than the carrier out-put amplitude. A double-emitter, chopper tran-sistor, Ql, achieves the required carriersuppression by reducing the carrier output froma modulator; this reduction takes place in thebridge circuit which balances the effects of the

carrier current as it switches the double-emittertransistor from cutoff to saturation. Voltagesdeveloped by the circuit are balanced, with re-spect to the load impedance and ground, bytying the collector of Ql and the base-trans-former combination to separate resistor bridges.The voltage developed across the emitter El tothe base junction B is equal and opposite to thevoltage developed across emitter E2 to the basejunction. Therefore, the sum of the emittervoltages is approximately zero with respect tothe load impedance. Any stray capacitance be-tween the collector and either emitter, or thebase and either emitter, is also nulled in thebalanced RC bridge network.

Source: C.F. Haist and A. Piscopo ofIBM Corp.

under contract toMarshall Space Flight Center

(MFS-12494)

Circle 1 on Reader Service Card

REMODULATOR FILTER

Three s imul taneous func t ions—demodula t ion ,f i l t e r i n g , and modulation—.are performed in asingle func t iona l remoduiator circuit which el im-inates spurious signal components at the output .The advantages are extreme simplicity, freedomfrom mismatch problems, and the ability to usepolarized capacitors which tend to be smaller forthe larger capacitance values used in other cir-cuits of this type.

The operation of the circuit is as follows:with the switch in position 1, a small positivevoltage builds up on Cl d u r i n g the positive halfof the input cycle, wi t . and appears at the out-put . During the next half cycle with the switchin position 2 and with a switching rate of u^t,C2 acquires a negative charge which also ap-

V i n

C1Vout

JC2

HlH T NHpears at the output . On succeeding half cycles,charges accumulate on the capacitors and theoutput rises accordingly. If wit and o>2t aresynchronous and in-phase, the envelope am-plitude of the resulting output signal exponen-t ial ly reaches the steady state amplitude of theinput signal. Upon removal of the input signal,

I

2 ELECTRONIC CIRCUITS FOR COMMUNICATIONS SYSTEMS

the discharge of the capacitors produces anexponential envelope decay, mirroring the inputbui ldup pattern. From the waveforms shown inthe figure, it can be seen that the input signal,when processed through a remodulator, willappear at the output without spurious signalcomponents when wit is synchronous and inphase with W2t.

Source: H. C. Vivian ofCaltech/JPL

under contract toNASA Pasadena Office

(NPO-10198)


RESISTANCE-CONTROLLED L I N E A R PRODUCT MODULATOR

A product modulator with a suppressed car-rier (see fig.) consists of a linear ampli tudemodulator that requires only an operational am-plif ier and a pair of field-effect transistors

Modulator

S 1 c V1fr Qvs ]Vc' \ '

<R2

1

(FET's). No inductors or transformers are re-quired. Ampl i tude modulat ion is produced byvarying the impedance of the FET'si, which, ineffect, control the closed-loop gain of t h e - a m -plifier. The carrier signal Vc is suppressed byuti l izing the common-mode rejection capabilitiesof the amplifier. Therefore, a very close approxi-mation to a true product modulator is obtainedwhen only the sum and difference frequencies ofthe two input frequencies Vs and Vc appear at

the output. Also, by adjusting the amount ofcommon-mode rejection, various degrees ofcarrier signal rejections are possible. The prod-uct modulator may be used in such applicationsas linear frequency translation, modulation anddemodulation techniques such as single sidebandAM, double sideband AM, with or wi thoutsuppressed carrier, quadrature modulation anddemodulation, and analog computation tech-niques.

Linear product modulation is equivalent tolinear amplitude modulation with a suppressedcarrier signal. Since the amount of common-mode rejection is adjustable, ampli tude modu-lation can be obtained with either full-carrieramplitude, reduced carrier, or suppressedcarrier. With suppressed carrier, the modulatoralso funct ions as an excellent frequency trans-lator in which a range of frequencies is shiftedto another frequency band.

Source: H. Brey and S. Gussow ofSperry Rand Corp.


(MFS-14391)


FERRITE ANTENNA MODULATOR

A mul t ip le antenna system located onboardan aircraft eliminates interference zones whichreduce the signal strength at the ground receiv-ing station dur ing critical f l igh t periods of takeoffand landing. The interference zones are reducedby inserting a ferrite attenuator into the appro-

priate an tenna waveguide and modula t ing theferrite attenuator to change the an tenna gainat the receive frequency. This permits groundtracking u n t i l the antenna is no longer required,at which time a fixed a t t enua t ion q u a n t i t y isinserted into the wavesuide.

MODULATION TECHNIQUES

VB+

The modulator schematic shown in the figureis composed of three sections: (1) a mult ivibra-tor, (2) an integrator, and (3) a driver for theferrite coil. The multivibrator is a free runningcircuit that establishes the frequency of themodulat ion wave. Q3 is a switch, in the Mil lerintegrator and is controlled by the mul t iv ibra toroutput. The charging and discharging of' thecapacitor between the base and collector of Q4determines the f i n a l output waveshape. Theemitter follower, Q5, drives the ferrite coil. Ablocking diode added to the emitter circuit

+ 29 Vdc

prevents feedback from the 29 Vdc applied tothe ferrite coil. R^, inserted in the emitter ofQ5, determines the amount of a t tenuat ion re-quired.

Source: S.G. Larson, F.H. Shorkley,J.C. Hooks, and B.T. Will iams of

Western Electric Company, Inc.under contract to

NASA Pasadena Office(NPO-12011)


Section 2. Frequency Control

FREQUENCY CONVERTER WITH HIGH SIGNAL-TO-NOISE RATIO

A novel frequency converter with two highlystable reference signals makes use of a verysimple phase detector circuit to el iminate theneed for .the complex and expensive amplif icat ionand f i l ter stages required in conventionalfrequency converters.

The reference frequency converter (see fig.) isbasically a conventional phase-lock loop con-figuration, except that a voltage-controlled

crystal oscillator (VCXO) replaces the moreconventional voltage-controlled oscillator (VCO).A VCXO is required to select the upper side-band (30 MHz + 256 Hz) and to reject thelower sideband (30 MHz - 256 Hz). For thispurpose, the frequency stabili ty of the VCXOmust be wi th in ±0.0008%.

The 256 Hz output of the mixer is ampl i f ied ,filtered by an active low-pass f i l ter , and applied

ELECTRONIC CIRCUITS FOR COMMUNICATIONS SYSTEMS

Reference Frequency Converter

To Mixer

to the phase detector which is a rsample-and-hold circuit. The sampling signal is a narrowpulse produced by a pulse generator from the256 Hz reference signal furnished by a stableoscillator. Each pulse turns on an FET in the

phase detector. .For this condit ion, the phasedetector output voltages assumes the value ofthe instantaneous input voltage. During the"off" stage, the holding capacitor dischargesvery slowly, thus main ta in ing the "holding" levelbetween sampling pulses. If the fixed intervalbetween sampling pulses coincides with theperiod of the sine wave input to the phase de-tector, the input wave is sampled at the samelevel each time, and the output level of the phasedetector does not change. If this is not thecase, an error voltage appears at the phasedetector output. This error voltage is appliedto the control input of the VCXO, causing thenecessary correction which brings the input ofthe phase detector into phase coincidence withthe 256 Hz sampling pulse train. The signals arenow phase-locked, and The output frequency isthe sum of the two input reference frequencies.

Source: G.B. Shelton ofSperry Rand Corp.


(MFS-14526)


FREQUENCY MULTIPLIER FOR REMOTE OPERATION

An oscillator frequency mult ipl ier can beadapted to harmonic mixer-receiver systems inwhich the mixer is located remotely from thereceiver. The remote oscillator frequency mult i-plier provides an increase in the sensitivity ofremote operated harmonic mixers and an operat-ing capability over a wider frequency range.The mixer is normally connected to the receiverby means of a single.coaxial cable, which con-ducts the local oscillator signal from the re-ceiver to the mixer, and the intermediatefrequency and crystal current from the mixer tothe receiver.

When the remote oscillator frequency mul t i -plier is used, the local oscillator frequency ismult ipl ied by a factor N, and the sensitivity ofthe mixer is improved because of the reductionof the harmonic mixing number. Also, the har-monic mixing number is reduced by mul t ip ly ingand f i l ter ing the local oscillator signal prior to

introducing this signal to the mixer diode. Thisprocess results in reduced mixer conversion loss.The remote oscillator frequency mul t ip l ie r isconnected at "A" and fed through the fre-

To —Receiver

r

A

BroadbandFrequencyMultiplier

FreqiSelecTee

C

encylive

High-Pass"Filter

B

IF and CrysCurrent to f

1D j To Mixer

tali/lixer*

j

quency selective tee to a broadband frequencymult ipl ier . The frequency selective tee isolatesthe B port from the local oscillator signal. Theoutput of the broadband frequency mul t ip l ie r isfiltered to remove the fundamenta l local oscil-lator signal and undesired harmonics; the mul t i -

FRE'QJUENCY CONTROL

plied local oscillator signal is available at portD for connection to the local oscillator inputport of a three-port mixer.

The intermediate frequency and crystal currentport of the three-port mixer is connected to port ,B of the remote local oscillator frequency mul-tiplier and these signals are coupled through thefrequency selective tee to port A of the remotelocal oscillator frequency multiplier. The fre-

quency selective tee isolates the IF signal andcrystal current from the broadband frequencymultiplier input C.

Source: C.W. Currie and W.H. Graham ofScientific-Atlanta, Inc.


(MFS-13291)

No further documentation in available.

VARACTOR FREQUENCY DIVIDERV5

cT

12 MHz

Z oOOi

/

/

ifK

Lr)f\

<

%L y

/

•

/N

M

V

jr vix )

^y

r- nrv -i

lry\

If11

/

86 MHz50 n

XB •!N /

n

/\

v _]

l)

Vniit

The varactor frequency divider shown in theschematic is designed to accept an input fre-quency of 172 MHz and to provide an outputfrequency of 86 MHz, with input and outputimpedances of 50 12. The varactor has a cutofffrequency of 22 GHz (well above the outputfrequency) and an output power of 1.5 W. Boththe input and output circuits of the divider usean "L" section matching network. This induct-ance, together with the variable capacitor, thequiescent point capacitance, and the reactanceof the tank circuits, forms a low-Q seriesresonant circuit at 172 MHz in the input and

at 86 MHz in the output circuit. The dividerhas a 3 dB bandwidth of 7 MHz and containsa very small harmonic content in the output.The varactor frequency divider offers a relativelysimple method of dividing frequencies withpassive elements.

Source: L.L. Vorel and M.A. Honnell ofAuburn University


(MFS-14345)


FREQUENCY DOUBLER WITH NEGLIBLE DISTORTION

Frequency mult ipl icat ion is achieved by a newcontrol circuit that locks a higher frequency to alower one. The input signal (see fig.) is appliedto the first stage Ql, a phase divider with two

outputs equal in magnitude but phase shiftedT rad (180°). These outputs are applied to thebases of emitter-followers Q2 and Q3 whichserve as sine wave shapers for driving the fre-


quency doubling stage Q4 and Q5. If Q4 andQ5 are matched, tne fundamental frequency ofthe input signals will cancel in the commondrain resistor Rl, and the second harmonicswill be added. R2 is used to make f ine adjust-ments for matching Q4 and Q5. Q6 is anemitter-follower output stage that provides alow output impedance. No filter circuits arenecessary, and the output waveform is sinus-oidal with neglible harmonic distortion within

the frequency range from 10 kHz to 10 MHz.Several of these doublers can be used in cas-cade if mult ipl icat ion factors greater than 2 arerequired.

Source: W.K. Wong ofLockheed Electronics Co.

under contract toManned Spacecraft Center

(MSC-13118)

No further documentation is available.

Section 3. FM Design Considerations

CIRCUIT REDUCES DISTORTION OF FM MODULATOR

This correction circuit reduces second har-monic and intermodulation distortion of avoltage-variable capacitor which modulates theoscillator. As shown in the schematic, the mod-ulating input signal is phase-inverted and ampli-fied by Ql and Q2. The amplified signal is thenapplied to the full-wave rectifier diodes Dl andD2 which generate the second harmonic withoutintroducing the fundamen ta l frequency com-ponent into the correction circui t . The outputfrom the rectifier-squaring network, applied tothe potentiometer R, provides a correction sig-

Voltage-VariableCapacitor

ModulatedInput

ModulatedSignal Output

DC Supply

-r*"* DC Supply

I'Voltage-Variable| Capacitor

FM DESIGN CONSIDERATIONS

nal to the voltage-variable capacitor C2 acrossthe tuned circuit of the oscillator. The correc-tion signal has the proper polarity to reducetotal tank circuit capacitance on both thepositive arid the negative peak swings of themodulat ing input signal.

Other applications for this correction circuitmight include its use in the master oscillator of

a radio transmitter or in a. subcarrier oscillatorof a telemetry system.

Source: Radio Corp. of Americaunder contract to

Goddard Space Flight Center(GSC-00257)

Circle 7 on Reader Service-Card

FREQUENCY OFFSET IN LINEAR FM/CW TRANSPONDERELIMINATES CLUTTER

Undesirable false echoes or reflections (clutter)of the signal used to interrogate a linearFM/CW transponder located on airborne ve-

ControlVoltage

TransponderI Signal

Coupler BalancedMixer

hides are eliminated by a technique which off-sets the frequency of the transponder signalwith respect to the interrogation signal.

As shown in the i l lustration, the interroga-tion signal from the master transmitter is radi-ated through a coupler to the airborne trans-ponder. The transponder receives this signal

through an offset control which generates a con-trol voltage by comparing the frequency of theinterrogation signal to that of the transpondersignal. This control voltage changes the fre-quency of the transponder signal so that itremains constantly offset from the frequency ofthe interrogation signal. At the master receiver,a portion of the interrogation signal is coupledto a balanced mixer. The transponder signalreceived by the master receiver is beat againstthe interrogation signal. The beat signal fre-quency is proportional to transponder range,provided the system is using an FM/CW alti-meter. Without frequency offset, the frequencyof the best signal in the master receiver is alsoa function of transponder time delay. Because ofthis, signals reflected from false targets whosetime delays are equal to that of the desiredsignals will be indistinguishable from the desiredsignals. These undesirable signals are eliminatedby offsetting the transponder frequency.

Source: Melpar, Inc.under contract to

Marshall Space FlightCenter

(MFS-00249)

Circle 8,011 Reader Service Card

OPTIMUM FM PREEMPHASIS: A CONCEPT

The spectral noise characteristics in the base-band of a receiver (without deemphasis) hasbeen measured as a Junction of rf input power.Experimental measurements of output noise vs

rf input power for the specific baseband centerfrequency are shown in the figure. A calibratedrf signal generator and precision rf attenuatorare connected to a wave analyzer (tunable Volt-


Base Band Signal Referenceat Se lected • Frequency.& Modulation Index

Thermal Level0

Signal Flow

F250 MHz Bandwidth K-lkiHz

No Deemphasis

Baseband Frequencies

30kHz"70kHz

100 kHz

SaturationLevel

120 110 100 90 80 70 60 50 40 30 20 10

meter). The wave analyzer measures the noise^in a small-frequency slot, centered at a specifiedbaseband frequency. The rf generator and fmreceiver are set to the operating carrier frequencyand the wave analyzer is then tuned to a fre-quency near the low end of the baseband fre-quency spectrum. As the rf signal generatoroutput power is varied in discrete incrementsover the dynamic range of the fm receiver, the

wave analyzer measures the noise in the selectedbaseband frequency slot for each discrete incre-ment of rf input power.

Source: K.W. MerzofThe Boeing Company

under contract toKennedy Space Center

(KSC-10151)

Circlet on Reader Service Card

Section 4. Filters

A SELF-TUNING FILTER

A self-tuning filter covers the range from 2kHz to 20 kHz, while maintaining constantbandwidth and center frequency gain. The

Capability of automatically adjusting centerfrequency to track the signal frequency permitsthe use of a filter with a bandwidth considerablysmaller than the range of input signal frequen-cies. The resistance-capacitance bandpass filteremployed in this design has a center frequencythat can be varied by-means o'f-a single.resistiveelement. An FET is used in place of an ordi-nary resistor to control the voltage of the tuningelement. The control voltage is derived from aphase detector which compares the phase of

Vjn

,n ir

1 Phaco n«ito^tr»r L

.^x^

+ /^ Vout

the filter input and output. If phase relation-ship between input and output is v rad (180°),the filter is tuned to the input frequency, if; lessthan TT rad, a correcting voltage, is applied to

FILTERS

the FET unt i l the phase is TT rad, at whichtime the fil ter is again in tune. A reference fre-quency is not required as in ordinary lock-inamplifier filters, and there is no phase change asthe signal frequency varies.

Source: G.J. Deboo and R.C. HedlundAmes Research Center

(ARC-10264)

Circle Wion Reader Service Card

COMPOSITE FILTER STEEPENS REJECTION SLOPESIN MICROWAVE APPLICATIONS

/u

65

RO

55

en

45

40

35

30

2b

20

15

10

5

i I i I i I i

-

BandpassBandrejectComposite

-

~

1

-I i

1Jil

- ^___^y0 ' 4 ' 8 ' 12 '

i 'i

•' ii * •i ii /i /• /• ///'

1 116

\V/

111

11\

\

1 1

-/"f

,'

fikar

BandpassBandrejectComposite

\\\

N.

"> 120 24

1 1 ' 1

Insertion Loss@f0 14 MHz

8.5 dB12.5 dB21.0dB

i I i I28 32

-

_

-

-

-

—Insertion Loss@f0 15 MHz

21.5dB28.5 dB50.0 dB

—

-

36

Frequency in MHz

A composite filter, consisting of a bandpassfi l ter that shapes the passband and band-rejectfi l ters on each edge of the bandpass, steepensthe rejection sjopes. The use of high, unloadedQ-filters prevents bandpass and band-reject filterinteraction that could result in spurious trans-missions. The bandpass filter (see fig.) exhibitsinsertion losses at f0 +.14 MHz and f0 + 15MHz. The band-reject f i l ter (tuned at f0 + 16.75MHz) exhibits sharper insertion losses at f0 +14 MHz and f0 + 15 MHz, while the compositef i l ter insertion loss is the algebraic sum of the

bandpass and band-reject losses. A typicalButterworth design would require 40 resonantcavities with unloaded Q's of 47,000, unat ta in-able in standard waveguide designs. The com-posite design contains only 27 cavities withunloaded Q values on the order of 8,000.

Source: Dome and Margolin, Inc .under contract to



TUNABLE BANDPASS FILTER WITH VARIABLE SELECTIVITY

This stable, RC bandpass fil ter main ta ins acontinuously noninteracting control of the center

frequency, the center-frequency gain, and thecircuit Q. The basic RC network is constructed

10 ELECTRONIC CIRCUITS FOR:COMMUNICATIONS SYSTEMS

CO

«/_ jfFrequency

R41

R2

from stages that achieve second-order transferfunctions using two integrators. The ready avail-ability of integrated circuits makes this methodattractive for general purpose tunable filters.

The highly stable RC bandpass f i l ter hasseparate controls that independently adjust

center frequency, center frequency gain, and Q.The center frequency is controlled by theganged variable resistors RO 'and the selectivityis controlled by R5. The gain at center frequen-cy is proportional to the value of R3 becausethe ratio R5/R2 is kept constant by gangingR5 and R2; the Q and gain controls are alsolinear. This technique should be particularlyuseful in processing low-frequency bioelectronicsignals where the center "frequency may varywith time.

Source: W.J. Kerwin and C.V. ShafferAmes Research Center

(ARC-10191)

Circlet 11 'on Reader Service Card

Section 5. Phase Detectors

CONTINUOUSLY VARIABLE, VOLTAGE-CONTROLLED PHASE SHIFTER

This phase shifter circuit adjusts the phaserelationship between a locally-generated refer-ence frequency and a received rf signal appliedto a phase-coherent detector. The phase shifter

(a)

Reference.—^ ,iInput ^

(0

M^<2>

+ 15 V«

Reference .. . . \Output > ]_ I ^

D5

eg71.'tf

K>7>5>80

'70

•9Vjj

~'.

\

J

— iA1itf1-1D3

iD4

i

1

**/1j41_

Kt.

•

I +E

P«§R5 S - E"o

— c

/

7^

X

^

checkout equipment. Important features include:(1) continuously variable de-voltage controlthrough 2^ rad ("360°); (2) rf circuit volumesmall enough to combine with a phase detector,

Sine/CosinePotentiometer

DO

(b)

(d)

> 180 270 31

Potentiometer Shaft Rotation, Deg.

i A\ /x\\X> '-

D1 D2 03 04 01i '"N

is small enough to be integrated into a receiversubassembly such as a phase detector module,and can-be operated ei ther j inanual ly or by com-mand from remote control panels of automatic

thereby e l iminat ing the phase shif ter subassem-bly; (3) phase stability wi thin ±r/18 rad (± 10°)over the range of 273 to 322 K; and (4) appli-cability to any reference input frequency from

PHASE DETECTORS 11

100 kHz to 10 MHz. The dc voltage applied todiodes Dl, D2, D3, and D4, (see fig.) is sup-plied from the armatures of the sine/cosinepotentiometer. As the potentiometer shaft isrotated through 2n rad, the conductance of eachdiode varies sequentially (see section d of fig.).

Reference signals, applied to the diodes atterminals W, X, Y, and Z, are vectoriallysummed across the load resistor RL. As thepotentiometer shaft is rotated from 0 to tr/2rad (0° to 90°), Dl goes from zero to ful l con-duction, and the signal appearing across RL isthat which is applied to terminal W (0 rad).Shaft rotation from jr/2 to TT rad (90° to 180°)turns Dl off and D2 on. As this occurs, the ef-fective resistance of Dl increases and that of D2decreases.

For shaft .rotation from tr/2 to K rad the re-

sultant signal across RL varies in phase from 0to JT/2 rad. For a shaft rotation from JT to 3-r/2rad (180° to 270°), the same action takes placebetween D2 and D3, and the signal across RLchanges phase from ?r/2 to TT rad. Between3*/2 to 2 TT rad (270° to 360°), the shif t is re-peated between D3 and D4. From 2^ to ?r/2rad (360° to 90°), it occurs between D4 and Dl.Since the potentiometer has no stops, the refer-ence signal phase can repeatedly be variedthrough 2v rad.

Source: C.E. Johns ofCaltech/J Pl-


(NPO-11129)

Circle 12 on Reader Service Caret

PHASE DETECTOR SYNTHESIZES OWN REFERENCE SIGNAL

A phase detection circuit synthesizes its ownphase reference signal from the phase-modulatedinput signal, and detects discrete-step signalswithout using an external reference.

Phase ModulatedInput Signal

IsolationAmplifier

The phase modulated input signal is appliedto an isolation amplif ier whose output is con-nected to both the frequency mult ipl ier and thesynchronous detector. The frequency multiplieruses the signal from the isolation amplifier as acontrol t iming signal to generate an output atN times the input frequency. The multiplicationfactor, N, for a particular system is determinedfrom the expression N = 2T/A0, where A<£ isthe phase change per step, in radians. A fre-quency multiplier is used because phase changesin the input signal do not appear in the mult i -plier output .

The output of the frequency multiplier is ap-plied to the frequency divider where, after divi-

SynchronousPhase

Detector

FrequencyMultiplier

• Output

FrequencyDivider

sion by N, a constant-phase signal is generatedas the reference signal in the synchronous de-tector. The synchronous detector compares thephases of the original signal from the isolationamplifier and synthesized reference signal. Theoutput of the detector is a dc voltage, thepolarity of which changes each time the inputsignal changes phase.

Source: Fairchild StratosCorp.under contract to

Marshall Space Flight Center(MFS-00247)



PHASE DETECTOR HAS HIGH EFFICIENCY IN PRESENCE OF NOISE

A phase detector circuit operating at 10 MHzis capable of mainta ining a balance 40 dB belowsignal in the presence of 30 dB noise. Thephase detector (see fig.) consists of two matched-

Vjn

T2

diode quads driven in opposite phase by thereference signal. The secondaries of T2, drivingthe quads, are separated to reduce the inter-action. An RC current- l imit ing circuit insertedbetween the two parts at rf zero eliminates theeffects of stray capacitance to ground. T3 es-

tablishes the ground reference at the midpointof the primary so that the ends of the pr imarywinding are equally off ground in the oppositerf phase. Thus, capacitance from primary tosecondary in T2 will not upset the secondarybalance. Each transformer is wound withtwisted t r i f i l a r wire to ensure that all windingsare as similar as possible. Since the quads areturned on out of phase, they rectify oppositehalf-waves of the signal to give full-wave detection.This action results in signal distortion cancella-tion which keeps the output balanced at zeroin the presence of large, random noise when thesignal source has even-harmonic distortion. Thetwo quad outputs are added in series to providevoltage doubling, and since the averaging ofrandom noise is just as efficient with a seriesarrangement as with a parallel, a 6 dB im-provement in dynamic range is attained. Thedynamic range (ratio of maximum linear outputto unavoidable drif t) without adjustment isnearly 80 dB, at a frequency of 10 MHz.

Source: General Dynamics Corp.under contract to


No further documentation is available

Section 6. Gain and Frequency Control

WIDE-RANGE AUTOMATIC GAIN CONTROL CIRCUIT

Radio receivers must use some form of auto-matic gain control (AGC) to prevent overload-ing of the f inal output stage. This AGC circuit(see fig.) is capable of handling input signals of1 V rms and can maintain a relatively constantoutput by attenuating the input signal.

The input signal is coupled through Cl to thevariable attenuator made up of diode tee-padnetwork Dl, D2, and D3. The attenuator is pre-biased in the low-attenuation condition by areference voltage coupled through rf chokes LIand L2. The attenuator control elements are Rl,C3, and Ql. Output of the attenuator is coupled

ReferenceVoltage

-15V

Output

orDCControl Voltage

GAIN AND FREQUENCY CONTROL 13

through C2 to reflex amplifier Q2. The ampli-fied rf output is taken from resonant tank cir-cuit L3-C6. The dc control voltage is fedthrough blocking filter L4-G7 to the dc.input ofreflex amplifier Q2. The amplified dc outputvoltage is developed across collector loadresistor R2 and fed to attenuator control tran-sistor Ql. Emitter resistor R3, in conjunctionwith R2, determines the dc gain of Q2, while

C8, C4, and C5 act as ac bypass capacitors tomaintain the rf gain of Q2 as high as possible.

Source: S.H. Black ofSperry Gyroscope Co.


(MSC-00166)

Circleil4ion Reader Service Card

SIMPLE, ACCURATE AUTOMATIC FREQUENCY CONTROL CIRCUIT

This simple, automatic frequency control(AFC) circuit designed for use with voltage-controlled variable-frequency oscillators (VCO's)operates with an accuracy comparable to that ofmore complex AFC circuits.

Output

The output frequency is not affected by thediscriminator tuning as long as the controlledvariable is within the relatively linear portion ofthe discriminator curve. Under this condition,the output frequency depends only on the sta-

Error Meter (Optional)

Modulation

The outputs of the VCO and the crystal oscil-lator are alternately switched into the discrimi-nator by electronic switch no. 1 at a raterequired for the specific application. Thedual-polarity outputs of the discriminator aredecommutated by switch no. 2, and the resultantsquare-wave, error-signal is integrated. The in-tegrated signal is summed with the modulationsignal, and the composite is applied to the VCOto reduce the output of the integrator towardzero. Center-frequency tuning of the discrimi-nator is relatively unimportant; the only require-ment is that the frequencies of both the VCOand crystal oscillator be within the passband ofthe discriminator curve.

Switch Driver

bility of the crystal oscillator. The sensitivity ofthe circuit is degraded at higher frequencies (aswith all discriminators), but this degradationcan be minimized by including more loop gain.At still higher frequencies, the circuit can beused at subharmonics of the output frequencyeither by a heterodyning process or by lockingto an unmultiplied frequency sample. Eitherway, the output frequency is not affected by anydiscriminator tuning uncertainty.

Source: F. ByrneKennedy Space Center

(KSC-10393)



GAIN CONTROL LIMITER CIRCUIT

A triple-ganged potentiometer consisting ofRl, R2, and R3 (see fig.) limits the audio gainof the circuit shown by controlling the output

+V Audio InputD1 ?

1H,< |

^ | ,

v T-v 6

1

TIT

— H —

D2

t L

IAudio Output

signal. Volume control section Rl controls thesignal level in a manner similar to a conven-tional audio control circuit. Diode Dl limits thepeak positive excursion of the signal. When

the signal is more positive than the voltages setby R2, Dl conducts and limits the output. Thenegative peaks are limited by D2 and set by R3.All three resistors are normally ganged togetherso that the limits are set as 'the control is ad-justed. Variations include using fixed .resistorsfor section Rl and using only limit controls.Important advantages afforded by this circuitinclude an ability to control the maximum leveland to limit any sudden upward changes in theoutput signal.

Source: G.D. Doland ofLockheed Electronics Co.


(MSC-11087)


Section 7, RF Oscillators

LOW-POWER, COMPACT, VOLTAGE-CONTROLLED OSCILLATOR

T+,12 VdcB1

.+5 VdcGate Voltage

64\> o Output

Control Voltage

A low power, compact, voltage-controlledoscillator (VCQ) has a wide control range, alinear control response, and an output whichis compatible with integrated circuitry.

The VCO (see fig.) is designed with an inte-grated circuit containing NAND gates Gl, G2,G3, and G4. The oscillator has a linear response

1 over the frequency range of 1 to 4 MHz for a

RF OSCILLATORS 15

control voltage of -1 to -6 V, respectively,. Thehighest frequency limit, with a linear response,is approximately 5 MHz. The l imit ing factor isthe propagation delay of the low power logic.

The oscillator section is a bistable multivi-brator consisting of G2, G3, D4, D5, R2, R3,Cl and C2. The output is buffered by .G4 forfanout purposes to additional circuits. The inputnetwork, consisting of Gl, D2, D3, and Cl,assures that the oscillator will start when poweris initially applied, and prevents severe powersupply transients from causing the oscillator to"latch up." CL1' is a current-limiting diode usedto conserve power.

Voltage regulation for the gates, provided byRl and zener diode Dl, prevents undesirablefrequency deviations which would otherwise

occur from ripple on the 5 V power supply. Cir-cuit power consumption is 65 mW at 4 MHz,when the control voltage power is a maxi-mum of 11 mW.

Assuming that a 5 V supply would have suf-ficient regulation to prevent undesirable frequen-cy deviation, Rl and Dl could be eliminatedand D2 changed to a 3 V zener. Subsequentpower consumption could be reduced to approxi-mately 25 mW, with no change in the controlcharacteristics.

Source: J.A. Exley ofIBM Corp.


(MFS-20136)

Circle 16 on Reader Service Card.

AUTOMATIC FREQUENCY CONTROL OF VOLTAGE-CONTROLLED OSCILLATORS

A unique feature of this: frequency-controloscillator circuit (see fig.) is the use of optical-capacitive coupling for isolation of the klystroncontrol electrode. The electrode voltage is con-

StabilizingNetwork

The circuit is designed to stabilize the fre-quency of an S-band reflex klystron. The errorinput, derived from a discriminator which com-pares the klystron frequency with a harmonic

ToControl

Cl ElectrodeHI—r 6

trolled by a voltage-divider photoresistor whosevalue approximates the potential of the controlelectrode. This circuit can be used for control-ling the frequency of laboratory oscillators andfor stabilizing the pump frequencies of para-metric amplifiers. •

To PowerSupply

of a stable crystal oscillator, is applied to astabilizing, lead-lag network and then amplifiedwith an operational amplifier. Transistors Qland Q2 form an emitter-coupled differentialpair for phase inverting the amplified signalwithout shifting the base operating point of Q3.


03 drives lamp LI and coupling capacitor Cl.The meter indicates the lamp current and actsas an error indicator. The increase in the cur-rent through the lamp decreases the resistanceof the photoresistor PR1 and changes the volt-age of the control electrode. Because the lamphas a relatively long time constant, the coupling

capacitor is used to transfer the high frequencycomponents of the error signal.

Source: R.B. Kolbly ofCaltech/JPL

under contract toNASA Pasadena Off ice

(NPO-11064)


A 225 MHz FM OSCILLATOR WITH RESPONSE TO 10 MHz

A frequency-modulated transistor oscillator(see fig.) designed for use in wideband televisiontransmitters, is an LC Colpitts tank-circuit con-figuration which provides sinusoidal outputwaveforms, even when excited by a nonsinus-

+VBo

VideoInput©-

L2

R5

If&

9+.28VX

1

Cl.. M

1225MHzModu laterOutputr

3

O2

^) ?

3

T i

i\

sC2

S ,

'R1

€'

R2

,C51

)Q1

;R3

oidal input. The circuit also provides a highrate of phase change at the resonant frequency,which has the effect of producing good frequen-cy stability.

A common-collector configuration "for tran-sistor Ql is used primarily because the collectoris internally connected to the case. The com-bined effects of case capacitance to ground

and heat-sink capacitance to ground provide aneffective rf ground for the collector.

The varactor Q2 is placed in series with thecoil to eliminate the modulation effects that canoccur when it is placed in a parallel configura-tion. At high frequencies, there is sufficientinternal inductance and lead inductance to causethe varactor to enter the series-resonant mode.If this condition is allowed to exist, a drastic-reduction in Q occurs, producing unpredictablemodulation response.

The problem of applying the modulation sig-nal and the bias voltage to the varactor issolved by using a blocking capacitor Cl and aseries-trap tuned to a 225 MHz,signal. Thetrap was designed with a low value ;Q to avoidundue attenuation of the, sidebands through thetrap and the video amplifier to ground.

The dc biasing network is Conventional, withQl biased at 40 mA collector'current and VB atapproximately 15 V. This biasing level enalilesthe transistor to operate at a' point which pro-duces small-signal, class-A oscillation.

Source:Auburn Universityunder contract to

Marshall Space Flight Center(MFS-14977)

[Circle tSjon Reader Service. Card.

VOLTAGE-CONTROLLED OSCILLATOR

The voltage-controlled oscillator outlined inthe block diagram represents a unique approachto the generation of FM subcarriers and FMmodulation. The VCO consists of an RC oscil-

lator, a modulator, and an automatic gain con-trol circuit. The basic oscillator is a phase shifterwhose output amplitude is always constant re-gardless of the amount of phase shift. Two of

RF OSCILLATORS 17

Vin

ModulationSignal

rI11

Modulator

Ampl#1

ModulatorAmplifier

ifier Amplifier

1 Oscillator Section

Automatic_ ^^ rcain

1

1

1

rii_i

Control

VCO Output

these phase shifters (amplifiers) are connectedin series and the output of the second phaseshifter is inverted and fed back to the input toproduce the oscillations. The feedback loop-gainis controlled by an AGC circuit so that theamplifiers operate in their linear regions with alow-distortion sinewave output. The frequency ofthe oscillator is primarily determined by thevalue of the RC time constant if the phase shift"in the amplifiers is small. The design of theoscillator requires a balanced, high input im-pedance, differential amplifier with matchedtransistors that have minimal dr i f t characteristics.

The modulator section generates a current atthe oscillator frequency, the magnitude of whichis proportional to the modulating signal. Themodulator can be considered as an analog mul-tiplier, since its output is the product of an in-

put signal at the oscillator frequency and themagnitude of the modulating signal. The modu-lator exhibits nonlinear characteristics whichserve to negate the opposite nonlinear modu-lating characteristics of the phase-shift oscillator.

Automatic gain control is accomplished byvarying the resistance in the feedback loop ofthe oscillator second stage. This resistance isvaried by changing the gate potential of a fieldeffect transistor. The AGC amplifier comparesthe output signal amplitude with a referencevoltage and supplies a current pulse.

Source: Spacelabs, Inc.under contract to

Manned Spacecraft Center(MSC-11707)


Section 8. RF Amplifiers

VARIABLE GAIN AMPLIFIER

The design of this amplifier combines im-proved input and output impedances with relativelylarge signal handling capability and an immuni tyfrom the usual adverse effects of automatic gaincontrol (AGC). These advantages are achievedthrough the use of two FET's, with sources anddrains in parallel, plus a resistive divider for thesignal, and bias to either of the gate terminals.

The ac signal is coupled into the circuit withCl. LI provides a high impedance to the acsignal and a low impedance path for the AGC

potential, while C2, C3, and C4 (bypass capaci-tors) provide low impedance ac-signal paths. Rland R2 divide the ac signal and the AGC poten-tial applied to the gate of Q1.\L2 and C5 forma tuned resonant circuit which is the loadimpedance at the ac operating frequency. Thetwo FET's, Ql and Q2, produce a remote cutofffeature that makes the circuit perform in amanner similar to that of a remote-cutoff vacuumtube. At low values of AGC bias, both Ql andQ2 contribute to the forward transfer admittance.


As the bias is increased, Ql approaches cutoffmore rapidly than Q2 since the bias applied toQ2 is also divided by Rl and R2. As Ql ap-

Vout

AGC

proaches cutoff, Q2 takes over control of theforward transfer admittance of the circuit. TheAGC bias required to cutoff Q2 is larger thanthat required to cutoff Ql, by a factor deter-mined by Rl and R2. With the proper selection

of the ratio of Rl and R2, the FET transfercharacteristic can be optimized for a smoothtransition.

The grounded-base configuration of Q3 pro-vides a low impedance drain load for Ql andQ2, fur ther reducing the reverse energy transferfrom drain-to-gate of Ql and Q2, and conse-quently reducing input impedance variations-pro-duced by the AGC.

The elimination of input impedance variations(as a function of AGC voltage) is especiallyimportant in rf and i.f. amplifiers where thesevariations could alter the bandwidth and centerfrequency of the previous stage.

Source: G. H. SpaidGoddard Space Flight Center

(GSC-10116)


LOW-NOISE, WIDE-BANDWIDTH I.F. PREAMPLIFIER

' ©

r "i, Input

-o—I Matching•j\ Network

® L __J

^

20 Vdc

vvout

i l l

260 nH

^ 2000 pF

(A) Shunt-L Input0 Matching Network

150 nH

Figure 1A. Two-Stage Amplifier

A comprehensive analytical evaluation of cir-cuit configurations, in terms of optimum overallwideband performance, resulted in the develop-ment of a wideband i.f. preamplifier (see fig.)which can be used in most types of communica-tions receivers. Its exceptional performancefeatures include a 20 dB power gain over the

(B) Shunt-C Series-LInput Matching Network

Figure 1B

passband of 10 MHz to 100 MHz, with a pass-band ripple of 1 dB. During the ini t ial design,the major problems which hindered the achieve-

RF AMPLIFIERS 19

merit of a high passband with a concurrentlow noise figure were found to stem from thedifficulty of achieving a source admittance thatwas optimum with respect to the noise figure overa wide bandwidth.

After a thorough analytical investigation, thebasic configuration chosen was a two-stage,common-emitter amplifier with series feedback inthe second stage only (see Fig. 1A). The designin Fig. IB provides a choice between the twoinput matching networks (A and B) that:optimize performance over different portions ofthe passband. Further computer analysis showed

that the simple shunt-L network meets the noisefigure of 1.5 dB at 60 MHz if the passband isfrom 20 to 120 MHz. The shunt-C, series-Linput matching network was found to providebetter overall performance from 10 to 110 MHz,but had ajriarginal noise figure.

Source: W. S. Jones and G. R. Pierson ofTexas Instruments

under contract toElectronic Research Center

(ERC-10321)

Circle 20ion Reader Service Card.

COMPLEMENTARY PAIR BROADBAND TRANSISTOR AMPLIFIER

A wideband distribution amplifier with abandwidth of 50 MHz can be used in commercialradio, FM, and television circuits. Addit ional ,applications may include pulse and timing cir-cuitry for computers.

The input signal is attenuated by the voltagedivider action of resistor Rl in series with a

Since the basic amplifier is a linear device,an automatic gain control (AGC) system is re-quired; 20 dB of AGC is obtained by varying

Output

Input R1

Ithermistor (see fig.). Transistor Ql acts as anisolation stage, and a voltage amplification of 10is achieved by the next stage, which includes Q2.The signal is then fed into a high-impedanceemitter follower Q3.

The complementary pair of transistors, Q4and Q5, functions as a driver for the final out-put stages. The output impedance of the com-plementary emitter follower output stages is verylow, allowing them to drive low impedance loadswith low distortion.

the resistance of the thermistor in the attenuatorcircuit.

Source: G. D. Thompson, Jr.and G. F. Lutes, Jr. of

Caltech/JPLunder contract to

NASA Pasadena Office(NPO-10003)



GAIN AND PHASE-TRACKING AMPLIFIER

«H(—1Signal L»Input

-12 Vdc

• Q1 *swvx*<

i

* V

i (\_z)

~~1 I . -LR i i«

+ 6 Vdc.

DC Return

AGC Output

A comprehensive analysis of i.f. amplifiersoperating at 40.8 MHz was undertaken to pro-vide criteria for selecting the appropriate matched

transistors. With the use of matched transistors,the adverse effects in a common-base amplifierstage are minimized. Included in this analysis isa mathematical model of the AGC circuit shownin the figure. The optimum AGC control voltagewas determined, and mathematical expressionswere derived to enable the design engineer toselect devices for optimum circuit performance.

Source: H. L. Slade ofRadio Corp. of America


(MSC-1227.7)

Circle 22 on Reader Service~Card.

Section 9. RF Measurement Techniques

DYNAMIC LINEARITY MEASUREMENT TECHNIQUE

This technique measures the dynamic linearityof a frequency modulated (FM) subcarrer os-cillator. An FM discriminator operating as a highgain nu l l detector produces an error signal corn-

square wave amplitude is directly proportionalto the frequency difference of the two oscillators,and its repetition rate is directly proportional tothe electronic switch speed. The amplitudes of

UnknownFrequencySource F1(Oscillatoror SCO)

OscillatorF2

F1

F2<X*^ ---o-1 Electronicv Switch

FrequencyCounter

Null DetectorDiscriminator Oscilloscope

iCamera

posed of two signals (one of known frequency,the other unknown) which are electronicallyswitched to the discriminator input. This analogerror signal becomes zero when both inputsignal frequencies coincide.

When both frequency sources are at the samefrequency (Fl = F2), the net output from thediscriminator will be zero. When the sourcesare at different frequencies (Fl * F2), a squarewave appears at the discriminator output; the

the signal generator outputs are unimportantbecause of the broad limiting action of the dis-criminator. The discriminator is fu l ly limitingwith an output of 5 m V or greater.

Source: K. Merz and L. Morrell ofThe Boeing Co.

under contract toKennedy Space Center

(KSC-10186)

Circle'23 on Reader Service Card.

RF MEASUREMENT TECHNIQUES 21

SIMPLIFIED METHOD FOR MEASURING THE IMPEDANCEOF RF POWER SOURCES

A simple test mettiod using the circuit shownin the figure measures the rf-sou;ree output im-pedance so that maximum power transfer canbe achieved. A conventional bolometer detector

rBolometer Detector

RFSource

Jo1 C' T S

I

•N C1

l IIJ "

1 <!

b

\

1J Y Osci11

11

111

AV Jand bridge circuit are used to measure the rfpower, and a bridge reference resistor is variedto achieve, the condition of maximum powertransfer.

With no rf power introduced into the bolom-eter detector, proper amounts of dc and af powercause the bolometer elements to be driven to theresistance which is equal to; the value of thebridge reference resistor. With the dc bias main-tained constant (to zero the meter), rf powerapplied to the input Jl changes the resistance

of the bolometer element, unbalances the powerbridge, and generates a voltage that reduces theaf oscillator power by an amount equivalent tothe rf power that has been added. The rf sourceresistance can be derived from the value of RR.

Source: E. C. Oakley ofCaltech/JPL


(NPO-10734)

on Reader Service Card.

LINEAR SIGNAL-NOISE SUMMER ACCURATELY DETERMINESAND CONTROLS S/N RATIO

A linear signal-noise summer accurately deter-mines and controls the signal-to-noise powerlevel ratio. This type of circuit should be applica-ble whenever it is necessary to mix backgroundnoise with the desired signal in known ratios andto measure detection as a function of varyingS/N ratio. The noise power is referenced tothe signal power, so that changes in the signallevel are not reflected as changes in S/N ratiobut as a change in absolute signal and noisepower levels. Noise is generated through a tem-perature limited diode and then amplified. Theeffective noise bandwidth of the system is es-

tablished by a Butterworth filter. The dc refer-ence is a zener, voltage adjustable in level andpreset to a desired automatic gain controloperating point on the noise amplifier.

The 1 kHz carrier-detected signal, used as areference signal, is applied to a constant amplitudephase-shifting network. Vectorial comparison ofthe reference signal and noise signal is made inan adder circuit and then fed to a linear-logamplifier.

The output of the amplifier is applied to thecontrol grid of the synchronous detector. Asecond signal, taken from the phase shift


Precision 0-50 dBAttenuator

network output, is applied through a limiterand square wave shaping urcuit to a push-pullamplifier, which in turn is used to drive the de-flection plates of the synchronous detector.The synchronous detector difference output isthen summed with the dc reference. The noisepower is therefore referenced to the carriersuch that changes in either of the two levels arenot reflected as changes in the S/N ratio.

With the noise power level accurately definedin bandwidth through the noise filter and ref-

S/NOutput

Precision 0-50 dBAttenuator

Angle-Modulated' Carrier

erenced directly to the angle modulated carrierthrough the feedback loop, accurate noiselevels and signal levels are' set through pre-cision 50-fi attenuators.

Source: J. L. Sundry ofWestinghouse Electric Corp.

under contract toJet Propulsion Laboratory

(JPL-SC-00152)


Section 10. Personal Communications Systems

SELF-CONTAINED MINIATURE ELECTRONICS TRANSCEIVER PROVIDESVOICE COMMUNICATION IN HAZARDOUS ENVIRONMENT

This voice communications system for use inprotective suits fulfil ls the following requirements:(1) The communications equipment does notimpede freedom of movement; i.e., attached cablesare eliminated. (2) Operation of the system isautomatic; i.e., the voice provides the inputenergy to^activate the system in the proper mode.(3) An acute awareness of the surroundings can

be maintained, and sounds which are advancewarnings of impending hazards (gas hissing,material cracking, voice warnings and the like)can be heard. (4) The communications systemdoes not introduce an additional hazard; i.e.,the rf output power does not activate electro-magnetic sensitive devices such as fuses or ex-plosives'.

PERSONAL COMMUNICATIONS SYSTEMS 23

OHeadphone

Microphone .

noises inside the protective 'suit, such as thehissing of flowing air. The system is normallyoperated in a receive condition except when

Inside theProtectiveSuit

When the operator speaks into the microphone,his voice is transmitted, external to the protectivesuit, by the combination speaker/receiver. Whenthe operator is not talking, the system is in thereceive position so that he can hear noisesabove a particular audio level in the immediatevicinity. The voice-operated amplifier must bebiased to a certain level to prevent activation by

the user speaks. In the transmit mode, the uni tbecomes a miniature audio amplifier with apower output of about 1 W.

Source: H. E. CribbKennedy Space Center

(KSC-10164)


PERSONAL COMMUNICATIONS ASSEMBLY

CH>No. 1

Microphone

Microphone

No. 2

Input Signal600 Q'

600ft ° $00 JOdBm±3dB

Earphone 600ftImpedance

,6000

600 n

,6001

Tone WarningInput Earphone 600 R

Isolation Network Isolation Network

Microphone Circuitry

This personal communications system shouldbe useful where work space requires miniaturizationand very high levels of noise rejection.

Earphone Circuit

The total integrated unit (see fig.) consists of ahousing, two-part acoustic tube assembly, micro-phone amplifier, isolation network, and wiring


harness. For redundancy, two microphonesoperate output level through an isolation networkto a balanced load (600 ± 60 ft) to provide anoutput level of 0 dBm ± 3 dB for an input of96 dB and 10 dB ± 3 dB for an input of 116dB.

Each earphone assembly includes 6 transducersto obtain a high level audio output of improvedquality in the presence of ambient noise. Eachassembly, adaptable for use in an earmuff ordirectly into the ear, consists of the transducers,isolation network, housing, mounting provisionsand wiring harness. A tone warning input signal

circuit is also provided to alert the operator toincoming intelligence. With two earphones con-nected to the isolation network (driven at 4 mWsinewave), the output sound pressure level ofeach earphone, measured at 1 kHz in a 6 cccavity, is 108 ± 3 dB.

Source: N.D. Atlas ofNorth American Rockwell Corp.


(MSC-720, 722)

Circle J27j on Reader Service Card

AUDIO SIGNAL PROCESSOR

The signal processing system shown in the dia-gram provides automatic volume control for anaudio amplifier or a voice communication sys-tem, without introducing noise surges during

AVC gain is nullified, and no noise surge occurs.When the signal resumes, the audio switch re-turns the amplifier gain to normal, at a rateapproximately equal to the decay of the AVCgain, in order to maintain overall gain at a con-stant level.

Automatic Volume Control Switch (AVC)

Microphone

pauses in the input, and without losing the initialsignal when the input resumes.

The preamplifier output is fed through a vari-able-gain amplifier to a conventional automaticvolume control (AVC) circuit, and also throughan audio switch which detects the presence of asignal and controls the variable-gain amplifier.When the system input halts for more than amoment, the audio switch reduces the gain of theamplifier at approximately the same rate as theAVC circuit gain increases. Thus, the increased

False noise triggering is controlled by settingthe detection threshold of the switch slightlyabove the ambient noise level. A short delay isprovided in the switch turn-on time to furnisheven greater protection against false triggering,by high-amplitude, short-duration impulses.

Source: R. HymerManned Spacecraft Center

(MSC-12223)


PHASE-LOCKED LOOP TECHNIQUES 25

Section 11. Phase-Locked Loop Techniques

HIGH NOISE IMMUNITY, WIDEBAND BI-PHASE DEMODULATION SCHEME

Vjn

IBalancedMixer

j Amp ]

1

NarrowBandpassFilterfo/2

V0ut

In communication systems where noise isa severe problem, the extremely narrow bandwidthof a phase-locked loop increases the effectivesignal-to-noise ratio when compared to othertypes of schemes. A wideband bi-phase de-modulator (see fig.) has been proposed as asubstitute for the phase-locked loop. The cir-cuit in most instances will yield the same per-

formance as a phase-locked-loop divider as faras signal detection is concerned. The basic re-generator is much less susceptible to noise thana digital divider because of the reduced band-width of the divider circuit. Because of the phasedetection process in a bi-phase demodulator,decreasing the clock circuit ry.'re by less than20 dB below the noise level in the widebandbi-phase circuitry does not appreciably increasethe S/N ratio out of the phase detector. There-fore, in many systems now using a phase-lockloop, this simpler regenerative divider is prac-tical. The major advantages of this system aresimplicity, low cost, light weight, and low volume,with a circuit performance that rivals a phase-lock loop.

Source: E. J. Mitchell ofRadio Corp. of America


(MSC-12102)


PHASE-LOCK LOOP PHASE MODULATOR WITH HIGH MODULATION INDEX,LOW DISTORTION

The phase-lock loop phase modulator shownin the schematic generates a 6.8 MHz carrier atmodulation indices as high as 2.5 with less than5% signal distortion. The modulation signalis applied to the noninverting input of amplifierZl, which provides a de-coupled modulatingsignal to Z2 so that the low frequency responseof the modulator is unrestricted. As the voltage-controlled oscillator Z3 is phase-locked to thereference signal at J2, the demodulated signaloriginating at the phase detector Z6 output can-cels any Z3 frequency changes that would becaused by the amplified modulation signalfrom Z2. This cancellation depends on theloop gain and the phase shift through the loop.As the modulation frequency is increased, thesignal increases in phase lead unti l the first break

point of the closed-loop response is reached.This circuit can have a theoretical phase de-

viation of TT rad (180°) (modulation index of3.14), but in actual practice is limited primarilyby the capability of the phase detector, Z6,to produce a properly damped, ful l square waveat the reference frequency. The large phasedeviations of this circuit are made possible bythe use of a digital-type (variable duty cycle)phase detector.

Source: C. G. Badstiibner ofRadio Corp. of America


(MSC-12247)



ModulationInput

6.8 MHzPhase Mod.Output

RF RECEIVING SYSTEM WITH IMPROVED PHASE-LOCK CHARACTERISTICS

Antennas

Phase-LockandCoherent AGC Looi

Signal

CombinedSignal Demodulator

Bandpass Filter]andAmplifier

ReferenceOscillatorSystem

Voltage 1Controlled 1Oscillator J

An improved receiving system automaticallycombines the output from two independent re-ceiving channels in a manner that optimizes signalreception. The 'combined signal output from thetwo receivers is applied to a primary phase-lockloop, shown by heavier lines in the block dia-gram. In this loop, the phase of the combinedsignal is compared with the phase of a referencesignal, and an error voltage .proportional to thephase difference is developed. This error voltageis used to vary the frequency of a voltage con-trolled oscillator, which in turn has its outputsignal heterodyned with the input signal in each

of the receiving channels. In this manner, theprimary phase-lock loop permits the system totrack changes common to both receiving channelsso that frequency variations, such as those due toDoppler shifts, are followed. Two secondary gain-control loops in the individual receiving channelscompensate for differential signal changes in thesechannels, and thus ensure phase coherence.

Source: C. R. Laughlin, Jr. and V. J. DiLosaGoddard Space Flight Center

(XGS-01222)


VIDEO CIRCUITS FOR TV APPLICATIONS 27

Section 12. Video Circuits for TV Applications

VARIABLE WORD-LENGTH ENCODER REDUCES TV BANDWIDTH REQUIREMENTS

An adaptable, variable resolution (variableword-length) encoding technique increases thetransmission efficiency of pulse code modulatedtelevision signals by reducing the required band-

which modulates the- transmitter. The FTPobtains analog-to-digital conversion by meansof a variable resolution analog-to-digitalconverter which is capable of converting input

PictureorData Source

ImageGenerator

Adaptive CompressionPseudo-Random NoiseProcessor(Pretransmission System)

RFTransmitter

VTransmitting System

RFReceiver

Adaptive CompressionPseudo-Random NoiseProcessor '(Postreception System)

Display andRecord ingEquipment

Receiver System

width. The electronic instrumentation includesa pseudo-random noise signal processor whichreduces the channel capacity required for signaltransmission. Complementary processors arerequired in both the transmitting and receivingsystems.

A pretransmission processor (FTP) is placedbetween the image generator and the transmitterin the transmission system. A postreceptionprocessor (PRP) is placed between the receiver andthe display equipment.

The basic function of the PTP is to add anadapted pseudorandom noise to the picture dataand encode this combined signal by means of avariable resolution encoder. This process gen-erates a pulse coded picture-pulse-noise signal

voltage into a 1-through-n bit word. The numberof bits per word is selected by a variable com-pression control generator.

After the signal has been received and de-coded, the PRP puts the signal through a variableresolution digital-to-analog converter which de-codes it into an analog signal. The adaptedpseudorandom noise from the combinedanalog signal is then removed and processed bya compensator to give an output equivalent tothe input signal the PTP was given.

Source: W. E. Sivertson, Jr.Langley Research Center

(LAR-00087)


MULTIPLEX TELEVISION

A time-multiplexing system (see fig.) enables -several cameras to share a single commercialtelevision transmission channel. In this multi-plex system, the output of each camera on astandard transmission link is controlled in anydesired sequence, and the sequence is repeatedonce each second.

TRANSMISSION SYSTEM

The number of frames allocated to each camerais decided on the basis of the expected or ob-served rates of change of the several scenesto be viewed. Each transmitted frame is identi-fied by digitally encoded signals added to thebasic camera signal. Automatic equipment at themonitoring end of the link decodes the camera


OperatorSelections

VisualInputs (Scenes)

mitted). In this way, flicker-free pictures areobtained with resolution and signal-to-noise asgood as in a multichannel camera system.

identification, and routes the successivelytransmitted frames from the various cameras to themonitors and corresponding magnetic di|S,krecorders. The recorded frames from 'each camerathat is not transmitting are reproduced andchanneled to the proper monitor at a rate of30 frames per second. Thus, every monitor re-ceives 30 frames per second ;regardless of howoften the content of the frame is changed (trans-

Source: W. R. ReedManned Spacecraft Center

(MSC-11595)


TV SYNCHRONIZATION SYSTEM FEATURES STABILITY AND NOISE IMMUNITY

The synchronization circuit shown in thediagram increases TV video presentation byeliminating ambiguous sync signals. An additionalsync level is introduced as a "back porch" on

O Input

from statistically separate sources. The noise isr.s.s.'d, and the sum is 3 dB greater than eitherseparately. The peak-to-peak signal is 6 , d'Bgreater, so that, for a given signal-to-noise

> Out put

the sync pulse. In the presence of rvnse, the se-quential signals at sync and porch frequenciesprovide a sync identification . from which acoincidence circuit can generate sync pulseshaving the required stability and noise immunity.

Basic components in the system include a syncfrequency bandpass' filter^ envelope detector,and low pass filter. A duplicate channel isintroduced to sense and translate the porch fre-quency. Noise at the summing point input tothe differential amplifier is derived from twoseparate equal-bandwidth frequency domains

condition, jitter due to noise vs rise-time is 3 dBless than that produced without the differentialamplifier. Output coincidence from the sync-channel one-shot and porch-channel Schmitttrigger causes an output from an AND gate whichtriggers the output one-shot to provide the actualsync pulse to the horizontal sweep.

Source: F. P. LandauerJet Propulsion Laboratory

(JPL-00915)

Circle/33lon Reader Service Card.

MIXERS 29

Section 13. Mixers

ADDED DIODES INCREASE OUTPUT OF BALANCED MIXER CIRCUIT

This balanced mixer circuit can be used toincrease the output signal level of conventionalcircuits commonly used in radio transmitt ingand receiving equipment. The addition of two.

D1

; :D3

;;D4

Improved Circuit ^2

diodes which form a half-wave carrier-switchbalanced modulator doubles the output signallevel and reduces spurious output and distortion.

The diodes, D3 and D4, are added to a con-

ventional balanced mixer (see fig.) to produce adifference frequency Fj at the output. The in-puts Fc and F'c + Fa represent inputs at thecarrier frequency and the carrier plus audiofrequencies, respectively. Addition of the center-tap-grounded diodes permits both legs of thecircuit to function throughout the ac cycle. Thiseffectively doubles the output frequency whilereducing the level of spurious signals at fre-quencies Fc + Fa and Fc - Fa 'by 50%. Dis-tortions caused by signais at the third harmonic ofFa are also reduced by 50%.

Source: G. B. RobinsonGoddard Space Flight Center

(GSC-00354)


COMPACT MICROWAVE MIXER HAS HIGH CONVERSION EFFICIENCY

Junction B

High Pass FilterOutput Terminal I.F. Input

Terminal

Primary-SignalInput Terminal

Junction A

Low Pass Filter

Voltage-Variable Capacitors

Tunable Shorting

Plugs

Low Pass Filter

IJF. InputTerminal

This compact, lightweight microwave mixerhas a relatively high conversion efficiency andpower output , A pair of back-to-back voltage-variable capacitors in a stripline network providesthe heterodyne action. A primary frequency signalapplied to the input terminal passes to branch1 of the branchline hybrid. The hybrid splitsthe input signal, providing two signals which are

T/2 rad (90°) out of phase. These out-of-phasesignals are applied to the parallel stripline cir-cuits 5 and 6. One of the parallel striplines is aquarter wavelength longer than the other so thatthe signals are in phase at the ends of thestriplines. Intermediate-frequency signals areapplied through the low pass filters to the volt-age-variable capacitors at the ends of the


parallel striplines. As a result of the heterodyneeffect produced by the voltage-variable capacitors,the signals are reflected back through the parallelstriplines. These signals include the high and lowsidebands of the primary input signal. The re-flected sidebands are passed by the high passfilters, but are rejected by the low pass filters,thereby reaching the branchline hybrid (1, 2,3, 4). Differences in electrical paths and phasescause the sidebands to be cancelled at junctionA. However, the sidebands are reinforced at

junction B, from which they pass to the output1 terminal. The tunable shorting plugs may bevaried over a selected frequency range to op-timize the output level of the voltage-variablecapacitors.

Source: N. J. Penque and H. A. Rosen ofHughes Aircraft Company

under contract toGoddard Space Flight Center

(GSC-00197)

Circle'34, on Reader Service Card.

SIGNAL MIXER PROVIDES OUTPUTS WITH MATCHED IMPEDANCECHARACTERISTICS

The novel circuit shown in the figure enablesthe routing of a single signal circuit simultane-ously into two or more circuits which havesimilar impedances. Immediate applications in-clude the connection of several radio receiversto the same antenna with no degradation tosystem performance.

The input signal at port PI is transformedinto two similar outgoing circuits of the same im-pedance at P2 and P3. The incoming signalfeeds into the primaries of transformers Tland T2, in series with the resonating networkL1-C1. A division circuit between the primariesof these transformers feeds directly through L2 tothe high side of output port P3. The transformerprimaries, having similar characteristics, rep-resent a signal divider which feeds in series withL2 to the output port. The value of L2, servingas a phase correction device, is determined by theinput-to-output impedance match. In this case,a 50-0 input is divided into two 50-0 outputcircuits.

The secondaries of Tl and T2 form a circuitdivider through an RC network, R1-C1, to theground side of both P2 and P3. The RC networkregulates the voltage delivered by the secondaryof T2 to P2, and load-tunes the secondary.

Source: R. C. Kinsel ofAvco Corp.

under contract toGoddard Space Flight Center

(GSC-00063)


NASA-Langley, 1972

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