35 CHAPTER 3 FREQUENCY SELECTIVE SURFACE FOR WIRELESS BAND 3.1 INTRODUCTION RF signal is affected by the reflection, refraction, diffraction and scattering. RF signals reach to any point with different amplitude and phase and theses combine together to produce the received signal. Wireless technologies are increasing finding place in our life. Due to several advantages over wired connections, like easy accessibility, faster re-configuration and drop in the price of the required hardware. Interference between neighboring wireless systems is an important issue because of the rapid growth in the use of wireless communications systems, especially, in unlicensed bands such as the ISM band. This interference is not only 2011), but also compromising the . In the past, this issues of interference have been addressed in various ways, including advanced signal processing techniques and antenna designs. Modern signal processing techniques can sometimes improve operation in low signal-to-interference ratio (SIR) environments and antenna technology can be used to enhance desired signals and reject interference to some extent. For instance, multiple-input multiple-output (MIMO) systems exploit the multipath phenomenon to increase the wireless system throughput with advanced signal processing and coding techniques (Jensen et al 2004) and a well designed ultra-wide-band
Transcript
35
CHAPTER 3
FREQUENCY SELECTIVE SURFACE
FOR WIRELESS BAND
3.1 INTRODUCTION
RF signal is affected by the reflection, refraction, diffraction and
scattering. RF signals reach to any point with different amplitude and phase
and theses combine together to produce the received signal.
Wireless technologies are increasing finding place in our life. Due
to several advantages over wired connections, like easy accessibility, faster
re-configuration and drop in the price of the required hardware. Interference
between neighboring wireless systems is an important issue because of the
rapid growth in the use of wireless communications systems, especially, in
unlicensed bands such as the ISM band. This interference is not only
2011), but also compromising the
. In the past, this issues of interference have
been addressed in various ways, including advanced signal processing
techniques and antenna designs. Modern signal processing techniques can
sometimes improve operation in low signal-to-interference ratio (SIR)
environments and antenna technology can be used to enhance desired signals
and reject interference to some extent. For instance, multiple-input
multiple-output (MIMO) systems exploit the multipath phenomenon to
increase the wireless system throughput with advanced signal processing and
coding techniques (Jensen et al 2004) and a well designed ultra-wide-band
36
(UWB) antenna can remove undesired frequencies to improve system
performance (Kerkho & Ling 2003, Kim & Kwon 2004, Suh et al 2005),
However, these solutions are frequently complex and costly, or require
cumbersome antennas (Hui-Hsia Sung 2006). At software level, security
specific measures consists of encryption of the signal, authentication when
connecting to the network and different signal hiding techniques. Although
these measures might be sufficient for most wireless networks, more
persistent malicious users may penetrate these barriers. Wireless DoS attacks
and signal pollution are also issues not accounted for through wireless
software security (Pal Johnsen Blakstad 2011).
A: Interference between adjacent WLAN systems
Figure 3.1 Use of FSS wall to prevent inter WLAN system interference
37
The very high user densities in typical indoor environments are
likely to require a multitude of strategies to reduce interference to acceptable
levels. Furthermore, the radio link is vulnerable to unauthorized access.
Although proper strategies employed in the medium access control (MAC)
layer (Mias et al 2001) can enhance data security, the radio link remains
vulnerable to eavesdropping or interception. Therefore, it is essential to
develop techniques that place a boundary on the radio link and accordingly
reduce the interference to improve system performance.
For indoor wireless systems, instead of employing methods to
cancel the effects of interference electronically within the radio receiver, an
alternative solution for interference control is to modify the physical indoor
propagation environment. Metal shields could be used to isolate an indoor
wireless system from all external electromagnetic signals. Unfortunately this
approach will also block desired external signals, e.g. broadcast radio and TV,
and cellular telephone transmissions. A better solution would be to transform
the building wall into a frequency- selective (FS) alter that intentionally filters
out unwanted interference, while still allowing desired radio services to pass
through, as illustrated in Figure 3.1. By transforming the wall into a
frequency-selective wall (FS-Wall), the undesired interference can be reduced
significantly in strength. This may improve system performance more
dramatically than other mitigation solutions, such as advanced signal
processing techniques or antenna designs (Hui-Hsia Sung 2006).
GSM-900 and GSM-1800 are the most popular frequency bands in
mobile communication which are commonly used in Asia, Europe, Africa,
Middle East and Oceania. GSM-850 and GSM-1900 frequency bands are
used in Canada, United States and some other countries of the world. In
United Kingdom, GSM-1800 is also called Digital Cellular System (DCS).
GSM-450 uses the same band of Nordic Mobile Telephone (NMT) system
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which is the first generation in Nordic countries, Benelux, Russia and Eastern
Europe. T-GSM which is also known as Trans European Trunked Radio GSM
(TETRA-GSM) is a two way transceiver mobile radio also known as Walkie
Talkie. In Asia, Africa, Middle East and Europe TETRA GSM system is used
by police forces, government agencies, transport services and military.
Utility of the mobile communication is well proven, however its
negative aspects are also noticed. The voice of ringing mobile in the places of
worship, hospitals and theaters can be very annoying. Also from security
point of view, a mobile phone signal can be used to detonate an explosive
device in an indoor environment like airports and other highly sensitive areas.
Most of these areas used jammers and reflectors to block GSM signals. But
their use can also disturb other personal communication within the
environment. So, there is a need of such type of reflector which can only
block desired communication signals. This can be done by designing
Frequency Selective Surfaces (FSSs) which can behave as a band-stop filter.
Therefore, an FSS may provide isolation or security (Umair Rafique et al
2012).
Electromagnetic radiation and related health risks are another
common concern associated with mobile communication. Therefore reflective
shielding consisting of wire mesh or metal foil is available to protect rooms
against electromagnetic radiation from GSM cell base stations. The
disadvantages of this technology are negative effects on the air exchange rate
and thus the room climate, complete opaqueness for broadcast frequencies
and difficult but nevertheless obligatory grounding measures. An FSS with its
individual elements does not show all these disadvantages (Wolfgang
Kiermeier & Erwin Biebl 2007).
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3.2 PERIODIC STRUCTURES
FSS is a periodic structure which is basically an assembly of
identical elements arranged in a one or two dimensional infinite array as in
Figure 3.2. Arrays can either be dipoles or slot types which are driven either
passively by an incident plane wave or actively by individual generators. The
main difference between the dipole and slot cases is that we excite electric
currents on the wires in the dipole case on the contrary we excite "magnetic
voltage distribution in the slots). The
two cases become quite similar and symmetric if the electric field in the
dipole case and the magnetic field in the slot case are compared. Depending
on the physical construction and element shape, when FSS elements are
excited by the incident electromagnetic wave they display different filtering
behaviors (Munk 2000).
During plane wave transmission, resonance will be induced if the
length of the elements is a multiple of half of the incident wavelength, i.e
g= 2. This array of elements acts as a spatial electromagnetic filter and
exhibits capacitive and inductive frequency characteristics. The frequency
response of these structures is determined by several factors, including the
periodicity along the X-axis and Y-axis, and the manner by which the
periodic surface is exposed to the electromagnetic radiation (for example,
incident angle etc).
In terms of functionality, these periodic structures can be classified
into four major categories; 1) low pass, 2) high pass, 3) band pass, and
4) band stop filters. In each of these four instances, the resonance
phenomenon remains the same.
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Figure 3.2 Periodic Structure in 2D
3.3 CLASSIFICATION OF FSS ELEMENTS
The physical shape of the FSS elements can be divided into four
different types (Munk 2000). Each element type exhibiting its own frequency
response characteristics. The combination of these types can be used to
generate new elements with a range of specific properties, such as elements
for multiband FSSs, polarization independent FSSs and miniaturized element,
and so on.
3.3.1 Center Connected Elements or N-Pole Elements
In this category are elements consisting in a connected union of
dipoles. Examples of such elements are the simple dipole given in Figure 3.3
a e, the tripole, the anchor, the Jerusalem cross, and the square spiral.
X Y
Z
p
q
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Figure 3.3 Examples of Centre Connected Elements
The main element of this group is a tripole array as shown in
Figure 3.3. These are formed by a combination of three poles of equal length
having the same center point. In general, tripole elements generate larger
bandwidth by reducing inter element spacing.
The cross-dipole can also be used as a dual polarized element,
dependent on the angle of incidence. The Jerusalem cross is the third most
important element in the N-poles class, and provides more tuning options for
obtaining the required response .
3.3.2 Loop-Type Elements
This kind of elements seems to be the most used, and it can provide
a wide range of bandwidths, depending on the element. Different structures
are shown in Figure 3.4 a e.
Figure 3.4 Examples of Loop Type Elements
(a) (b) (c) (d) (e)
(a) (b) (c) (d) (e)
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The looped tripole, the ring, and the square loop are the main
members of this family; as shown in Figure 3.4(b), (d) and (e). The length of
the two orthogonal poles should be equal in order to provide for
proportionality in the loop type structures.
Since the last decade, these structures have attained significant
attention from researchers due to their better performance in angle stability,
ease of fabrication, and higher bandwidths. Transmission line theory can be
used to analyze the basic characteristics of loop type elements. The half-wave
dipole exhibits the property of a shorter dipole which is reactance loaded (ZL).
The property of reactance is to absorb the radiation of shorter wavelengths.
When referring to the total impedance of the dipole, it should have a value of
zero at resonance. As the half wave dipole shows capacitive characteristics at
the resonant frequency, the load ZL should in turn have inductive
characteristic, in order to make the overall impedance equal to zero. However,
inductance can be induced by shorting the transmission line.
3.3.3 Solid Interior/Plate Type
The solid interior/plate types are a patch consisting of a metallic
array which can be either a circular disk, square, rectangular, or hexagonal in
shape and having a g = 2 element length. This class of FSS element
captured the attention of designers, as reported in an early study. The circular
elements of this class are generally reflecting arrays, while the behavior of the
square patch tends to be transparent to radiation. These elements are not
recommended for general purpose filter designs because of their poor angle
stability and early onset of grating lobes. This type exhibits characteristics
which are more useful for the design of miniaturized FSS elements.
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3.3.4 Combinations Elements
As can be expected, all the above-mentioned elements have been
suitably integrated to form a new element in order to improve the
performance of the purely center connected element, loop-type element, and
solid-interior-type element FSSs. Examples of such combinations are shown
in Figure 3.5 a e.
Figure 3.5 Examples of Various Combined Elements
The design of the combination type of element is intended to
overcome the performance shortcomings (angular stability, bandwidth, and so
on) of FSS structures. They are formed from a combination of: 1) solid
interior shape, 2) loop, or 3) center connected and typically have a wide range
of elements. A combination of any two elements from the first three types
makes a new element which is categorized as a combination type.
3.4 PERFORMANCE ANALYSIS OF FSS ELEMENTS
The choice of suitable FSS depends on the desired characteristics
and the designers experience. Table 3.1 presents a comparison of the different
shapes for various standard FSS elements.
(a) (b) (c) (d) (e)
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Table 3.1 Performance analysis of different shapes of FSS elements (Reproduced from Wu 1995)
Element Shape
Angular stability
Cross polarization level
Larger bandwidth
Small band separation
Dipole 4 1 4 1
Cross dipole 3 3 3 3
Loaded dipole 1 2 1 1
Tripole 3 3 3 2
Jerusalem 2 3 2 2
Ring 1 2 1 1
Square loop 1 1 1 1
Rating: 1 = best, 2 = second best, 3 = third best,4 = fourth best
The square loop shows the best performance characteristics in
terms of obtaining larger bandwidth, angular stability, polarization, and
sensitivity in comparison to all other elements. If the side length of the loop is
equal to a multiple of half of the wavelength, the element behaves as a dipole.
In the case of a square loop, we can different resonance points at the stage that
the lengths of two sides of the square reach a multiple of .There are factors
which influence the performance of a FSS. These factors are discussed below.
3.4.1 Incidence Angle
In real life scenario the signal may travel in multiple path and
arrived at the FSS at oblique angles. The oblique incident wave induces a
current distribution upon coming into contact with the surface of a periodic
structure. The response is significantly different to that caused by a normal
incident wave, and depends on two primary factors, the separation between
the elements, and the thickness of the elements. Due to these factors the drift
in the resonance frequency may be observed when the angle of incidence
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changes. This drift can be minimized by keeping the distance between the unit
cell to minimum and keeping the FSS between the dielectric substrate layers
(Hui-Hsia Sung 2006).
3.4.2 Substrate Permittivity
Dielectric used to provide support to FSS elements arranged in two
dimensions. The dielectric constant of the support influence the resonant
frequency of the FSS. This happens because the capacitive component of the
FSS depends on the dielectric constant of the dielectric support. The
transmission coefficient of the FSS with the unit cell depends on relative
permittivity values of the substrate at normal incident angle. Larger
bandwidth within the first pass band was achieved by reducing the
permittivity value. The resonance point changes with a large change of the
permittivity. Hence, permittivity plays an important role in deciding the
transmission characteristic of an FSS, particularly at higher frequencies.
3.4.3 Substrate Thickness
The transmission characteristic of the FSS also depends on the
thickness of the substrate. Following are the effects of the substrate thickness
on the dielectric constant of FSS.
i)
elements are sandwiched between the substrate, then the
effectiveness dielectric constant equals dielectric constant of
the substrate ( r).
ii)
elements are on one side of the substrate, then the
effectiveness dielectric constant equals ( r +1)/2.
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iii) If , then the
effectiveness dielectric constant become non linear function
of the substrate thickness therefore effective dielectric
constant become very sensitive to the thickness of the
substrate (Munk 2000).
3.5 DESIGN OF FSS
There are few important parameters which are to be considered
while designing FSS. Following section discusses them in details.
3.5.1 Bandwidth
Bandwidth is one of the important design parameter of FSS. It is
known that closer FSS element spacing leads to larger bandwidth. This can be
most easily explained by remembering that the FSS will act as reasonably
good ground plane when the impedance is very low. One way to achieve low
impedance is to have all the capacitive components of the FSS more-or-less
cancel all of the inductive components. Another method is to simply minimize
the total possible impedance.
The increase in bandwidth when the element spacing is reduced can
be explained indirectly by the fact that an element in an array has lower
impedance than the same element isolated in free space. This can be
conceptually explained by the fact that while a single element has the ability
to store charge between the edge of the element to infinity, this same element
placed in an array is only able to store charge from the edge of the element to
half the distance to its nearest neighbor.
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If either inter-element spacing Dx, or Dy is increased by a relatively
small amount, say l0%, the bandwidth will be reduced by about 10%. If they
are both increased by the amount, the bandwidth is reduced by about 20%.
3.5.2 Finite FSS Arrays
To meet periodicity requirements, true FSS will be infinite .The
infinite FSS will be not realizable. Therefore, it will be worthwhile to discuss
the properties of infinite FSS arrays and apply this knowledge to the closest
realizable approximation, namely finite FSS arrays. Finite FSS introduces two
major additional considerations to the design of an array, namely edge
diffraction and radiating surface waves. Edge diffraction causes the stop band
bandwidth to increase slightly.
In frequency range where radiating surface waves exist, the finite
array contains two surface waves which propagate along the array in opposite
directions. The currents associated with these waves can be quite strong,
many times stronger than the Floquet currents which are induced in both finite
and infinite FSS arrays. They also travel with a different phase velocity than
the Floquet currents, causing current fluctuations across the array. As a note,
Floquet currents are the currents induced by an incident wave used to excite
the array and have the same amplitude and phase as the incident wave.
These surfaces waves cause particularly high currents near the
edges of the array. While the currents associated with surface waves are quite
strong, they do not radiate as efficiently as Floquet currents. However, while
these currents do not significantly affect the main beam, they can raise side
lobe levels by 10 dB or more.
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3.5.3 Grating Lobes
One of the important consideration in designing an FSS array is the
onset of grating lobes. They are undesirable secondary beams occurring at
angles with high order constructive interference (Wu 1995). Grating lobe
emerges when the element spacing periodicity become electrically large
compared to wavelength in air at the operating frequency. Grating lobe cause
the dispersion of the desired signal and it should be avoided. Rays from two
different collinear point sources are delayed in phase by
sin ( ) cos( )r (3.1)
If d in phase and create
a grating lobe. The smallest spacing will occur when,
sin ( ) cos( ) 1 (3.2)
2 2mrad
rr (3.3)
mr (3.4)
The presence of grating lobes for many applications has the potential to
significantly degrade the performance of an antenna. For example if an
antenna is being used as a receive antenna, it will receive signals from both
the desired direction and also the direction in which the grating lobe is
present, where the FSS should be transmissive. Since grating lobes are only a
function of frequency and element spacing, it is not possible to avoid grating
lobes forever. It is mostly important to be aware of their presence (Dana C
Kohlgraf 2005).
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To determine the reflection and transmission coefficients of the
FSS, the near electric fields are computed at a large distance away from the
FSS (i.e. at ten wavelengths). In the absence of grating lobes, only the main
beam contributes to the near-fields. However, as the frequency is increased,
grating lobes will form that will also contribute to the near-fields. The
interference between these two plane waves causes the oscillation in the
amplitude of the near-field, also for large distances. So when grating lobes
exist, one cannot simply compute the reflection or transmission coefficients
by looking at the near-fields at a single point (https://www.feko.info/
A quality element should have a stable resonant frequency with
angle of incidence. Interelement spacing must be kept small in terms of
wavelength. Spacings larger than will lead to early onset of grating lobes
which always will push the fundamental resonance downward with angle of
incidence, irrespective of the element type.
3.6 NUMERICAL ANALYSIS OF FSS
Reflection and transmission characteristics of electromagnetic
waves through FSS structures have been analyzed and evaluated extensively
by using different numerical methods. Numerical methods assume that the
FSS behaves as a planar double periodic structure, that the FSS has an infinite
number of arrays of equal dimensions, that the unit cell can be simulated by
applying boundary conditions. However, all methods have their advantages
and limitations, which are given in the following sections.
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3.6.1 Equivalent Circuit Method
This method(Langley & Parker 1982) treat FSS as the filter as
shown in Figure 3.6.This method is ideally applicable for those periodic
structures whose thickness, periodicity, and dimension of inductive patch and
capacitive gaps are less than the incident signal wavelength .This method
cannot be used to calculate cross polarization and wide angle response, but
the oblique incident angle of less than 450 has previously been evaluated .
(a) (b) (c) (d)
Figure 3.6 A-Band stop, B-Band pass, C-Low pass, D-High pass Frequency selective surface, their frequency response and equivalent circuit
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3.6.2 Mode Matching
This procedure (Henderson 1983) is associated with the growth of
boundary conditions and scattering from cavities. Initially this technique was
implemented to solve the demanding waveguide scattering problems, where
each side of a broken field diverge in a wave guide mode. The Mode
Matching (MM) technique was developed to solve this problem. This
technique uses test functions to minimize the integral equation into a matrix
form. Consequently it can be used for evaluation of multilayer FSSs. In the
FSS periodic structures, the field is expanded into Floquet modes on both side
of the unit cell . The procedure is associated with the growth of boundary
conditions and scattering from cavities.
3.6.3 Finite Difference Time Domain Method
Time domain (TD)(Taflov & Umashakar 1983) is suitable for
determining the broadband response of periodic structures; but the TD method
is not suitable for oblique excitation of periodic scattering, because of the
required phase shift among adjacent periodic boundaries. Phase shift in the
frequency domain transforms to time delay, by storing data of all time
intervals along appropriate periodic boundaries. This method is not well
suited for oblique incident angles, as it requires an independent FDTD run for
each unique frequency point.
3.6.4 Finite Element Method
This method (Bardi,Remski etl 2002), divides the element structure
into smaller elements, and reconnects them back through nodes (which hold
the elements). This method was originally used for simplifying closed domain
problems, as it is appropriate for evaluating the eigenvector of random
structures. The Finite Element Method (FEM) performance becomes complex
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in three-dimensional periodic scattering, with diminishing accuracy at oblique
incident angles. To overcome these issues, a hybrid of FEM and the boundary
element or boundary integral can be utilized. The computational methods of
this technique require substantial computer time to evaluate the structures.
3.6.5 Finite Integration Technique
The Finite Integration Technique (FIT) method( Weiland T 1977) is
pretty much similar to FDTD and FEM. Some researchers have attempted to
promote FIT, though FEM has established better solver techniques. In FIT,
the solver domain is separated into two grids. The space between grids is
designed in such a way that corner of one grid is placed in the middle of a cell
in the other grid.. The FIT circuit model does not have coupling for
connecting separate branches, whereas these coupling relations exits in FEM.
However, two dimensional structure coupling design can convert into a
coupling free model equivalent; but in the case of three dimensional FIT,
equivalent FEM does not seems effective.
This tool is based on Method of moment technique.
3.7 FSS FOR 5 GHz WLAN SYSTEM
3.7.1 Design of Unit Cell
5 GHz WLAN system works in Unlicensed National Information
Infrastructure (UNII) Band. UNII covers following frequency ranges.
5.15-5.25 GHz indoor operations
5.25-5.35 GHz indoor or outdoor operations
5.725-5.825 GHz outdoor operations
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Physically, when illuminated by incident waves, a unit cell of a FSS
can be treated as a resonance circuit, in which the resonant frequency is
determined by the formula 1/ 2f LC , where L and C represent
equivalent inductance and capacitance of the unit cell, respectively.
Therefore, to decrease the resonant frequency, it is required to increase the
values of inductance or capacitance of the unit cell. Structure of a unit cell is
designed as shown in Figure 3.7. The proposed structure is the modification
of conventional crossed dipole. In which each arm is rotated into 35° and
extended within the limited boundary in order to increase its electrical length.
Better area utilization within the unit cell gives compactness and smaller size
to the unit cell.
Figure 3.7 Proposed unit cell geometry
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A compact FSS design is proposed with stopband performance. It
acts as an indoor shield for 5 GHz in order to protect the WLAN signals from
the intruders and prevent interference between the adjacent network. The
geometrical symmetric nature will provide polarization independent
operation. While designing a unit cell, effort is made to increase the
inductance in limited space. Inductive patches are etched on a single side of
the FR4 substrate, and the distance between the adjacent cells (g) is kept as
0.4 mm. Table 3.2 gives dimensions of the proposed geometry.
Table 3.2 Geometrical details of the proposed FSS
Symbol Parameter Values D Unit cell Width 7 mm
L1 Flat arm length 5.2 mm
L2 Slant arm length 3.8mm
h Substrate height 1.6 mm
w Strip width 0.4 mm
3.7.2 Simulated Structure and Results
To assess the angular stability of the design the unit cells was
illuminated with different incident angles of 0 , 30 , 45 , 60 and the TE
mode transmittance is plotted in Figure 3.8. It is observed variation in the
tuning frequency for different incidence angles was almost negligible. For
different incident angles ,there was slight change in the attenuation offered in
the stop band. Results for TM mode illumination were similar to TE mode.
TM mode transmittance is plotted in Figure 3.9. The angular stability is
retained but slight variation in attenuation in the stop band was noticed.
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Figure 3.8 TE mode response for normal and oblique incident angles
Figure 3.9 TM mode response for normal and oblique incident angles
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The length of the arm and its width were varied to study the
co-relation of them with the performance of the FSS. Table 3.9 lists various
unit cell dimension and line-width values and their effect on resonant
frequency.
Table 3.3 Parametric sweep
Unit Cell dimension (mm)
Line width (mm)
Resonant frequency (GHz)
6 0.4 6.3
6 0.3 6.22
6 0.2 6.1
5 0.4 7.92
5 0.3 7.74
5 0.2 6.8
7 0.4 5
With the change in the dimension of the unit cell and width of the
arm, resonant frequency can be varied as shown in the table above and in the
Figure 3.10. By changing the width of the arm, gradual change in the resonant
frequency can be achieved as evident from the Table 3.3. Therefore the width
of the arm is an important tuning parameter for this FSS.
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Figure 3.10 Parametric sweep for various strip width and dimension
Figure 3.10 shows parametric sweep for strip width (w) and patch
dimension (D) against resonant frequency. D=7mm and w=0.4 is chosen for
the final design due to better angular stability.
3.7.3 Measurement
In principle the unit cell must be replicated in both dimension to
infinite extent. Due to practical reasons only finite number of unit cells will be
replicated in both dimensions. The designed unit cell is fabricated on FR4
lossy substrate of 30cm × 30cm area containing 42 × 42 elements. The
substrate is of 1.6 mm height with the loss tangent value of 0.025. The
fabricated prototype is shown in Figure 3.11.
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Figure 3.11 Fabricated proto-type
Figure 3.12, Figure 3.13 show the line diagram of the test setup used
to measure the transmission coefficient of the designed FSS.
Figure 3.12 A view of the test setup for FSS characterization
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Figure 3.13 Front view of the test setup for FSS characterization
Measurements were carried out inside semi anechoic chamber as
shown in Figure 3.14. Transmitting and receiving antennas were kept at 1m
distance from the FSS. FSS screen is fixed on the frame and surrounded by
the microwave absorbers to avoid the diffraction from edges of the FSS
screen. Microwave absorbers were kept on the ground to avoid reflections.
Horn antennas are placed on either side of the FSS at a 1.5m distance. The
transmission characteristics (S21) are measured using the Vector Network
Analyzer.
microwave absorber Window to
hold FSS
1.5m
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Figure 3.14 Measurement setup
3.7.4 Results and Discussion
Measured transmittance (S21) is shown in Figure 3.15. Transmittance shows the stop band around 5 GHz. Due to FSS geometrical
symmetric nature, it gives identical response for TE and TM polarization. It is observed that the simulated and measured S21 are in good agreement. The
small deviation is due to the losses associated with FR4 and scattering from
the stand used to hold FSS.
Figure 3.15 Comparison of simulated and measured results
Tx Antenna Rx Antenna
FSS
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3.7.5 Comparison with Other Structures
FSS designed for the 5 GHz WLAN is smallest among the available
FSS for the similar band. The size of Square loop FSS used by Hui-Hsia Sung
(2006) for 5.8 GHz WLAN was 21×21 mm. Ghaffer I Kiani (2008) used cross
dipole of 16.6×16.6 mm size for 5 GHz WLAN system. Small size of the unit
cell will results in better angular stability performance and onset of the grating
lobe will be shifted to higher frequency range.
3.8 FSS FOR UWB BAND
3.8.1 Design of Unit Cell
The unit cell geometry of the proposed FSS is illustrated in
Figure 3.16. It resembles the shape of the garland and it is printed on both
side of the dielectric substrate. The side view of the proposed unit cell design
is given in Figure 3.17. The dielectric substrate used is FR4 with dielectric
constant, r = 4.3 and dielectric loss tangent of 0.025. The dimensions are
detailed in Table 3.4.
Figure 3.16 Top View of Unit Cell Geometry
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Figure 3.17 Side View of the proposed FSS
The proposed garland design shown in Figure 3.18, is derived from
the circular ring geometry which provides stable response for various angles
compared to other geometries as shown by Taylor et al (2012). The design
also exhibit polarization independent operation. The smaller unit cell of
dimension 8mm × 8 mm is chosen to avoid early onset of the grating lobes.
To increase the bandwidth, the FSS is printed on both sides of the FR - 4
Substrate.
Table 3.4 Dimensions of Unit cell
Parameters Dimension (mm)
Unit Cell Dimension, D 8
Patch thickness, t 0.035
Substrate height, h 1.6
Width, d 0.6
Inner Distance, L1 3.25
Outer Distance, L2 3.85
Radius, R 3.55
Inter cell Gap, g 0.3
3.8.2 Simulated Structure and Results
The simulated results are shown in Figure 3.19, Figure 3.20 and
Figure 3.21 which gives the transmission characteristics (S21) of the proposed
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FSS. It is observed that the design provides stop band characteristics for a
broad band of 3.5 GHz with its start and stop band frequencies at 7.04 GHz
and 10.55 GHz respectively for -20 dB point. The design provides a relative
band width of 39.89 % centered around 8.8 GHz. The proposed design gives
identical response for both TE and TM mode of polarization.
Figure 3.18 FSS realized as an array of Unit Cell
Figure 3.19 S21 Characteristics of the Unit Cell
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The results clearly show that the proposed design provides highly stable
response for oblique angular incidence for TE and TM mode of polarization.
Figure 3.20 TE Mode Characteristics
Figure 3.21 TM Mode Characteristics
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3.8.2.1 Parametric analysis
3.8.2.1.1 Inner distance L1
The Transmission Characteristics (S21) of the proposed cell
geometry obtained by varying distance L1 while maintaining other parameters
undisturbed is given in Figure 3.22. The stop band range shifts from 7.04GHz
-10.55GHz to 7.34GHz -11.97GHz and to 7.85GHz-14.70GHz for varying
distance L1 of 3.25mm, 3.0mm and 2.6 mm respectively. It is evident from
the results that the change in distance L1 eventually affects the width (d) of
the design ultimately shifting the curve to the right covering the higher
frequencies. This is expected since as L1 increases, the width of the FSS
increase, which in turn reduces the inductance. Therefore the resonances
frequency shifts upwards.
Figure 3.22 S21 of the FSS for varying L1 distance
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3.8.2.1.2 Outer distance L2
Keeping all other parameters constant, outer distance L2 is varied
and its simulated results are illustrated in Figure 3.23. It is observed that by
varying the L2 from 3.85mm to 4.2 mm the curve gets shifted towards left,
indicating that the value of the inductance increases thereby decreasing the
frequency of operation. It is to be noted that increasing distance L2 ultimately
increases the unit cell size.
Figure 3.23 S21 of the FSS for varying L2 distance
3.8.2.1.3 Radius R
It is demonstrated from the transmission characteristics reported in
Figure 3.24 that the entire UWB range can be covered by varying the radius
R. The change in radius R either increases or decreases the unit cell
dimension. It is illustrated in Table 3.5 that range of frequencies covered
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varies with varying R ultimately affecting the unit cell size and the bandwidth
performance.
Figure 3.24 S21 of the FSS for varying R distance
Table 3.5 Effect of radius R on Unit cell size
Unit Cell Area (mm2)
Distance R (mm)
Frequency range at -20 dB (GHz)
8 × 8 3.55 7.04 - 10.55
10 × 10 4.55 5.80 - 9.54
20 × 20 9.55 2.71 - 6.60
3.8.2.1.4 Substrate height h
The Transmission Characteristics (S21) obtained by varying the
substrate height is given in Figure 3.25. Height of the substrate plays a
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dominant role in deciding the attenuation in stop band of the FSS. More the
attenuation better the performance will be.
Figure 3.25 S21 of the FSS for varying h
3.8.3 Measurements
Fabricated FSS screen is shown in Figure 3.26. Dimensions of this
FSS are 33 cm × 31.2 cm with 35 × 35 unit cell in both directions. The
prototype is tested inside in an anechoic chamber and transmission
characteristics were measured. The measurement setup is shown in
Figure 3.27. Test setup and measurement procedure as given in section 3.7.3
was used.
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Figure 3.26 Schematic of Fabricated FSS
Figure 3.27 Experimental setup
3.8.4 Results and Discussion
The measured transmission characteristic (S21) of the fabricated
FSS screen is compared with the simulated result in Figure 3.28. It is evident
that the measured results are in good agreement with those of the simulated
results. The stop band characteristic was observed from 7.2 GHz to 10.5GHz
at -20dB band providing a wide stop band of 3.3GHz, which lies within the
Tx Antenna FSS
70
UWB range. The small deviation from the simulated transmission
characteristics may be due to the losses in dielectric substrate used for
fabricating the prototype. Some ripples observed in the transmittance curve
may be due to the reflections from the anechoic chamber walls and
diffractions from the edges of the stand used to hold FSS. The proposed
design can be tuned to cover the entire UWB range spanning from 3.1GHz to
10.6GHz. Hence the proposed structure can be tuned to whole UWB
frequency range.
Figure 3.28 Comparison of the Measured and Simulated Results
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3.8.5 Comparison with Other Structure
Figure 3.29 A novel polarization selective design
Figure 3.30 Compact FSS Design
R
Y
D
W
L3
L1
L2 y
g
72
In polarization selective design Wan-lu Li et al (2010) shown in
Figure 3.29 limits the performance to one polarization. Therefore it will not
meet the performance requirement for both polarizations. Whereas proposed
design is polarization independent. In the compact FSS design of Wan-lu Li
et al (2012) shown in Figure 3.30, unit cell size is 8.5mm×8.5 mm. This
design is tunable from 3.47 GHz to 6.98 GHz by varying the length of
extruded arms. This can cover the remaining band in UWB by varying the
length of extruded arms.
In the proposed design the unit size cell is 8mm ×8mm. This size is
tunable from 7.04 GHz -10.55 GHz. To cover the lower frequency range
(3.1GHz-7.04 GHz) the unit cell size must be 10×10mm.Therefore it is
concluded that proposed design is better for high frequency, whereas the
extruded arms FSS is better for the lower frequency range. The bandwidth of
extruded length varies from 49% to 41%, whereas for the proposed design
bandwidth is 40%. Shape of the proposed FSS is simpler than the extruded
arm FSS of Wan-lu Li et al (2012). Simpler FSS will be more convenient to
fabricate for the large scale deployment
3.9 FSS FOR GSM 1800 BAND
3.9.1 Design of Unit Cell for GSM 1800
Following Figure 3.31 shows the unit cell of Crossed like design
(CLD). The unit cell is obtained by rotating the alphabet V one to the other by
90o to get a cross like design (CLD) and is printed on either side of the
dielectric substrate. Dimensions of the CLD are given in the Table 3.6.
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Figure 3.31 Crossed Like Design (CLD)
Table 3.6 Dimensions of Unit cell
S.No. Parameter Dimension 1 Resonant Frequency 1.82 GHz
2 Substrate Details (FR4) r = 4.3
3 Thickness of the Patch (mm) 0.035
4 Unit Cell Area (mm2) 63 × 63
5 Diameter of the Vent (mm) 20
6 Length of each V element, L (mm) 75.4
7 Width of the line, W (mm) 2
8 Gap between the elements, G (mm) 1.2
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3.9.2 Simulated Structure and Results
Figure 3.32 Unit Cell Geometry
The Figure 3.32 shows the simulated structure of the Unit cell
.There are four dipole elements and four circular apertures. The length of each
0/2 and the radius of the circular aperture is 20 mm. To
increase the bandwidth, the FSS is printed on both sides of the FR - 4
Substrate. Structure is symmetrical along the x and y axis to ensure same
response for both TE and TM mode of operation.
Figure 3.33 FSS realized as an array of Unit Cell
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The designed FSS in Figure 3.33 produces a -20dB bandwidth of
133 MHz and functions as band stop filter for the frequency range of
1.76 GHz to 1.89 GHz, which exactly falls in line with the downlink range
of GSM 1800 MHz band. The proposed FSS gives the identical S21
characteristic for TE and TM mode of polarization.
For TE polarization, the -10 dB stop-band bandwidths at 1800
MHz is more than 200 MHz, while for TM polarization the -10 dB
bandwidths at 1800 MHz is more than 200 MHz (Figure 3.34).
Figure 3.34 S21 Characteristics of the Unit Cell
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3.9.2.1 Angular stability
The stability of the design for various incident angles is simulated
for both TE and TM modes of polarization, and their responses are plotted in
Figure 3.35. From the results it is clear that the proposed design offers
excellent angular stability response.
Figure 3.35 Simulated TE and TM mode responses for various incident angles
3.9.3 Comparative Analysis with Existing Structure
Comparative analysis was carried out with the FSS structure used
in the past. Coaxial ring resonator (Figure 3.36), Double-square-loop (DSL)
and double-ring (DR) (Figure 3.37) and are the structure used in the past.
These structures are dual band structure, operating at GSM 900 and GSM
1800. These structures are compared with our proposed crossed like design
structure in terms of angular stability, onset of grating lobes and bandwidth.
