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116
CHAPTER 4
POWER QUALITY ANALYSIS USING VARIOUS
CLOSED LOOP REFERENCE CURRENT
ESTIMATION TECHNIQUES
4.1 INTRODUCTION
Previous chapter analysed the SHAF based VSI for various PWM
techniques to reduce the %THD which didn’t consider the variation in load
current. Hence this chapter deals with the performance of closed loop control
which takes into account load current variation by various current estimation
techniques.
As shown in Figure 4.1, the reference signal to be processed by the
controller is the key component that ensures correct operation of AF. The
reference signal estimation is initiated through the detection of essential
voltage / current signals to gather accurate system variables information given
by Peng et al (1998). The voltage variables to be sensed are AC source
voltage, DC-bus voltage of the AF and voltage across the interfacing
transformer. Typical current variables are load current, AC source current,
compensation current and DC-link current of the AF proposed by Soares et al
(1997), (2000). Based on these system variable feedbacks, estimation of
reference signals in terms of voltage / current levels are carried out in
frequency-domain or time-domain.
117
Numerous publications on the theories related to detection and
measurement of the various system variables for reference signals estimation
are reported. Figure 4.1 illustrates the considered reference signal estimation
techniques proposed by Motano et al (2002) and Gobrio et al (2008). This
section presents the considered reference signal estimation techniques,
providing for each of them a short description of their basic features.
Figure 4.1 Topology of reference signal estimation techniques
4.2 FREQUENCY DOMAIN APPROACHES
Reference signal estimation in frequency-domain is suitable for
both single and three phase systems. It is mainly derived from the principle of
Fourier analysis as follows:
Fourier Transform Techniques
In principle, either conventional Fourier Transform (FT) or Fast
Fourier Transform (FFT) is applied to the captured voltage / current signal.
118
The harmonic components of the captured voltage / current signal are first
separated by eliminating the fundamental component. Inverse Fourier
Transform is then applied to estimate the compensation reference signal in the
time domain. The main drawback of this technique is the accompanying time
delay in the sampling of system variables and computation of Fourier
coefficients. This makes it impractical for real-time applications with
dynamically varying loads. Therefore, this technique is only suitable for
slowly varying load conditions. In order to make computation much faster,
some modifications were proposed and implemented. In this modified
Fourier-series scheme, only the fundamental component of current is
calculated and this is used to separate the total harmonic signal from the
sampled load current waveform.
4.3 TIME DOMAIN APPROACHES
Time-domain approaches are based on instantaneous estimation of
reference signal in the form of either voltage or current signal from distorted
and harmonic polluted voltage and current signals. These approaches are
applicable to both single-phase and three-phase systems except for the
Synchronous Detection Method and Synchronous Reference Frame theorem
which can only be adopted for three-phase systems.
4.3.1 Instantaneous Reactive Power Theory (p-q theory)
The instantaneous reactive power theory otherwise known as p-q
theory is proposed by Akagi et al (2007). This theorem is based on -0
transformation which transforms three-phase voltages and currents into the
-0 stationary reference frame. From these transformed quantities, the
instantaneous active and reactive power of the nonlinear load is calculated,
which consists of a DC component and an AC component. The AC
119
component is extracted using HPF and taking inverse transformation to obtain
the compensation reference signal in terms of either current or voltage. This
theorem is suitable only for a three-phase system and its operation takes place
under the assumption that the three-phase system voltage waveforms are
symmetrical and purely sinusoidal. If this technique is applied to
contaminated supplies, the resulting performance is proven to be poor. In
order to make the p-q theory applicable for a single-phase system, some
modifications in the original p-q theory was proposed and implemented by
Czarnecki (2004), (2006).
4.3.2 Synchronous Detection Method (SDM)
The Synchronous Detection Method is very similar to the p-q
theory. This technique is suitable only for a three-phase system and its
operation relies on the fact that three-phase currents are balanced. It is based
on the idea that the AF forces the source current to be sinusoidal and in phase
with the source voltage despite load variations. The average power is
calculated and divided equally between the three phases. The reference signal
is then synchronized relative to the source voltage for each phase as discussed
by Chen et al (1993). Although this technique is easy to implement, it suffers
from the fact that it depends to a great extent on the harmonics in the source
voltage.
4.3.3 Synchronous Reference Frame Theory (d-q theory)
This theorem relies on the Park’s Transformations to transform the
three phase system voltage and current variables into a synchronous rotating
frame. Active and reactive components of the three-phase system are
represented by direct and quadrature components respectively as given by
Bhattacharya et al (1995). In this theorem, the fundamental components are
120
transformed into DC quantities which can be separated easily through
filtering. This theorem is applicable only to a three-phase system. The system
is very stable since the controller deals mainly with DC quantities. The
computation is instantaneous but incurs time delays in filtering the DC
quantities.
4.4 REFERENCE CURRENT ESTIMATION
The control scheme of a SHAF must calculate the current reference
waveform for each phase of the inverter, maintain the dc voltage constant, and
generate the inverter gating signals. The block diagram of the control scheme
of a SHAF is shown in Figure 4.2. The current reference circuit generates the
reference current required to compensate the load current harmonics and
reactive power, and also try to maintain constant the dc voltage across the
electrolytic capacitors.
There are many possibilities to implement this type of control and
the most popular of them will be explained in this section. Also, the
compensation effectiveness of an active power lter depends on its ability to
follow with a minimum error, time delay and the reference signal calculated
to compensate the distorted load current. Finally, the dc voltage control unit
must keep the total dc bus voltage constant and equal to a given reference
value. The dc voltage control is achieved by adjusting the small amount of
real power absorbed by the inverter. This small amount of real power is
adjusted by changing the amplitude of the fundamental component of the
reference current.
121
Figure 4.2 Overall control system of the proposed SHAF
4.4.1 p-q CONTROL TECHNIQUE
The p-q theory formally known as “The Generalized Theory of the
Instantaneous Reactive Power in a Three-Phase Circuit” was first developed
by Akagi (1984). It is based on instantaneous values in three phase power
systems with or without a neutral wire, and is valid for steady state or
transitory operations, as well as for generic voltage and current waveforms.
The p-q theory consists of an algebraic transformation known as a Clarke
transformation of the three phase input voltages and the load harmonic
currents in the a-b-c coordinates to the – – 0 reference frame followed
by calculation of real and reactive instantaneous power components.
122
0
1 1 1
2 2 2
2 1 11
3 2 2
3 30
2 2
a
b
c
v v
v v
v v
(4.1)
0
1 1 1
2 2 2
2 1 11
3 2 2
3 30
2 2
a
b
c
i i
i i
i i
(4.2)
0 0 00 0
0
0
p v i
p v v i
q v v i
(4.3)
' ' '
' ' '' 2 '2
'1
'
v i i p
v i i qi i (4.4)
'
'
'
'
'
1 0
2 1 3
3 2 2
1 3
a
b
c
vv
vv
v
(4.5)
Based on equations (4.1) to (4.5), the algebraic formula for
determining the instantaneous zero sequence power, instantaneous real power
and instantaneous imaginary power is shown in equation (4.6).
0 0 0p v i
123
p v i v i (4.6)
q v i v i
Figure 4.3 shows the interactions of each of the power components
within the power system and how each relates to one another.
‘ ’ is the average value of the instantaneous zero sequence power.
This corresponds to the power which is transferred from the power supply to
the load through the zero sequence components of voltage and current.
‘ ’ corresponds to the alternating power of the instantaneous zero
sequence power. This relates to the exchanged power between the power
supply and the load through the zero sequence components of voltage and
current. The zero sequence power only exists in three phase systems with a
neutral wire.
‘p’ is the mean value of the instantaneous real power. This
corresponds to the energy per unit time unity which is transferred from the
power supply to the load.
‘ ’ is an alternating value of the instantaneous real power. This
corresponds to the power which is exchanged between the power supply and
the load.
‘q’ is the instantaneous imaginary power. This corresponds to the
power that is exchanged between phases of the load. This component is not
constructive to the system and is accountable for undesirable current which
circulates between the system phases. The reactive power does not transfer
power from the supply to the load nor does it exchange power.
124
Figure 4.3 Power components of the p-q theory in – – 0 coordinate
From Figure 4.3, the only component of the power obtained
through the p-q theory that is desirable and constructive is the average real
power and the average zero sequence power. This is because power is
transferred from the supply to the load. The other components of the power
are less desirable and this can be compensated by the SHAF.
The control diagram for the SHAF controller is shown in Figure 4.4. An
important component to note is that the high- pass filter with a cut off
frequency of 50Hz. This filter receives the instantaneous real power from
equation (4.6) and filters all frequencies of power greater than the
fundamental. The output waveform is thus the harmonic power which is
recognized as containing only current harmonics.
2 2
1c
c
i v v p p
v v qv vi (4.7)
0
0
0
11 0
2
2 1 1 3
3 2 22
1 1 3
2 22
ca
cb
cc
i i
i i
i i
(4.8)
125
The harmonic power output from the high pass filter together with
the reactive power is used in equation (4.7) to determine the alpha reference
and beta reference of the currents. These currents are then input to equation
(4.8) where the instantaneous current references to the PWM current control
are determined.
Figure 4.4 Control diagram for SHAF using p-q control theory
Since the SHAF is designed predominantly for current harmonic
mitigation, the harmonics present in the power waveform can be assumed to
be attributed solely by the current harmonics demanded by the non-linear
load. If one assumes that the voltage waveform is perfectly sinusoidal and
free from all harmonics, then this condition becomes true. If the three phase
voltage input to the controller is unbalanced or highly distorted, the reference
currents calculated would not completely filter the current harmonics
demanded by the non-linear load. This situation gives rise to the need of a
positive sequence voltage detector.
126
4.4.2 Positive Sequence Voltage Detector
The positive sequence voltage detector shown in Figure 4.5 derives
the positive sequence fundamental signal from a three phase voltage signal
carried by the power line. The PLL control circuit tracks the positive
sequence voltage at the fundamental frequency of highly distorted and
unbalanced three phase signals. The synchronizing circuit determines
accurately the fundamental frequency of the system voltage and phase angle
of the measured signals which may be unbalanced and contain harmonics.
The fundamental frequency is used as an input to a sine wave generator that
produces three auxiliary signals, namely (ia', ib', ic') to be used as ‘fundamental
positive sequence currents’ along the detector. These currents together with
the line voltages are then input to a Clarke -0 transformation algorithm and
power calculation. Equation (4.1) shows the transformation matrix which
converts the phase voltages and phase currents into an appropriate reference
frame. Equation (4.3) determines the power values composed from the
fundamental positive sequence voltage and auxiliary currents.
The – - 0 voltage reference box of Figure 4.5 calculates the
alpha and beta reference voltages given by equation (4.4). Finally, the a-b-c
instantaneous values of the fundamental positive sequence voltage are
determined by the –0 inverse transformation box, without errors in the
amplitude or phase angle as shown in equation (4.5). The voltages calculated
from equation (4.5) are now considered as inputs to the main control circuit.
127
Figure 4.5 Block diagram of the fundamental positive sequence voltage
detector
Thus, the purpose of the positive sequence voltage detector is
justified as the active filter controller compensates the load current as if it
were connected directly to a perfectly balanced sinusoidal voltage source,
irrespective of whether the source is in fact unbalanced or highly distorted.
4.4.3 Synchronous Reference Frame Theory (d-q theory)
The d-q theory is based on a synchronous rotating frame derived
from the mains voltages with the use of a Phase Locked Loop (PLL). In this
theory, active filter currents are obtained from the instantaneous active and
reactive current components (iLd and iLq) of the nonlinear load in a two-step
procedure. In the first step, the load current in the a-b-c reference frame is
transformed to the reference frame. In the second step, these stationary
reference frame quantities are then transformed into synchronous reference
frame quantities based on the Park’s Transformation.
128
The relationship of the real and imaginary components of the
current space vector in the original stationary two-axis reference frame and
the new rotating reference frame is shown in Figure 4.6. In fact, the
transformation is the subset of d-q transformation.
A rotating coordinate system can be defined to enable the vector
representation to become a constant without any time variations. Thus, a d-q
coordinate system has been defined such that the‘d’ and ‘q’ axes rotate at an
angular frequency ‘ ’ in the plane. A balanced three-phase vector
representation in this rotating d-q coordinate system will now be constant
over all times and the angle ‘ ’ is a uniformly increasing function of time.
This transformation angle is sensitive to unbalanced and distorted main
voltage conditions so its change with respect to time may not be constant over
the main period.
Figure 4.6 Space vectors representation in the stationary and
synchronous frames
From Figure 4.6, the direct and quadrature current components can
be written as in equation (4.9):
129
Ldq Ld Lqi i ji . (4.9)
which can also be written in matrix form given by equation (4.10),
cos sin
sin cos
Ld L
Lq L
i i
i i (4.10)
where,
1tanv
v
The real component of the current space vector in this new
reference frame is the direct axis component (id) while the imaginary
component is called the quadrature axis component (iq). With vector rotation,
the direct voltage component and the quadrature voltage component are given
by equation (4.11):
2 2 , 0dq q
v v v v (4.11)
By using simple geometry, equation (4.11) is written in terms of the
stationary reference frame load voltage vectors given by equation (4.12):
2 2
1Ld L
Lq L
i v v i
i v v iv v (4.12)
The block diagram of this theory is given in Figure 4.7. In the
nonlinear load case, the instantaneous active and reactive load currents can
also be decomposed into oscillatory and average terms. Since the d and q axes
rotate at an angular frequency ‘ ’ (=2* *ffundamental) in the plane; the first
harmonic positive sequence current is transformed to a dc quantity and other
130
current components constitute the oscillatory parts. After removing the DC-
component of iLdq by using low pass filters, the compensation current is
obtained given by equation (4.13),
2 2
1c Ld
c Lq
I v v i
I v v iv v (4.13)
Figure 4.7 Block diagram of the instantaneous active and reactive
current method
4.4.3.1 Phase Locked Loop (PLL) Circuit
The PLL circuit tracks continuously the fundamental frequency of
the measured system voltages. The appropriate design of the PLL should
allow proper operation under distorted and unbalanced voltage waveforms
given by Arruda et al (2001). The PLL synchronizing circuit is shown in
Figure 4.8, which determines automatically the system frequency and the
phase angle of the fundamental positive sequence component of a three phase
input signal.
131
Figure 4.8 Block diagram of PLL circuit
4.4.4 Synchronous Detection Method (SDM)
Figure 4.9 shows the control circuit of an AF system using SDM.
The control circuit consists of an outer voltage control loop and two inner
current control loops. The outer control loop is used to maintain the capacitor
voltage constant and to determine the amplitude of the mains currents
required in an AF system. The SDM method is basically used for the
determination of amplitude of the source currents. In this algorithm, the
three-phase mains currents are assumed to be balanced after
compensation. The real power p(t) consumed by the load could be
calculated from the instantaneous voltages and load currents as given by
equation (4.14).
( )
( ) [ ( ) ( ) ( )] ( )
( )
la
sa sb sc lb
lc
i t
p t v t v t v t i t
i t
(4.14)
where, vsa(t), vsb(t), vsc(t) are the instantaneous values of supply voltages and
ila(t), ilb(t), ilc(t) are the instantaneous values of load currents.
132
Figure 4.9 represents the block diagram of the system. The real
power p(t) is sent to a low-pass filter to obtain its average dc value Pdc. The
required expressions are explained in the following section:
Figure 4.9 Block diagram of Synchronous Detection Method
The average value Pdc is determined by applying p (t) to a low pass
filter. The real power is then split into the three phases as given by the
equation (4.15),
dc sma
a
sma smb smc
P VP
V V V
dc smb
b
sma smb smc
P VP
V V V (4.15)
dc smc
c
sma smb smc
P VP
V V V
Thus for purely sinusoidal balanced supply voltages, Pa=Pb=Pc
given by equation (4.16),
3
dca b c
PP P P (4.16)
133
where,
22 2
sma sma sma smaa sa sa
V I V IP V I
2 a
sma
sma
PI
V
Thus the reference source currents for all the phase are given by
equation (4.17),
2
2 ( )( ) sa a
sa
sma
v t Pi t
V
2
2 ( )( ) sb b
sb
smb
v t Pi t
V (4.17)
2
2 ( )( ) sc c
sc
smc
v t Pi t
V
where, Vsma , Vsmb and Vsmc are the amplitudes of the supply voltages. The
compensation currents are then calculated using equation (4.18),
( ) ( ) ( )ca sa lai t i t i t
( ) ( ) ( )cb sb lbi t i t i t (4.18)
( ) ( ) ( )cc sc lc
i t i t i t
This method can be extensively used for compensation of reactive
power, current imbalance and mitigation of current harmonics. It can be
134
inferred as the simplest method as it requires minimum calculations.
However, this method suffers a drawback from individual harmonic detection
and its mitigation.
4.4.5 Perfect Harmonic Cancellation (PHC) Technique
The Perfect Harmonic Cancellation (PHC) technique can be
regarded as a modi cation of the three previous theories. Its objective is to
compensate all the harmonic currents and the fundamental reactive power
demanded by the load in addition to eliminating the imbalance. The source
current will therefore be in phase with the fundamental positive-sequence
component of the voltage at the Point of Common Coupling (PCC).
The reference source current will be given by as in equation (4.19),
1refi K v (4.19)
where, v1+ is the PCC voltage space vector with a single fundamental
positive- sequence component given by equation (4.20),
1 1 1( )s
P v Kv K v v v v (4.20)
The constant ‘K’ will be determined with the condition that the
above source power equals the dc component of the instantaneous active
power demanded by the load given by equation (4.21),
0
2 2
1 1
La Lp pK
v v (4.21)
Finally, the reference source current will be given by
equation (4.22),
135
0
0
12 2
1 1
1
0ref
La L
ref
ref
ip p
i vv v
i v
(4.22)
where, and are the fundamental components of the load voltages and
can be obtained from the original voltages by means of two simple Band Pass
Filters (BPF). Also, Pdc is filtered from p(t) using a simple Low Pass Filter
(LPF). The block diagram of Perfect Harmonic Cancellation strategy (PHC) is
shown in Figure 4.10. Rational term in equation (4.22) is a constant and it can
be seen that after compensation, the currents will have the same shape as the
fundamental components of voltages in and axes. The desired currents
calculated using this method is symmetrical and sinusoidal.
Figure 4.10 Block diagram of Perfect Harmonic Cancellation Method
136
4.5 SIMULATION RESULTS FOR VARIOUS CLOSE LOOP
REFERENCE CURRENT ESTIMATION TECHNIQUES
Simulation results for various close loop reference current
estimation techniques are given below in Tables 4.1 and 4.2 for two loads
namely Resistive load (R) and DC Motor load:
Table 4.1Simulation parameters for R-load
System Source voltage 230V
System frequency 50Hz
Source resistance(Rs) and Inductance(Ls) 0.1 , 0.03mH
Impedance upstream of the rectifier Rc and Lc 0.3 0.07mH
Load(three phase diode bridge rectifier) Rdc, Ldc
and Cdc
0.5 ,
0.3mH,470µF
DC link capacitor Cdc(ref) 1000µF
Reference voltage Vdc(ref) 500V
Active filter output inductance , Lf 4.5mH
Table 4.2 Simulation parameters for DC Motor Load
Power 5HP
Supply voltage 240V
Speed 1750rpm
Field excitation 300V
Torque 10 m
137
4.5.1 Simulation results of p-q technique for R-load and DC motor
load
The simulation model for p-q algorithm for R load is shown in
Figure 4.11. Figures 4.12 to 4.14 show subsystem model for pulse generation,
reference current generation and active filter. Figure 4.15 shows the
simulation waveforms of source voltage, source current and load current
variations. Figure 4.16 shows the phase displacement between source voltage
and source current. Figures 4.17 and 4.18 give the DC link capacitor voltage
and %THD of VSI-SHAF based p-q technique for R-load.
Figure 4.11 Simulation Model for VSI-SHAF using p-q technique for
R-Load
138
Figure 4.12 Subsystem for pulse generation of p-q technique
Figure 4.13 Reference current generation circuit for p-q technique
139
Figure 4.14 Active Filter circuit for p-q technique
Figure 4.15 Simulation waveforms for source voltage, source current
and load current using p-q technique for R-load
140
Figure 4.16 Simulation waveform showing Phase displacement between
source voltage and source current for R-load
Figure 4.17 DC link capacitor voltage using p-q technique for R-load
141
Figure 4.18 %THD for VSI-SHAF using p-q technique for R-load
Figure 4.19 shows the simulation diagram of VSI-SHAF based p-q
technique for DC motor load. Figure 4.20 shows the waveforms of source
voltage, source current and load current and Figure 4.21gives the phase
displacement between source voltage and source current. Figures 4.22 and
4.23 show the DC link capacitor voltage and % THD.
Figure 4.19 Simulation Model for VSI-SHAF based p-q technique for
DC Motor Load
142
Figure 4.20 Simulation waveforms for source voltage, source current
and load current using p-q technique for DC Motor load
Figure 4.21 Simulation waveform showing Phase displacement between
source voltage and source current using pq technique for
DC motor load
143
Figure 4.22 DC link capacitor voltage using p-q technique for DC Motor
load
Figure 4.23 %THD for VSI-SHAF using p-q technique for DC Motor
Load
4.5.2 Simulation results of d-q technique for R-load and DC motor
load
The simulation model for d-q algorithm for R-load is shown in
Figure 4.24. Figure 4.25 shows pulse generation circuit and Figures 4.26 and
144
4.27 show subsystems for d-q and inverse d-q transformations. Figures 4.28 to
4.30 show source voltage, source current and load current waveforms, DC
link capacitor voltage waveform and %THD.
Figure 4.24 Simulation Model for VSI-SHAF using d-q technique for R-
Load
Figure 4.25 Subsystem for pulse generation
145
Figure 4.26 Simulation diagram for d-q transformation (subsystem1)
Figure 4.27 Simulation diagram for inverse d-q transformation
(subsystem2)
146
Figure 4.28 Source voltage, source current and load current waveforms
for VSI-SHAF using d-q technique for R-Load
Figure 4.29 DC link Capacitor voltage for VSI-SHAF using d-q
technique for R-Load
147
Figure 4.30 %THD for VSI-SHAF using d-q technique for R-Load
Figure 4.31 shows the simulation diagram of VSI-SHAF based d-q
technique for DC motor load. Figure 4.32 shows the waveforms of source
voltage, source current and load current, Figures 4.33 and 4.34 show the DC
link capacitor voltage and % THD.
Figure 4.31 Simulation Model for VSI-SHAF using d-q technique for
DC Motor Load
148
Figure 4.32 Source voltage, source current and load current waveforms
for VSI-SHAF using d-q technique for DC Motor Load
Figure 4.33 DC link Capacitor voltage for VSI-SHAF using d-q
technique for DC Motor Load
149
Figure 4.34 %THD for VSI-SHAF using d-q technique for DC motor
load
4.5.3 Simulation results of SDM technique for R-load and DC motor
load
The simulation model for SDM algorithm for R-load is shown in
Figure 4.35. Figures 4.36 and 4.37 show subsystem for pulse generation and
subsystem model. Figures 4.38 to 4.40 show source voltage, source current
and load current waveforms, dc link capacitor voltage and %THD.
150
Figure 4.35 Simulation Model for VSI-SHAF based SDM technique for
R-load
Figure 4.36 Simulation diagram for Pulse generation of SDM
151
Figure 4.37 Simulation diagram of Subsystem block in the pulse
generation for VSI-SHAF based SDM technique
Figure 4.38 Source voltage, source current and load current waveforms
for VSI-SHAF based SDM technique for R-load
152
Figure 4.39 DC link capacitor voltage for VSI-SHAF based SDM
technique for R-load
Figure 4.40 %THD for VSI-SHAF based SDM technique for R-load
Figure 4.41 shows the simulation diagram of VSI-SHAF based SDM
technique for DC motor load. Figure 4.42 shows the waveforms of source
voltage, source current and load current, Figures 5.43 and 5.44 show the DC
link capacitor voltage and % THD.
153
Figure 4.41 Simulation Model for VSI-SHAF based SDM technique for
DC motor Load
Figure 4.42 Source voltage, source current and load current waveforms
for VSI-SHAF based SDM technique for DC motor Load
154
Figure 4.43 DC link capacitor voltage for VSI-SHAF based SDM
technique for DC Motor load
Figure 4.44 %THD for VSI-SHAF based SDM technique for DC motor
Load
4.5.4 Simulation results of PHC technique for R-load and DC motor
load
The simulation model for PHC algorithm for R-load is shown in
Figure 4.45. Figures 4.46 and 4.47 show pulse generation circuit of PHC
155
technique and simulation diagram for subsystem1 in the pulse generation
circuit. Figures 4.48 to 4.50 show source voltage, source current and load
current waveforms, dc link capacitor voltage and %THD.
Figure 4.45 Simulation Model for VSI-SHAF based PHC technique for
R-Load
Figure 4.46 Pulse generation circuit of PHC technique
156
Figure 4.47 Simulation diagram for subsystem1 of pulse generation
circuit of PHC technique
Figure 4.48 Source voltage, source current and load current waveforms
for VSI-SHAF based PHC technique for R-Load
157
Figure 4.49 DC link capacitor voltage for VSI-SHAF based PHC
technique for R-load
Figure 4.50 %THD for VSI-SHAF based PHC technique for R-Load
Figure 4.51 shows the simulation diagram of VSI-SHAF based
PHC technique for DC motor load. Figure 4.52 shows the waveforms of
source voltage, source current and load current, Figures 4.53 and 4.54 show
the DC link capacitor voltage and % THD.
158
Figure 4.51 Simulation Model for VSI-SHAF based PHC technique for
DC motor load
Figure 4.52 Source voltage, source current and load current waveforms
for VSI-SHAF based PHC technique for DC motor Load
159
Figure 4.53 DC link capacitor voltage for VSI-SHAF based PHC
technique for DC Motor load
Figure 4.54 %THD for VSI-SHAF based PHC technique for DC motor
Load
Tables 4.3 and 4.4 show the overall comparative results for various
closed loop reference current estimation techniques showing the variation of
real and reactive powers, power factor and %THD for Resistive load and DC
Motor load.
160
Table 4.3 Overall comparative results for various closed loop reference
current estimation techniques showing the variation of real
and reactive powers, power factor and %THD for Resistive
load
R-load
parameterWithout
filter
Close loop technique
p-q d-q SDM PHC
Real
power(P)2.18e5 2.371e5 2.417e5 2.46e5 2.563e5
Reactive
power(Q)4.45e4 4.404e4 2.676e4 1.342e4 9800
pf 0.92 0.9931 0.9941 0.9986 0.9992
%THD 13.15 9.28 8.95 7.46 4.89
Table 4.4 Overall comparative results for various closed loop reference
current estimation techniques showing the variations of real
and reactive powers, power factor and % THD for DC Motor
load
DC Motor load
parameterWithout
filter
Close loop technique
p-q d-q SDM PHC
Real
Power(P)8.099e4 8.22e4 8.5e4 1.054e5 1.56e5
Reactive
Power (Q)7.841e4 6.665e4 2.535e4 2.321e4 2.15e4
Power
factor (pf)0.65 0.8153 0.8339 0.8713 0.8865
% THD 95.89 9.03 7.19 5.95 4.15
161
Figures 4.55 to 4.58 show the graphical representation of
comparative analysis of various closed loop current estimation techniques
based on % THD and real and reactive power variation for Resistive load
and DC Motor load.
Figures 4.55 Graphical representation of % THD for closed loop
reference current estimation techniques for Resistive load
Figures 4.56 Graphical representation of % THD for closed loop
reference current estimation techniques for DC Motor load
162
Figures 4.57 Graphical representation of Real and Reactive power
variation for closed loop reference current estimation
techniques for Resistive load
Figures 4.58 Graphical representation of Real and Reactive powers for
closed loop reference current estimation techniques for DC
Motor load
163
4.6 CONCLUSION
Among the various closed loop reference current estimation
techniques, namely p-q, d-q, SDM and PHC, %THD for PHC is the least
(4.89% for R-load and 4.15% DC Motor load). Also it gives better load
balancing, better power factor and good reactive power compensation.