IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY. VOL. 32, NO. 2, MAY 1990

t h e new model ’ = o n e solid shel l model

0 0.5 1.0 Comparison of amplitude-frequency characteristic. Fig. 4.

effects on the shielding effectiveness of braided cables and the induced voltage between the core and the shield caused by the lightning cur- rent. We take braided cable SYV50-3 as an example for calculation. Its parameters are c = 1.989 mm, b = 1.689 mm, d = 0.150 mm, p = 47~ x lop7 H/m, and U = 5.8 x lo7 1/Rm.

A . Amplitude-Frequency Characteristic of Zi, Z,, and Z d

From Fig. 4, it can be seen that the values of Z d ( w ) of the new model are greater than those of one solid shell model, especially in high frequencies. Compared with ones of one solid shell model, the values of Z i and Z, are less in low frequencies and greater in high frequencies.

B . The Shielding Effectiveness Factor f o r Braided Cables

The shielding effectiveness factor is a measure of the shielding effect of the braided shield. When a current I , flows on the outer shield, the shielding effectiveness factor S is the ratio of E z ( b ) to E,(c) :

s = E z- ( I $ )

E, (c ) . By using ( 6 ) , ( 7 ) , (8), and the condition Z b = 0, we can get

Z

The absolute value of S varies from one to zero. If the frequency is higher, S will be less, and the shielding effect will be better. Compared with the values of /SI for the new model, the values for a solid shell model decrease (see Fig. 5).

C. Terminal Core-Shield Voltage Induced by the Lightning Current on the Outer Shield

s = L! z,, .

The cable is 1-m long, and its core wire at one end is connected with its shield. At the other end, it is open. The parameters of the lightning are as follows: amplitude = I A; wave front = 1.5 ps; wave tail 40 p s . As the cable is short, the terminal core-shield voltage due to the current diffusion can be approximated as follows:

v ( W ) = z d ( C d ) I o ( W )

where Z,(w) is the Fourier transformation of the lightning current. The voltage can be obtained by using the FFT algorithm (see Fig. 6). It is seen that after modification, the wave front and the wave tail decrease. The former decreases from 3.8 to roughly 1.6 ps, and the latter decreases from 17 to nearly 8 ps. The amplitude of the voltage, however, changes little. This is because they have the same dc resistance.

V. C ~ N C L ~ J S I O N S

As Schelkunoff’s formulas for a solid shell are not suitable for the shield of braided cables, a new model for the braided cable shield is

Fig. 5 . Comparison of the shielding effectiveness factor S

163

1 _ _ p one solio s h e l l nodel I t (MS)

’3 L 8 12 16

Fig. 6. Comparison of the induced voltage due to the current diffusion

developed, and new formulas for Zi , Z , , and Zd are then derived. The new formulas are different from those of a solid shell in the forms of expressions and the numerical calculation results. The new formula for Z d conforms with Vance’s assumed one, which was verified in the experiments, so that the correctness of the new model is then verified.

REFERENCES

[I] S . A. Schelkunoff, “The electromagnetic theory of coaxial transmis- sion lines and cylindrical shields,’’ Bell Syst. Tech. J.. vol. 13, pp. 532-579, 1934. E. F. Vance, “Shielding effectiveness of braided wires shields,” IEEE Trans. Electrornagn. Cornpat., vol. EMC-17, pp. 71-77, May 1975. K. F. Casey and E. F. Vance, “EMP coupling through cableshields,” IEEE Trans. Electrornagn. Cornpat., vol. EMC-20, pp. 100-106, Feb. 1978. L. M . Wedepol, “Transient analysis of underground power transmis- sion systems,” Proc. Inst. Elec. Eng., vol. 120, pp. 253-260, Feb. 1973.

[2]

131

[4]

Characterization of a Novel Passive RF Filter for Frequencies of 4-225 MHz

Abstract- The design and experimental characterization of a novel passive RF (4-225 MHz) filter pertinent to industrial and military avi- ation environments is presented here. The filter utilizes three discrete low-pass stages and a simple dc discriminating circuit. A detailed the- oretical analysis (lumped parameter modeling) of the filter’s frequency response is provided. Even though it is not part of the initial design cri- teria, the filter also effectively blocks both out-of-band high-frequency (f > 225 MHz) and low-frequency harmonics (f > 4 MHz).

Manuscript received February 1, 1989; revised October 15, 1989. This work was supported by the Hazards of Electromagnetic Radiation to Ordnance (HERO) group at the Naval Surface Warfare Center.

The authors are with the Electrical Engineering Department, Auburn Uni- versity, Auburn, AL 36849.

IEEE Log Number 8934188.

OO18-9375/90/05OO-0163$01 .OO 0 1990 IEEE

164 IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. 32. NO. 2. MAY 1990

F F l a m n i a b l e Mix

l . T :r -a-

I - - - < ‘ I I . :_ - ‘ I I . _ _ - - i .I

L 2 -2- - I * _ , , -

.nfiResislive Element (bridgewire) Heallng, _‘J

TYPICAL RF VOLTAGES MEASURED ON A6 AIRCRAFT

Fig. 1. Structural diagram of typical EED

TABLE I

Frequency (MHz)

1. 4.04

2. 4.805

3. 5.385

4. 6.400

5. 6.970

6. 7.595

7. 7.990

8. 8.022

9. 9.050

10. 9.259

11. 9.803

12. 11.065

13. 12.045

14. 13.530

15. 16.060

16. 17.048

17. 18.036

18. 19.270

19. 20.510

20. 21.460

21. 23,180

22. 24.450

23. 26.875

Voltage Irrns)

28

44

69

108

89

66

119

75

180

127

168

126

118

136

380

410

370

208

148

160

142

198

68

I. INTRODUCTION

One widely used electronic component located on Naval surface ships is the electro-explosive device (EED). Typical EED’s utilize a 1-0 resistive bridge wire (see Fig. 1) with an all-fire current rating of 1 A. The all-fire current is defined as the minimum sustained dc current that guarantees ignition of the device. In the past, EED’s have been found to be highly sensitive to electromagnetic interference when subjected to field strengths defined in MIL-STD-1385B (Navy) t l l .

In addition to being able to operate in the field strengths spec- ified in the previously mentioned standard, the ordnance utilizing the EED’s must also be able to withstand RF-induced arc over (dis- charge) in the Naval environment. This event can occur whenever an aircraft (or similar object such as a cart) is excited by high-power antennas. If the aircraft or object is not properly grounded to elim- inate charging, fields sufficient to cause electrical breakdown of the

b & D

. -J

IGNITER CAN

FIRING BAND

Fig. 2. Schematic equivalent of 2.75-in folding fin aircraft rocket (FFAR).

air between the aircraft and ordnance can develop. The resulting RF-induced arc over can couple power levels of the order of several thousand watts. Typical RF voltages measured on a test aircraft with respect to the surface ground plane are shown in Table I.

A Mark I EED (1-0, 1-A, 1-W all-fire rating) was selected for the characterization due to its typical firing characteristics and past record of sensitivity to EM1 [l] . The test vehicle used in conjunc- tion with the EED was a 2.75-in-diameter folding fin aircraft rocket (FFAR). This rocket has demonstrated a high susceptability to EM1 due to the fire-control circuitry (see Fig. 2).

The ignition system consists of an exposed metal firing band that surrounds the motor assembly at the rear of the rocket, a wire that connects the exposed band to the lead of the EED, and another wire, which connects the second lead to rocket case. Firing of the EED is accomplished by passing a dc current from the firing band to the rocket case, which acts as ground.

During handling, the exposed firing band may contact an excited aircraft with the rocket case grounded to deck via a person or cart. This results in RF current flowing directly through the EED, which may cause ignition. In addition, the rocket may act as an antenna that couples RF power to the EED and likewise may result in ignition.

11. THEORETICAL ANALYSIS

A lumped parameter model of the igniter and rocket will serve as the basis for the theoretical and numerical analysis for frequen- cies ranging from 1-30 MHz (bandwidth chosen in accordance with MIL-STD-138SB). Use of lumped parameter modeling in this range of frequencies is justified since the maximum electrical feature size of the system considered is much less than the minimum source wave- length (i.e., maximum rocket feature size -1 m, Xmin = 10 m).

The electrical characteristics of the 2.7s-in rocket used here are well documented [2]. The input impedance of the rocket (as seen by the ignition system) has a frequency dependence that is shown in Fig. 3. An approximate electrical equivalent circuit of the rocket is constructed using passive elements and is shown in Fig. 4 (CB = 27 pF, and L , = 1.8 pH). These elements are included in the total lumped-parameter model of the rocket and filter.

The theoretical analysis and simulation of the ignition system’s response with and without the filter network installed are conducted under the following condition: A direct contact is assumed between the rocket firing band and the ungrounded aircraft with the rocket case connected to the external ground (i.e., an aircraft carrier flight deck).

The frequency response of the EED to an ideal external source connected between the firing band and rocket case is given as

where R, is the resistance of the EED (1 O), and L , is the equiv- alent inductance of the rocket ignition wire (1.8 pH). The rocket’s capacitance Ce is considered effectively in parallel with the external voltage source approximating the EM1 signal at the point indicated in Fig. 4 (a worst-case model of the voltage source approximating the EM1 signal is used, and therefore, its series impedance is assumed to be zero), and hence, CB does not influence the voltage across Rs .

The software program “Microcap” is used in the analysis of all

IEEE TRANSACTIONS ON ELECTROMAGhErlC COMP4TIBILITY VOL 3 2 . NO 2 MAY 1990 I65

I O 0

50

FREOUENC’r (MHzl I O 0

60

20

-20

-60

I O 0

Fig 3 Input impedance of 2 75-111 rocket

“In +T cE I ” CB = 27pf Lw = 1.8pH R, = IQ

Fig. 4. Equivalent electrical circuit of 2.7s-in rocket

14.5 30 FREOUENCY (MHzl

RF attenuation of unfiltered 2.75-in rocket Fig. S .

equivalent L-R-C models presented in the study. Fig. S shows the frequency response of the rocket with no filter inatalled.

After examining the resulting simulations, it is apparent that there exists a significant frequency range where failure occurs. At fre- quencies ranging from 10-20 MHz, the voltage attenuation is ap- proximately 40 to SO dB. According 1.0 Table I, the relative aircraft voltage with respect to ground may reach 170 to 410 V rms at the respective frequency limits. The predicted voltage reduction of this signal at the EED without filtering is at best two orders of magnitude. The minimum excitation voltage at the EED for a frequency of 10 MHz is therefore -1.7 V (developed across the EED), resulting in -1.7 A of current at the EED; this level far exceeds the all-fire cur-

l

;- r

Fig. 6. Photographs of filter

rent of the EED. This predicted failure of the system to RF-induced EM1 was confirmed experimentally (which is discussed in a later section). The use of filtering as the first measure of protection from EM1 in this case is obvious.

Photographs of the filter designed to eliminate the relative EM1 problem are shown in Fig. 6. The first problem addressed is the excessive input voltage between firing band and rocket case, which can be present during typical flight deck operations. Measurements have shown that potential differences of approximately 10 kV rms (f = 18 MHz) have been measured between the landing gear and ground plane (the experimental data has been provided by the Naval Surface Warfare Center). A solution to the phenomenological prob- lem is to provide an approximate electrical short circuit across the firing band and rocket case for the RF frequencies of interest, thereby reducing the input voltage to the rocket. The circuit element selected to perform this function is a low-dissipation multilayer ceramic ca- pacitor C , placed between the firing band and rocket case.

The geometric design and material properties, especially of the dielectric, are crucial to proper operation of the device. Besides the requirement that the device be resistant to nonelectrical environmen- tal effects (e.g., temperature variance, corrosion, and possible hand- ling damage), the constraints placed on the dissipation factor (DF) of the dielectric become critical.

In order to understand the significance of the dielectric’s electrical characteristics in this design, a simple planar capacitor’s operation should be understood. A lossy capacitor having a given geometry will dissipate heat internal to the device via joule heating. The RF impedance of the capacitor will decrease as the relative permittiv- ity increases ( Z - 1 / e r ) . Increasing the relative permittivity usually increases the loss factor of the dielectric and therefore, for a given geometry, the thermal losses within the capacitor. This implies that for given geometric limits, selecting the dielectric material will be based on tradeoffs between the relative shorting effect of the capac- itor (large value of er preferred) and the internal electrical losses (small value of E , preferred for low internal heating).

Considering the above discussion and the availability of capacitors that meet military specifications, three different categories of capaci- tors were carefully considered: NPO, stable K, and high K (dielectric material). Fig. 7 shows the typical frequency dependence of the DF and dielectric constant e , for the different classes of capacitors. The stable K class of capacitors have the largest ratio of c,/DF for the frequency range considered in this design and were therefore chosen. A 0.8-pF capacitor (stable K) was selected for the design ( C , shown in Fig. 8). The parasitic inductance of the capacitor was measured at 2 nH ( L , shown in Fig. 8) using an HP 481SA impedance meter. The capacitor effectively shorts high-frequency EM1 to the rocket case.

166 IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY. VOL. 32. NO. 2, MAY 1990

g STABLE-K CERAMIC OIEI-ECTRIC 3o B z

2 :o" 2 e 0.0 0 6 E -20.0 g 5 -10.0

5 -30.0 IKc IOKc IOOKC IMc IOMc IOOMc 2 s FREOUENCY s

g STABLE-K CERAMIC OIEI-ECTRIC 2 1 1% DlSsP$lKb f A C T Q B - - t - F a z 2 3o B % CAPACITAW CHANGE

z - 1 o " c d - E 5 -j6:@ ' ' I 1 I I

IKc IOKc IOOKC IMc IOMc IOOMc 2 s FREOUENCY s

" . - I

s FREOUENCY s

(c)

ceramic dielectric materials. Fig. 7. Frequency characteristics of (a) NPC), (b) stable-K, and (c) high-K

_ _ . .

Fig. 8. Equivalent electrical circuit ot filter.

Several additional measures against EM1 were also incorporated in this phase of the design to ensure proper operation. In order to minimize the possibility of unwanted resonance effects and pro- vide greater low-frequency attenuation, ferrite beads were added [4]. Thirty resistive ferrite beads were slid over the wire connecting the firing band to the input of the EED. The wire and ferrite bead as- sembly of the filter was inserted into a hollow stainless tube (stabi- lizer rod) with the tube mounted along the axis of the rocket. The impedance of the length of wire with the 30 ferrite beads was deter- mined (using the HP 4815A impedance meter) for low frequencies (4--32 MHz) and found to be 3000 R (resistive).

A second capacitor C2 was included in the design for additional high-frequency attenuation (shown in Fig. 8). The parasitic induc- tance of C2 and attached wire Lz was determined using the HP 4815A impedance meter (80 nH).

Another dissipative element was put in series with the above stage. The element consists of a ferrite toroidal core wound with 60 turns of 26-gauge magnet wire. The impedance of this structure was measured (using an HP 4275 LCR meter) and is approximated by a 30-KQ resistor RB2 with a parallel parasitic capacitance C B z of 0.2 pF for the previously mentioned frequency range (4-32 MHz). The use of an element that appears resistive Rg2 was incorporated into the design to eliminate series resonances with the EED [SI. Another 0.1- p F shunting capacitor C3 and connecting lead that has approximately 80 nH of series parasitic inductance L i was connected to the rocket case (which is shown in Fig. 8). Two resistive ferrite beads were slid over the leads of the EED to provide a resistive impedance of

9

6 Fig. 9. dc discriminating circuit

t ~60.00

5 -120.00 >o -180.00

-240 00

15.5 3c -300.00

F R E O U E N C Y (MHz)

Fig. 10. RF attenuation of filter

approximately 200 12 (shown as Ru3 in Fig. 8). A 50-0 resistor was connected in parallel with Ci to provide a discharge path for any charge that may accumulate on the capacitors.

The above circuit configuration was designed to provide significant filtering and thereby protect the EED from the previously discussed RF hazard. Due to imperfect solder joints, connections, etc., some rectification of the RF may occur [6]. Since the fire-control signal used was dc, it was necessary to include a series circuit, which will discriminate between the fire-control signal and the component of a rectified arc.

The circuit that accomplishes this function is referred to as a limiter and consists of two Zener diodes (Fig. 9) connected in series (anodes connected together). Until the threshold voltage of either Zener diode is exceeded, the circuit conducts only negligible levels of current. The circuit therefore serves to compare the magnitude of the input signal with V z (the Zener threshold voltage) and switch on only for the fire-control signal.

The Zener diodes possess a junction capacitance C D . At RF fre- quencies, the diodes appear as capacitors. The equivalent series impedance of both junctions was measured as 600 pF (using an HP 4275 LCR meter). These elements were added in series to the RF filter to produce the equivalent circuit shown in Fig. 8. A parasitic lead inductance of 40 nH LD was included in the equivalent model of C O . The frequency response of the circuit (Fig. 8) was calculated using the program "Microcap" (Zener diodes represented as C,) and is shown in Fig. 10.

111. FIELD MEASUREMFNT~

In order to verify operation of the filter, a series of tests were performed on the ground plane facility at the Naval Surface Warfare Center. An A6 aircraft was placed 10 m from a communication whip antenna.

IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY. VOL. 32. NO. 2. MAY 1990 167

Fig. 11, Instrumentation system to measure bridge wire current

A Mark I EED (1-0, 1-A, 1-W) was instrumented with a small thermocouple in order to infer the rms current in the bridge wire (Fig. 11). The thermocouple was connected to a high gain voltage amplifier. The amplifier was used to drive a fiberoptic cable. The optical signal from the cable was used to drive a high gain amplifier, and the signal was used to drive a strip chart recorder. The response time of the entire system was determined to be approximately 20 ms. In all field tests, the coupling was maintained from several seconds to several minutes to avoid any problems with response time. The minimum detectable bridge wire current was 5 mA.

The instrumented EED was placed in a rocket. Shielded fittings were used for all electrical connections. Low, intermediate, and high RF frequencies of approximately 9, 18, and 24 MHz were used.

The testing done was fairly straightforward. The firing band of an instrumented rocket was touched to the aircraft to initiate an arc. The current flowing through the bridge wire was measured during this event.

Three versions of the rocket were tested: unfiltered, low-pass filter only, and low-pass filter plus diodes. Frequencies used were 9, 18, and 24 MHz at field strengths of 2 0 , 150, and 200 V/m, respec- tively.

To induce a signal (arcing), the firing band was touched to the wing tip of the aircraft. This resulted in a current being induced in the bridge wire, which immediately burned the bridge wire into two pieces (frequencies of 9 and 18 MHz were used). The instrumentation system used to monitor the bridge wire current was calibrated for a maximum current of 1 A. The current induced in the bridge wire caused the instrumentation to saturate (bridge wire current exceeded 1 A at 9 and 18 MHz).

Next, a rocket with only the low-pass stage was tested. The largest current measured during contact to the aircraft was a 30-60-mA spike at 9 and 18 MHz. Zener diodes were added in series with the EED (Fig. 8), and when the test was repeated, no bridge wire current was observed. No bridge wire current was detected at 24 MHz with the low-pass or the low-pass with diodes stages in place.

Additional measurements were obtained at frequencies of 225 and 450 MHz for power densities of 20 mW/cm2. The firing band of an unfiltered rocket was touched to a metal cart on the ground plane (f = 225 MHz). A variety of corners and edges were contacted. The largest current induced in the bridgewire was 30 ma. The low- pass filter with diodes was installed in the 2.75-in rocket. During all subsequent testing, no bridge wire current was detectable. The frequency was next increased to 450 MHz. No bridge wire current was detected in either the unfiltered or filtered rocket.

A second series of tests that would determine the performance of the rocket filter when exposed to 200 Vim field strengths of frequen- cies of 9, 18, and 24 MHz were initially planned. However, when a standard 2.75-in rocket (no filter included) was tested according to

the procedure described by MIL-STD-l385B, no bridge wire current was detected. A rocket with the filter installed was tested according to MIL-STD-l385B, and no bridge wire current was detected. The improvement in the EED’s immunity to EM1 with the addition of the filter was therefore impossible to determine.

A precise quantitative determination of the RF signal attenuation with the low-pass filter plus diodes installed was not possible since the exact value of neither the attenuated nor the unattenuated cur- rent was able to be determined. However, the bridge wire current measured without the filter was sufficient to evaporate part of the bridge wire. The current flowing through the bridge wire with the filter installed was below the detectability limit of the instrumen- tation system. A qualitative comparison of the two extreme values substantiates the filters effectiveness.

IV. S U M M A R Y

A high-attenuation RF filter has been described. The filter has been tested in an arcing environment at frequencies of 9, 18, 24, 225, and 450 MHz. The filter severely attenuates RF signals. In addition, the filter removes any spurious dc, which results from rectification of a coupled RF signal.

The proposed filter has proven its capability to withstand RF exci- tation. It utilizes inexpensive, standard components, which can eas- ily be integrated into any ordnance. As such, the design provides a simple, low-cost, high-attenuation filter for a specific military appli- cation.

REFERENCES

[l]

[2]

[3]

[4]

W. S . Rose, “EM1 tests of electro-explosive devices,” NSWC Memo. F 52/WSR: Ijb 8020, Sept. 1978. R. I. Gray, “Electromagnetic near field coupling to aircraft systems,” NWL Tech. Rep. TR-2482, Oct. 1970. A. W. Thompson, “Micro-Cap 11, Microcomputer circuit analysis pro- gram,” Spectrum Software, Sunnyvale, CA. I . W. Ha and R. B. Yarbrough, “A lossy element for EMC fil- ters,” IEEE Trans. Electromagn. Compat., vol. EMC-IO, no. 4,

H. W. Denny and W. B. Warren, “Lossy transmission line filters,” IEEE Trans. Electromagn. Compat., vol. EMC-IO, no. 4, Dec. 1968. H. A. Schwab, “Electrical properties of radio frequency glow dis- charges in air at atmospheric pressure,” NWL TR 2124, Nov. 1967.

pp. 141-148, NOV. 1976. [5]

[6]

On A Subclass of Welti Codes and Hadamard Matrices

Abstmct-A derivation is presented to show the relationship between a subclass of the Welti D codes and Hadamard matrices of order F .

I. INTRODUCTION

When Welti introduced the binary D codes in 1960 and Golay published his paper on Complementary Series in 1961, there was no apparent relationship between the two types of codes [ l ] , [2]. In 1977, Wilson showed that binary D codes were composed of inter- leaved complementary sequences [3]. We now show that Hadamard

Manuscript received November 28, 1988; revised August 25, 1989. The authors are with the Naval Surface Warfare Center, Dahlgren, VA

IEEE Log Number 9034194. 22448.

018-9375/90/0500-0167$01.00 0 1990 IEEE