Construction of a digital-TV receiver for the second-generation satellite broadcasting, DVB-S2

Department of Science and Technology Institutionen för teknik och naturvetenskap Linköpings universitet Linköpings universitet SE-601 74 Norrköping, Sweden 601 74 Norrköping

ExamensarbeteLITH-ITN-ED-EX--07/016--SE

Construction of a digital-TVreceiver for the

second-generation satellitebroadcasting, DVB-S2

Anders JonassonNedim Ramiz

2007-06-20

LITH-ITN-ED-EX--07/016--SE

Construction of a digital-TVreceiver for the

second-generation satellitebroadcasting, DVB-S2Examensarbete utfört i Elektronikdesign

vid Linköpings Tekniska Högskola, CampusNorrköping

Anders JonassonNedim Ramiz

Handledare Sven JansonExaminator Qin-Zhong Ye

Norrköping 2007-06-20

RapporttypReport category

Examensarbete B-uppsats C-uppsats D-uppsats

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Svenska/Swedish Engelska/English

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ISBN_____________________________________________________ISRN_________________________________________________________________Serietitel och serienummer ISSNTitle of series, numbering ___________________________________

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Avdelning, InstitutionDivision, Department

Institutionen för teknik och naturvetenskap

Department of Science and Technology

2007-06-20

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LITH-ITN-ED-EX--07/016--SE

Construction of a digital-TV receiver for the second-generation satellite broadcasting, DVB-S2

Anders Jonasson, Nedim Ramiz

Digital television is one of the biggest broadcasting media available. All over the world televisioncompanies are rearranging their broadcasting from analogue to digital transmission. Former standarddisagreements in the analogue era have lead to an agreement of one common European standard fordigital television. Countries like USA and Japan have their own similar standards.

The report consists of two objectives; a survey of the most commonly used standards for digitaltelevision today and the construction of a prototype receiver for the second generation satelliteDVB-standard.

A thorough literature study and careful design resulted in a fully functioning system. Measurementsperformed on the DVB-S sections gave exemplary results. Comparing these results with correspondingmeasurements performed on the DVB-S2 section showed much better performance for DVB-S2 with thesame code rates. This shows some of the advantages of the new standard and proving the coding theoryright. New coding algorithms make it possible to transmit more information on noisier channels ofinferior quality. In laymen´s words; DVB-S2 gives a better picture and more television channels on thesame satellite compared to DVB-S.

modulationstekniker, digital-tv, dvb-s2, elektronikkonstruktion

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© Anders Jonasson, Nedim Ramiz

Abstract

Digital television is one of the biggest broadcasting media available. All over the world

television companies are rearranging their broadcasting from analogue to digital transmission.

Former standard disagreements in the analogue era have lead to an agreement of one common

European standard for digital television. Countries like USA and Japan have their own similar

standards.

The report consists of two objectives; a survey of the most commonly used standards for digital

television today and the construction of a prototype receiver for the second generation satellite

DVB-standard.

A thorough literature study and careful design resulted in a fully functioning system.

Measurements performed on the DVB-S sections gave exemplary results. Comparing these

results with corresponding measurements performed on the DVB-S2 section showed much better

performance for DVB-S2 with the same code rates. This shows some of the advantages of the

new standard and proving the coding theory right. New coding algorithms make it possible to

transmit more information on noisier channels of inferior quality. In laymen’s words; DVB-S2

gives a better picture and more television channels on the same satellite compared to DVB-S.

Acknowledgements

A number of people have helped us making this thesis possible. First of all, we would like to

thank everybody at Zenterio, especially our supervisor Sven Jansson, who was very helpful. We

would also like to thank Kent Lundberg for giving us opportunity and recourses for this thesis.

We would also like to thank the following people who helped us in various ways: Erna

Mehmedagic and Kam Hung To, our opponents for their constructive criticisms. Qin-Zhong Ye,

our examiner at Institution of Science, University of Linköping, for his support and help.

Last but not least, we would like to thank our families and friends for their love and support.

Table of contents

1 Introduction ............................................................................................................................ 1

1.1 Preface .......................................................................................................................................................... 1

1.2 About Zenterio AB ....................................................................................................................................... 1

1.3 About the authors ......................................................................................................................................... 1

1.4 Background .................................................................................................................................................. 1

1.5 Purpose ......................................................................................................................................................... 1

1.6 Goal .............................................................................................................................................................. 1

1.7 Method ......................................................................................................................................................... 2

1.8 Target group ................................................................................................................................................. 2

1.9 Reading guide .............................................................................................................................................. 2

2 Analog Modulations techniques ............................................................................................ 5

2.1 Amplitude Modulation (AM) ....................................................................................................................... 5

2.2 Frequency Modulation (FM) ........................................................................................................................ 6

2.3 Phase Modulation (PM) ............................................................................................................................... 7

2.4 Fading .......................................................................................................................................................... 7

2.5 Multipath propagation .................................................................................................................................. 7

3 Digital modulation techniques .............................................................................................. 9

3.1 Amplitude Shift Keying (ASK).................................................................................................................... 9

3.2 Frequency Shift Keying (FSK) .................................................................................................................... 9

3.3 Phase Shift Keying (PSK) ............................................................................................................................ 9

3.4 Binary Phase Shift Keying (BPSK) ........................................................................................................... 10

3.5 Quadrature Phase Shift Keying (QPSK) .................................................................................................... 10

3.6 Differential Phase Shift Keying (DPSK) .................................................................................................... 11

3.7 Offset QPSK (OQPSK) .............................................................................................................................. 11

3.8 Quadrature Amplitude Modulation (QAM) ............................................................................................... 11

3.9 Code Division Multiplexing (CDM) .......................................................................................................... 12

3.10 Orthogonal Frequency Division Multiplexing (OFDM) ............................................................................ 13

4 Digital television systems ..................................................................................................... 15

4.1 ISDB .......................................................................................................................................................... 15

4.1.1 ISDB-Satellite ................................................................................................................................... 15

4.1.2 ISDB-Terrestrial ................................................................................................................................ 16

4.2 ATSC ......................................................................................................................................................... 16

4.2.1 ATSC-Terrestrial ............................................................................................................................... 17

4.2.2 ATSC-Satellite .................................................................................................................................. 18

4.3 DVB ........................................................................................................................................................... 18

4.3.1 DVB-Satellite .................................................................................................................................... 19

4.3.2 DVB-Satellite 2nd generation ........................................................................................................... 20

4.3.3 DVB-Cable ........................................................................................................................................ 22

4.3.4 DVB-Terrestrial................................................................................................................................. 23

4.3.5 DVB-Handheld .................................................................................................................................. 25

5 Satellite reception ................................................................................................................. 27

6 Receiver functions ................................................................................................................ 29

6.1 Tuner .......................................................................................................................................................... 29

6.2 Demodulator............................................................................................................................................... 29

6.3 LNB Driver ................................................................................................................................................ 29

7 Available chipsets for DVB-S2 reception ........................................................................... 31

7.1 ST STB6100 ............................................................................................................................................... 31

7.1.1 Features ............................................................................................................................................. 31

7.2 ST STB0899 ............................................................................................................................................... 31

7.2.1 Features ............................................................................................................................................. 31

7.3 Conexant CX24118A ................................................................................................................................. 32

7.3.1 Features ............................................................................................................................................. 32

7.4 Conexant CX24116 .................................................................................................................................... 32

7.4.1 Features ............................................................................................................................................. 32

7.5 Intel® CE 5038 .......................................................................................................................................... 33

7.5.1 Features ............................................................................................................................................. 33

7.6 Broadcom BCM4501 ................................................................................................................................. 33

7.6.1 Features ............................................................................................................................................. 33

7.7 Selected chipset .......................................................................................................................................... 33

8 Circuit design considerations .............................................................................................. 35

8.1 RF component selection ............................................................................................................................. 35

8.1.1 Capacitors .......................................................................................................................................... 36

8.1.2 Inductors ............................................................................................................................................ 38

8.1.3 Wires ................................................................................................................................................. 38

8.2 RF energy developed in PCB ..................................................................................................................... 39

8.2.1 Magnetic flux and cancellation ......................................................................................................... 40

8.2.2 Common-mode and differential-mode currents ................................................................................. 41

8.2.3 Input power consumption .................................................................................................................. 42

8.2.4 Component packaging ....................................................................................................................... 43

8.2.5 Inductance reduction ......................................................................................................................... 44

8.2.6 Transmission line analysis ................................................................................................................. 45

8.3 Power supply design elements ................................................................................................................... 46

8.3.1 DC/DC converters ............................................................................................................................. 46

8.4 Power loss .................................................................................................................................................. 49

8.5 Heat removal .............................................................................................................................................. 49

9 Circuit design ........................................................................................................................ 51

9.1 LNB driver and interface ........................................................................................................................... 53

9.2 The LNB boost converter ........................................................................................................................... 53

9.2.1 Output capacitor ripple current .......................................................................................................... 55

9.2.2 Output ripple voltage ......................................................................................................................... 57

9.2.3 Input capacitor ripple current ............................................................................................................ 58

9.3 Voltage requirements by the demodulator chip.......................................................................................... 58

9.4 The 1.2V Buck Controller .......................................................................................................................... 59

9.4.1 Output capacitor ripple current .......................................................................................................... 60

9.4.2 Output voltage ripple ......................................................................................................................... 61

9.4.3 Input capacitor rms current ................................................................................................................ 62

9.5 Voltage requirements by the tuner chip ..................................................................................................... 64

10 PCB layout ............................................................................................................................ 65

11 Drivers ................................................................................................................................... 67

12 Measurements and testing ................................................................................................... 69

12.1 Front-end .................................................................................................................................................... 69

12.1.1 Voltage levels and ripple measurements ........................................................................................... 69

12.1.2 Tuner and demodulator functionality tests ........................................................................................ 71

12.2 LNB driver ................................................................................................................................................. 72

12.2.1 Measurements .................................................................................................................................... 72

13 Result ..................................................................................................................................... 77

14 Conclusion ............................................................................................................................. 79

14.1 The future of digital television ................................................................................................................... 79

15 Abbreviations ........................................................................................................................ 81

16 References ............................................................................................................................. 83

1

1 Introduction

1.1 Preface

This report describes a final year project conducted during spring 2007 at Zenterio AB. The

project goal was to build and test a receiver for the new satellite standard for digital television

broadcasting, DVB-S2.

1.2 About Zenterio AB

Zenterio AB was founded in March 2002 by 15 employees from Nokia Home Communications.

Due to the dot-com crash, Nokia had to liquidate its R&D department in Linköping. Kent

Lundberg, a born entrepreneur at Nokia, had a vision to keep 15 top experts and continue with

Nokias mission. Major downsizing and cutting of salaries were the solutions to stay interactive.

Today, with 28 employees, Zenterio is an established software company in digital TV and Set-

Top-Boxes domain.

1.3 About the authors

The authors of the report are Anders Jonasson and Nedim Ramiz. Anders has completed his final

year as a student in the Master of Science Program in Applied Physics and Electrical

Engineering at University of Linköping. Nedim is currently completing his final year as a student

in the Master of Science Program in Electronics Design Engineering at University of Linköping.

1.4 Background

Digital television is one of the biggest broadcasting media available. All over the world

television companies are rearranging their broadcasting from analogue to digital transmission.

Former standard disagreements in the analogue era have lead to an agreement of one common

European standard for digital television. Five of the broadcasting standards are covered in this

report. Those five standards are satellite S/S2, cable C, terrestrial T and handheld H. Countries

like USA and Japan have their own similar standards.

1.5 Purpose

The main purpose of this thesis was to acquire a greater knowledge on digital television

transmission systems and to construct a receiver for the second generation satellite standard,

DVB-S2. The new standard is in many ways an improvement over the present DVB-S standard,

and in the next couple of years more and more DVB-S2 channels will pop up. The receiver was

built for testing and educational purposes.

1.6 Goal

Several different approaches will be covered in this thesis. The theoretical approach will discuss

current broadcasting ability between different standards. Different broadcasting and modulations

2

techniques will be presented and analyzed. The practical approach consists of learning the design

techniques and applying them later in the real product. The empirical approach will show the

examined product after testing and measuring. The final goal will be to understand the biggest

differences between broadcasting standards and keeping the knowledge in the case of future

work in the area.

1.7 Method

For the theoretical knowledge of this thesis, a literature study of different broadcasting standards

and modulation techniques has been performed. Comparison was made between the standards

and modulation techniques to consider the advantages and disadvantages. The starting point was

to understand the present DVB-S system and decide which components were compatible and

available in order to construct a receiver for DVB-S2.

The most demanding task of this thesis was the Printed Circuit Board (PCB) layout due to its

complexity in Electro Magnetic Compliance (EMC) and Radio Frequency (RF) design. The

finishing point was to construct and install drivers and later on measure and test the finished

product.

1.8 Target group

To fully understand the facts presented in this thesis a prior knowledge in electronics, physics,

and hardware development is valuable. Even though the reader is presumed to have previous

knowledge in the mentioned areas, most of the information should be relatively easy to consume.

1.9 Reading guide

This section contains a short description of each chapter and appendix in the report.

• Chapter 1 contains a short introduction to the thesis. The goal, purpose, background,

method and target group are discussed.

• Chapter 2 shows the reader a quick overview of analogue modulation techniques,

required to understand important terms in Chapter 3.

• Chapter 3 gives the reader an introduction to digital modulation techniques, describing

important terms in digital transmission and digital television.

• Chapter 4 describes digital television systems all over the world. It also contains deeper

and detailed information on second generation DVB satellite broadcasting.

• Chapter 5 contains speculations on the future of digital television.

• Chapter 6 briefly describes how satellite reception works.

• Chapter 7 describes the different subsystems needed to build a satellite receiver.

3

• Chapter 8 contains information about several different chipsets to build a receiver for

second generation satellite Set-Top-Boxes.

• Chapter 9 explains the physical and electronic behavior of the components on the PCB,

covering important issues of EMC compliance and showing the development of RF

energy in circuits.

• Chapter 10 presents circuit design considerations such as calculations of voltages and currents between interactive components.

• Chapter 11 shows the actual final product and briefly explains its functions.

• Chapter 12 briefly describes the software drivers needed by the STB.

• Chapter 13 contains information about measurement and testing.

• Chapter 14 presents the report results.

• Chapter 15 contains a final conclusion, discussion and ideas about future work in the

field of digital television.

• Chapter 16 lists the books, articles and doctorate works used in the thesis as literature

framework.

4

5

2 Analog Modulations techniques

2.1 Amplitude Modulation (AM)

Amplitude modulation is characterized by that the message m(t), equation 2.1, decides the

amplitude of a carrier frequency , see eq. 2.2

· cos (2.1)

· cos · cos · cos (2.2)

The envelope of the resulting modulated signal is an approximation of the message. A problem

appears when the message m(t) changes sign. When this happens the modulated signal s(t) will

also change sign which results in that the carrier frequency changes phase 180 degrees. This

problem is solved by the adding of a constant voltage to the message m(t). The constant voltage

A in equation 2.3 will always, if A/M>1, result in a positive amplitude of the modulated signal

s(t).

· cos · 1 · cos · cos (2.3)

The constant in equation 2.3 is called the modulation index and is an indication of how much

the modulated carrier varies around its original level. Equation 2.3 can also be described in a

different way such as in equation 2.4.

· cos · · cos · · cos (2.4)

Equation 2.4 contains three parts: The carrier frequency with amplitude C, one lower sideband in

which the frequency varies between and , and an upper sideband where the

frequency varies between and . This property characterizes Amplitude Modulation

with Double SideBands (AM-DSB).

Amplitude modulation with double sidebands and carrier is inefficient in terms of power usage in

relation of how much information is being sent. The quote between useful and the total signal

effect is at most 33%. Most of the power is concentrated in the carrier signal which conveys no

information and suppression of the carrier (double sideband suppressed carrier DSBSC) makes it

more efficient. The sidebands are mirrors of each other, one of the sidebands can be suppressed

without information loss (Single SideBand Suppressed Carrier, SSBSC). This will also make it

more power efficient and lowers the required bandwidth by half. One more advantage of SSBSC

is protection of selective fading. Fading occurs when the channel adds a non wanted phase

difference between the upper and lower sideband. The result can be disastrous if the relative

phase difference between the channels approach 180 degrees, which can cause the components to

almost disappear. The foremost advantage of AM-DSB is the use of simple modulators and

demodulators.

6

Vestigial Sideband Modulation (VSB) is a modulation scheme used by the ATSC DTV system.

Traditional amplitude modulation generates a double sideband RF spectrum about the carrier

frequency. The sidebands are mirrors of each other and one is redundant and can be discarded

without information loss, see figure 2.1. Strategy is employed to some degree in VSB, in which

the lower sideband has been partly suppressed. Using suppression of the lower sideband lowers

the required bandwidth of the channel. The drawback is the demodulation becoming more

troublesome. VSB is not as effective in terms of power usage as SSB but demodulation is more

straightforward. Generally, when modulation is more directed towards the ideal SSB in terms of

power and bandwidth efficiency, more advanced techniques are required for both modulation

and demodulation [1].

Figure 2.1 - AM-DSB, SSB and VSB described in the frequency plane

2.2 Frequency Modulation (FM)

Amplitude modulation was one of the first modulation techniques used. When radio

transmissions became more and more used, the interest for new modulation techniques grew

rapidly. To get more traffic in the air, the seek for a modulation form that could squeeze more

information into a specified bandwidth begun. FM was proposed and was in the beginning

thought to have a lower bandwidth than AM. This was not the case and the interest for FM

shrunk rapidly. Nowadays FM is used frequently in a wide spectrum of radio transmission areas.

Due to the constant amplitude, noise and interference sensitivity is much lower for FM than for

AM. Another feature with FM is the lower sensitivity for nonlinearities. If FM is used on a

nonlinear channel this will add components with multiples of the carrier frequency which easily

could be filtered out.

When using frequency modulation the instantaneous frequency will be dependent on the message

m(t), which will set the phase of the cosine.

· cos (2.5)

7

The phase will be set by equation 2.7.

· cos (2.6)

!" " # · (2.7)

· cos # · · $ cos % · cos &·'() · sin (2.8)

The instantaneous frequency of the resulting modulated signal is varying in a sine like fashion in

the interval , · #, · #.. / !0" " · # · cos (2.9)

The FM spectrum can be found by Fourier transform and the use of Bessel functions. The

interested reader could for example see [1].

2.3 Phase Modulation (PM)

Phase Modulation uses different phase states to carry information. PM in the simplest form has

never been used much. It requires more advanced hardware to be used, and if for example two

phase positions are used there is hard to see the difference between 0 and 180 degrees phase

shift.

2.4 Fading

Fading is a multipurpose name for mathematical models of the distortion when sending over

various channels.

2.5 Multipath propagation

Whenever a wireless RF signal is sent (ground wave) the multipath propagation phenomenon

appears. The signal takes multiple paths from the transmitter to the aerial, see figure 2.2. These

different signals interfere at the aerial causing the amplitude of the received signal to fluctuate.

Signals can be reflected and phase disturbed by terrestrial objects such as mountains and

buildings among other things. Ionospheric reflection and refraction, and atmospheric ducting are

other things causing multipath propagation. OFDM is a technique (explained later in this report)

to get rid of the artifacts of multipath propagation. This is done by sending small information

packets with delays in between. The reflected signals can then pass by the aerial before the next

information packet is sent, thus getting around the problem of multipath propagation.

8

Figure 2.2 - Multipath propagation

9

3 Digital modulation techniques

3.1 Amplitude Shift Keying (ASK)

Amplitude Shift Keying is the digital counterpart of analog AM. As the name implies the

envelope of the modulated signal is varied. Especially on radio channels the damping is of great

concern when using ASK. Big and fast variations of the total channel damping affect the

amplitude of the transmitted signal and can in the worst case lead to a false symbol received. The

symbol rate has an upper limit that varies with the bandwidth of the channel. The preceding

symbol must have time to decay to a low level before the next symbol is sent. If the time allowed

for decay is too short an effect called inter symbol interference will make it hard to separate

symbols following each other. With the use of several discrete amplitudes, for example 8 levels,

more information can be sent at the same time as three bits can be coded as one level. This is for

example done in 8-VSB used by the ATCS digital television system [14].

3.2 Frequency Shift Keying (FSK)

Frequency Shift Keying is the digital counterpart of analog FM. The frequency is shifted

between two or more predetermined values. Minimum Shift Keying (MSK) is a subgroup of

FSK where the frequencies used differs by half the data bit rate. This will assure that the

transitions between the different frequencies will be smooth. FSK is used by many of all the

worlds’ telephone companies.

3.3 Phase Shift Keying (PSK)

Phase Shift Keying is a modulations technique that shifts period of the wave, the so called wave

carrier. There are three major examples of PSK, Binary Phase Shift Keying (BPSK) which uses

two phases, Quadrature Phase Shift Keying (QPSK) and Differential Phase Shift Keying (DPSK)

which depends on the difference between successive phases. There are also extended versions of

PSK, such as Offset QPSK (OQPSK) and π/4–QPSK.

The modulation schemes are representing digital data by using a finite number of distinct signal

constellations. Every single phase is made by a unique pattern of binary bits; where usually each

phase encodes an equal number of bits. The symbol that is formed by a certain bit pattern is then

represented by a particular phase. The demodulator, which is an electronic circuit that recovers

information content from the carrier wave of the signal, determines the phase and decodes the

symbol it represents so that the original data is recovered. This system is termed coherent,

meaning the receiver needs to be able to compare the phase of the received signal to a reference

signal.

There are various examples of PSK applications in several existing technologies. The most

popular is wireless Local Area Network (LAN) that uses a variety of different PSK depending on

the required data-rate. If the data rate is 1 Mbit/s, DBPSK is used. Should the data rate increase

to 2 Mbit/s, it would be appropriate to use DPSK. At data rates over 5.5 Mbit/s to 11 Mbit/s

QPSK is employed, but with some complementary code keying. Higher speed wireless LAN use

other coding techniques.

10

Radio Frequency Identification (RFID), a small tag, which is used in credit cards, passports,

product tracking and automotive, is using BPSK because of its simplicity for low-cost passive

transmitters. The Bluetooth technique is using π/4–QPSK for low data rates and 8-DPSK for

higher data rates when the link between the two devices is satisfactorily robust. ZigBee also

relies on PSK, but operates in only two frequency bands: 868–915 MHz where it employs BPSK

and at 2.4GHz where it uses OQPSK.

3.4 Binary Phase Shift Keying (BPSK)

BPSK is the easiest form of PSK, which uses two phases separated by 180 degrees. The coding

can be pictured as constellation points in the complex plane. In BPSK the constellation points are

positioned on the real axis, see figure 3.1. Because of the 180 degree phase difference it has

robust characteristics which make decoding easy. It’s not suitable for high data rate applications

for the reason that is only able to send 1 bit/symbol.

Figure 3.1 - Illustration of the BPSK constellation

3.5 Quadrature Phase Shift Keying (QPSK)

In QPSK there are four constellation points placed with equal distance to center, see figure 3.2.

QPSK enables the use of either double data rate, compared to a BPSK system, while maintaining

the same bandwidth, or the same data rate as BPSK but with less bandwidth.

11

Figure 3.2 - Illustration of the QPSK constellation

3.6 Differential Phase Shift Keying (DPSK)

This method is an alternative method of demodulation and it is used for signals that have been

encoded differentially. By ignoring carrier-phase ambiguity and demodulating, the phase

between two received signals is compared and used to decode the data.

3.7 Offset QPSK (OQPSK)

This method is another variant of QPSK which uses 4 different phase values for coding. In

QPSK there is a possibility for the phase to jump 180 degrees at a time. When the signal is low

pass filtered, as it is in for example transmitters, this causes large amplitude fluctuations. In

OQPSK this phenomenon is prevented by offsetting the changes in the I- and Q-channels by one

bit-period, which results in a maximum phase jump of 90 degrees.

3.8 Quadrature Amplitude Modulation (QAM)

QAM uses both phase shift and amplitude modulation. By using the amplitude as an extra degree

of freedom every symbol can contain more information. Information is thus stored both in the

phase and length of the transferred vector. There are several types of QAM modulation, such as

16-QAM that can assume 12 phases and three amplitude sizes, see figure 3.3. The most common

type used for digital TV is 64-QAM which can transfer 64 different symbols, each containing 6

bits. Distribution of the 6-bits is done with Gray Coding, changing only one bit at a time for

adjacent symbols. Should the decoder miss one bit group, the error would probably only be one

misread bit [2], [3].

12

Figure 3.3 - 16-QAM (4x4) constellation pattern

3.9 Code Division Multiplexing (CDM)

CDM is one of the techniques used in the ISDB standard for frequencies at 2.6 GHz and it is

used as an access technology named Code Division Multiple Access (CDMA). Its main

application is in the Universal Mobile Telecommunications System (UMTS) which is the

standard for third generation mobile phone systems. Another field is within Global Positioning

Systems (GPS) where it is regularly used. Other multiple access techniques are Time Division

Multiple Access (TDMA) and Frequency Division Multiple Access (FDMA).

CDM transmits bits in each channel as a coded channel specific sequence of pulses. The coding

is accomplished by transmitting a certain time-dependent series of short pulses, which later are

placed in a chip within larger bit time, see figure 3.4. Each channel has a different code and can

be sent on the same fiber and be asynchronously de-multiplexed.

3.10 Orthogonal Frequency Division Multi

OFDM is mainly used in digital communications transmitting data on multiple carriers over one

single communications channel. This technique is used in wireless applications, such as cable

modems, digital radio, Digital Subscriber Line (

radio and television, wireless LAN and many more.

In OFDM, the stream of bits to be transmitted is split up into multiple data streams and is

transmitted using multiple data BPSK carriers at

wireless communications channel.

data stream subcarriers. The available frequency spectrum is split up into several sub channels in

form of cosine and sine waves and each

converting a sub-carrier using a modulation technique such as PSK, QAM. The frequencies are

chosen so the modulated data streams are orthogonal to each other,

send several signals over the same medium without any interference

13

Figure 3.4 - Code Division Multiplexing

Orthogonal Frequency Division Multiplexing (OFDM)



Digital Subscriber Line (DSL), power line technology, terrestrial digital

radio and television, wireless LAN and many more.


transmitted using multiple data BPSK carriers at a certain frequency over the same wired or

wireless communications channel. However, any digital modulation method can be used for


form of cosine and sine waves and each symbol is transmitted over one sub

carrier using a modulation technique such as PSK, QAM. The frequencies are

data streams are orthogonal to each other, which results in the ability to

the same medium without any interference [3].



power line technology, terrestrial digital


over the same wired or

ny digital modulation method can be used for the


is transmitted over one sub-channel by

carrier using a modulation technique such as PSK, QAM. The frequencies are

which results in the ability to

14

15

4 Digital television systems

Digital television transmission systems differ around the world. In Japan the Integrated Service

Digital Broadcasting (ISDB) system is standard [15]. In USA, Advanced Television Systems

Committee (ATSC) is the main standard [14] and in Europe the Digital Video Broadcasting

(DVB) standard [16] is most widespread. The next few chapters will compile a survey of the

mentioned standards.

4.1 ISDB

ISDB were created to transmit signals from radio and television stations in the new digital format

era. The ISDB standards are ISDB-S (satellite), ISDB-T (terrestrial), ISDB-C (cable), ISDB-Tsb

(terrestrial digital sound broadcasting) and a mobile broadcasting standard for the 2.6 GHz band.

For mobile reception in TV bands ISDB-S and ISDB-Tsb formats are often used. This concept

allows multiple channels of data to be transmitted at the same time, which is very similar to the

digital radio system Eureka 147, which calls each group of stations on a transmitter an ensemble

[15]. This system is also similar to European multi channel digital TV standard DVB-T.

Compression of video and audio in ISDB is done with MPEG-2. The same compression system

is used in some of the American and European standards. Other compression methods which are

used in both DVB and ISDB are Joint Picture Experts Group (JPEG) and Moving Pictures

Experts Group (MPEG-4). ISDB is split up in different modulations, due to different

requirements of different frequency bands and media. PSK-modulation is used in the satellite

standard ISDB-S for the 12 GHz band. CDM-modulation is used in the 2.6 GHz band for digital

sound broadcasting and the terrestrial standard ISDB-T is using COFDM modulation with

PSK/QAM. The most used interaction that ISDB uses, besides audio and video, is with Internet

as a return channel over several other medias, such as mobile phones, wireless LAN, telephone

line modem etc.

4.1.1 ISDB-Satellite

Japan has been using the DVB-S standard since 1996, but some of the Japanese broadcasters

were not satisfied. The requirements were HTDV capability, interactive services, network access

and effective frequency utilization. The DVB-S standard can only transmit 34 Mbit/s with a

single satellite transponder, which means the transponder only can send one High Definition

Television (HDTV) channel. The Japanese satellite has four vacant transponders. The unused

transponders were used to try out new transmitter techniques which led to the development of

ISDB-S, a standard that can transmit as much as 51Mbit/s with a single transponder. That means

the efficiency is increased 1.5 times and that one transponder can transmit two HDTV channels,

which would be a groundbreaking achievement for future companies and businesses.

Satellite digital broadcasting is using phase shift keying modulations such as QPSK and BPSK.

For audio and video coding and multiplexing, MPEG-2 encoding technology is applied. The

frequency range for BS digital broadcasting is 11.7 to 12.2 GHz and for wide band CS digital

broadcasting the frequency is between 12.2 and 12.75 GHz. Transmission bit rate are 51 Mbit/s

and 40 Mbit/s respectively. However, the 34.5MHz transmission bandwidth is the same.

16

4.1.2 ISDB-Terrestrial

HDTV began its journey and development around 1960s but the standard was suggested around

mid 1970s. The television camera, high definition Cathode-Ray Tube (CRT), video tape receiver

and other editing equipment came at 1980s. At the same time a company named Nippon Hōsō

Kyōkai (NHK) had developed MUltiple sub-nyquist Sampling Encoding (MUSE), which was the

first HDTV video compression and transmission system. MUSE implements a digital video

compression system, but one of the drawbacks is that the digital signal must be converted by a

digital-to-analog converter in order to modulate the signal. Both Europe and USA were

impressed by the technology which led to the development of the ATSC standard for USA and

the DVB standard for Europe.

There are several characteristics that distinguish ISDB-T from other standards. For example both

a HDTV channel and a mobile phone channel can be transmitted within the 6 MHz bandwidth

usually reserved for standard TV. The bandwidth is enough for sending one HDTV channel or

two, maybe three, multiplexed Standard Definition Television (SDTV) channels. It provides

interactive services with data broadcasting and Electronic Program Guides (EPG). ISDB-T also

provides robustness to multipath interference called “ghosting”, co-channel analog television

interference and to impulse noises coming from power lines and motor vehicles in suburbs.

ISDB-T first commercial use was adopted in Japan at the end of 2003, taking over a market of

about 100 millions television sets. Brazil has also transferred from the analogue TV system

(PAL-M) to the ISTB-T standard, calling it SBTVD-T. For development of technology in Latin

America, other countries such as Argentina, Venezuela are thinking of cooperating with Brazil in

order to prevent import from other standards and countries. In addition, reception tests have

shown better results using ISDB-T compared to both ATSC and DVB. The tests showed that

ISDB was the most flexible solution for better answering the necessities of mobility and

portability.

Terrestrial digital broadcasting uses amplitude modulation like 64QAM-OFDM, 16QAM-

OFDM, 16QAM-OFDM and DQPSK-OFDM. For audio coding, video coding and multiplexing,

MPEG-2 encoding technology is mostly used, but MPEG-4 is used for some mobile phone

segments [15].

4.2 ATSC

The ATSC group [14] helped develop the new digital television standard for the United States of

America. Several other countries such as Canada, Korea and Mexico have adapted to this

standard. The ATSC standard covers both terrestrial, cable and satellite transmissions. For

terrestrial use the 8-VSB modulation form is applied. When using cable the signal-to-noise ratio

is higher, and more advanced forms of modulation can be used, such as 16-VSB and 256-QAM,

to capsule more information on the same channel bandwidth. As the ATSC terrestrial standard is

almost identical to the cable standard just ATSC-T will be described in greater depth in this text.

Generally the video content is digitalized by MPEG-2 encoding, error correction added and

finally the modulation form and RF up-converter are used to transmit the information.

17

4.2.1 ATSC-Terrestrial

Digital information in VSB is transmitted exclusively in the amplitude envelope unlike other

modulation formats where also the phase can contain information. 8-VSB used by standard

ATSC handles eight amplitude levels. The spectrum of digital 8-VSB is contained in a 6 MHz

channel as in the National Television Standard Committee (NTSC) television standard. With the

use of data encoders the data is transformed and will get almost noise-like characteristics.

Therefore the digital spectrum is, in contradiction to the NTSC spectrum, flat throughout most of

the band, making efficient use of the channel bandwidth.

From the MPEG-2 video output stream to the data stream it is required to add error correction

codes among other things, see figure 4.1. Every block in the figure will be explained in the

following text.

Figure 4.1 - Functional block diagram of the ATSC-T modulator and encoder

The MPEG-2 package from the video converter contains 187 bytes plus a packet sync byte. The

packet sync byte is removed and the data is further processed. The Data randomizer is used to

get a random data stream that will be almost noise-like to make efficient use of the channel

space. For example, if our data contained repetitive patterns some of the parts of the spectrum

would be overused, leaving holes in other parts of the spectrum. The data randomizer is built up

by a Pseudo Random Binary Sequence (PRBS) generator. The code used for randomization is

also stored in the receiver for proper recovery of the data values. The Reed-Solomon encoder

adds redundant information for error correction purposes. Adding redundant information in this

way is in general terms called Forward Error Correction (FEC). The extra information is used if

the data stream sent in any way is corrupted by for example atmospheric noise, multipath

propagation, signal fades and transmitter nonlinearities. The error correction bits can compensate

for this up to a certain point. All the 187 bytes from the MPEG-2 packet is used by the Reed-

Solomon encoder. The encoder adds an extra 20 parity bytes to the tail of the packet. The

receiver compares the MPEG-2 packet with the parity bytes in order to determine the validity of

the received data. Up to 10 error bytes can be found and corrected before the entire packet must

be discarded. The Data interleaver incorporates a buffer and combines bytes from different

packages (according to a specified pattern, which is known to the receiver) to account for burst-

type interference effects in the channel. If a sent segment is lost due to burst interference this

means that only a part of the original packet is lost and this can often be restored by the Reed-

Solomon correction code. The Trellis code is an evolving error correction code and as the Reed-

Solomon code it is another FEC. Every incoming 2-byte word is compared to the past history of

2-byte words to generate a 3-byte word describing the changes from the foregoing transitions.

18

The receiver can correct potential errors by looking at the past history of received 3-byte words.

Help signals such as ATSC pilot, segment sync and frame sync signal must be applied before

modulation, helping the receiver to place the received packets in the correct order. In the 8-VSB

modulator the 3-byte words from the Trellis encoder are converted to an 8-level discrete

amplitude and modulated according to AM-VSB. The signal is amplified and adjusted for the

used channel and sent to the receivers.

4.2.2 ATSC-Satellite

The modulation forms used in the satellite standard are QPSK, 8PSK and 16QAM. With the use

of more advanced modulation techniques the consumed bandwidth is less, for a constant bit rate,

than for QPSK. The bit rate is increased as well as overall performance. The main drawback is

the use of more power to achieve the same level of performance. For a functional block diagram,

see figure 4.2.

Figure 4.2 - Functional block diagram

The standard relies on previous work done within the DVB project, especially [21] and [22]. The

main distinction between DVB-S and the ATSC-S system is the use of arbitrary data streams.

The ATSC-S system can use MPEG data streams or an arbitrary constant rate data stream. In the

case of the arbitrary data stream, the modulator packetizes 187 bytes together with a 0x47hex

sync byte to form a MPEG-like package. In the demodulator end this sync byte is discarded to

deliver the arbitrary data stream to the output. Modulation and coding is done in the same way

according to the DVB standards [21] and [22]. See the DVB-S section for more information.

4.3 DVB

The DVB project is a cooperation of about 250-300 companies worldwide. It is an open standard

of European origin but now spreading over the world. The specifications proposed by the DVB

alliance are passed on to the European Broadcasting Union (EBU/GENELEC/ETSI) Joint

Technical Committee for approval and later standardization. With the close cooperation with the

industry the DVB specifications has been market driven and the development has been done with

the finished product in mind. This has probably contributed greatly to the DVB success.

19

There are several digital television standards developed by the DVB project group, among which

are:

• DVB-Satellite

• DVB-Cable

• DVB-Terrestrial

• DVB-Handheld

• DVB-Satellite 2nd Generation

4.3.1 DVB-Satellite

The DVB-S standard was published in 1993 by the EBU commission and it became standard for

broadcasting of digital television over satellite. Its main purpose is to prepare a digital MPEG

transport for satellite transmission. The standard for DVB-S is similar to the standard for DVB-

T, except from using the modulation technique QPSK instead of COFDM. Quaternary PSK

modulations technique is more suitable for satellite transmission due to higher bandwidth with

small and weak noise transferring channel. The key features in DVB-S are: Changing encoding

parameters in real time, Variable Coding and Modulation (VCM) and Adaptive Coding and

Modulation (ACM). For a functional block diagram, see figure 4.3.

Figure 4.3 – DVB-S functional block diagram

4.3.1.1 Signal coding and channel adaption

Video, audio and data information, so called bit waves, are received in the program MUltipleXer

(MUX). Different packages form a transport stream together with Program Specific Information

(PSI). Then the transportation MUX combines the different TV-channels transportation streams

to a common Transport Stream (TS), where each stream is supplied with its own identification, a

transport- ID, “TS-id”. A device for energy spread is used for evening out the sequences with

binary ones and zeros, evening out the energy located in the transmission channel. The signal

move on to the device called Reed-Solomon encoder. This type of encoding is called RS (204,

188 t=8), meaning 16 additional bytes is added to the 188 bit package which can correct up to

eight incorrect bytes. Burst type interference can be managed with the interleaving encoder,

where streams permute byte by byte into 12 different streams. The first has no delay time; the

second is delayed with 17 bytes, the third with 34 etc. Viterbi encoding, which is a FEC, gives

protection against random noise. Least safety is given by 7 8⁄ which adds 1 extra control bit for

every 7 bits. Highest safety gives 1 2⁄ which doubles the number of bits. In satellite-TV context a

FEC with value of 1 2⁄ is very unusual. After Viterbi encoding the data stream is sent for

encoding into Gray coded QPSK, where only one bit is changed between adjacent pair of bits.

4.3.1.2 Applications and Market

The DVB-S standard is mostly used in broadcasting applications, but has other purposes like

point-to-point transmission. The reason why it became such a great success in the broadcasting

market is due to inexpensive silicon used in the receivers. The market for DVB-S is divided in to

20

two segment, DTH (Direct-to-home) and Contribution and Distribution. The DTH-segment is

referred to TV for home reception, Satellite Master Antenna Television (SMATV), Internet

Access/Content delivery, Interactive TV and many more. Contribution-and-Distribution-segment

includes Satellite News Gathering (DSNG), cable feeds, flight entertainment and Internet content

distribution/trunking.

4.3.2 DVB-Satellite 2nd generation

DVB-S2 was developed in the DVB Project in 2003 and became the second generation

specification for satellite broadcasting. Combining a variety of modulation formats such as

QPSK, 8PSK, 16APSK and 32APSK with the latest developments in channel coding, it became

a great benefit in interactive applications. Broadcasting services are managed with DVB-S and

with the flexible VCM. Broadcasting Services (BS) offers great levels of protection for both

robust SDTV and less robust HDTV. Along with the existing DVB Return Channel Standard

(DVB-RCS), Interactive Services (IS) is designed to operate in both CCM and ACM modes.

ACM mode is here used to enable receiving station to control the protection around the traffic

addressed around it. DTV and DSNG uses either CCM or ACM modes for facilitating point-to-

point or point-to-multipoint communications of single or multiple MPEG transport streams.

ACM is implemented for optimization of transmission parameters for each individual user

depending on the path conditions. Even backwards-compatible modes are used for DVB-S Set-

Top-Boxes (STB) for continuous work during the transitional period.

There are three concepts in the DVB-S2 standard: best transmission performance, total flexibility

and reasonable receiver complexity. Using the recent techniques in channel coding and

modulation, DVB-S2 can achieve the best performance complexity trade-off, 30 % gain capacity

over DVB-S. Due to its flexibility, DVB-S2 can cope with any existing satellite transponder

characteristics with a large variety of spectrum efficiency and associated C/N requirements.

Being not limited to MPEG-2 video and audio coding, it is designed to handle a variety of

advanced audio-video formats. It can accommodate any input stream format, single and multiple

Transport Streams, continuous bit-streams, ACM and IP packets.

4.3.2.1 Modulation

DVB-S2 is using an advanced FEC as the key subsystem for achieving excellent performance, in

the presence of high levels of noise and interference. Low Density Parity Check (LDPC) is

selected as a coding technique, and offers a minimum distance from Shannon limit on the linear

AWGN channel. QPSK in combination with some additional codes have shown that the signal

can be retrieved even when the level is below the noise floor. Avoiding the error floors at low Bit

Error Rates (BER) is done by concatenated Bose-Chaudhuri-Hocquenghem (BCH) outer codes.

Four different modulations modes are used for the transmitted payload, as shown in figure 4.4.

QPSK and 8PSK are used in broadcasting applications due to its constant envelopment and usage

in non-linear satellite transponders driven near saturation. 16APSK and 32APSK which mainly

are used at professional applications can be used for broadcasting, but needs a higher level of

available C/N and adoption of advanced pre-distortion methods. The spectrum efficiency is

much greater, whilst the power-efficient modes are not. By placing the constellations points on

circles it optimizes the constellation to operate over a non-linear transponder, even though their

performance on a linear channel is comparable with performances of both 16QAM and 32QAM.

21

Spectrum efficiencies from 0.5 to 4.5 bit/symbol are available by selecting specific modulation

and code rates, and can be chosen dependent on the capabilities and restrictions of the satellite

transponder used [7]. To determine the spectrum shapes DVB-S2 can choose from three “roll-

off-factors”, 0.35 as in DVB-S or 0.25 and 0.20 for tighter bandwidth restrictions.

Figure 4.4 - Four possible modulation constellations.

From left to right and up to down: QPSK, 8PSK, 16APSK and 32APSK.

DVB-S already installed receivers are giving some major problems for many broadcaster,

especially where the receivers are subsidiary and for free-to-air public service. In that case, DVB

receivers need to be backward compatible in order to allow receivers to continue operating while

still providing additional capacity and services to new, advanced receivers. After the migration

process when most users have migrated to DVB-S2, the transmitted signal can be modified to a

non-backward compatible mode, exploiting the full potential of DVB-S2. Backwards

compatibility is implemented by high priority and low priority transport streams. They are

combined at modulation symbol level on a non-uniform 8PSK constellation. The low priority

signal is coded with BCH and LDPC. The high priority signal is defined by QPSK modulation,

where the single bit from the DVB-S2 LDPC encoder sets an additional rotation before

transmission. The result of having quasi-constant envelopment makes it possible to transmit on a

single transponder driven near saturation.

4.3.2.2 ACM

ACM is used in wireless communication to decide the matching of modulation, coding, signal-

and protocol-parameters, especially in radio links. That includes interference caused by signals

coming from other transmitters, the sensitivity of the receiver and the available transmitter power

margin. Considering interference distribution, ACM uses the entire C/I range. The interference

together with other channels, noise and background effects have a direct influence in the quality

of the received signal. In atmospheric conditions, ACM maximizes instantaneous data-rate as a

function of time versus location. Depending on the location and modulation, such as 8PSK,

QPSK, 16APSK or 32APSK, the system can on its own select which optimum couple is most

22

convenient. ACM render it possible for a return channel to be available from each receiving site

to the transmit site, making it possible to modify the coding rate and modulation scheme for

every single frame according to the measured channel conditions where the frame must be

received. The return channel dynamically reports the receiving conditions at each receiving site.

All depends on the weather. Should the probability of clear weather be 95 %, the C/No-ratio will

vary between 84.3 and 88.1 dBHz. These values give information on which modulation and

coding will be used, such as 16APSK 3 4⁄ or 16APSK 5 6⁄ . These coding and modulations are

ranged from 8PSK 3 4⁄ to 16APSK 5 6⁄ . Compared with DVB-S, efficiency is improved with

130 % with ACM in the DVB-S2 system [8].

4.3.2.3 VCM

VCM transmits different services on the same carrier using their own modulations scheme and

coding rate. VCM is stronger when protection levels of other services are not needed, for

example in case of fade rain it is acceptable to lose a secondary channel. The same principle can

be used when different services are intended for different stations with different average

receiving conditions. If the probability of clear weather is 99.85 %, the Co/No-ratio will be

between 6.0 and 10.6 dB. These coding and modulations are ranged from QPSK 5 6⁄ to 16APSK 2 3⁄ . Compared with DVB-S, efficiency is improved by 65.7 % with VCM in DVB-S2 system

[8].

4.3.2.4 CCM

In CCM all frames are modulated and coded with fixed parameters and it is the simplest mode in

DVB-S2. LDPC code is used as inner error correction code, compared to DVB-S which uses

Reed Solomon. In comparison with DVB-S, the efficiency is improved by 29 % with CCM in a

DVB-S2 system [8].

4.3.2.5 Performance of the DVB-S2 system

The system has the characteristics to operate at C/N-ratios from -2.4 dB (using QPSK 1/4) to 16

dB (using 32APSK), depending on the selected code rate and modulation. The distance from the

Shannon limit is 0.7 dB to 1.2 dB. Under the same transmission conditions as for DVB-S, DVB-

S2 has 2-2.5 dB more robust reception for the same spectrum. A DVB-S2 system can be used in

both “single-carrier-per-transponder” and “multi-carriers-per-transponder” FDM configurations.

4.3.3 DVB-Cable

After a year, in 1994, specifications for digital cable networks system were proposed. It uses the

more sophisticated modulation technique QAM to squeeze in more information at a specified

bandwidth.

The carriers can be modulated according to the following constellations:

• 16-QAM

• 32-QAM

• 64-QAM

• 128-QAM

• 256-QAM

23

In Sweden the 16-QAM and 64-QAM modulation forms are mostly used. None or very few

receivers are equipped with modulators that can decode 128- or 256-QAM.

When comparing different channels, transportation of information in a cable is least prone to

addition of errors. Noise, damping and reflections are the main problem areas when transmitting

over a cable network. The Viterbi algorithm, used in for example DVB-T and DVB-S, is

therefore excluded from the DVB-C standard, see figure 4.5. The use of randomization, Reed-

Solomon encoding, and interleaving fulfills the need for protection of the data stream. Protection

of burst errors is achieved by byte interleaving. Differential encoding of the modulation

constellations is used to get rotational invariant reception of the phase differences used in QAM.

Figure 4.5 - DVB-C functional system block diagram

4.3.4 DVB-Terrestrial

In 1998 the terrestrial system was standardized. Due to harder environment, like multipath

propagation and different noise characteristics, the terrestrial system needed to be more complex.

The use of the existing VHF and UHF spectrum allocation used by the old analogue system

places constraints on bandwidth and protection against Co-Channel Interference (CCI) and

Adjacent Channel Interference (ACI) with the existing analogue transmissions of

PAL/SECAM/NTSC. To cope with multipath propagation, COFDM is used with a guard

interval, chosen such that the interference from multiple terrestrial signal paths are of less

concern. The guard interval can be flexible chosen to allow for different network topologies and

frequency efficiency. For a functional block diagram, see figure 4.6.

Figure 4.6 - DVB-T functional block diagram

24

The transmission system has two modes of operation, either the 2K mode or the 8K. The two

modes actually specify the FFT length of the modulator, and thus the number of carriers used by

COFDM. The 2K mode uses 1705 carriers and the 8K mode 6817 carriers. For synchronization

purposes some carrier positions always contain pilot carriers. Other pilot carriers are also present

but are scattered in frequency and time in a predefined way. Some carriers are Transmission

Parameter Signaling (TPS) carriers. The TPS carriers are always modulated by differential 2-

PSK, and these convey information on:

• Modulation used, including the 9 (spacing of the different QAM “points”) value of the

QAM constellation pattern

• Hierarchy information

• Guard interval

• Inner code rates

• 2K or 8K transmission mode

• Frame number in super-frame (one super-frame contains 4 · 68 COFDM frames)

• Cell identification

Several different modulation forms can be chosen for the video transport, trading between bit

rate and ruggedness.

Carriers can be modulated in the following constellations:

• QPSK

• 16-QAM 9 1, non-hierarchical and hierarchical

• 64-QAM9 1, non-hierarchical and hierarchical

• Non-uniform 16-QAM 9 2, 4 • Non-uniform 64-QAM 9 2, 4

9 is the minimum distance separating two constellation points carrying different HP-bit values

divided by the minimum distance separating any two constellation points.

With the use of 16-QAM or 64-QAM additional information can be buried in QPSK like fashion,

shown in figure 4.7.

25

Figure 4.7- QPSK information buried in QAM

Two MPEG streams can be sent simultaneously, one low- and one high priority stream. The high

priority stream (low bit rate) is mapped as QPSK on the low priority stream modulated as either

16-QAM or 64-QAM. The high priority stream is thus more rugged against noisy environments

and the broadcaster can choose to send the same program with both a high and a low bit rate. A

receiver in very noisy environments, which has problem receiving the low priority stream, allows

switching to the high priority stream. The drawback of this implementation is found on the

receiver end. The receiver must adapt to the different transmissions by the broadcaster. The

adaption to new coding and mapping when switching between one layer and another takes some

time to complete and thus instantaneous switching cannot be done. Usually video and sound

freezes a short amount of time (around 0.5s) before lock on the new data stream have been

accomplished.

4.3.4.1 Comparisons between ATSC-T and DVB-T in Taiwan

The standard for analog television in Taiwan has been NTSC. They originally planned to directly

adapt to the new ATSC-T standard but when Sinclair Broadcasting demonstrated COFDM

(DVB-T) vs. 8-VSB (ATSC) reception in USA 1999, this raised deep concerns about the

qualities of the ATSC system. In-door, out-door and mobile reception was tested in Taiwan and

in almost all cases the DVB-T system outperformed ATSC-T. After the test trials, the Taiwanese

Directorate General of Telecommunications proposed the DVB-T as the new DTV system, and

this was adopted by all terrestrial broadcasters in 2001. The main reason for the excellent

reception of DVB-T originates from the fact that it uses OFDM and resists multipath propagation

[9].

4.3.5 DVB-Handheld

DVB-H is essentially the same as DVB-T, but a 4K transmission mode is added. The 4K mode

offers a trade-off between transmission cell size and mobile reception capabilities. DVB-H uses

the exceptional features of the DVB-T standard, as the possibility to receive broadcasting

services with portable devices and even in cars. The standard was published by ETSI in

November 2004.

26

One of the biggest problems with handhelds is the limited available power. Low power

consumption is therefore necessary to obtain reasonable usage and standby cycles. Another fact

which makes transmission to handhelds a very demanding task is the high mobility of the

handheld. Access to services must be possible whether located indoor, outdoor or in a moving

vehicle. The high mobility, small antenna and interference from for example GSM mobile radio

signals transmitted and received from within the same device, places hard constraints on the

transmission/receiver. The close affinity between DVB-T and DVB-H makes re-use of the same

transmission equipment an attractive possibility.

The power-saving technique used is called time slicing. The algorithm is based on time-

multiplexed transmission of different services, which can be used to turn off the front end in time

slots where no relevant data are sent, see figure 4.8. The power saving may be more than 90 %,

compared with conventional DVB-T front ends [10]. Another advantage of time-slicing is the

ability to seamlessly move between different adjacent radio-cells, which can be done in the

power-saving period.

Figure 4.8 - DVB-H time slicing compared to DVB-T transmissions

An enhanced error-protection scheme, Multi-Protocol Encapsulation – Forward Error Correction

(MPE-FEC), is used for reliable transmission in poor reception conditions. To be backwards

compatible with DVB-T (for transmission purposes) time slicing and MPE-FEC is put on the

protocol layer above the DVB transport stream. The system is based on the Internet Protocol

(IP). The IP data are embedded in the MPEG-2 transport stream used as the base layer by means

of the Multi-Protocol Encapsulation (MPE). This makes effortless migration to other IP-based

networks.

27

5 Satellite reception

To use the available frequency spectrum in satellite transmissions in a more effective way,

several satellites transmit at the same frequencies. This places constraints on the reception tool,

the aerial, on earth. The aerial must have a high directivity to receive just the right signal. This is

achieved by using a parabolic reflector, often called dish, concentrating the received signal to

one point. To properly retrieve the signal in the focal point a Low Noise Block down-converter

(LNB) is used [2].

Satellites transmit at fairly high frequencies (in the multiple gigahertz regions). When receiving

radio signals at this high frequencies it must be done out-doors as the signal is easily damped out

when sent through different materials. Coaxial cables are used for transportation from the dish to

the set-top-box. Because of the high attenuation of gigahertz frequencies in coaxial cables the

received signal must be down-converted to a lower Intermediate Frequency (IF). This also has

the effect that the Radio Frequency (RF) front end of the STB is easier to construct. The LNB

serves multiple purposes:

• Collect the signal at the focal point

• Amplify the received signal

• Down-convert the received spectrum to lower frequencies

The satellite transponder can transmit at two different frequency bands with either horizontal or

vertical polarization of the microwaves. This means that one LNB can receive four different

signals placed in the same position. In the early beginning of satellite transmissions one LNB

could just receive one band and polarization switching had to be done with an outer mechanical

or electrical switch. Nowadays there exist a so called universal LNB that has all these features

built in. Switching between the different modes can easily be done with the universal LNB by

varying a voltage applied on the same coaxial cable that transports the content to the set-top-box.

The different frequency bands and the polarization are switched according to table 5.1.

Voltage applied 13V 18V 13V + 22kHz 18V + 22kHz

Low band, vertical. pol. X

Low band, horizontal pol. X

High band, vertical pol. X

High Band, horizontal pol. X Table 5.1 - LNB switch modes

28

29

6 Receiver functions

The Radio Frequency (RF) converted to Intermediate Frequency (IF) by the LNB must be

processed in some way to decode the channel adapted signal. In most receiver systems this is

done by a front-end tuner converting the IF to I- and Q-baseband channels. The I- and Q-

channels are then decoded in the demodulator and converted to the transport stream known by

the MPEG video processor. In figure 6.1, a general overview of a receiver is presented.

Figure 6.1 - General receiver overview

6.1 Tuner

The main purpose of the tuner circuit is to lock on a frequency of interest and transmit I- and Q-

channels to the demodulator. The tuner has a LNA with Automatic Gain Control (AGC) to

amplify the weak signal to a level suitable to the mixers and output circuitry. The tuner

communicates with the demodulator and outer world, often through an I2C interface. Through

the interface common settings such as frequency band, gain, filtering or lock detect can be sent

or read.

6.2 Demodulator

To cope with noise, signal strength loss and other channel artifacts the transmitter added lots of

coding to the data stream. The task of the demodulator is to decode the data stream and forward

the MPEG transport stream to the MPEG video and applications processor. The LDPC decoder

used in DVB-S2 is a highly complex design taking up lots of area and computational effort

which can be seen in the current consumption.

6.3 LNB Driver

The LNB receives the signal that is gathered with the parabola liked antenna and focused into the

LNB. Inside the LNB the converter takes the incoming signal with high frequency,

approximately 12 GHz, and converts it to about 1.5 GHz, called the first signal conversion. The

signal is carried to the receiver where the signal is improved by a low noise amplifier. The other

purpose of the LNB is to change polarization type, either horizontal or vertical, or change

operative band by command signals controlled by the receiver. Other ways to steer the LBN is

by use of the bidirectional DiSEqC standard which is a communication protocol for frequency

and motor position adjustment. The DiSEqC signal is a pulse width modulated 22 kHz signal

with amplitude between 0.3 and 0.6V aligned to the supply voltage. The LNB driver is a

component or system that handles the communication and powering of the LNB.

30

31

7 Available chipsets for DVB-S2 reception

To make a cost efficient and physically small design the needs for integrated circuits apply. In

this chapter the available chipsets for DVB-S2 reception (spring 2007) is presented in short form.

7.1 ST STB6100

The STB6100 is an 8PSK/QPSK direct conversion tuner Integrated Circuit (IC) which is a highly

integrated, low-cost tuner for DVB-S2 broadcasting satellite applications. The device has an

LNA, down-converting mixers, gain control, on-chip VCO and a low noise Phase Locked Loop

(PLL). The tuner is controlled by two interfacing wires designed to minimize external Bill Of

Material (BOM) to reduce manufacturing costs and simplify board layout. The reduced number

of components gives system stability and reliability. The capacity is increased with 30% by the

DVB-S2 standard and renders possible high data rate applications such as HDTV and broadband

Internet.

7.1.1 Features

• Direct conversion

• Input range 950 to 2150 MHz

• 2-wire I2C serial interface

• Low power consumption

• Differential I/Q outputs

• 8PSK/QPSK support

7.2 ST STB0899

The STB0899 is an advanced demodulator which enables set-top boxes to produce base-band

DVB-S2 signals into digital video transport stream data. The demodulator supports QPSK in

DIRECTV and DVB-S up to 45 Msymbols/s in legacy transmission, plus 8PSK in DVB-S2

transmissions (up to 30 Msymbols/s). The demodulator is using the modulation technique 8PSK

with LDPC/BHC and Reed-Solomon/Viterbi decoding. The FEC has integrated BER and

Package Error Rate (PER) computation to evaluate reception quality. It accepts both baseband

differentials I and Q signals using AD-converters.

7.2.1 Features

• Compatible with direct conversion tuners

• QPSK, 8PSK, 16 and 32 APSK

• Legacy DVB-S and DVB-S2

• I2C serial bus interface

• Multi-tap equalizer for RF reflection removal

• Power-saving features

• DiSEqC 2.0 22 kHz to 100 kHz interface

32

7.3 Conexant CX24118A

The CX24118A is digital satellite tuner RF IC which is used for high-volume digital video, audio

and data receiving. The device increases phase noise performance, has low implementation loss

and is also using the modulation techniques required for DVB-S2.

7.3.1 Features

• Single-chip RF-to-baseband satellite receiver

• Zero-IF architecture

• Variable baseband filters

• Low power consumption

• Auto-tuning machine for elimination of software calibration

• Lead-free package

7.4 Conexant CX24116

The CX24116 is used both for demodulation and FEC decoding. The demodulator is using the

modulation technique 8PSK with LDPC/BHC forward error decoding. It can demodulate and

decode transmitted signals with both the new DVB-S2 and old DVB-S specification. It

automatically searches and acquires the carrier within a 10MHz range during initial acquisition

and performs a smart search to reacquire under fade condition. The CX24116 has integrated

Signal-to-Noise Ratio (SNR) and BER monitors for performance measurements.

7.4.1 Features

• 8PSK/H-8PSK/QPSK/BPSK

• Internal microcontroller

• DiSEqC 2.x compliant

• DVB-S/DSS backward compatible

• Power-down mode

• 1.2-V core voltage, 3.3-V I/O voltage

33

7.5 Intel® CE 5038

All Intel front-end chips originally come from Zarlink Semiconductor Limited, acquired by Intel

in 1995. Since this is a single tuner chip, it must be combined with some kind of DVB-S2

demodulator chip for correct function.

7.5.1 Features

• Supports 8PSK and 16QAM modulations by high signal-to-noise and inter-modulation

performance

• Low power modes for saving energy

• RF loop-through

• Guaranteed maximum of 2 degrees integrated phase jitter over the full operational

temperature range

• Automatic and manual tuner modes

• Consumes less than 1.5 W

• Excellent immunity for adjacent channel interference

7.6 Broadcom BCM4501

This is a fully integrated single-chip solution for DVB-S and S2 reception. The single-chip

solution reduces the external component count and increases performance.

7.6.1 Features

• Single-chip solution

• I2C interface

• Dual tuner and demodulator sections

• Supports QPSK and 8PSK modulation

• Demodulators DVB-S and S2 compliant

• 208-pin MQFP package

7.7 Selected chipset

The ST microelectronics companion chip solution (STB0899 [17], STB6100 [18]) was selected

with the aid of our supervisor. The availability, price and contacts with ST made the choice easy.

Interfacing the chips is done with the I2C communication protocol, with the demodulator chip

forwarding the commands to the tuner chip. This way the communication with the tuner chip can

be turned off when not used thus reducing the risk of interference. There are in addition several

registers available for BER, Modulation Error Ratio (MER) and Carrier-to-Noise Ratio (CNR)

estimates, making tuning and reception quality evaluation of new satellites easy.

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35

8 Circuit design considerations

Before the design start there are several considerations that must be taken into account. Are there

RF sections where RF design methods must be addressed?

The following chapters present issues to consider when working with RF. The main issues are:

• EMC- Electromagnetic Compliance

EMC refers to the ability of a product to coexist in its intended electromagnetic

environment without causing or suffering functional degradation or damage.

• EMI – Electromagnetic Interference

EMI is the process where electromagnetic energy is transferred from one electronic

device to another in a non wanted manner.

Unintentional radiation is regulated by international EMC requirements and the PCB with

components must fulfill these requirements for product certification. To avoid redesign of the

PCB, EMC requirements must be thought of under the whole design phase.

8.1 RF component selection

When dealing with RF design there are numerous new parameters to take into account. The

nearly ideal passive component behavior at DC will transform to new behavior at high

frequencies. For example, lead length inductance, which is not recognized when dealing with DC

or low frequencies will greatly alter a capacitors behavior. Every component contains hidden

parasitic elements such as inductance and capacitance. In figure 8.1 the RF behavior, in terms of

ideal circuit elements, of some of the most used passive components are listed.

Figure 8.1 - Component RF characteristics

36

To successfully complete a RF design without having to redesign for EMC compliance one must

have these component behaviors in mind under the whole design process.

8.1.1 Capacitors

Capacitance is defined as the measure of the amount of electrical charge stored for a given

electric potential.

:; (8.1)

Negative and positive charges attract each other. Two metal plates, separated with an insulating

material (dielectric), connected in a circuit, will accumulate charge as a result of charge

attraction. This will result in a capacitance between these metal plates, as defined in equation 8.1.

The closer the metal plates get to each other more and more capacitance will appear. This will

also apply if the area of the plates is enlarged. As such capacitance will appear between every

conductor where charges freely can move. Even if there is just a tiny capacitance between for

example two leads of an inductor this will influence the behavior at high frequencies. As seen in

equation 8.2 the reactance of the capacitor will affect the circuit more and more as frequency

rise.

< =(· (8.2)

Even if the capacitance is very low it can affect circuit behavior if the frequency is high enough.

Designing RF circuits including capacitors is a more delicate art than choosing capacitors for DC

circuits. Different dielectric materials and capacitor sizes must be chosen for correct circuit

behavior. In a forthcoming chapter the materials used for the capacitors in this design is

presented, listed with their respective characteristics. Several characteristics within a capacitor

are temperature dependent, such as capacitance, ESR (Equivalent Series Resistance) and leakage

current (sometimes called insulation resistance). To describe losses in a capacitor the dielectric

loss tan @ is often listed.

tan @ ABCDE (8.3)

In the capacitor body leads and electrodes inductance is found. As the inductance and

capacitance are in series a series resonance occur at a frequency where the two reactances are

equal. At this frequency the impedance of the capacitor is equal to ESR. As seen in figure 8.2 the

capacitor only works as a capacitor up to the series resonance frequency. This means that

capacitors must be chosen carefully for the application they are meant to be in.

37

Figure 8.2 - Capacitor behavior as frequency rise

As the size of the capacitors shrink the lead inductance is lowered so therefore smaller surface

mount capacitors are useful at higher frequencies then bigger. This is also one of the reasons that

surface mount components are used.

8.1.1.1 Dielectrics

There are quite a few different dielectrics that can be used for capacitor design. There are

tradeoffs between for example capacitance per area, loss and stability. The capacitors used in the

design in this report use ceramic and electrolytic dielectrics.

Electrolytic capacitors have large capacitance per area ratio. They are therefore used where the

need for a big reservoir and small footprint are the main objectives. The main drawbacks are low

stability and large leakage current. Ceramic capacitors often have better characteristics but can’t

be made with as large capacitance values.

Ceramic capacitors are divided into three different classes [13]:

Class 1 materials have low dielectric constant but are highly stable and have very low losses at

high frequencies. Because of the low dielectric constant these capacitors cannot be made with

high capacitance. They are often manufactured with capacitance values in the range 0.47 pF

(single layer) – 0.1uF (multi layer). Capacitors of class 1 used in this design are of dielectric type

NP0.

Class 2 materials have higher dielectric constant but have nonlinear temperature, frequency and

voltage dependencies. They are often manufactured with capacitance values in the range 10 pF

(single layer) – 10uF (multi layer). Capacitors of class 2 used in this design are of dielectric type

X7R and X5R. For comparison NPO has a dielectric constant FG of 60 and X7R a constant of

1500. Class 2 capacitors are used as output capacitors for the buck converter later described in

this report.

38

Class 3 materials have almost the same or slightly poorer specifications as class 2 materials. The

voltage ratings are rather low but as the dielectric constant of these materials are in the range

10000 – 50000 it is possible to make small capacitors with large capacitance. Class 3 dielectrics

are not used in this report.

8.1.1.2 Capacitor ripple current

Ripple current capability for capacitors has it origin in the losses accompanied with charge and

recharge. The charge and recharge currents dissipate power in the ESR of the capacitor, which in

turn generates heat and temperature rise in the capacitor. High temperatures reduce the capacitor

reliability, and could lead to failure. ESR decreases with frequency thus increasing the current

capability of the capacitor. The reliability of capacitors depends mostly on applied voltage and

core temperature. For good reliability it is therefore mandatory keeping the capacitor voltage and

temperature down [11].

8.1.2 Inductors

Whenever a current is conveyed in a conductor a proportional magnetic field forms around the

conductor. A current change will then create a change in the magnetic field that, in turn, will try

to oppose the current change by introducing an electromotive force in the inductor. Inductance is

a measure that tells us how much the current change will be opposed by the inductor. To increase

the inductance, materials that make the magnetic field coupling more confined to the windings

are introduced. This approach has the drawback that the material has a certain point when it can’t

handle more magnetic flux (in a linear way) and the material goes into saturation. At the

saturation point and beyond the inductor changes it’s approximately linear relationship in a

highly non linear manner which in most applications is a not wanted characteristic. Therefore the

current, that creates the magnetic flux in the core material, must be held below the saturation

point specified by the manufacturer.

Most of the inductors used in this design are situated in connection with rather low frequency,

high current circuits, such as DC/DC converters. Therefore the capacitance that occurs between

the windings of the inductor is of less importance then if it would be situated in a complex RF

environment. The parameters that are of great importance are the losses in the core and leads,

and the saturation current of the core material.

Small inductors may be etched right onto the printed circuit board. In prototypes as in this report

this can be rather dumb as the designer can have been making a mistake calculating the inductor,

or some properties not fully known at the design phase may alter the behavior of the circuit.

Therefore the only RF inductor included in this design is made up of a physical component.

EMI filters are also placed on the input power lines to suppress RF and noise interference picked

up on the lines from the regulators to the circuit board.

8.1.3 Wires

In DC circuits the primary wire characteristic is resistance. At higher frequencies other

characteristics will play an important role. Parasitic elements, such as inductance and

capacitance, can in special cases interact with other components and self resonance can occur. If

the PCB trace that this resonance phenomenon occurs in is long enough, it can form an efficient

39

radiating antenna. Even if active components aren’t driving the trace into resonance, incoming

radiation may do so, thus injecting unwanted RF energy into components. The physical

dimensions of the traces are also of great concern when the wavelength becomes short in relation

to the trace length. Analysis with lumped circuit parameters cannot be used anymore. The trace

assumes the characteristic of a transmission line and for optimal signal transfer transmission

lines must be terminated in their characteristic impedance. The primary concern is to preserve

signal integrity but it also helps to control EMI [4].

8.2 RF energy developed in PCB

One of the main problems in PCB design is the unwanted RF energy developed by both passive

and active digital components [5]. To understand this phenomenon it is required to have some

basic knowledge of Maxwell’s equations. Based on the physics of electromagnetics they describe

the relationship of electric and magnetic fields. These equations express the root cause for how

EMI is developed, i.e. time varying currents. Static charge distributions create static electric

fields, but not magnetic fields. Constant current sources create magnetic fields, but not electric

fields. Time varying currents produce both electric and magnetic fields. For deeper insight,

Maxwell’s equations are related to Ohm’s law. In time domain Ohm’s law is expressed as

H I · J (8.4)

where V is voltage, I is current and R is resistance. Switching to the frequency domain

transforms the equation to

HGK IGK · L (8.5)

where Z is impedance (R+jX) and subscript rf refers to radio frequency. If RF current is present

in a PCB trace with some impedance value, an RF voltage is created proportional to the RF

current present. R is substituted by Z, a quantity containing both resistance (DC-real component)

and reactance (AC-complex component). The quantity is the impedance in both time and

frequency domains, which is resistance to the flow of energy. There are several different ways to

describe impedance. For a wire or a trace in PCB, the impedance can be expressed as

L J M<N =ODP J MQ =O( (8.6)

where <N Q, < =( and 2RS. The magnitude change of impedance versus frequency

in equation 8.7 must be taken in consideration, when a component with known inductance,

capacitance and resistance is applied.

|Z|VJ M< VJ M<N < (8.7)

For higher frequencies the inductive reactance exceeds R, forcing the current to take the path of

least impedance. The path for low frequencies is resistive and for higher the path of least

reactance is dominant. The correct current path for RF currents is always through the smallest

impedance. For the reason that most circuits operate at frequencies higher than a few kHz, the

40

idea that current takes the path of smallest amount of resistance, provides an wrong theory of

how current flow occurs within a transmission line structure.

Each PCB trace has some impedance and that is the reason why RF energy is developed inside

the PCB. A connection between silicon dies and mounting pads also develop RF energy. Traces

on a PCB can be highly inductive, especially traces that are electrically long. Traces which don’t

return fast edge events to the source before the next edge-trigged event are called electrically

long. In the frequency domain this occurs when the length exceed approximately λ/10 of the

frequency present inside the trace.

A circuit must be closed in order to function otherwise the RF current will take the path through

any accessible impedance and emit electromagnetic energy. Both a signal- and return-path is

needed, otherwise the signal would never travel through the trace from source to load. The

absence of a path between load and source would force RF current to return through an

alternative return path causing EMI to occur and develop RF energy, which disturbs other

adjacent components.

Figure 8.3 -Closed and open circuitry

In figure 8.3, the left picture shows the path direct for both signal and return current, where total

impedance is small. In the right picture, the return path is non-existing. Since current needs to

obey Ampere’s law, the RF energy will return through the lowest impedance path present. When

the impedance of the return path is greater than 377 ohms, free space becomes the return channel

and is seen as radiated EMI.

8.2.1 Magnetic flux and cancellation

Why is the Ampere’s law so important in PCB design?

It is because a magnetic field is created around the transmission line whenever there is a closed

loop present and current is flowing freely from one point to another. These fields are radiated

through free space as unwanted electromagnetic energy and are the main source of

electromagnetic disturbance. To prevent RF energy inside the PBC, the concept of flux

minimization needs to be considered. By bringing the RF return path adjacent and parallel to its

source trace, magnetic flux lines in the return path (clockwise field), relative to the source path

(counterclockwise) will be in the opposite direction.

achieved by combining the clockwise

with the counterclockwise magnetic field caused by the return current.

loop or a turn will cancel out the magnetic flux if the traces are positioned close to each other.

Unwanted magnetic flux between the

Regardless how great the design is, magnetic and electric fields will always be present. The

several different techniques for flux minimization. Some techniques

• Proper PCB layer stack up assignment and impedance control

• Routing a clock trace adjacent to a RF return path

• Reducing RF currents by

• Reducing ground noise voltage in the power and ground plane structure

• Proper termination of clock and signal traces to prevent ringing, over

8.2.2 Common-mode and differential

Common mode (CM) and differential mode (DM) currents exi

determine how much of the RF energy

of data or signal of information. The common mode current is just a side effect of differential

mode currents and is the EMC

great levels of radiated electric fields, despite their smaller size compared to differential mode

currents. Even small sizes of wires can produce significant radiated emission levels. The

troubling part is that small amount

RF energy as a larger amount of differential mode current because common mode currents do

not cancel out magnetic flux.

The origin of common and differential mode current

current loops. Two sources aligned in series with each other providing power for example to an

amplifier makes it possible for both currents to exist

Figure

41

l be in the opposite direction. Cancelation of magnetic flux will be

achieved by combining the clockwise magnetic field caused by the forward propagating current

magnetic field caused by the return current. The trace forming a small

t the magnetic flux if the traces are positioned close to each other.

nwanted magnetic flux between the source and the return path will be cancelled or minimized.

Regardless how great the design is, magnetic and electric fields will always be present. The

several different techniques for flux minimization. Some techniques [5] are listed below:

stack up assignment and impedance control

Routing a clock trace adjacent to a RF return path

Reducing RF currents by decreasing RF voltage

educing ground noise voltage in the power and ground plane structure

Proper termination of clock and signal traces to prevent ringing, over

mode and differential-mode currents

Common mode (CM) and differential mode (DM) currents exist in every circuit. These currents

RF energy that propagates. The differential mode signals are carriers


mode currents and is the EMC compliance problem cause. Common-mode signals can produce



g part is that small amounts of common mode current will produce the same amounts of

a larger amount of differential mode current because common mode currents do

The origin of common and differential mode current comes from erroneous grounding and


amplifier makes it possible for both currents to exist, shown in figure 8.4.

Figure 8.4 - Common mode and differential mode currents

Cancelation of magnetic flux will be

caused by the forward propagating current

The trace forming a small

t the magnetic flux if the traces are positioned close to each other.

return path will be cancelled or minimized.

Regardless how great the design is, magnetic and electric fields will always be present. There are

are listed below:

educing ground noise voltage in the power and ground plane structure

Proper termination of clock and signal traces to prevent ringing, over- and undershoot

st in every circuit. These currents

propagates. The differential mode signals are carriers


mode signals can produce



of common mode current will produce the same amounts of

a larger amount of differential mode current because common mode currents do

comes from erroneous grounding and


42

Usually there are some small resistors and ground inside the amplifier where the current flow

through. In picture 8.4, common mode currents propagate in two separate directions, in the

upper- and lower-side of the amplifier. The current propagating in the upper and the lower side

of the amplifier is called differential mode current. Differential mode current is always present if

the path of the current is forming a loop. Common mode current propagates in a one-way

direction. The common mode current is created by poor differential mode cancelation. The best

way to reduce common mode current is proper grounding and to avoid having too many loops.

Another important aspect that needs consideration is the differential mode radiation caused by

the flow of RF current loops with in a system structure. In PCBs, emission sources are created

from currents flowing between components and in the power and 0V planes. These sources can

be seen as small loop antenna carrying interference RF currents. A small loop is defined as one

whose dimension is smaller than a quarter wavelengths (λ/4) at a particular frequency.

Common mode radiation comes from unintentional voltage drops in circuits which cause some

ground parts to rise above the real ground potential. Radiation of common mode potential comes

from cables connected to the affected ground system where they act as an antenna. With a

constant current and antenna length, the electric field is proportional to the frequency. Common

mode radiation is a more difficult issues to solve, unlike to differential mode radiation which is

easy to reduce using proper design techniques. The common path impedance for common mode

current is the only variable available to the designer. Elimination of common mode radiation is

done forcing the common mode fields to approach zero, which is made by using a sensible

grounding scheme.

8.2.3 Input power consumption

One major contribution to noise generation on the board is power supply transition currents to a

transistor gate, which is either conveyed by a power- or ground-plane. The ground reference

refers to the source that is at ground potential relative to the power source. It is usually referred

to as 0V reference or the power return plane, image plane or ground plane. The main source of

differential currents and RF energy are transition currents, which is causing an EMI spectral

profile to exist if the transmission time is short or edge rate is fast. The power supply transition

current demands during component switching can be quite large. These currents have no

relationship with the same current needed to establish a “1” or “0” signal state in digital logic.

The inrush of surge current is created due to partial conduction overlap of output drive transistors

for TTL and some CMOS technologies. The power bus is virtually shorted to ground through

two partially saturated transistors during the crossover time from logic high to logic low.

Manufacturers provide Schottky barrier diodes to prevent the output transistors from going into

saturation in order to avoid crossover conduction currents. There are other techniques for

fabrication of the component that includes “output edge rate control”. This is done by replacing

one big output transistor with several smaller ones. Only when voltage swing is between logic

high and logic low RF voltages and capacitive crosstalk can exists. The current that is needed to

change logic state from low to high and opposite is larger than the quiescent current and it can be

estimated to be

43

I" ; " (8.8)

where C is the sum of distributed capacitance of the load and trace capacitance to ground.

Single-sided boards have 0.1 to 0.3 pF/cm and multilayer boards have values from 0.3 to 2pF/cm

[5]. The peak current combines nonsymmetrical current usage with the power supply transition

current.

Other problems related to radiated EMI emissions is the difference between manufacturers’

active components. Differences do exist in the manufacturing process even though digital

devices are form, fit and function compatible. Not every manufacturers design the components in

the same way, since they don’t assume the same problems from similar components related to

functionality of EMC.

8.2.4 Component packaging

Placement of the components on a PCB must be considered to their interconnect traces, bus

structures and decoupling. The designers do not always consider design parameters like how

digital components are packaged or if their functionality is correct. Its often speed to get to the

market that is important in high-technology designs. What they do forget is that the component

packaging plays an important role in the development or suppression of RF currents. One of the

major problems is the lead-length inductance, which allows several abnormal operating

conditions to exist. These conditions concern creations of small loop antenna and ground bounce

that may radiate RF currents that exist between source and load. Whenever the devices output

switches from one logic state to another ground bounce causes glitches to occur in logic input

circuitry.

The loop area of the die, the bonding wire and components leads to the PCB can become great

contributors to the creation of EMI. In components like VLSI components and high speed

components mounted on multilayer PCB’s, radiating loops become small which causes IC leads

to become relatively large and being equivalent to a radiating antenna. As mentioned in previous

chapters, differential mode currents generate radiated emission. These currents are set up by the

existence of the loop between components and a plane on a multi-layer board. For inductance to

exist, a loop must exist, and for minimization of inductance the loop area must be kept small.

These inductances include everything from the lengths of the bond wires internal to components,

internal bond leads for capacitors, resistors and other passive components.

Figure 8.5 - Loop areas between components

44

Major emissions sources are established from currents flowing between components. As seen in

the figure 8.5, radiated emissions can be seen as a small-loop antenna carrying interference

current. If dimensions are smaller than λ/4, a small loop is present and it exist for frequencies up

to several hundred Mhz. Common mode currents are harder to manage and control. They usually

are observed from cables affixed noise to the unit, where the RF energy is determined by

common mode potential. The common mode current can be seen as a monotonic antenna driven

by ground–noise voltage.

The best way to control differential mode radiated emissions is in the design and layout of the

PCB. That is done by minimizing loop areas, where logic components are selected with power

and ground pins located in the center of the package or physically adjacent to each other. Power

pins in the center gives optimal placement of decoupling capacitors. This configuration also

minimizes trace length connections between the device and decoupling capacitor. It also

minimizes the inductance from the power and ground pins internal to the silicon wafer of the

package. Since vias are needed to bring power and ground to the device, these same vias can be

used for heat removal when placed around the active components.

Looking at the creation of RF currents, surface mount technology components has an advantage

over through-hole components due to a smaller loop area. By minimizing loop areas of adjacent

power and 0V reference it can prevent RF currents to develop and allow for improved flux

cancellation between power and ground. Large RF currents exist on power input pins due to

differential-mode switching currents created by switching all pins under maximum capacitive

load. The RF flux must find an alternative path to cancel out internally generated RF currents by

minimizing differential ground noise voltage. Knowing this, it can be established that surface-

mount technology is preferred to through-hole technology for minimized RF emissions. The

characteristic difference is due to shorter lead-length inductance from the die of the component

to the circuit trace on the PCB, where Surface Mount Technology (SMT) has smaller loop size

areas. RF currents can also be created by lead length inductance, which cause induced RF

currents.

Another important issue concerning RF emissions is how IC package leads are bonded to the

PCB. Even though the loop areas are small, bonding wires along the interconnect leads permit

RF paths to be developed. The RF currents can be minimized by keeping the loop area of the

traces in multilayer PCB’s smaller than the lead-length inductance. Even bond wires become

small antennas at high frequencies, making it possible for RF path to exits and interfere with

electromagnetic compliance. In today´s high technology products there are great problems with

bond wiring, for example Dual In-Line Package (DIP) components is one to mention. By

choosing SMT components over DIP components performance is increased and package size is

reduced by approximately 40%. Lead-length inductance is decreased between the die and the

mounting pad because of the smaller board space with less trace length in between SMT

components.

8.2.5 Inductance reduction

One of the important issues in PCB design is to reduce inductance. Traces inside the PCB can

produce a large amount of EMI depending on the size and frequency. Designers who prefer

45

smaller width of the traces have to cope with inductance, voltage ripples and high voltage drops.

Using separate power and ground planes can also reduce inductance, where the inductance of a

plane is much lower than a PCB trace. This solution reduces the parasitic elements because the

parasitic inductance is very small due to a very large contour and a good magnetic coupling

factor. For higher frequencies, the plane technique is efficient if the distance between the source

and the load is short. These planes are inefficient at lower frequencies because of the small

equivalent capacitor and the impedance existing across the circuit.

Another technique in reducing inductance is selecting a proper and small package for the

components. Impedance and resistance are present inside the package where pins are connected

to dies. The die has some internal impedance and the package is in series with the die, and can be

seen as an RLC circuit which has some resonance frequency. Even the quality factor can be

reduced by selecting the smallest package.

8.2.6 Transmission line analysis

A transmission line is a material medium that form a path for directing transmission of energy,

such as electromagnetic waves or acoustic waves. It can include wires, coaxial cables, dielectric

stubs, micro strips or optical fibers. In this particular design micro strips were used as

transmission line. Important characteristics when designing a transmission line is the

characteristic impedance, dielectric constant, operating frequency, height of the substrate, width

of the strip and the thickness of the strip metallization. Characteristic impedance, Z0, is the ratio

of the complex voltage of a given wave to the complex current of the same wave at any point on

the line and it is defined by equation 8.9.

LW XCYO(NZYO( (8.9)

The importance of Z0 will be comprehended due to Kirchhoff’s laws no longer can be applied.

Should impedance be present in the circuit and the average size of discrete components is more

than a tenth of the wavelength, transmission line theory must be applied.

[\ ] =W (8.10)

At this length the phase delay and the interference of any reflections on the line become

important and can lead to unpredictable behavior in the system. Traveling voltages and current

waves expressed as exponential functions are used instead of Kirchhoff laws. The voltage wave

can be expressed as in equation 8.11

H_ HY`abc Ha`Ybc (8.11)

and the current wave as in equation 8.12.

I_ IY`abc Ia`Ybc (8.12)

Another important thing is to decide whether the transmissions line is lossless or not. When the

elements R and G are relatively small, the transmission line is considered as an idealized lossless

structure. For lossless transmissions lines characteristic impedance is

The design requirement for

determined due to characteristic impedance, such as width, height and thickness. The dielectric

constant for frequencies over 1 GHz is approximately 4.34. To determine the width of

wave impedance in free space

where Zf = 377Ω and factor A = 2.1862

with the knowledge from the PCB

substrate needs to be h = 1.6mm and t = 35

required value of width, which is 1.47mm.

When all these parameters are determined the PCB design can begin to take form.

8.3 Power supply design elements

8.3.1 DC/DC converters

Traditionally used linear regulators are often made up of

circuit that tries to keep the output voltage at a des

46


ments R and G are relatively small, the transmission line is considered as an idealized lossless

structure. For lossless transmissions lines characteristic impedance is

the characteristic impedance is 75Ω. Other characteristics are


constant for frequencies over 1 GHz is approximately 4.34. To determine the width of

wave impedance in free space, Zf, is needed and a factor A given by equation

and factor A = 2.1862. Substituting A in equation 8.15

the PCB manufacturer that both thickness of trace and height of the

substrate needs to be h = 1.6mm and t = 35µm, see figure 8.6. Solving equation

required value of width, which is 1.47mm.

Figure 8.6 - Microstrip line

ameters are determined the PCB design can begin to take form.

Power supply design elements

Traditionally used linear regulators are often made up of a more or less

circuit that tries to keep the output voltage at a desired value. When the load draws large amounts


ments R and G are relatively small, the transmission line is considered as an idealized lossless

(8.13)

Other characteristics are


constant for frequencies over 1 GHz is approximately 4.34. To determine the width of the trace,

uation 8.14 [6].

(8.14)

(8.15)

that both thickness of trace and height of the

. Solving equation 8.15 gives the

ameters are determined the PCB design can begin to take form.

more or less sophisticated linear

ired value. When the load draws large amounts

47

of current and the voltage drop over the regulator is high the power loss in the regulator is large.

High power loss generates heat which in turn must be conveyed off the regulator to prevent

malfunction. The regulator will be destroyed because of the too much heat.

A DC/DC converter takes care of two of the main drawbacks of linear regulators. The first is that

the converter can be made highly efficient thus radically reducing the power loss in the

converter. The second advantage is that voltage can be converted in a large number of ways, for

example both down- (Buck) and up-converted (Boost). Only voltage down-conversion can be

made with linear regulators. The main drawback of DC/DC converters is the less well regulated

output voltage and the noise emissions due to the high switching frequency. In comparison linear

regulators have well regulated output voltage and low noise.

DC/DC converters uses the fact that transistors can be used as switches and that these when

switched in either position (on or off) have very low power dissipation. High voltage and low

current in off position makes low power dissipation. In the on position the current is high and the

saturation voltage low which again give low power dissipation. In the design in this report both

Buck and Boost converters are used. First the function of the down-converting buck topology is

presented.

8.3.1.1 Buck down-converter

If the waveform, with 50% duty cycle, in figure 8.7 would be time averaged the value would be

half the top voltage.

Figure 8.7 - Solid line input voltage, dotted line time averaged voltage.

If the duty cycle (D) is varied, the time averaged voltage will vary as in equation 8.16.

Hdef Hgh · i (8.16)

The time averaging of a Buck down-converter is done as in figure 8.8. When the transistor is

fully on then current is flowing through the inductor, partly to the load and partly to capacitor.

When the transistor is turned off, the current through the inductor cannot suddenly disappear, due

to the energy accumulated in the magnetic field. The current continue to flow, transferring

energy to the capacitor and the load. As the transistor is cut off the current therefore takes the

way through the diode. This goes on several times per second, called the switching frequency.

48

Figure 8.8 - Buck step down converter

The ripple current of the output capacitor is small compared with the input capacitor. As the

inductor obstructs changes in current, it facilitates the task for the output capacitor. The capacitor

current has small deviations from the load current. The input capacitor has a more demanding job

delivering more current in portions determined by the switch frequency. It must be capable of

more ripple current, and thus a more expensive capacitor must be selected. The opposite applies

to a Boost converter, where the output capacitor has the more demanding job.

8.3.1.2 Boost step-up converter

The principle for Boost step-up conversion, see figure 8.9, is similar as the principle for the Buck

converter. Energy is stored in the inductor magnetic field when the transistor conducts. When the

transistor is turned off, the current continues to flow through the diode to the load and the

capacitor. The input voltage and the inductor induced electromotive force are also in series

resulting in a higher output voltage then input voltage. Equation 8.17 relates input voltage and

duty cycle to output voltage.

Hdef Hgh · 1 j=aj (8.17)

Figure 8.9 - Boost step-up converter

49

As the input current is filtered through the inductor, sudden changes in input current cannot

occur. The smoothing function of the inductor makes the task for the input capacitor easier than

for the output capacitor. When the switch is turned on, current is flowing through the inductance

to ground. The output capacitor is then the only load current supplier, and no current flows

through the diode. When the transistor is turned off, the output capacitor is recharged through the

diode. This means that the output ripple is proportional the load current consumption. More load

current means more ripple current and thus affects the selection of output capacitor. In other

words, a more expensive capacitor with greater ripple current capability must be used.

8.4 Power loss

Theoretically, the conversion between voltages with the two switching topologies can be made

100 % efficient. Some converters come close with efficiencies of 96-97% at low currents but the

efficiency gets lower when load current increases. The main power loss factors are the series

resistance of the inductor, losses in the switching transistor and the diode. The transistor can’t

switch off and on infinitely fast and thus there is power loss at the moment of every switching.

The saturation voltage of a bipolar transistor is also a limiting factor. One solution to get higher

efficiency is to replace the bipolar transistor with a MOSFET with low on resistance. However,

with such an advantage there is always a drawback. The gate of the MOSFET acts as a capacitor

and at every switching instant charge must be moved to or removed from the capacitor. This

charge transport introduces new power losses. This can in fact also be found in the bipolar case

but not in the same order of magnitude.

Theoretically, a Boost step-up converter could step-up the voltage as high as wanted.

Unfortunately power loss and the forward voltage of the diode set stop and limit the available up-

conversion factor to a few times. Sometimes the diode is replaced with a MOSFET transistor,

which makes the forward diode voltage problem disappear. If the MOSFET doesn’t switch at the

right time the electromotive force from the inductor can increase dramatically. Fortunately the

MOSFET has a parallel body diode, and often a built in Schottky diode, conducting for a short

period of time before the switching has occurred. In other case, the transistor could easily break

down by the high reverse voltage applied by the inductor.

8.5 Heat removal

As virtually all components heat up, some more than others, this heat must be removed for

correct functionality, or to maintain a low component temperature. If the temperature is not held

within specification, the increased temperature can break or potentially decrease the component

length of lifetime. All of the chips in this design needs more or less cooling. As these chips are

situated on top and near the PCB one easy way to remove heat is to put a copper plane right

under the chip, reaching out on the sides, which will aid temperature removal. Some of the chips

have an exposed thermal pad right under the chip package. This pad is soldered to the PCB, and

with the aid of thermal vias to the underlying ground plane this makes a rather efficient heat sink,

see figure 8.10. More information about thermal via size and solder mask techniques for efficient

heat removal and soldering can be found in [12].

50

Figure 8.10 - Thermal pads for heat removal

51

9 Circuit design

The tuner, demodulator and LNB driver are only a small part of a complete STB, see figure 9.1.

The transport stream comes out of the demodulator and must be processed to filter the needed

packets that belong to a certain TV-channel, text-TV or maybe subtitling. Afterwards the filtered

data must be decoded by a MPEG decoder if there is a video content and an audio decoder if

there is audio content. A great part of all the processing is done in a special STB processor. The

processor contains special hardware for most of the task which implies much computational

effort.

Figure 9.1 - System overview

A working system for terrestrial DVB were given with the task of rebuilding it for the second

generation satellite DVB-S2 standard. This might seem like an easy task but rebuilding a system

for new functionality without altering the existing base hardware can make things difficult. The

constraints on the new design were:

1. Use the existing base hardware as far as possible

2. Fit the new front end at the same place as the old

3. If possible, do not make the new front end larger than the old (44 x 70 mm)

4. To make manufacturing easy, do not place components on the back side on the PCB

5. Power the new front-end on the same power supply as the old (5.7V, 3.3V, 2.5V, 1.1V)

Some things that made the conversion difficult were the fact that the tuner and demodulator were

on different PCBs. The original system had a stand-alone tuner on a separate shielded PCB

which was soldered to the main PCB. The demodulator chip was soldered right on the main

PCB. Packaging and pin configuration of the new and old demodulator was not equal and

therefore the new demodulator was forced to be on the tuner PCB. By making this decision the

base hardware had to be altered with a contact for the transport stream. This was interfering with

52

constraint 1 but as the main PCB was under revision the contact could easily be added in. At this

point the tuner, demodulator and LNB driver were planned to reside on the same PCB at the old

tuner position. Trying to fit all this functionality on such a small space as 44 x 70 mm without

interfering with constraint 4 could not be done. The decision was then to lift out the LNB driver,

placing this on an own PCB, and see if the remaining components could fit.

A power converter for the demodulator had to be added to cope with the current consumption.

This was placed on the same PCB as the demodulator (see figure 9.2) as the feedback loop had to

be held small and the response fast. Unfortunately the size constraint could not be held anymore

and the PCB had to be prolonged form 70 to 110 mm. This was just 10 mm less than the hard

‘fit-in-the-box’ constraint and the PCB could not be much longer.

Figure 9.2 - Front-end PCB overview

An overview of the LNB PCB functionality can be seen in figure 9.3. As can be seen the LNB

driver must be powered by around 12V. This interfered with constraint 5 but could be solved by

switching wall socket ac/dc adapter.

53

Figure 9.3 - LNB driver PCB overview

9.1 LNB driver and interface

The integrated circuit LNBH21 [19] from ST Microelectronics were chosen to control and power

the LNB. It has a built in DC/DC converter that generates 13 and 18 V from a single supply

source and a continuous 22 kHz tone for selection of the high frequency band.

9.2 The LNB boost converter

The rectifying diode in the DC/DC converter dissipates a lot of power. For a normal diode the

power loss can be quite large. The reverse recovery, or charge removal, will also add some loss.

Schottky diodes will partly solve this problem, as they have a relatively low forward voltage and

good reverse recovery. For the LNB driver circuit a Schottky diode is thus selected. A dual

MOSFET switch would have been even better but the LNBH21 IC is not equipped with dual

driver circuits, and is therefore forced to use a Schottky diode. For cost and area efficiency an

integrated Schottky and MOSFET combo in a SO-8 package has been selected. Thermal pads

have been applied as in chapter Heat removal to maintain a low chip temperature. A compromise

between switch loop area and thermal pad area has been made in order to keep both EMI and

temperature down. As this is a Boost converter the ripple current at the output is larger than at

the input. This places constraints on the output capacitor ripple current capability. Load current

measured with several different LNBs shows a maximum of 200mA steady state current. At the

low 13V output the DC/DC converter must keep an output value of 13+2.2V. The 2.2V voltage

drop has its origin in the post regulator within the LNBH21. The post regulator is used to get an

almost ripple free output. To calculate the rms ripple current before the regulator, the mode in

which the converter works must be determined. The inductor value 22µH used in the design is

chosen according to the recommendation in the LNBH21 documentation. To maintain

continuous mode operation the average ripple current through the inductor must equal the mean

output current, see figure 9.4.

54

Figure 9.4 - Current waveform for continuous mode operation

The minimum output current for continuous mode operation is then

Ikl,mn ∆gp (9.1)

Disregarding losses in the circuit, the current increase during ql is

∆IN ;rsN · ql (9.2)

During qKK the decrease in current must match the prior increase in current

∆IN ;tuva;rsN · qKK (9.3)

Combining these two equations leads to equation 9.4.

"wx"wyy ;tuv;rs 1 =z.= 1 0.27 (9.4)

The off- and on-time at the typical switch frequency 220 kHz lead to

ql qKK =W·=W ~k / (9.5)

0.27 1 · qKK =W·=W ~k / (9.6)

qKK ==.·W·=W 3.58 ~k / (9.7)

ql 0.27 · qKK 0.97 (9.8)

By combining the equations above the minimum average current can be calculated, see equation

9.9.

55

Ikl,mn = · ;rsN · ql 265 (9.9)

The minimum current for continuous operation is thus larger than the maximum measured

current drawn by the LNB, and the converter will work in discontinuous mode. It can easily be

seen from the equation that the minimum current increases with increasing on-time (corresponds

to a larger output voltage), and the converter will thus never be in continuous mode with the

200mA load.

9.2.1 Output capacitor ripple current

As the inductor current will reach zero during qKK, due to the discontinuous mode operation, this

will alter the previous equations. The off-time can be divided into two different parts, qKK= with

inductor current and qKK when current has decreased to zero, see figure 9.5.

Figure 9.5 - Discontinuous mode operation

The current increase during ql is the same as before but equation 9.3 is altered slightly and

become

∆IN ;tuva;rsN · qKK= (9.10)

Setting the two inductor current equations equal yields

qKK= ;rs;tuva;rs · ql (9.11)

The charge supplied to the load during ql, qKK= and qKK must equal the charge supplied by the

inductor at qKK=.

Iq" · ;rs·N · ql · qKK= (9.12)

Combining equations 9.11 and 9.12 creates an expression for the output voltage.

Hdef Hgh ;rs ·"wx·N·gtuv·f (9.13)

56

Recombining the variables also leads to an expression for ql,

ql,=; Xgtuv·;tuva;rs··N;rs ·K0E 0.94 (9.14)

and with the use of equation 9.11, qKK= can be calculated for a output current of 200mA.

qKK= 3.54 (9.15)

By knowing ql the peak inductor current can be calculated.

IN,mb ;rsN · ql 514 (9.16)

During qKK= this current decreases according to

IN IN,mb ;tuva;rsN · (9.17)

The inductor current refills the capacitor with charge through the diode until the inductor current

has decreased to 200mA. Thereafter the capacitor current reverses and adds with the inductor

current to sum to the 200mA output current, see figure 9.6. The equation for IN can therefore be

used to calculate the rms current flowing through the capacitor if Idef is subtracted to account

for the output current.

Figure 9.6 - Current waveform for discontinuous mode operation

With all this information the output capacitor rms current can be calculated.

IG/ S/k" · $ IN,mb Idef ;tuva;rsN · %"wyyW $ Iq" %"wxY"wyyW 227 (9.18)

57

Due to the fact that the design uses two capacitors this value is halved and the rms current

capability of each capacitor must be at least 113mA for the low voltage level. The calculations

for the high level are not showed but results in an rms ripple current of 276mA and thus 138mA

for each of the capacitors.

9.2.2 Output ripple voltage

As the LNBH21 IC is equipped with a linear regulator after the Boost converter the ripple

voltage at the output is of less importance. However, if the ripple is big there is a chance of

malfunction as the linear regulator has a 2.2V voltage drop. By the well known equation · ; " , the drop due to current supplied to the load can be calculated. By using the equations in the

previous chapter ql, qKK= and qKK can be calculated for the high voltage output. Then the

output voltage drop can be calculated as

∆H gtuv·"wxY"wyy = $ ;tuva;rsN · %"wyy∆" (9.19)

∆ N·gp,agtuv;tuva;rs (9.20)

Where ∆ is the time from when the capacitor begins to recharges to the time where it starts to

supply current over again.

Increasing the output current or time where the capacitor supplies all the current, increase the

output voltage ripple. Increasing the capacitance helps lowering the ripple and thus for a

specified output voltage and maximum ripple voltage a minimum capacitance value can be

calculated. Recommendations in the datasheet suggest a 220µF and a 100µF with a ferrite bead

filter in between. The ferrite bead is almost a short for dc current and therefore the ripple voltage

can be calculated by adding the capacitance values. A 320µF value corresponds to around 1,6mV

output voltage ripple. Unfortunately this is not the only voltage ripple source. The voltage I · J in the capacitor adds to the final ripple. When the capacitor discharges with 200mA the

ESR voltage is 6.8mV (for an ESR of 34mΩ). Thus the negative peak value is 8.3mV. The

positive peak ripple value can be determined by

∆H = $ IN% ∆"W = $ IN,mb Idef ;tuva;rsN · % ∆"W 1.6H (9.21)

As can be seen the delta voltages are exactly the same. This is no surprise since the same amount

of charge is added or subtracted from the capacitor plates. From the definition of capacitance :; it can be seen that a charge change corresponds to a direct proportional voltage increase

or decrease.

The final peak-to-peak voltage ripple thus become 10mV (equation 9.22) which is acceptable as

the linear regulator makes the output voltage almost ripple free.

HGk,a 2 · ∆H Idef · J 10H (9.22)

58

9.2.3 Input capacitor ripple current

The converter exhibits a triangular waveform at the input. The main function of the input

capacitor is to smoothen out the current supplied by the input power circuitry. This also leads to

minimized areas of high frequency currents which reduce RFI. As most of the current is directly

supplied by the input power circuitry the ripple current of the input capacitor will be several

times less than for the output capacitor. With unknown powering circuit the rms ripple current

for the capacitor is impossible to calculate with acceptable accuracy. However, a good estimate

or guess might be a calculation of the rms current with subtracted mean current value.

Ikl,ml S/k" · $ ;rsN · %"wxW $ ;rsN · ql ;tuva;rsN · %"wyyW (9.23)

Ikl,G/a/" S/k" ·

¡¡¡¢

$ ;rsN · Ikl,ml %"wxW $ ;rsN · ql ;q"a;klN · Ikl,ml%"wyyW £¤¤¤¥ (9.24)

Calculating the input capacitor rms current estimate shows a lower current, as expected. For

comparison, table 9.1 shows input and output rms ripple currents for the LNB Boost converter

with a 200mA load.

Output voltage Input capacitor rms ripple

current (estimate)

Output capacitor rms

ripple current (estimate)

Input/output rms

current ratio

15.2V (13+2.2V) 147mA 227mA 0.65

20.2V (18+2.2V) 225mA 276mA 0.82 Table 9.1 - Capacitor ripple current

As the converter works in discontinuous mode the input/output ripple current ratio is quite high.

This ratio would have been much lower in continuous mode as the inductor ripple current would

not have been so high.

9.3 Voltage requirements by the demodulator chip

The LDPC FEC in the demodulator chip consumes lots of power when activated. According to

the demodulator datasheet [17] the maximum current consumption is 1.7A. To keep power

dissipation on a low level the LDPC runs at a voltage of 1.2V. See table 9.2 for a complete

overview of the required voltages and current consumption.

Supply voltage Maximum current consumption Power dissipation

3.3V 30mA 99mW

2.5V 100mA 250mW

1.2V 1.7A 2.04W Table 9.2 - Supply voltage, current consumption and power dissipation of the demodulator chip

59

As the available voltage at the input of the STB is 5.7V the use of a linear regulator for the 1.2V

would result in a voltage drop of 4.5V over the regulator. A power dissipation of 7.65W in the

regulator would then be the result. This requires a special package and lots of PCB heat sink area

in order to maintain a relatively cool chip. As there are restrictions on the dimensions of the

PCB, and so much power dissipation would be a waste of energy, another voltage conversion

method must be considered. A DC/DC Buck converter could do an excellent job of voltage

conversion. With an efficiency of 90%, the power dissipation in the converter decreases to

around 0.8W which is more manageable with the area restrictions of the design.

The demodulator chip requires two more voltages, 2.5V and 3.3V. The hardware driven by these

voltages draws considerable less current than the LDPC and thus the use of linear regulators will

easily accomplish the goal. The use of linear regulators has the advantage of low noise and a well

regulated voltage, needed by for example the ADC converter for I- and Q-channel decoding.

Linear regulators also keep the price and complexity of the design down. The regulator can often

be made with a single integrated circuit and two additional capacitors. In comparison a DC/DC

converter often need at least one or two more, quite bulky components.

The 2.5V digital and analog sections share the same supply but are separated by an EMI filter.

The more sensitive analog section is thus isolated from the more noisy digital section.

9.4 The 1.2V Buck Controller

The TPS40190 is a synchronous buck controller which operates from 4.5V to 15V and

implements fixed frequency voltage mode power supply [20]. By using an adaptive anti-cross

conduction scheme it prevents both the high side and rectifier MOSFET to be turned on at the

same time. In that manner it prevents shoot through current in the two MOSFET´s. The

controller is also protected against short circuit with a threshold that is selectable between one of

three values. Should the controller sense an output short circuit, both MOSFETs will be turned

off and a timeout period would be observed before attempting to restart. It also provides drivers

to minimize switching losses in the power stage, reducing heat building up in the MOSFETs and

permitting larger MOSFETs to be used without undue switching time penalty. In figure 9.7, a

simplified application diagram is presented. Values for the inductor and capacitors have been

determined by iteration of the equations in the next chapters. The inductor was set to 10µH and

the output capacitor to 2x22µF + 10µF in near proximity to the demodulator chip. The input

capacitor was chosen to be 100µF.

60

Figure 9.7 - TPS40190 simplified application diagram

9.4.1 Output capacitor ripple current

To calculate the currents the on- and off-periods must be determined. It is assumed that the

converter operates in continuous mode. The following calculations will determine if this is true.

Disregarding losses in the circuit, the current increase during ql is

∆IN ;rsa;tuvN · ql (9.25)

During qKK the decrease in current must match the prior increase in current

∆IN ;tuvN · qKK (9.26)

Combining these two equations leads to

"wx"wyy ;tuv;rsa;tuv =.z.a=. 0.27 (9.27)

The off- and on-time at the typical switch frequency 300 kHz lead to

ql qKK =WW·=W ~k / (9.28)

0.27 1 · qKK =WW·=W ~k / (9.29)

qKK ==.·WW·=W 2.63 ~k / (9.30)

61

ql 0.27 · qKK 0.7 (9.31)

The peak delta current is then

INmb∆ ;rsa;tuvN · ql 316 (9.32)

As can be seen in equation 9.32 the mean delta peak current is much lower than the load current.

The assumption that the converter worked in continuous mode was true. With this knowledge the

input- and output-rms current can be calculated. The inductor current for continuous mode

operation is depicted in figure 9.8.

Figure 9.8 - Inductor current for continuous mode operation

The output capacitor will deliver current when the inductor current is less than Idef and be

refilled when the current is higher than Idef. The rms capacitor current can thus be calculated as

in equation 9.33.

IG/ S/k" ·

¡¡¡¢$ ;rsa;tuvN · gp∆ %"wxW $ ;tuvN · gp∆ %"wyyW £¤

¤¤¥ 91 (9.33)

As this is an estimate of the output capacitor rms current, measurements of the real

implementation will most probably be a bit higher. The equation would still help to pick a

capacitor as the rms current would be in the same order of magnitude.

9.4.2 Output voltage ripple

As the inductor current, the capacitor will exhibit a triangular current waveform. The waveform

mean value is zero as depicted in figure 9.9.

62

Figure 9.9 - Inductor- and output capacitor-current

At the instances when the current reverses direction the voltage drop caused by the capacitor

ESR is at its maximum value. The voltage drop can be calculated with the already determined

delta peak current in equation 9.32.

HABC,mb gp∆ · J 0.8H (9.34)

The ESR of the output capacitor will ultimately set the limit on how low the output ripple can be.

Increasing the capacitance beyond a certain level will not be of any use since the ESR voltage

ripple always will be there. The ESR voltage ripple will be at its peak value when the capacitor

current is at its maximum, as in equation 9.34. The capacitor contribution can be calculated by

integration of the current in the grey area in figure 9.9. The charge accumulated will raise the

voltage level as described in equation 9.35.

Hmb = · $ ;rsa;tuvN · %wxW $ ;tuvN · %wyyW 3H (9.35)

The capacitor ripple voltage is a few times higher than the ESR voltage ripple. This means that

the ESR ripple voltage is only a small part of the total ripple and the capacitor ripple can be

taken as a measure of the total ripple. The capacitor ripple peak will also occur at a different time

instant and if the capacitor is large enough and the ESR is small enough, the capacitor ripple

peak is the true peak.

9.4.3 Input capacitor rms current

As the converter is powered through long interconnects with intrinsic inductance there is no way

for the power supply to deliver current as fast as the converter needs. The input current can then

be assumed nearly constant. Therefore the input capacitor has to act as a supply during these

moments.

63

During ton the total input current must be the same as the inductor current in steady state. The

current supplied to the input capacitor during qKK must then be (assuming constant input

current):

Igh "wx"wyy · Idef (9.36)

Figure 9.10 depicts the behavior of the input current, the capacitor current and the inductor

current.

Figure 9.10 – Input capacitor current, input current and inductor current

Positive capacitor current is defined as flowing into the capacitor. During ton the input current

and capacitor current therefore adds up to the total inductor current during the same instance.

With this information an estimate of the capacitor rms current can be calculated.

IG/ S/k" · $ ;rsa;tuvN · Igh gp∆ %"wxW $ Igh %"wyyW 455 (9.37)

The true input capacitor rms current will differ from the calculated values but equation 9.37

presents an indication of how to select the input capacitor with regards to the current capability.

Most probably will the RMS current be smaller as the real input current is not constant.

64

9.5 Voltage requirements by the tuner chip

The tuner [18] power supply could be made less sophisticated than for the demodulator chip.

One single, low current, 5V power supply is all that is needed but care has to be taken when

placing bypass capacitors. The voltage regulation could therefore be made by a simple linear

regulator.

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10 PCB layout

The PCB layout has been made with a trial version of Altium designer as both of us had a little

experience from earlier software releases from the same company. As there were size constraints

on the finished tuner and demodulator PCB it had to be made with four layers. The first and the

fourth layer were chosen as signal layers, the second layer as a dedicated ground plane and the

third layer as a power plane for the different voltage levels.

In figure 10.1, the finished tuner and demodulator card can be seen. At the left, with predrilled

holes for a shield, is the tuner chip located. The zero IF baseband I- and Q-channels are

transferred from the tuner chip to the demodulator chip which is located in the low center of the

card. This chip handles all the tasks required to put out a decoded transport stream to the white

PFC contact to the right. In the upper right the 1.2V buck converter is located, as can be seen by

the bulky input capacitor and the inductor. There is also a 2.5V linear regulator located in the

same area.

Figure 10.1 - Populated tuner and demodulator PCB, top side

Decoupling capacitors for the demodulator chip are located right below the power pins, on the

back side, to make the connection length as small as possible, see figure 10.2.

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Figure 10.2 - populated tuner and demodulator PCB, back side

The LNB driver was originally planned to reside on the same PCB as the tuner and demodulator

but there was not enough room and the solution was to do one more PCB. The size (44x100mm)

was chosen so that it could easily be attached to some unused area in the finished STB. In figure

10.3, the two layer LNB driver PCB is depicted.

Figure 10.3 - Populated LNB driver PCB

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11 Drivers

To get anything useful out of the hardware some software drivers were needed. As the base

platform with the st7100 CPU/MPEG-decoder was running STLinux, drivers for the Linux

system had to be written.

The hardware was also tested on the PC with a ST developed Graphic User Interface (GUI) with

their own drivers. All measurements in the next chapter have been done with the aid of the ST

GUI in figure 11.1 to lock on different frequencies. In the GUI all demodulator and tuner

registers can be read and programmed.

Figure 11.1 - ST GUI for front-end measurements

Communication between the PC and the front-end has been done with an I2C-interface. The

measurement setup can be seen in figure 11.2.

Figure 11.2 – Front-end measurement setup

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69

12 Measurements and testing

In order to check compliance with specifications some basic measurements had to be performed.

The measurement equipment used was an Agilent digitizing oscilloscope, DSO6034A, and a

DVB-S and S2 digital television test transmitter from Rhode & Schwarz. As discussed earlier the

ST GUI was used to read valuable data from the tuner and demodulator registers.

12.1 Front-end

In this section the most basic performance measurements for correct function of the front-end is

presented. Of course there are several more measurements that have to be done in order to fulfill

a full test plan for the finished product, but as the time were running short only the most

important measurements were finished.

12.1.1 Voltage levels and ripple measurements

When the LDPC FEC starts there is a sudden current surge from the power supply. ST specifies a

voltage drop limit of 40mV for correct functionality at this instant. As can be seen in figure 12.1

this requirement is fulfilled by a margin of approximately 10mV. When the LDPC stops working

the voltage rises (the left yellow top) but soon settles down on 1.2V. It can be seen on the lower

MOSFET drive voltage that the converter stops switching until the output voltage has decreased

before continuing switching. Then the FEC starts and the voltage dips for a short while (the right

yellow dip). At this instance the upper MOSFET is turned on for a longer period in order to

supply more current.

Figure 12.1 – LDPC supply voltage dip (yellow) and low MOSFET drive voltage (green)

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The 1.2V output voltage ripple can be seen in figure 12.2. It is around 5 - 10mV as predicted in

the calculations and satisfactory for our application. All other voltage levels are within the

tolerances of specification.

Figure 12.2 - 1.2V buck converter output ripple voltage (yellow) and lower MOSFET switch voltage (green)

For a basic input capacitor current test a 0.1 ohms resistance was connected in series with the

input capacitor. In this way the voltage over the resistance could be measured and an indication

on how great the ripple current was could be determined. One of the measurements could be seen

in figure 12.3.

Figure 12.3 - basic input capacitor ripple current measurement

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The measurement is not exact but it can be seen by the RMS voltage that the current is smaller

than the calculated in chapter 9.4. The ripple is still high for a normal electrolytic capacitor. For

the purpose of reducing the ripple current of the big input capacitor a small ceramic capacitor

had been added in the schematic design phase. This capacitor was placed as close to the

switching MOSFET input as possible to reduce inductance. In figure 12.4 the difference with the

ceramic capacitor can be seen. The RMS current ripple was reduced by approximately three

times, and the electrolytic capacitor could now be replaced with a less ripple current capable

capacitor (cheaper).

Figure 12.4 - basic measurement of input current with both electrolytic and ceramic capacitor

12.1.2 Tuner and demodulator functionality tests

Quasi Error Free (QEF) is a term in DVB context which defines a post-Viterbi BER (DVB-S) or

PER (DVB-S2) threshold for acceptable performance. For DVB-S, BER values higher than the

QEF limit of 2E-4 is defined as unacceptable. In table 12.1, the theoretical and measured C/N

ratios for QEF operation are presented. In the DVB-S case the acceptable implementation loss is

0.8 dB [23]. As the DVB-S2 FEC doesn’t use Viterbi, direct comparisons between DVB-S and

S2 is impossible. For DVB-S2 the QEF is defined as a PER of 1E-7. The DVB-S2 measurements

are presented in table 12.2.

Theory Measured

Code rate C/N (dB) C/N (dB) 1 2⁄ 3,3 3,3 2 3⁄ 5,1 5,1 3 4⁄ 6,1 6,13 5 6⁄ 7,1 7,1 7 8⁄ 7,7 7,79 Table 12.1 - DVB-S performance measurements

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Measured QPSK Measured 8PSK

Code rate C/N (dB) C/N (dB) 1 2⁄ 1,05 - 3 5⁄ 2,5 5,7 2 3⁄ 3,09 6,7 3 4⁄ 4,04 7,96 4 5⁄ 4,67 - 5 6⁄ 5,2 9,41 8 9⁄ 6,23 10,78 9 10⁄ 6,45 11,09 Table 12.2 - DVB-S2 performance measurements

The DVB-S measurements show exemplary performance. All measurements are within the 0.8

dB implementation loss limit, and some even touch the theoretical limit. The DVB-S2

measurements, see table 12.2, show a much better performance for a given code rate making it

possible to send data on even noisier channels. Two 8PSK code rate values could not be output

with the instrument so they were not measured. With 8PSK the data rate is doubled while the

C/N limit is roughly deteriorated by 3 dB in comparison to DVB-S. With the increased data rate

it is possible to send several more TV channels on the same frequency.

12.2 LNB driver

At first the chip was unable to put out the high voltage level. Fairly quick an error was found in

the schematic. One pin that should be AC coupled to the output had mistakenly been directly

coupled to the output. This made the chip go into current protection mode. After correcting the

mistake the design worked as planned.

12.2.1 Measurements

In the following text measurements on the design are presented. The LNB driver must handle

two voltage levels with and without 22 kHz added (tabulated in chapter 6). As can be seen in

picture 12.5 the driver handles the crossing between no tone and tone smoothly. The tone

frequency is also within specification.

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Figure 12.5 - LNB driver, right; high voltage, left; high voltage (22 kHz added)

Figure 12.6 show switching between the high and low voltage output while the tone is on. The

slope becomes a bit jagged but this is a consequence of the chip design (when the 22 kHz is on)

and nothing that will alter the driver function.

Figure 12.6 - LNB driver, switching between high and low voltage (22 kHz added)

In figure 12.7 the switching behavior from high to low without tone can be seen. The transition is

smooth and there are no real remarks.

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Figure 12.7 - LNB driver, switching between high and low voltage

The transition from low to high takes a great deal more time than from high to low. This can be

seen in picture 12.8 and 12.9. The behavior is a consequence of the step up converter that takes

some time to get the voltage up to the high level.

Figure 12.8 - LNB driver, switching between low and high voltage (22 kHz added)

75

Figure 12.9 - LNB driver, switching between low and high voltage

The measurements on the LNB driver card show good behavior with no remarks. The output

ripple was also measured but is not shown as this is almost immeasurable with the equipment

available and therefore nothing to worry about.

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77

13 Result

The purpose that was set up in the beginning of this report has been fulfilled. The result of this

thesis is a fully functionally DVB-S2 system. The advantages of the DVB-S2 system, is the best

transmission performance, total flexibility and reasonable receiver complexity. The

disadvantages are pretty obvious because of the systems short age. The conversion from DVB-S

to DVB-S2 will take approximately 10 – 15 years like the entrance of HDTV. The testing and

measurements that has been made during this master thesis has strengthened the thesis and the

theories about the DVB-S2 system. The measurements performed on the DVB-S sections of the

chip gave exemplary results. Comparing these results with corresponding measurements

performed on the DVB-S2 section showed much better performance with the same coding rates,

which made our expectations from the literature study fulfilled. Applications of the DVB-S2

system can be found in broadcasting television programs, interactive services, data content

distribution and Internet trunking.

The future seems pretty bright for the DVB-S2 standard. Many companies are manufacturing

their products according to the DVB-S2 standard. The sought-after characteristic on the market is

HDTV where high data-rate is a requirement. The higher data-rate, the greater quality will

HDTV have. Separate set-top-boxes are soon going to be taken off the market. Placing them into

the television set, forces the manufacturers to come up the new solutions for media

entertainment, which is an important aspect we find our self in the information era. People are

getting more and more aware of television and entertainment. Almost every household have a

desire to one day own a HDTV with a built in set-top-box, where the receiver might be a DVB-

S2 receiver.

78

79

14 Conclusion

This report has gradually been developed to the present version, part by part from different

scientific areas. From the beginning this thesis was said to be too difficult to even be a master

thesis due to several difficult moments. All these moments could be structured down to own

fields and give base for several master theses. The literature study was the most important part

since both of us had limited experience and knowledge of digital television. Studying all the

modulation techniques, both analogue and digital gave us an idea what was expected from us,

and how much effort and time this thesis would require.

The second part of the literature study was to understand how digital television works and why

different standards are needed for different broadcast media. Most important was to understand

the differences between the first and second generation of the satellite DVB standard. The

following moments were the most challenging and the most enjoyable to do, the actual design of

both schematics and PCB layout. Even though neither of us had much previous experience in

design; one could say that it went pretty well. The knowledge and experience we got after the

prototypes were made was priceless. Now we know how much effort and knowledge that are

needed in order to come even close to a functioning prototype. Fortunately the errors in the

design were few.

When the components were mounted, the testing of the prototype appeared to be the easy part

since there were not many mistakes in the design. One critical error of the LNB driver design

was found during the measurement phase but could be temporarily corrected. In the future, a

smart thing would be to have extra control checkpoints on the schematics level before PCB

layout. This will prevent small and annoying errors.

14.1 The future of digital television

Looking at the past evolution of radio and television the next area of exploration will most

certainly be DTV reception among handhelds. With the DVB-H standard already set, field trials

have been performed in 2006 and several more countries will begin in 2007. Mobile media of

different forms has seen an up going trend and many new service providers will eventually pop

up, making handhelds more and more attractive to the consumers. There are a few more

standards for mobile purposes but DVB-H is the most used so far.

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81

15 Abbreviations

For the purposes of the present document, the following abbreviations apply:

ACI Adjacent Channel Interference

ACM Adaptive Coding and Modulation

AM Amplitude Modulation

ATSC Advanced Television Systems Committee

BCH Bose-Chaudhuri-Hocquenghem

BER Bit Error Rate

BOM Bill Of Material

BS Broadcasting Services

CCI Co-Channel Interference

CCM Constant Coding and Modulation

C/I Carrier to Interference

CM Common mode

C/N Carrier to Noise ratio

CNR Carrier-to-Noise Ratio

COFDM Coded Orthogonal Frequency Division Multiplex

CRT Cathode-Ray Tube

CS Communication Satellite

DM Differential mode

DTH Direct-to-home

DSB Double Sideband

DSL Digital Subscriber Line

DSGN Digital Satellite News Gathering

DTV Digital Television

DVB Digital Video Broadcasting

DVB-RCS DVB-Return Channel Standard

EBU European Broadcasting Union

EMC Electro-Magnetic Compliance

EHF Extremely High Frequency

EPG Electronic Program Guides

ETSI European Telecommunications Standards Institute

FEC Forward Error Correction

FFT Fast Fourier Transform

FM Frequency Modulation

FSK Frequency Shift Keying

GUI Graphic User Interface

HDTV High Definition Television

IC Integrated Circuit

IF Intermediate Frequency

IP Internet Protocol

IS Interactive Services

ISDB Integrated Services Digital Broadcasting

JPEG Joint Picture Experts Group

LAN Local Area Network

82

LNB Low Noise Block down converter

LDPC Low Density Parity Check

MER Modulation Error Ratio

MPEG Motion Pictures Experts Group

MPE-FEC Multi-Protocol Encapsulation – Forward Error Correction

MSK Minimum Shift Keying

MUSE MUltiple sub-nyquist Sampling Encoding

MUX Multiplexer

NHK Nippon Hōsō Kyōkai

NTSC National Television Systems Committee

PAL Phase Alternation Line

PCB Printed Circuit Board

PER Package Error Rate

PLL Phase Locked Loop

PRBS Pseudo Random Binary Sequence

PSI Program Specific Information

PSK Phase Shift Keying

QAM Quadrature Amplitude Modulation

QEF Quasi Error Free

QPSK Quaternary Phase Shift Keying

RF Radio Frequency

RFID Radio Frequency Identification

SBTVD Sistema Brasileiro de Televisão Digital

SECAM Séquentiel couleur à mémoire

SDTV Standard Definition Television

SHF Super High Frequency

SMT Surface Mount Technology

SSB Single Sideband

S/N Signal to Noise ratio

SMATV Satellite Master Antenna Television

TPS Transmission Parameter Signaling

TS Transport Stream

UHF Ultra High Frequency

VCM Variable Coding and Modulation

VHF Very High Frequency

VSB Vestigial Sideband

83

16 References

[1]. Svärdström, Anders. Modulation och teleteknik, 2nd

Ed. 1996. ISBN: 9789144001395

Publisher: Studentlitteratur.

[2]. Röjne, Mats. Digital TV , 1st Ed. 2003. ISBN: 9789144030548. Publisher:

Studentlitteratur.

[3]. Beasley, J & Miller, G. Modern Electronic Communications 8th

Ed. 2004. ISBN-10:

0131130374. Publisher: Prentice Hall.

[4]. Montrose, Mark I. EMC and the printed circuit board. 1st Ed. 1998. ISBN-10:

078034703X. Publisher: Wiley-IEEE Press.

[5]. Montrose, Mark I. Printed circuit board design techniques for EMC compliance, 2nd Ed.

2000. ISBN-10: 0780353765. Publisher: Wiley-IEEE Press.

[6]. Ludwig, Reinhold & Bretchko, Pavel. RF circuit design – theory and applications,

Har/Cdr edition 1999. ISBN-10: 0130953237. Publisher: Prentice Hall. Publisher: EBU

Technical review

[7]. Morello, Alberto & Mignone, Vittoria. DVB-S2, ready for lift off, October 2004.

Publisher: EBU Technical review

[8]. Breynaert, Dirk. Analysis of the bandwidth efficiency of DVB-S2 in a typical data

distribution network. March 2005. Maximilien d’Oreye de Lantremange, Newtec.

[9]. Huang, Chi-Fang, Chang, The Nan & Hsu, Chau-Yun. Field evaluations in Taiwan of the

DVB-T COFDM and ATSC 8-VSB digital TV system, July 2003. Publisher: EBU

Technical review.

[10]. Kornfeld, Michael & Reimers, Ulrich. DVB-H, the emerging standard for mobile data

communication, January 2005. Publisher: EBU Technical review.

[11]. Reed, Erik. New ripple current guidelines for very low ESR tantalum capacitors. CARTS

2004 conference, San Antonio.

[12]. Application notes for surface mount assembly of Amkor’s MicroLeadFrame (MLF)

packages. Rev E, December 2003. Publisher: Amkor Technology.

[13]. Faktasidor, ELFA. 2007.

[14]. Welcome to ATSC. URL: http://www.atsc.org. February, 2007

[15]. Wikipedia. URL: http://en.wikipedia.org. February, 2007.

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[16]. DVB – Digital Video Broadcasting. URL: http://www.dvb.org. February, 2007

[17]. STB0899 datasheet, ST Microelectronics.

[18]. STB6100 datasheet, ST Microelectronics.

[19]. LNBH21 datasheet, ST Microelectronics.

[20]. TPS40190 datasheet, Texas Instruments.

[21]. DVB standard EN 300 421. V1.1.2 August 1997. European Standard

(Telecommunications series).

[22]. DVB standard EN 301 210. V1.1.1 March 1999. European Standard

(Telecommunications series).

[23]. Discussions and information from Sven Jansson, our supervisor at Zenterio AB.

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Construction of a digital-TV receiver for the second-generation satellite broadcasting, DVB-S2

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