Dc Series Motor Control

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Simulation and construction of a speedcontrol for a DC series motor

J. Santana a,*, J.L. Naredo a, F. Sandoval a, I. Grout b,O.J. Argueta c

a Centro de Investigaci oon y Estudios, Avanzados del IPN, Prol. Loo pez Mateos Sur # 590,

Guadalajara 45090, Mexicob Department of Electronic and Computer Engineering, University of Limerick, Limerick, Ireland

c Universidad Autoonoma de Guadalajara, Av. Patria #1201, Guadalajara 44100, Mexico

Abstract

DC series motors are preferred for mechatronic applications requiring high torque/speed

ratios. This paper describes the design and implementation of an open loop DC motor speed

control that is based on a micro-controller and on IGBTs. Trial and error designs are ex-pensive and time consuming. This problem is solved here by using simulation tools which can

predict the dynamic behavior of systems consisting of mechanic and electronic modules. The

simulations provided along the paper show a satisfactory agreement with laboratory mea-

surements.

Ó 2002 Elsevier Science Ltd. All rights reserved.

1. Introduction

DC series motors usually are selected for traction applications requiring hightorque/speed ratios. Examples of these are wheel chairs, golf carts, hoists, cranes,

actuator arms, etc. [8]. A typical application consists in a human operator driving a

DC motor by means of an accelerator pedal or a lever. The electronic system reg-

ulating the electric power fed into a motor, in accordance with a pedal or lever’s

position, customarily is referred to as speed control. Such a system can be either in

closed or in open loop configuration [8]. While closed loop systems are required for

high accuracy applications, there are many situations for which an open loop system

will suffice. This paper is concerned with the latter ones.

Mechatronics 12 (2002) 1145–1156

* Corresponding author. Fax: +52-33-3134-5579.

E-mail address: [email protected] (J. Santana).

0957-4158/02/$ - see front matter Ó 2002 Elsevier Science Ltd. All rights reserved.

PII: S 0 9 5 7 - 4 1 5 8 ( 0 2 ) 0 0 0 1 9 - 3

http://mail%20to:%[email protected]/





A typical DC motor speed control often has its internal signals generated and

processed by analog circuitry and has its power driving stage made of several

MOSFET modules in parallel [1]. This typical control can be improved by replacing

or complementing its analog functions with digital ones and, in addition, by sub-stituting each paralleled arrangement of MOSFETs with a single IGBT module [9].

The improved control would thus result more reliable, less costly and much simpler

to produce. An open loop digital speed control based on IGBTs is thus proposed and

developed here. The main purpose of this paper nevertheless is to present the

methodology being employed by these authors for developing a working prototype

of the proposed speed control.

Mechatronic prototypes usually are implemented by a trial and error process

which, most of the time, ends up being very expensive and time consuming. It is

proposed here that this drawback can be alleviated substantially by properly com-

bining computer simulations and laboratory tests. The conjunct simulation of anelectric motor and its speed control has been done before by applying very simple

models for the motor and/or for the electronic control modules [1,8,12].

For the work reported here, a detailed motor model is first developed. Then, this

model is tuned up for attaining a very close match between simulations and the

actual performance of the motor being employed as test bench. Later on, the model

of the proposed speed control is constructed using the manufacturers provided

models of its constituent electronic devices. Finally, the conjunct simulations of the

motor and its speed control are applied at refining and debugging the proposed

design, even before its construction.

Nowadays, various available software packages permit simulating mechatronicprototypes; e.g., MATLAB-SIMULINK from MathWorks [11], INTUSOFT

SPICE, Mentor Graphics Mechatronics Library, Saber from Analogy, PSPICE from

OrCad [7], etc. While some of these packages provide generic models for various

power electronic devices and/or mechatronic components, others facilitate the use of

manufacturer provided models of specific devices as well as of user developed

models. Generic models usually permit a fast simulation development; on the other

hand, however, these models often cannot be modified by the user for attaining a

closer match between simulations and real performance. PSPICE permits both, the

use of manufacturer provided models and the inclusion of user developed models.

The latter is through an extension called analog behavioral modeling (ABM) [7]. Due

to this reason and to its relatively low cost, PSPICE is being used for the work re-

ported here. In a previous work by other author ABM has been applied in the

construction of a detailed IGBT model [3]. In this paper, while ABM is employed in

the construction of a detailed DC series motor model, the IGBTs and other elec-

tronic devices are represented by means of manufacturers supplied models.

2. Principal functions of a DC motor speed control

The accelerator pedal for an electric motor usually is coupled mechanically to a

rheostat that translates its position into the voltage signal fed to the speed control. A

1146 J. Santana et al. / Mechatronics 12 (2002) 1145–1156



first function for this control is thus the conditioning of the pedal signal for removing

rheostat noise and smoothing sudden changes at speed ups and slow downs.

The speed control then uses this conditioned signal for its second function con-

sisting in the generation of the switching waveforms that would drive the outputpower electronic devices. Pulse width modulation (PWM) has become a widely

adopted method for generating these waveforms [1]. A third function for the speed

control is to deliver the required power to the DC motor. This is performed by the

control’s power output stage usually made of power semiconductor devices: diodes,

MOSFET, SCR, IGBT, etc [6]. The fourth and last function being considered here is

the provision of protection mechanisms for the control itself, for the motor and for

the user. A list of the main protection functions is as follows:

• Protection against start up at full voltage.

• Protection against overcurrents in windings.• Protection against loss of pedal signal.

• Protection against low battery voltage.

• Protection against high temperature.

• Protection against switching signals absence at the power drive gates.

3. Proposed design for a DC motor speed control

Current digital technologies provide several clear advantages over the analog ones

for the functions of generating the PWM switching signals and of processing theprotection signals of the DC motor speed control. One single processor replaces

several analog modules and their associated discrete components [1]. The control’s

physical dimensions and manufacturing costs can thus be reduced while, at the same

time, its reliability can be increased. Fig. 1 shows a block diagram for the proposed

speed control. The micro-controller used here is the MC68HC811 which incorpo-

rates eight A/D 8-bit converters.

The conditioning of the pedal signal is performed by the circuit shown in Fig. 2.

The rheostat voltage first is fed into emitter follower OpAmp 1 (TL084). Shunt re-

sistance Rpp ensures that V cond goes to zero if the rheostat connection is lost. This thus

Fig. 1. Proposed layout for a DC series motor speed control.

J. Santana et al. / Mechatronics 12 (2002) 1145–1156 1147



implements the function of protection against loss of pedal signal. OpAmp 1 output

then goes into an RC circuit where diode D1 allows for two different time constants,

one for accelerating and the other for decelerating. With the values shown in the

figure the latter is 47 ms, while the former is at least 738 ms depending on the valueof Radjust. Finally, the RC circuit output is injected through emitter follower OpAmp

2 into the micro-controller’s first A/D converter.

To generate the PWM switching signals, V cond from Fig. 2 circuit is fed into the

micro-controller’s first A/D converter input. V cond is thus discretized into 256 levels,

each one of them is taken as an address for the micro-controller’s E2PROM. The

corresponding memory cell contains the intended duration of the pulse. The pulse

width is thus varied in 256 steps according to V cond. A discrete linear variation is

adopted here for the prototype being implemented. This linear characteristic, how-

ever, could be easily changed if this is deemed convenient.

Before their injection into the power driver gates, the PWM signals are first passedthrough a gate driver stage consisting in an optoelectronic isolator and a buffer. This

is shown in Fig. 3. As for the power electronic stage, a full DC to DC H-bridge

converter made with four IGBTs and their corresponding parallel diodes is highly

recommended [6]. For the work reported here, however, only one branch of this

bridge is implemented in the form shown in Fig. 4. This branch is made to perform

as a step down converter [6].

In addition to the protection against loss of pedal, the other protection functions

listed above are implemented using voltage sensors as well as current and temper-

Fig. 2. Pedal signal conditioning circuit.

Fig. 3. Gate driver.




ature transducers as needed. The analog signals delivered by these devices are then

fed into the micro-controller via its additional A/D converter inputs and theirmonitoring is made by the micro-controller’s program.

4. DC series motor modeling using analog behavioral modules

4.1. Motor equations

The equations that describe the electromechanical behavior of a DC series motor

are given as follows. The electrical equilibrium equation is [2,8]:

V s ¼ E a þ Rt I s þ Lt

d I s

dt ð1Þ

where V s is the voltage at the two windings connected in series, E a is the induced

electromotive force (emf), Rt is the total series resistance, I s is the current through the

windings and Lt is the total series inductance. The relation of E a with magnetic flux /

and angular speed x is [2,8]

E a ¼ K a/xðt Þ ð2Þ

where K a is a motor constant. The electromagnetic torque developed by the DC

motor depends on I s and on / in the following manner:T emðt Þ ¼ K a/ I s ð3Þ

The torque balance equation is [2,8]:

T emðt Þ ¼ T L þ Bxþ J dx

dt ð4Þ

where T L is the load torque, B is the viscous friction constant and J is the motor’s

rotor and shaft inertia. The magnetic flux and the windings current are related

through the machine’s magnetization curve which is denoted as follows [2,5]:

/ ¼ f ð I sÞ ð5Þ

Function f ð I sÞ in general includes saturation and hysteresis effects.

Fig. 4. Power output stage.




4.2. Analog behavioral modeling of a DC series motor

Eqs. (1)–(5) provide a mathematical model for a DC series motor suitable for

the purposes of this paper [2,5]. The ABM implementation of this model is shown inFig. 5.

Module 1 of Fig. 5 is a series branch of elements that corresponds to Eq. (1). Note

that the emf term ‘‘ E a’’ is injected into this branch from module 2 by a voltage

controlled voltage source of unit gain. Included in this branch also is a current

controlled voltage source of unit gain which acts as a current sensor. The sensed

current is required by modules 3 and 5 in the calculation of both, the magnetic flux /

and the electromagnetic torque T em.

The magnetic flux actually is an intermediate variable for the calculation of E a. On

combining (5) and (2)

E a ¼ K ax f ð I sÞ ð6Þ

The magnetization characteristic of an electric motor usually is provided as a set of

points of E a vs. I s obtained at a fixed value x0 ¼ 180:6 rad/s of the angular speed.

This value usually corresponds to the motor’s nominal or nameplate speed. Let E a0

denote this characteristic at x0. According to (6), for a different value of x [2]:

E a ¼ E a0

x0

x ð7Þ

Module 5 of Fig. 5 actually is the specified or measured non-linear curve E a0 vs. I s.An example of this is provided by Fig. 6. It should be mentioned that for the im-

plementation reported here the hysteresis was neglected. Module 2 actually imple-

ments Eq. (7). Its output is the emf fed back into module 1.

In addition to E a0, the other input to module 2 is the angular speed calculated by

module 4 that corresponds to Eq. (4). On neglecting Bx, the viscous friction term,

this equation yields [8]:

Fig. 5. ABM implementation of the motor.




x ¼1

J

Z t À1

ðT em À T lÞ dt ð8Þ

It is clear from Fig. 4 that module 4 implements this last equation. On combining

Eqs. (2) and (3) into (7) the following expression is obtained for the electromagnetic

torque

T em ¼ ð E a0 I sÞ=x0 ð9Þ

Module 3, finally, implements this last equation.The above-described motor model was used to reproduce the operation of an

available 1 hp DC series motor. First, the parameters Lt, Rt and J of this motor were

measured. Then, the magnetization characteristic E a0 vs. I s was obtained as a col-

lection of 80 points which is plotted in Fig. 6. Next, all these data were applied to

Fig. 5 model. Finally, various experiments were performed on both, the motor and

its ABM model. These experiments were devised so as to permit the refinement of

the measured parameters. The ABM motor model, in addition, actually supplied

the lack of certain special instruments, such as a dynamometer. The values of the

variables and parameters used for the simulations are as follows: Rt ¼ 55 mX,

J ¼ 0:06 kgm2, V s ¼ 12 V (tension between A and A0), C p ¼ 10000 lF and theparasitic inductance Lc was neglected. The wound inductance Lt is a value that varies

with frequency [10]. From the wound’s step response the following two values of Lt

were estimated experimentally: Lt ¼ 75 lH at 2100 Hz and Lt ¼ 150 lH at 2500 Hz.

At intermediate frequencies Lt is calculated using linear interpolation. These values

are listed in Table 1.

4.3. SPICE model of the proposed speed control

Fig. 1 schematizes the SPICE model of the proposed DC series motor speedcontrol. It consists of four basic building blocks: pedal signal conditioner, micro-

controller, gate driver and power output stage. Except for the micro-controller, all

Fig. 6. Measured values for the magnetization curve of an 1 hp DC series motor.




these blocks are represented in great detail using manufacturer provided SPICE

models for the semiconductor devices being used.

A detailed representation of the micro-controller is not deemed necessary and,besides, it is unrealistic for the desktop computer being used in this work. It was

opted instead for representing the PWM function only by means of a rectangular

wave generator with its pulse width being varied in 256 steps according to the output

of the pedal signal conditioner. Before being injected into the motor model, these

PWM signals are passed through the gate driver and the power output models. It

should be mentioned here that, apart from the protection against loss of pedal sig-

nals, the other protection functions of the micro-controller and their associated

circuitry were not included in the simulation.

The detailed diagram for the pedal signal conditioner is the one provided in Fig. 2,

while the diagrams for the gate driver and for the power output stage are provided inFigs. 3 and 4. An advantage of using SPICE for the modeling of these blocks is the

access to a vast library of models for commercially available devices [4]. Only the

model for the IGBT module was not in this library, but it could be easily down-

loaded from the manufacturer’s web information site. Before its physical imple-

mentation, the speed control SPICE model was tested along with the ABM motor

model. This permitted the debugging and fine-tuning of the preliminary design even

before its implementation.

5. Results

Fig. 7a shows the steady state current being injected by the speed control into the

DC series motor as obtained from a simulation considering a 70% duty cycle and a

frequency of 2250 Hz. Fig. 7b shows this same current as obtained from an exper-

imental measurement. Note that the agreement between these two figures is satis-

factory. Fig. 8a shows the plot of the steady state current injected into the DC motor

obtained by simulating a 50% duty cycle and a frequency of 2100 Hz. Fig. 8b pro-

vides the plot of the corresponding experimental measurement. The agreement be-

tween these two figures also is satisfactory.The transient current injected into the motor at the starting up now is obtained

through a simulation considering a 90% duty cycle and a frequency of 2500 Hz. This

Table 1

Name plate and measured values of the DC motor

Symbol Name plate Measured

Nominal frequency x0 180.6 rad/s – Rated power – 1 hp –

Rated voltage – 12 V –

Rated current – 60 A –

Wound – Series –

Wound resistance Rt – 55 mX

Wound inductance Lt – 150 lH at 2100 Hz, 75 lH at 2500 Hz

Rotor inertia J – 0.06 kg m2




current is plotted in Fig. 9a and the corresponding measurement is provided by Fig.

9b. These two figures coincide in showing that the transient peaks of current can be

as high as two times the motor’s nominal current. Fig. 10a provides the transient

current at start up when it is simulated assuming a 50% duty cycle and a frequency of

2100 Hz. This figure should be compared with 10b that corresponds to the transient

current obtained experimentally for the same duty cycle and frequency.

Current transients as the ones in Fig. 9a and b can cause serious damage to the

speed control power output stage. The simulations help determining the adequate

Fig. 7. Steady state current at 70% duty cycle and 2250 Hz frequency: (a) simulated, (b) measured.

Fig. 8. Steady state current at 50% duty cycle and 2100 Hz frequency: (a) simulated, (b) measured.

Fig. 9. Transient state current without load at 90% duty cycle and 2500 Hz: (a) simulated, (b) measured.




starting conditions and the appropriate power electronic devices. Another advantage

of the simulations is the possibility of observing various internal variables of thesystem. Fig. 11 for instance shows the simulated electromotive forces for various

Fig. 10. Transient current without load at 50% duty cycle and 2.1 kHz: (a) simulated, (b) measured.

Fig. 12. Simulated angular velocities at startup for various duty cycles and frequencies.

Fig. 11. Simulated electromotive force at startup for various duty cycles and frequencies.




frequencies and duty cycles, while Fig. 12 shows the obtained angular velocities. Fig.

13a and b show, finally, the simulated electromagnetic torque obtained for a 2 Nm

load and at 90% and 50% duty cycles, respectively.

6. Conclusions

A methodology for developing complex electronic–electromechanical prototypes

has been presented in this paper. It consists essentially in combining computer

simulations with laboratory tests. This methodology has been further applied in thedevelopment of a speed control for an 1 hp DC series motor. The simulation tools

adopted here are PSPICE from OrCAD and its ABM utilities.

While PSPICE has permitted a detailed representation of the electronics and

power electronics modules which include manufacturers’ supplied modules, the

ABM utilities have enabled the accurate description of the motor’s electromechan-

ical features. The conjunct simulation of the motor and its speed control has thus

been possible. The agreement attained here between simulations and laboratory tests

has been satisfactory.

Previous works by others (Chee-Mun, INTUSOFT, HDLA Mentor) have sim-

ulated speed controls coupled to very simple (one branch) motor models. The ABM

model provided here accounts for more realistic motor features. It can be further

adopted by other developers and the paper provides the necessary data for repro-

ducing the results presented here. Third parties, in addition, may even modify this

model with relative ease to suit the particular needs for other developments. The

simulations have permitted the testing and the debugging of the speed control even

before it was constructed. For instance, the simulations have helped to establish that

the system should not be started up with a duty cycle above 50%.

The overall experience with the methodology presented is that it has helped re-

ducing development time and costs substantially. The simulations, when combinedwith experimental work, even supplied the lack of certain specialized instrumenta-

tion, like dynamometers. They also enabled the monitoring of inaccessible variables

Fig. 13. Simulated torque at startup with a 2 Nm load: (a) 90% duty cycle at 2500 Hz, (b) 50% duty cycle

at 2100 Hz.




like the electromotive force. A remarkable fact is that not a single power electronic

device was burned during this project.

Finally, ongoing work is in the development of a full H-bridge speed control, as

well as on applications of this to DC motors of different dimensions where the ABMmodel provided here may have to be modified to include additional features, like

viscous friction, hysteresis and the frequency dependence of wound inductance.

References

[1] Castagnet T, Nicolai J. Digital control for brush DC motor. IEEE Tran Ind Appl 1994;30(4):883–8.

[2] Ong Chee-Mun. Dynamic Simulation of Electric Machinery Using MATLAB/SIMULINK. New

Jersey: Prentice Hall; 1998.

[3] Cotorogea Marııa. Using analog behavioural modeling in PSPICE for the implementation of subcircuit-models of power devices. IEEE Proceedings of CIEP, International Power Electronics

Congress; 1998. p. 158–63.

[4] Goody RW. MicroSim PSPICE for Windows. New Jersey: Prentice Hall; 1998.

[5] Krause PC, Wasynczuk O, Sudhoff SD. Analysis of Electric Machinery. New York: IEEE Press; 1995.

[6] Mohan N, Underland TM, Robins WP. Power Electronics Converters, Applications and Design.

New York: John Wiley and Sons; 1995.

[7] OrCAD, 1999. Reference Manual for PSPICE ABM Modules.

[8] Sen PC, MacDonald ML. Thyristorized DC drives with regenerative braking and speed reversal.

IEEE Trans IECI 1978;25(4):347–54.

[9] Argueta OJ. Desarrollo e Implementacioon de un Control de Velocidad e Motores de CD en Serie,

MSc Thesis, Centro de Investigacioon y Estudios Avanzados del IPN, Unidad Guadalajara, 1999.

[10] DeWolf FT. Measurement of inductance of DC machines, IEEE Trans. Power Apparatus andSystems 1979; PAS-98(5):1636–44.

[11] MATLAB, 1999. Reference Manual.

[12] Muhammad HR. Power electronics circuits, devices and applications. Prentice Hall Inc. 1993.


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