Deliverable 1.1. Report on active Self-interference cancellation...

The Marie Currie Action Global Individual Fellowship“FUDACT” No. 656940

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Deliverable 1.1.Report on active Self-interference cancellation model

(parameters, usage and prospective topologies)


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ContentsAcronyms and Abbreviations ..................................................................................... 3Introduction................................................................................................................ 41. Self-interference cancellation limits ..................................................................... 5

1.1 SIC in wireless propagation domain .................................................................. 61.2 SIC in RF domain ............................................................................................. 81.3 SIC in analog domain...................................................................................... 12

2. Comparison of active feed-forward SIC solutions .............................................. 143. Proposed SIC solution ........................................................................................ 15References ............................................................................................................... 16


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Acronyms and Abbreviations

4G Fourth generation5G Fifth generationADC Analog to digital convertorBW BandwidthCG Common gateCM Common modeCMOS Complementary metal oxide semiconductorEBD Electrical balance duplexerENOB Effective number of bitsFBAR Thin film bulk acoustic resonatorFBS Femto base stationFDD Frequency division duplexingIBFD In band full duplexIC Integrated circuitIL Insertion lossLNA Low noise amplifierLTE Long Term EvolutionLTE-A Long Term Evolution – AdvancedNF Noise figureM2M Machine to MachinePA Power amplifierPAPR Peak to average power ratioPCB Printed Circuit BoardRF Radio FrequencyRX ReceiverSAW Surface acoustic waveSDR Software defined radioSI Self-interferenceSIC Self-interference cancellationSINR Signal to interference and noise ratio


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IntroductionModern mobile handsets are efficiently using the spectrum due to the use of modern modulationtechniques that bring the current systems (LTE-A) within a bit of Shannon limit. However, theutilization of the frequencies is based on frequency-division or time-division duplexing, Fig.1,which comes with its own efficiency limitations. To further improve spatial spectral utilization 5Gsystems are expected to deploy more densely wireless infrastructure thus paving the way for theutilization of the 27GHz and higher frequency bands. Dense deployment will inevitably increase theinterference between the users. Development of an effective cancellation technology would enablefurther density increases leading to higher system capacities. By taking the cancellation techniqueeven further, it would be used for cancelling self-interference caused by our own transmitters in thefrequency-duplexed systems. At the extreme, this technology would enable simultaneous operationof a transmitter (TX) and a receiver (RX) in the same frequency band (also known as in-band full-duplex (IBFD), Fig. 1) thus offering the potential to double the spectral ef ciency, as measured bythe number of information bits reliably communicated per second per Hz. Beyond spectralefficiency the in-band full-duplex feature can be advantageously used in wireless communicationsystems in multiple ways such as enabling introduction of novel and efficient channel accessmechanisms, reduced air interface delay, more flexibility in spectrum usage etc. [1]. All thesefeatures show a great potential of IBFD to enable various 5G mobile network targets. However,whether IBFD will fulfill its potential and show widespread use in the future depends strongly onsolving its main limiting issue - self-interference (SI).

(a) (b)

(c)Fig. 1. Uplink and Downlink in (a) frequency-division duplexing (FDD) (b) time-division duplexing

(TDD) (c) in-band full duplex (IBFD)Self-interference is a phenomenon that appears in a number of contexts, even in half duplexsystems. In an FDD system, when a radio transmits and receives on different channels (uplink anddownlink), transmit interference still overwhelms the receiver unless it is filtered out at the front-end of the transceiver. Current 4G LTE systems in Europe are based on frequency-divisionduplexing. Uplink and downlink can be separated by as little as 40 MHz, which makes filtering the

Time

Freq

uenc

y

Uplink

Downlink

Time

Freq

uenc

y UplinkDownlink

Time

Freq

uenc

y

& UplinkDownlink


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self-interference from the transmitter to the receiver very challenging. Sufficient attenuation of theTX signal, by 45-55dB is commonly accomplished by the use of thin-film bulk acoustic resonators(FBAR) and surface acoustic wave (SAW) filters. Unfortunately, these filters are not flexible, and aseparate filter is required for each frequency band. Since there are presently approved 43 differentLTE bands worldwide and there are 96 known band combinations for carrier aggregation, making atruly-world phone is very challenging. State-of-the-art solutions (Qualcomm RF360 and Intel XMM7160) support about 10 bands, where these bands are selected from 43 available through aninterposer. However, it is not feasible to efficiently support a growing number of bands with thisapproach. Besides the obvious size and substantial cost implications, this frontend partitioning iscontrary to the hardware sharing concept provided in software defined radios (SDRs). If self-interference cancellation/TX-RX co-existence is achieved it will in case of FDD relax duplexer-isolation requirements and enable integration of compact/tunable duplexers thus paving the waytowards reconfigurable or multi-band FDD radios.In this deliverable, prospective techniques for self-interference isolation/cancellation that are/can beapplied in IBFD or/and FDD will be presented. As necessary level of self-interference cancellation(SIC) in more demanding case- IBFD- can only be achieved by compilation of antenna, RF andbaseband SIC techniques overview of various SIC scenarios in different signal propagation domainswill be given.

The typical transceiver topology that will be used is direct-conversion transceiver as it enables thelowest power consumption, maximum hardware sharing both in RF and base-band sections, easyand exible frequency planning and the use of minimal external components.

1. Self-interference cancellation limitsThe main task of the receiver is to capture a signal coming from the distant source and thus at least1/d2 attenuated (d distance between source and RX). In the absence of any isolation and/orcancellation of the local TX signal, strong transmitter signal that occupies the same or closefrequency band (reflected from the environment and/or coming from the local TX) interferes withthe reception and if not taken care of can compromise RX sensitivity and thus link throughput.Therefore, to prevent RX performance degradation IBFD/FDD radio has to completely cancel (6dBbelow RX noise floor to get just 1dB increase in SINR) the significant self-interference signal andits TX/RX generated distortion and noise products.

In case of femto base station (FBS) for assumed NF=4dB and BW=20MHz receiver noise floor is -107dBm. For self-interference cancellation the worst case is FBS TX power of 20 dBm. Ifassumed isolation between RX and TX signal paths is 15dB, the base station self-interference willbe 20-15-(-107-6) dB=118dB above the RX noise floor. For systems with larger cells and coveragearea situation is even more severe as they will require higher transmit power levels and thus moreself-interference suppression. Even in the ideal case where self-interference is perfectly cancelled indigital domain, analysis [2, .] shows that residual errors due to quantizationand present noise will still be 6.02(ENOB-2) dB below the level of the self-interference signal at theinput of ADC, where 6.02(ENOB-2)dB is effective dynamic range of ADC that has preventedclipping (PAPR=5dB) and quantization noise limiting( = /4). If for example AD 9683 (ADCdesigned for communication application) with ENOB=11 is used, for given case where self-interference power at ADC input is 20dBm-15dB=5dBm, the residual interference floor will remainat 5dBm-6.02(11-2)dB = -49dBm which is still 58dB above receiver noise floor (-107dBm) and64dB above wanted power level (6dB below RX noise floor). As the ENOB in commercial ADCs isimproving only by about 1bit/decade [3] it is necessary to reduce self-interference before theADC. However, cancelling self-interference completely in RF/analog domain is not feasible sinceSI contains delayed TX components reflected by the environment. Cancelling these would require


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impractically large amount of tunable analog delay. Consequently, self-interference channel throughthe air is modelled and its influence cancelled in digital domain.

To satisfy tough self-interference cancellation requirement demanded by IBFD to maintain thesimilar SNR as for FDD operation or to obtain better isolation in FDD case and relax duplexrequirements we need to prevent RF signal generated by the local transmitter from leaking onto itsown receiver (isolation) and subtract any remaining self-interference from the RX path using theknowledge of the TX signal and TX to RX leaked signal (cancellation). Though the system alreadyknows what TX is sending the knowledge of TX leakage is necessary. It is not sufficient just tosubtract TX replica signal based on information from TX digital domain. Once the TX signal isconverted from digital to analog domain and up-converted it looks different from its basebandrepresentation due to various linear, non-linear distortion and noise (phase, thermal, quantization)mechanisms present in TX chain and unknown coupling from the transmitter to the receiver, whichcouples both on-chip and off-chip through the package, board, and antenna interface. TX leakagegives us in lumped fashion information about those phenomena.

There are three major types of self-interference:

direct electrical crosstalk on the transceiver IC,self-interference due to limited isolation of the chosen antenna solution,self-interference due to reflections of the transmitted signal through the environment(cancelled in digital domain).

To achieve required SIC (higher than 100dB, for FBS 118dB) the combination of isolation andcancellation techniques in all the domains signal is going through needs to be used, Figure 2.Typical commercial ADC have 8-12 ENOBs representing SIC of 36-60 dB which leaves 58-82dBSIC to be achieved in wireless propagation, RF and analog domain, which is in agreement with thedata stated in [4], Fig. 2. The amount of the SIC achieved in different domains will dictate therequirements for the TX and RX performance such as receiver linearity, phase noise, andtransmitter noise. Higher wireless propagation/RF/analog isolation relaxes the transmitter noiserequirement, and the linearity and phase-noise requirements of the receiver.

Fig. 2. Wireless propagation, RF and analog cancellation (shown in blue) is necessary to preventreceiver saturation (~50–80 dB). Once within the dynamic range of the receiver, digital cancellation

(shown in purple) can handle the remaining distortions (~60dB). [4]

1.1 SIC in wireless propagation domainIn wireless propagation domain isolation/cancellation is achieved with the use of two or moreantennas and by combining antenna directionality, cross-polarization and transmit beamforming to


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generate constructive and destructive interference patterns over the space [5, 6]. Though thesemultiple antenna designs achieve up to 40dB of isolation they have several drawbacks. Thesetopologies generate null regions of destructive interference in far-field regions thus spoiling far-field coverage. Maximum isolation (40-70dB) is limited by the reflections from the surroundingenvironment and imperfections in hardware realization [6]. According to [1] it is necessary to haveantenna distances beyond 160 mm to obtain an interference reduction of 40dB. Around 50dB SIC isobtained in the wireless propagation domain by antenna separation of around 40cm between the TXand RX antennas, while two separate TX/RX antennas distanced by 20cm can achieve 30dB SIC[7]. As a consequence, required physical distance between antennas prevents dense integration andmakes this technique unsuitable for small form-factor devices.

From Fig. 3 (a) can be seen that mobile device evolution is heading towards portable devices withsmaller form factor such as smart phones and M2M devices [8]. Moreover, research studies haveforecast almost exponential growth in mobile data traffic, Fig. 3 (b) [8], with estimate [9] thatalmost 80% of that traffic will come from indoor location. Consequently, there is a high demand forhigh-capacity indoor wireless solutions such as pico and femto base stations. To preserve the formfactor of these radios, the dimension of the wireless propagation/analog/RF self-interferencecancellation solution should typically not exceed 10% of the sizes listed in Table I [10].

(a) (b)Figure 3. (a) Global Mobile Devices and Connections Growth (b) Mobile Data Traffic [8]

Table I Form factor of wireless communication devices [10]

BASE STATIONS

Femto-cell 236 x 160 x 76 mm

Pico-cell 426 x 336 x 128 mm

Macro-cell 1430 x 570 x 550 mm

ACCESS POINTS/USEREQUIPMENT

Netbook 285 x 202 x 27.4 mm

Tablet PC 241 x 186 x 8.8 mm

Smartphone 124 x 59 x 7.6 mm

M2M Sensor nodes 50 x 50 x 50 mm

An alternative design approach towards more compact SIC solution was taken by building a two-port dual-polarized antenna where TX and RX signals, fed into two ports, have orthogonalpolarization [7, 10]. Theoretically there should be no energy transfer between TX and RX, andantenna presents a TX to RX isolation beyond 50 dB in frequency range from 2.3GHz to 2.6GHz[10], Fig. 4 (a), value probably limited by the parasitic coupling between antenna feed points.Additional performance degradation down to 40dB of SIC is noticed when different external objects(metal wall, wooden wall, hand) were in antenna proximity (from 10cm to 35cm) [7], Fig. 4 (b).Given results do not take into account case where reflected TX signal has the same/similarpolarization as the RX signal that will cause further isolation degradation. Authors suggest buildinga cancellation stage to increase robustness of their solution to the environmental effects.


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The antenna design dimensions are 90mm x approx. 90mm which are comparable to 100mmx100mm dimensions of the PCB for off-the shelf SIC solution [11]. According to 10% rule anddimensions in Table I it can be seen that this solution can be adequate just for macro cell basestations.

(a) (b)Fig. 4 (a) Measured and simulated S21 for the dual polarized antenna w/ and w/o active

cancellation, (b) Effects of environmental change on the antenna SI- suppression [7]

1.2 SIC in RF domainIn most systems, cancellation should be applied at the very front of the receiver, since the unwantedSI signals are very strong (in given FBS example 5dBm) and can easily saturate the receiver chain.Recently investigated solutions for single antenna FDD/IBFD transceivers that can providealternative to SAW filters or FBAR are based on either circulator or electrical balance duplexerstructures [11, 12]. However, compared to passive RF SIC structures, active RF SIC [13-17] arepotentially more compact, reconfigurable and tunable but other issues such as noise, distortion andpower consumption must be addressed.

Circulator is a three port non-reciprocity device that provides limited isolation between port 1 andport 3 while letting signals pass through consecutive ports, as seen in Fig. 5 (a). They are used as atype of duplexer, to route signals from the transmitter (port 3) to the antenna (port 1) and from theantenna to the receiver (port 2), without allowing signals to pass directly from transmitter toreceiver. Currently these devices are expensive, bulky off the shelf components with limited amountof isolation (15dB [11]). Typical efficient solutions are almost exclusively based on magnetic/opticmaterials which are not compatible with Si CMOS technology. Other monolithic compatiblesolutions are limited by noise and non-linearity performance of transistors [18]. Recent researchresults [19] illustrate magnetic free passive based circulator with isolation from 40 - 60dB, Fig.5.

(a) (b) (c)Fig. 5 (a) Circulator symbol (b) measured isolation for different varactor DC voltage values (c)

schematic representation of the non-magnetic passive circulator [19]


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The circuit is based on the parametric modulation of three identical, strongly and symmetricallycoupled resonators (LC resonant circuits with tunable/variable capacitances), Fig. 5. Modulationgenerates resonant modes that add constructively at one port or destructively at another thusinducing non-reciprocity. However, off the shelf components were used with quality factors notavailable in submicron monolithic technologies suggesting that this solution is still not applicable insilicon as isolation depends on inductor quality factor.

Another potential solution for high TX to RX RF isolation and single antenna transceiver that canreplace commercial duplexer for FDD and improve TX-RX isolation for IBFD has emerged in aform of electrical balance duplexers (EBDs) [7, 20-23]. The low quality factor of CMOS passivecomponents makes it impossible to achieve stringent filtering requirements demanded by standardsand fulfilled by commercial duplexers (for eg. in ACMD-6007, 4G/LTE Band 7 duplexer, RX noiseblocking is min 50dB while TX interference blocking is min 55dB). By relying on electrical balancerather than frequency selectivity EBD integration along with the rest of the CMOS RF integratedcircuit (IC) is highly possible, as matching and resolution are easier to achieve than high Q passivesin a CMOS process. Electrical balance duplexing is by no means new [20]. It has been used in thehybrid transformer which is, at its heart, a bridge circuit (4 port structure, Fig. 6(a)) with certainuseful null, or conjugacy, properties. If impedance match exist at one port a match will exist at allports. The relation between port impedances in matched case is defined by transformer ratios andimpedances seen at non-conjugate ports, Fig 6. Power ratio transfer (insertion loss, IL) from one toother two non-conjugate ports can be designed by choosing transformer ratios. In case of hybridtransformer IL from antenna to TX (ILTX) and RX (ILRX) are given in Fig. 6(b), which directlyillustrate the trade off that exist between RX NF and TX power consumption/efficiency [20].Commercial duplexers and literature [21, 22] results show slight preference towards TX withILTX<ILRX and autotransformer ratio r slightly higher than one.

The EBD architecture given in Fig. 6 is built of single-ended PA and differential LNA structure. Itenables wideband operation, where PA signal appears as common-mode at the input of differentialLNA. However, at high PA output powers, this common-mode signal can be as high as 10Vpp, so itwill limit the LNA linearity performance and may cause breakdown of the LNA transistors. In [23]a fully differential version of the duplexer is proposed to enable high power operation by cancellingthe common-mode capacitive coupling at the LNA input with more than 50dB differential modeand 60dB common mode TX band isolation and more than 50dB RX band isolation.

For conjugacy and impedance match:

= (1 + ) ( + )

=1

(1 + )

=

= 1 +

(a) (b)

Fig. 6 (a) Hybrid transformer (b) TX and RX IL as a function of autotransformer turn ratio rBesides previously mentioned issues electrical balance duplexers need to be continuouslyelectrically tuned to ensure accurate impedance balancing with the antenna across differentenvironmental conditions [7, 23]. As it is dif cult to retune the network and keep it balanced acrossa wide bandwidth, EBD are still insuf cient for an integrated, tunable, software-de ned radioapplication. Additional issue is distortion in EBD and generation of additional signal that needs tobe suppressed down the RX chain. PA power is not only delivered to antenna but lost at balance


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network. For r close to 1 almost half of the PA power gets dissipated on balance network. As it ispart of antenna impedance tracking and calibration network the non-linearity of the CMOS switchescauses distortion that gets delivered to RX. In case this distortion is not taken into account andsubtracted from the RX signal down the RX path it will limit the headroom for SIC in subsequentstages.Feedback action in RF is active SIC technique that is able to minimize the error signal in FDD,which is dominated by TX leakage, and reduce the interference signal seen at the input of the LNA.Filtering of the RX signal is commonly done by down-converting the LNA output to the baseband,

ltering, then up-converting, thus creating an effective high-Q, high-order band-stop filter. Thistechnique is insensitive to LNA nonlinearity and can handle multiple blockers, but a majordrawback is the sensitivity of the loop to gain and phase shift variations, which can cause the loopto go unstable. Any residual RX signal fed back to the LNA input incurs insertion loss for thereceived signal and directly adds to noise gure.Feed-forward techniques are mostly active SIC techniques that do not suffer from instabilitybecause no loop is created, but feed-forward path mismatch directly limits maximum transmittersignal attenuation [11, 13-17]. An adaptation loop can be used to tune the gain, phase, and possibleFIR taps for the feed-forward network [11], but care must be taken to provide very ne mismatchadjustments, given the large TX leakage signal and required attenuation. Additionally, feed-forwardnetworks require a unilateral path between input and output, otherwise feedback paths formed fromthe reverse gain path can cause unintended ltering or worse, destabilize the network.

In feed-forward IBFD one antenna solution [11], Fig. 7, one taps the TX chain to obtain a smallcopy of the transmitted signal just before the circulator that provides just 15dB of TX to RXisolation. This copy includes the transmitter distortion and noise introduced by the TX chain thatcan only be cancelled in RF and analog domain. The copy of the signal is then passed through acircuit which consists of parallel xed lines of varying delays (essentially wires of different lengths)and tunable attenuators. The lines are then collected back and added up, and this combined signal isthen subtracted from the signal on the receive path. In effect, the circuit based on sampling andinterpolation technique is providing copies of the transmitted signal delayed by different xedvalues and programmatically attenuated by different variable amounts.

(a) (b)

Fig. 7 (a) Full duplex radio block diagram (b) the requirements of RF/analog and digital SIC [11]The circuit that was tested for IBFD was WiFi transceiver with 20dBm power level. Initial analysisshowed that in order to suppress all components (linear, non-linear and transceiver noise) below RXnoise floor (-90dBm) SIC needs to be as large as 20dBm-(-90dBm)=110dBm, where full duplexdesign needs to provide 110dB of linear cancellation, 80dB of non-linear cancellation, and 60dB ofanalog cancellation, Fig. 7 (b). The key challenge is to pick the xed delays, as well as todynamically program the tunable attenuators appropriately so that the self-interference cancellation


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can be maximized. In [11] dynamic adaptation of analog cancellation was implemented that onlytakes 1% overhead for circuit tuning. Depending on the number of sampling and interpolation taps(8 vs. 16), RF/analog cancellation can range from 45dB to 63dB in the tested 80MHz bandwidth.After 62dB of analog cancellation, digital cancellation cleans up additional 48dB and 16dB of linearand non-linear self-interference components, respectively. The circuit was designed by using off-the-shelf components mounted on the PCB with 10 cm x 10 cm dimensions. To implement full-duplex in small form-factor devices (20-30 sq.mm, 10% of dimensions given in Table 1) SIC needsto be implemented as RF IC. The major bottlenecks of given solution are: delay lines (key spaceconsumers) that are realized as board via traces and narrow-band nature of several analogcomponents used in analog cancellation circuit (eg. attenuators). For truly multimode compactsoftware defined radio these two issues are essential.Another feed-forward FDD SIC active solution is presented in [13-15] where challengesassociated with noise, distortion and SI were addressed by employing noise-cancelling self-interference cancelling receiver architecture, Fig. 8 (a) [13, 15]. First, noise cancelling widebandtopology was used for cancelling noise from the matching (common gate- CG) device (shows up asCM for differential signal) while adding constructively the desired signal. Second, appropriatelyscaled TX replica signal is driving the gate of CG device and thus enables SI cancellation right atthe input of the LNA. An additional benefit is that entire noise from the active canceller (seen in theFig. 8 (a) as phase shifter and variable gain amplifier) is completely cancelled thorough noise-cancelling property of the circuit and thus does not increase NF of the RX. However, large ITXreplica current will create significant voltage swing at the CG output and degrade severely RXlinearity. This current limits the applicability of the topology only to FDD and not IBFD case as itcan be filtered only in former case where it is outside of the RX band. Typical solution is to designGm noise cancelling differential amplifier followed by current-mode down-conversion stage withimpedance transformation from the baseband consisting of passive mixers and basebandtransimpedance amplifiers. As TX noise and TX leakage in FDD are in different bands, former inRX and latter in TX band, and due to frequency selective nature of antennas and wireless channel,cancelling TX leakage (circuit is designed to cancel primarily this signal) does not imply that TXnoise in RX band will also be cancelled. Therefore, additional injection point at the output of the CSdevice is added to help cancelling remaining TX noise in baseband domain. As this signal was notadded at CG device its noise will contribute to increase in RX NF, Fig. 8 (b).In FDD/IBFD feedforward active design [14, 15] issue related to the selectivity of the antennainterface was addressed together with enabling IBFD and FDD operation. Conventional SIC activecancellers usually have flat amplitude and phase characteristic which provide narrow-band SICsolutions (exact SIC at just one frequency point), resulting in for eg. 8MHz SIC BW for more than20dB SIC [13]. By employing two 2nd order band-pass filters implemented as N-path Gm-C filterstwo additional degrees of freedom are obtained. SIC circuit is now able to set not only phase andamplitude of the TX replica but also slope of its amplitude and phase thus enhancing thecancellation bandwidth. The difference in isolation bandwidth can be seen in Fig. 8 (d) for fourcases: without SIC, with conventional SIC, with one 2nd order filter and with two 2nd order filtersemployed where improvement in SIC BW from 8MHZ for conventional SIC, to 15MHz for one 2nd

order and 25MHz two 2nd order filters can be seen [14]. As tunable, reconfigurable 2nd order high-QRF band-pass filters are implemented as linear N-path Gm -C filters they provide additional RXband TX noise filtering in FDD case.


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(a) (b)

(c) (d)Fig. 8 (a) Noise- cancelling self-interference cancelling RX (b) TX noise in RX band cancellation[13] (c) RX with reconfigurable wideband SIC [14] (d) Cancellation measurements across the 1.4

GHz narrowband dipole-antenna pair: TX/RX isolation after theoretical flat amplitude- and phase-based SIC as well as measured SIC with one and two filters enabled [14]

1.3 SIC in analog domainSome IBFD feed-forward perform active cancellation in baseband [16], which is reasonable forweak signal but not for practical implementations where higher TX power levels and self-interference will desensitize RF front-end. In [17] for better linearity no amplifiers are used in RFpath up to cancellation point, just low-ohmic switches in series with poly resistors to realize 50and build two (RX and VM (vector modulator)) passive down-conversion mixers, Fig. 9 (a). TXsignal is tapped as close as possible to the antenna to include transmitter imperfections in thecancellation path. Analog SI cancellation for a varying antenna near-field is implemented as a 31-slice vector modulator down-mixer to subtract a phase-shifted, amplitude-scaled TX replica in theanalog BB. Each slice is followed by static multiplexer switches (4 bit MUX) that can rotate the 4-phase output currents through the four virtual ground nodes and enable cancellation/absorption ofthe SI currents by VM before amplification. The 31-slice VM covers a square constellation of 32 by32 phase/amplitude points with resolution that enables 28.5dB SIC. To obtain further SIC theRX/VM should have very high linearity under cancellation of strong SI in order not to introducedistortion that raises the RX noise floor and masks the desired RX signals. The linearity for residualSI is improved by employing tunable negative conductance at the baseband amplifiers input [24](IIP3 +8/+16.2dBm for negative conductance off/on) which gives a slight NF penalty. Though mosttesting was performed for 802.11g and 2.5GHz, the cancellation technique is essentially broadband,Fig. 9 (b) with NF in range from 10.3 to 12.3 dB vs. 6.3dB in half-duplex. Proposed solution issuitable for mass production (2mm2 in 65nm CMOS) and digital/analog/RF co-integration.


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(a) (b)

(c) (d)Fig. 9. (a) IBFD transceiver (b) IBFD receiver (c) Implementation of one slice of the VM andnegative-conductance compensated BB (d) Performance over range of LO frequencies [17]


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2. Comparison of active feed-forward SIC solutionsISSCC 2014 [13] ISSCC 2015 [14] ISSCC 2015 [17]

Architecture Noise cancellingSI cancelling RX

RX with wideband SICbased on RF frequency

domain equalization

Mixer-first RX +SI-cancelling VMdown-mixer

No of antennas 1/2 1/2 2

RX Frequency 0.3-1.7 GHz 0.8-1.4 GHz 0.15-3.5 GHz(@2.5GHz)

RX NF 4.2-5.6 dB12dB (noise

cancellation off)

4.8dB 10.3-12.3dB (6.3dBin FDD)

Gain 19-34 dB 27-42 dB Max 24 dB

20dB cancellation BW 3MHz FDD 17MHz one filter,24MHz two filters

SC-FD 15MHz one filter25MHz two filters

NA

BB BW 2-76 MHz N/R 24MHz

OOB IIP3 +12dBm SIC off+33 dBm SIC on

+17dBm SIC off+25-27 dBm SIC on*

22dBm

OOB IIP2 N/R +61dBm SIC off+90dBm SIC on FDD

N/A

IB IIP3 N/R -20dBm SIC off+2dBm SIC on SC-FD

8/16.2 dBm(neg. cond off/on)

IB IIP2 N/R +10dBm SIC off+68dBm SIC on FD

N/A

Max handled peak SIpower

+2dBm(6dB back-off)

-4dBm SIC on FDD-8dBm SIC on SC-FD

>1.5 dBm

NF degradation due toSIC

0.8dB FDD 0.5dB one filter0.6 dB two filters

SC-FD 0.9-1.2 dB one1.1-1.5 dB two filters

(4-6)dB

RX Power Consumption 75-83 mW 63-69 mW 23-56 mW (RXincl. LO tree)

Canceller PowerConsumption

13-72 mW 0-47mW Gm cells 1 filter44 mW LO @1.35GHz

for 1 filterTX/RX isolation 25dB 33dB 27dB

SC-FD No Yes Yes

Technology 65nm CMOS 65nm CMOS 65nm CMOS

Active Area 1.2 mm2 4.8 mm2 2 mm2

*FDD, from TB measurements N/A not applicable, N/R not reported


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3. Proposed SIC solutionTo perform active cancellation a replica PA is introduced in order to cancel the TX signal at theinput of the LNA in such a way that replica PA does not interfere with the transmission at theantenna, Fig. 10 (a). The network works as follows: If the current the PA would deliver to theantenna in the absence of the LNA could be measured, and circulated in the secondary of thetransformer by a replica current source, then none of the TX current would ow in the receiver, andno voltage swing would be seen across the receiver. Accordingly, the receiver would be isolatedfrom the transmitter. Additionally, as a zero ohm impedance has been created across the LNA forthe PA signal, the PA is ideally unloaded by the LNA. For the receive signal, the network exploitsthe asymmetry in impedance between the PA and the replica. If the PA, in series with the LNA, isdesigned to have a low output impedance, while the replica, in shunt with the LNA, maintains ahigh output impedance, then the LNA can receive signal from the antenna relatively unloaded.In order to create the replica lter, the replica current DAC can be broken into a number of sub-DACs weighted by the coef cient taps, and driven with delayed copies of the TX data to becancelled, Fig. 10 (b). The coef cients of this lter can be adapted through a LMS feedback loopcontrolled through the digital baseband. This loop would sense the residual signal at the output ofthe receiver, correlate against the TX data at the corresponding time sample, and adjust thecoef cients of the lter in order to whiten this correlation to minimize the residua.

(a) (b)Fig. 10 Single antenna SIC transformer network (b) actual realization


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[20] E. Sartori, “Hybrid Transformers”, IEEE Transaction On Parts, Materials and Packaging, Vol.4, No. 3, pp.: 59-66, 1968[21] M. M. Mikhemar, A. A. Abidi, “A multiband RF antenna duplexer on CMOS: design andperformance”, Journal of Solid State Circuits, Vol. 48, No. 9, pp.: 2067-2077, 2013.[22] M. Elkholy et al., “A 1.6-2.2GHz 23dBm low loss integrated CMOS duplexer”, IEEEProceedings of Custom Integrated Circuits Conference, pp.:1-4, 2014[23] S. H. Abdelhalem, P. S. Gudem, L. E. Larson, “Tunable CMOS integrated duplexer withantenna impedance tracking and high isolation in the transmit and receive bands”, IEEETransactions on Microwave Theory and Technique, Vol. 62, No. 9, pp.:2092-2104, 2014[24] D.H. Mahrof et al, “Cancellation of OpAmp Virtual Ground Imperfections by a NegativeConductance applied to improve RF Receiver Linearity”, IEEE Journal of Solid State Circuits, Vol.49, pp.: 1112–1124, 2014

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