Design and Comparison of High Performance Stationary-Frame Controllers for DVR

602 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 22, NO. 2, MARCH 2007

Design and Comparison of High PerformanceStationary-Frame Controllers for

DVR ImplementationYun Wei Li, Member, IEEE, Frede Blaabjerg, Fellow, IEEE, D. Mahinda Vilathgamuwa, Senior Member, IEEE, and

Poh Chiang Loh, Member, IEEE

Abstract—The performance of a dynamic voltage restorer(DVR) is determined solely by its controller. The design of highperformance control algorithms for DVR control with improvedrobustness and desirable steady-state and transient characteris-tics is therefore an important area of study. In this paper, twovoltage controllers are proposed in the stationary frame for DVRvoltage regulation. A P+resonant controller is first designed toachieve good positive- and negative-sequence fundamental voltagecontrol with the virtue of having high gains around 50 Hz.Stationary-to-synchronous frame transformations carried out intraditional synchronous propotional–integral regulators are nolonger required with this method. However, with the purposeof achieving explicit robustness in face of parameter variations,an controller is also designed. Detailed design procedureis presented to show how an controller with high gainsaround 50 Hz can be synthesized through careful selection of itsweighting functions. A thorough discussion and performance com-parison of these two controllers in both transient and steady-stateconditions are also carried out. Finally, both controllers are ex-tensively tested on a laboratory 10-kV medium voltage level DVRsystem with various voltage sags and loading conditions.

Index Terms—Dynamic voltage restorer (DVR), control,nonlinear load, resonant controller, stationary-frame controller,unbalanced voltage sag.

I. INTRODUCTION

THE dynamic voltage restorer (DVR) is a series custompower device intended to protect sensitive loads from the

effects of voltage sags at the point of common coupling (PCC).A typical DVR connected system circuit is shown in Fig. 1,where the DVR consists of essentially a series connected injec-tion transformer, a voltage source inverter (VSI), inverter outputfilter and an energy storage device connected to the dc-link. The

Manuscript received October 17, 2006; revised March 13, 2006. This paperwas presented at the 21st Annual IEEE Applied Power Electronics Conferenceand Exposition (APEC’06)Dallas, TX, March 19–23, 2006. Recommended forpublication by Associate Editor H. du T. Mouton.

Y. W. Li was with the Center for Advanced Power Electronics, School ofElectrical and Electronic Engineering, Nanyang Technological University,Singapore 639798 and is now with the Electrical and Computer EngineeringDepartment, Ryerson University, Toronto, ON M5B 2K3, Canada (e-mail:[email protected])

F. Blaabjerg is with the Institute of Energy Technology, Aalborg University,Aalborg DK-9220, Denmark (e-mail: [email protected]).

D. M. Vilathgamuwa and P. C. Loh are with the Center for AdvancedPower Electronics, School of Electrical and Electronic Engineering, NanyangTechnological University, Singapore 639798 (e-mail: [email protected];[email protected]).

Digital Object Identifier 10.1109/TPEL.2006.890002

Fig. 1. Typical DVR circuit topology (single-phase representation).

power system upstream to DVR is represented by an equiva-lent voltage source and a source impedance. The basic operationprinciple of the DVR is to inject an appropriate voltage quantityin series with the supply through an injection transformer whena PCC voltage sag is detected. Loads connected downstream arethus protected from the PCC voltage sag.

The performance of a DVR is determined solely by itscontroller. The design of high performance control algorithmsfor DVR control with improved robustness and steady-state andtransient performances is therefore an important area of study.The majority of faults on a power system are unbalanced innature which results in unbalanced voltage sag at PCC (usuallywithout zero-sequence components due to the widely useddelta-star distribution transformers and three-wire mediumvoltage level in most countries [1]). Appropriate generation ofunsymmetrical compensation voltage components by a DVR tocompensate both the positive- and negative-sequence compo-nents of unbalanced voltage sags is therefore also an importantarea of research.

In this paper, two voltage controllers are designed in thestationary frame with embedded inner current loops for DVRvoltage control. A P resonant controller is first designed toachieve good positive- and negative-sequence fundamentalvoltage regulation with the virtue of having high gains aroundpositive- and negative-sequence fundamental frequencies.Complex stationary-to-synchronous frame transformationscarried out in the traditional synchronous proportional–inte-gral (PI) regulators are not required by this method [2]–[4].Then, with the purpose of achieving explicit robustness in faceof parameter variations, an controller is also designed.Detailed design procedure is presented to show how ancontroller with high gains around 50 Hz can be synthesized

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Fig. 2. Proposed DVR control scheme in the stationary frame.

through careful selection of its weighting functions. A thor-ough discussion and performance comparison of these twocontrollers in both transient and steady-state conditions is alsopresented. Finally, both controllers are extensively tested ona laboratory 10-kV medium voltage level DVR system withvarying voltage sag (balanced and unbalanced) and loading(linear and nonlinear) conditions.

II. PROPOSED DVR CONTROL SCHEME

The proposed control scheme for the DVR is illustratedin Fig. 2. As shown, the voltage at the PCC, , is mea-sured and used for voltage sag detection and for DVR injectionvoltage reference generation (at DVR side of the injection trans-former). For the DVR injection voltage control, a multiloopcontrol scheme similar to [13] is implemented as illustratedin Fig. 2, where an inner filter inductor current feedbackloop is shown embedded within an outer filter capacitor voltage

(or voltage at DVR side of injection transformer)feedback loop. Both the voltage and current controllers areimplemented in the stationary frame to avoid reference frametransformations. An algorithm is also implemented to takenecessary actions for reference voltage change rate limit, trans-former saturation limit, and overmodulation protection (shownin Fig. 2 as “inverter voltage reference limit” [1]).

After a voltage sag is detected, the difference between thereference PCC voltage and measured is divided bythe injection transformer turns ratio to determine the referenceDVR injection voltage in the stationary – frame. TheDVR injected voltage feedback is compared with its referenceand the error is fed to a voltage controller. In this work two typesof voltage controllers are investigated, namely a P resonantcontroller and an controller. Both controllers are imple-mented in the stationary – frame and with significant gainsin the vicinity of positive and negative fundamental frequen-cies ( 50 Hz) to ensure almost zero steady-state error. A de-tailed discussion and comparison of these two controllers arepresented in Sections III–V.

The output of the voltage controller is then transformed to thea–b–c frame to generate the reference current which is com-pared with the filter inductor current feedback . The currenterror is fed to a proportional controller whose output gives thedesired voltage to be generated by the inverter, and issubsequently passed to the pulsewidth modulation (PWM) gen-erator. Note that the inner current loop mainly functions to im-prove the dynamic stability of the voltage control scheme, andas such the proportional controller used will not affect the ac-curacy of voltage tracking of the outer voltage loop irrespectiveof whether the P resonant or controller is being used. In

Fig. 3. Bode diagram of an ideal P+resonant controller.

the next few sections (Sections III–V), the proposed two voltagecontrol schemes will be investigated.

III. REVIEW OF P RESONANT CONTROLLER

The ideal P resonant controller, expressed in (1), can bemathematically derived by transforming a synchronous-framePI controller to the stationary frame without consideration ofthe redundant cross coupling terms [2], and has an infinite gainat the controller’s resonant frequency , which in this case ischosen to be the line fundamental frequency (2 50 rad/s)

(1)

In [3]–[5], it is shown that by transforming PI controllers inboth positive- and negative-sequence synchronous frames of athree-phase system to the stationary frame (using either a fre-quency domain or time domain technique), the final stationarycontroller of (1) can be obtained, and the cross coupling termsgenerated from positive- and negative-sequence synchronousframes cancel each other if the same PI parameters are employedin both synchronous frames. Therefore, the P resonant con-troller has infinite gains at both the positive and negative fun-damental frequencies and in principle will achieve zero steady-state error for both positive- and negative-sequence regulation.A typical Bode plot for the ideal P resonant controller is shownin Fig. 3. As illustrated, the ideal P resonant controller has in-finite gains at 50 Hz. Note that the controller gains at otherfrequencies are quite small and are determined by the value of

(gains at frequencies around 50 Hz are also related to ).Although theoretically the ideal P resonant controller

achieves zero steady-state error by having infinite gains at50 Hz, there can be practical problems during its implemen-

tation, particularly as it is sensitive to frequency variations.More realistic forms are therefore proposed in [2], [6], and the


practical P resonant controller in [2] is adopted as expressedin

(2)

where is the controller cutoff frequency. The frequency re-sponse of (2) is drawn in Fig. 7(a) (together with the proposed

controller bode plot), which shows that the controller hasa wider bandwidth around the resonant frequency, which mini-mizes the sensitivity to slight frequency variation, at the expenseof a reduced resonant peak. However, the resulting 44-dB reso-nant peak is still sufficient for fundamental tracking error elim-ination. This practical P resonant controller has three parame-ters, proportional gain for overall gain regulation, resonantgain for the resonant peak adjustment and cutoff frequency

for resonant bandwidth control. An extensive discussionabout these parameters and their effects on the overall systemperformance is presented in Section V.

IV. DESIGN DESCRIPTION OF THE CONTROLLER FOR DVR

Developed in the later last century, robust control has beingwidely used in industry plants. With explicitly specified degreeof robustness in face of plant uncertainties and disturbances,the control method has been implemented in dc–dc con-verter [7], high frequency resonant inverter [8] and single-phaseUPS inverter [9]. In the mixed-sensitivity loop shapingproblem, the design specifications such as tracking performanceand robustness performance are expressed as constraints on thesingular values (or gain responses) of transfer functions of dif-ferent loops (from input to error or closed-loop transfer func-tion). Therefore those loops can be shaped by choosing properweighting functions for those constraints, and the con-troller can be synthesized by using a software package (RobustControl Toolbox in Matlab [10]) which finds the suboptimal oroptimal solution (by iteration) of the mixed-sensitivity problemin state space by solving two algebraic Riccati equations.

The standard configuration for control is shown inFig. 4, where is the nominal plant; is the desiredcontroller; , , and are the controlled output, the measuredoutput, the exogenous input and the control input respectively.

, and are the weighting functions for the trackingerror performance, robust performance and the weight on thecontroller transfer function respectively. The controllersynthesis is conducted by singular value loop-shaping usinga mixed sensitivity approach, such that the norm of thetransfer function from to is less than unity. This can beexpressed as

or equivalently (3)

Where 1is the complementary sensitivity transfer function, and

1 1 is the sensitivity transferfunction. It can be seen from (3) that the mixed sensitivityapproach is simply the shaping of (transfer function fromreference to output, or closed loop transfer function) and(transfer function from reference to error), achieved by properlyselecting their respective weighting functions and . Notethat a small value of 0.1 is assigned to to ensure the

Fig. 4. StandardH1 configuration used for DVR voltage control.

matrix of the augmented plant is of full rank as required by theMatlab augtf and hinf functions [10].

A. Weighting Function Selection for Robust Performance

The nominal plant, with the inner closed-loop current con-troller, is expressed as

(4)

where is the proportional gain for the current loop. Due tothe difficulties in making parameter variations to the laboratoryDVR prototype, the controller is designed based on 2/3 ofthe known system parameter values and then implemented onthe original system to test its performance. System parameteruncertainty is chosen to vary from 80% to 150% of the nominalDVR model values (which is quite representative for the param-eter variations due to aging, saturation and thermal effects, etc.),so that the synthesized controller is expected to performwell with the original system (note that load is not consideredas a part of DVR model and the load current can be treated asan additional disturbance source). This parameter uncertainty istransformed to multiplicative output uncertainty [11], and theresultant relative plant uncertainty with respect to the nominalplant is expressed as

(5)

where is the plant uncertainty, is the disturbed plantand stands for the singular value of .

The relative plant uncertainty with a combination of dif-ferent parameter values (varying from 80% to 150% of the nom-inal values) is plotted in Fig. 5 (with the dashed line indicatingthe worst-case uncertainty boundary). For the robustness mea-sure with 1, is selected as above the worst-caseboundary as shown in Fig. 5, where 0.000266 s 0.8.

B. Weighting Function Selection for Tracking ErrorPerformance

The tracking error performance can be expressed in terms ofthe constraint as 1. For properly shaping thesensitivity function , and therefore achieving satisfactorytracking error performance, the weighting function shouldbe carefully determined. Since the reference for the DVR is a si-nusoidal command only at the line frequency, it would be prefer-able to use a controller that exhibits high gains at vicinity ofthe line frequency while having reduced gains at the other fre-quencies similar to the P resonant controller. Tracking errorsat the line frequency can be made small by properly shaping


Fig. 5. Singular values of �(s) and W .

Fig. 6. Singular values of weighting functions of 1=W and S(s).

the sensitivity transfer function to have very small gain atthe line frequency. This can be achieved by using a standardsecond order weighting function for as suggested in [9].This weighting function is shown in

(6)

where the natural frequency is selected to be the line fre-quency, in the numerator gives a freedom for adjustmentof the tracking error over the whole frequency range and thedamping ratio provides another degree of freedom for regu-lating the tracking error around . Since the resultant sensi-tivity function is shaped in frequency according to the pro-file specified by 1 , as shown in Fig. 6, a smaller will givea larger peak response of at the natural frequency and thusguarantee a smaller steady-state error at the line frequency. As

Fig. 7. Bode plot of (a) controllers and (b) open-loop system using differentcontrollers.

0, the resultant controller would behave like an idealP resonant controller, which has theoretically infinite gain andzero steady-state error at 50 Hz. The tracking error at otherfrequencies can be regulated by tuning in (6).

Having defined the necessary weighting functions, a mixedsensitivity optimization control design can be conducted to syn-thesize an controller such that the norm of thetransfer function from to is less than unity, as expressed in(3). By using Matlab hinf function, it can be shown that the re-sultant controller would have the same order as the plantmodel augmented with the weights. The controller is sub-sequently reduced to a third-order controller as shown in (7)for implementation convenience. A bode plot of (7) is shownin Fig. 7(a)

(7)


Fig. 8. Robustness analysis for the H1 controller: (a) (b) injected and loadvoltages with 2/3 system parameters and (c) (d) injected and load voltages withnominal system parameters.

Fig. 9. Robustness analysis for the P+resonant controller: (a) (b) injected andload voltages with 2/3 system parameters and (c) (d) injected and load voltageswith nominal system parameters.

V. PERFORMANCE AND ROBUSTNESS COMPARISON

As discussed in the previous sections, both of the P resonantcontroller and the synthesized controller exhibit signifi-cant gains at the vicinity of 50 Hz with very good voltagetracking. However, characteristics deviate at other frequencies[see Fig. 7(a)] resulting in differences in system dynamics androbustness.

A. Performance Discussion

The practical P resonant controller in (2) has three parame-ters for tuning according to specific control objectives. The tran-sient performance of the DVR system with P resonant con-troller mainly depends on the proportional gain . As it can

Fig. 10. Experimental 10-kV DVR system setup.

Fig. 11. PCC voltage: (a) with P+resonant controller and (b) with H1controller.

be seen from Fig. 7(a), the gains at frequencies other than theneighborhood of the controller resonant frequency remain con-stantly low and are related to . With a larger , the systemwill have a wider bandwidth and thus faster transient response toa change in reference signal. A larger also means high am-plification of the DVR filter resonance [for the new plantwith inner current loop closed, this resonant frequency isshifted to a higher frequency as shown in Fig. 7(b)], which willcause transient resonant oscillations or even affect the systemstability. To ensure a system with reasonably fast dynamics, thevalue of should not be too small. The system steady-stateerror is mainly governed by the controller resonant peak. Asthe resonant gain determines the peak value, a large gainvalue can be assigned to without substantially affecting thetransient and system stability. The cutoff frequency of the


Fig. 12. Injected voltages: (a) with P+resonant controller and (b) with H1controller.

Fig. 13. Load voltage: (a) with P+resonant controller and (b) with H1controller.

P resonant controller determines the resonant bandwidth of thecontroller. For a DVR system, where a voltage sag is usually ac-companied with a phase angle jump, a wider resonant bandwidthwould be preferable, since the angle jump will affect the PLLperformance and produce effects similar to a frequency change.A large would also improve the transient dynamics, but thesystem would be driven into filter resonance if the controllerbandwidth is extended beyond the filter cutoff frequency.

The third order controller, on the contrary, would exhibitvery good resonance attenuation while maintaining a rea-sonable transient response by the virtue of having an additionalpole with fast roll-off at high frequency [see Fig. 7(a)]. As men-tioned in Section IV, the controller is tuned by selectingtwo weighting functions for the robustness and error trackingperformance. The robust performance design is intended to pro-duce fast high frequency roll-off characteristics for the

Fig. 14. Zoom-in view of injected voltages: (a) with P+resonant controller and(b) with H1 controller.

Fig. 15. Frequency spectra of injected voltage: (a) with P+resonant controllerand (b) with H1 controller.

controller and make it immune to the filter resonance. Theerror tracking performance is determined by the weight func-tion in (6). As mentioned, error tracking at 50 Hz can beaccurately maintained by having a small . Similar to the pro-portional gain of P resonant controller, the overall gain ofthe controller on the frequency spectrum can be regulatedby tuning the parameter in (6). To ensure the existence of

controller, the weighting functions cannot be chosen ar-bitrarily. A basic criterion is the 0 dB crossover frequency of

must be sufficiently below the 0 dB crossover frequency of.

For the sake of consistency, control parameters of both con-trollers are chosen to give the same resonant width and peak(and thus same steady-state performance) as shown in Fig. 7(a)(with , , ). Moreover, the sameinner current gain is used in each approach (a large

would flatten the filter resonant peak, but is alwayslimited due to practical considerations such as amplifications of


Fig. 16. Frequency spectra of load voltage: (a) with P+resonant controller and(b) with H1 controller.


capacitor current noise, measurement noise and dc offset). Theresultant controller and system open loop bode plots are illus-trated in Fig. 7(a) and (b). It can be seen that both the systemshave similar bandwidths. However, compared to the P resonantcontroller, which is sensitive to the filter resonance, thecontroller would make the system more immune to the filterresonance by having a steeper high frequency attenuation.

It can be shown in Fig. 7(a) and (b) that the P resonantcontroller has higher low frequency attenuations below 50 Hz,which is due to the small (0.4) limited by the considerationat filter cutoff frequency as discussed earlier. This low fre-quency attenuation would filter out dc offsets present in most in-tegrated circuit (IC) chips (as used in analog interfacing boards)and other sub-harmonics or sub-resonance disturbances. It wasobserved in experiments that the relatively higher low frequencygains of the controller can cause stability problems. Such

Fig. 18. Injected voltage: (a) with P+resonant controller and (b) with H1controller.


a situation may arise due to imperfect calibrations of measure-ment, ADC conversion effects or distorted DVR references gen-erated from the PCC, where the presence of low frequency com-ponents can be attributed to phenomena such as ferro-resonanceof the distribution transformer. To reduce low frequency gains,

in (6) should be kept small enough ( 1.1 in this work).

B. Robustness Comparison

As illustrated in Fig. 7(b), the system with the con-troller has much better attenuation of high frequency distortions.The immunity to high frequency disturbance especially aroundthe filter resonance frequency is responsible for the systemrobustness.

A robustness comparison of the P resonant and thecontrollers was carried out based on simulations. As discussedin Section IV, the controller was designed based on 2/3 ofthe nominal system parameter values with an estimated range of


Fig. 20. Zoom-in view of injected voltages: (a) with P+resonant controller and(b) with H1 controller.

Fig. 21. Frequency spectra of injected voltage: (a) with P+resonant controllerand (b) with H1 controller.

parameter variations from 80% to 150%. Therefore, when im-plemented on the original system there should be no degradationof performance due to its specified robustness. This is confirmedin Fig. 8, where waveforms (a) and (b) are the DVR injectionvoltage and load voltage with 2/3 of filter inductance and filtercapacitance values, and waveforms (c) and (d) are the injec-tion voltage and load voltage with original system parameters.Therefore, it can be seen that the DVR compensation perfor-mance with controller remains very robust despite systemparameter variations.

For the purpose of comparison, a P resonant controller wasspecifically tuned with 2/3 of the filter inductance and capac-itance values. Its performance in terms of DVR compensationvoltage and load voltage are shown in Fig. 9(a) and (b), respec-tively. As has been discussed, the P resonant controller maynot have adequate robustness for parameter variations. This lack

Fig. 22. Frequency spectra of load voltage: (a) with P+resonant controller and(b) with H1 controller.


of robustness is verified in Fig. 9(c) and (d), where the DVR in-jecting voltage shows serious transient resonance under param-eter variations. This transient resonant response, if not dampedeffectively, may hamper the functioning of the load or evencause the system to become unstable.

VI. EXPERIMENTAL INVESTIGATIONS

The proposed and P resonant controllers have been im-plemented on a laboratory 10-kV DVR system and extensivelytested in experiments. The 10-kV DVR hardware prototype con-figuration is illustrated in Fig. 10. The ac supply is a 15-kVACalifornia Instrument Supply programmed at 380 V. A 50-kVAstar-delta transformer steps up the voltage to 10 kV. The loadvoltage is stepped down using another 50-kVA delta-star trans-former. The DVR is connected at the 10-kV level through three67-kVA single-phase series injection transformers. The dc-busvoltage of the DVR can be charged up to 600 V using two dc sup-plies, which provide a maximum of 4680-J energy in the 26 mF


Fig. 24. Injected voltage: (a) with P+resonant controller and (b) with H1controller.


of dc capacitors, for transient use in the DVR sag ride-throughcapabilities.

Control of the DVR is implemented on a dual digital signalprocessor (DSP)-Microcontroller system, with an Analog De-vices AD21026 floating-point Sharc DSP for implementation ofcontrol algorithm and a Siemens SAB 80C167 Microcontrollerfor PWM signal generation for the six IGBT phase-legs. A DualPort RAM unit (DPRAM) is used as a communication link be-tween the DSP and Microcontroller, where the results from theDSP are transferred to DPRAM and are read by the Microcon-troller. Analog signals are converted to digital signals using twoeight channel AD 7891 A/D converters and read by the DSPonce in each switching cycle.

Once the measured PCC voltage magnitude drops below 90%of its nominal value, a voltage sag is detected. DVR compen-sates for the voltage sag with a reference generated by the errorbetween PCC voltage reference (calculated using a pre-set ref-

Fig. 26. Negative-sequence components of the PCC and load voltages: (a) withP+resonant controller and (b) with H1 controller.

erence voltage magnitude and a PLL [1], [12]), and the mea-sured PCC voltage, expressed as (see Fig. 2). TheP resonant controller is tuned based on the system parametersgiven in Table I in the Appendix. However the designedcontroller based on 2/3 of the system parameter values is im-plemented in the experiment for robustness verification. Bothvoltage controllers are transformed to discrete form using bi-linear transformation before implemented on the DSP controllerboard.

A. Balanced Voltage Sag and Linear Load

The first experiment is carried out for a balanced voltage sagand a linear load (67 ). The PCC voltage drops to 70% of itsnominal value from 40 ms to 140 ms (five fundamental cycles).The PCC voltage, DVR injected voltage (at high voltage side ofthe injection transformer) and load voltage with both controllersare shown in Figs. 11–13. It can be seen that both controllers per-form load voltage restoration effectively during a PCC voltagesag. However there is still a slight difference between the DVRinjected voltages during the startup transient, which can be seenin the zoomed-in view of the injected voltage in Fig. 14. TheP resonant controller gives slight startup oscillations, this isdue to the relatively high gains of the controller at high frequen-cies and therefore high amplification of the filter resonance,which makes the system oscillate at the resonant frequencyonce it is excited. The controller, on the contrary, has verysmooth transient during startup as it has fast high frequencyroll-off characteristics. This can also be verified by comparingresults shown in the frequency spectra in Figs. 15 and 16. TheDVR injected voltage with controller has less THD (1.9%)compared to the P resonant controller (THD 2.1%), due tothe increased high frequency attenuation of the controllerFig. 15(b). This difference is seen in load voltage too. As shownin Fig. 16, the load voltage with controller has a little betterTHD (0.77%) with lower high frequency harmonics comparedto that with P resonant controller (with THD of 0.8%). In gen-eral, both the P resonant and controllers have good per-formance with the linear load. There is only slight differenceduring start up.


TABLE ISYSTEM PARAMETERS

B. Balanced Voltage Sag and Nonlinear Load

The DVR performance with a nonlinear load and a balancedPCC voltage sag is also investigated. The nonlinear load is adiode rectifier bridge with a capacitor bank (200 F) and resis-tive load (80 ) connected in parallel. The PCC voltage dropsto 70% of its nominal value from 40 ms and lasts for five cy-cles. The PCC voltage, DVR injected voltage and resultant loadvoltage are shown in Figs. 17–19.

It can be observed that with the nonlinear load connected atdownstream, the PCC voltage, DVR injected voltage and loadvoltage are distorted. The distortion of PCC voltage is due to thevoltage drop across the source impedance (mainly the step-uptransformer in this experiment). The distorted PCC voltagegives rise to a distorted DVR reference voltage ( ,where is a sinusoid but has harmonic components)and subsequently a distorted injected voltage. However, furtherdistortion in the injected voltage is observed in Fig. 18(a) withP resonant controller compared to that with controller [inFig. 18(b)]. The same effect is shown clearly at the zoomed-inwaveforms in Fig. 20(a) and (b). The increased distortion withP resonant controller is due to the fact that the nonlinear loadwill continue to introduce high frequency harmonic excitations.The controller in this case is a better option due to its highfrequency disturbance rejection capability. As can be seen fromFig. 21(a) and (b), the injected voltage with controllerhas much less 7th, 11th, and 13th harmonics and less THD(3.7%) compared to the voltage with P resonant controller(with THD of 5.8%). These different injection voltages causethe load voltage with the controller to have smaller highfrequency harmonics and smaller THD of 2.5% (compared to2.9% THD with P resonant controller). It can be concludedthat with nonlinear load, the advantages of controller

TABLE IISUMMARIZED PERFORMANCE FEATURES OF

P+RESONANT AND H1 CONTROLLERS

become more obvious with its effective attenuation of highfrequency harmonic distortions (see Fig. 22).

C. Unbalanced Voltage Sag

The last experiment is carried out with an unbalanced voltagesag. The PCC voltages shown in Fig. 23 are programmed with apositive-sequence voltage drop of 20% and an injection of 10%negative-sequence voltage for five cycles starting from 40 ms.The corresponding DVR injection voltages and load voltages areshown in Figs. 24 and 25. It can be seen that both the P resonantand controller would effectively inject appropriate unbal-anced voltages and symmetrically restore the load voltage to itsnominal value with the sufficiently large gains at 50 Hz.

A sequence analysis of the PCC voltages and load voltagesis shown in Fig. 26(a) and (b). With both controllers, thenegative-sequence component of the load voltage is reducedto 1.6%. Note that the waveform oscillatory transients inFig. 26(a) and (b) are a fluke and are introduced by the lowpass filtering effect (averaging data over one fundamental cycleto filter out the negative/positive-sequence components in posi-tive/negative synchronous frame) of the sequence componentsanalysis tool used in offline analysis.

VII. CONCLUSION

In this paper, two voltage controllers, namely a P resonantcontroller and an controller, are proposed in the stationaryframe for DVR voltage regulation. Both controllers have goodsteady-state sinusoidal error tracking of the positive- and nega-tive-sequence components with the virtue of having high gainsaround positive- and negative-sequence fundamental frequen-cies. A detailed analysis of parameter tuning for the P resonantcontroller and weighting function selection for the con-troller design is presented. A thorough discussion and perfor-mance comparison of these two controllers in both transient andsteady-state conditions are also carried out. Finally, both con-trollers are extensively tested on a laboratory 10-kV medium


voltage level DVR system, with varying voltage sag and loadingconditions. It is shown that both controllers are equally effec-tive in unbalanced voltage regulation. With linear loads, bothcontrollers perform well with slight difference during startuptransient. However, the advantages of the robust controllerbecome more obvious with nonlinear loads due to its better at-tenuation of the high frequency harmonic distortions. Perfor-mance comparisons of the proposed controllers are summarizedin Table II in the Appendix.

APPENDIX

See Tables I and II.

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[12] J. G. Nielsen, M. Newman, H. Nielsen, and F. Blaabjerg, “Control andtesting of a dynamic voltage restorer (DVR) at medium voltage level,”IEEE Trans. Power Electron., vol. 19, no. 3, pp. 806–813, May 2004.

[13] D. M. Vilathgamuwa, A. A. D. R. Perera, and S. S. Choi, “Performanceimprovement of the dynamic voltage restorer with closed-loop loadvoltage and current-mode control,” IEEE Trans. Power Electron., vol.17, no. 5, pp. 824–834, Sep. 2002.

Yun Wei Li (S’04–M’06) received the B.Eng degreein electrical engineering from Tianjin University,Tianjin, China, in 2002, and the Ph.D. degree fromthe School of Electrical and Electronic Engineering,Nanyang Technological University, Singapore, in2006.

From February to August 2005, he was attachedto the Institute of Energy Technology, AalborgUniversity, Aalborg, Denmark, as a Visiting Scholar.Currently, he is a Postdoctoral Fellow in the Elec-trical and Computer Engineering Department,

Ryerson University, Toronto, ON, Canada.Dr. Li is a member of the IEEE Industrial Application Society.

Frede Blaabjerg (S’86–M’88–SM’97–F’03) wasborn in Erslev, Denmark, on May 6, 1963. Hereceived the M.Sc.EE. and Ph.D. degrees fromAalborg University, Aalborg, Denmark, in 1987 and1995, respectively.

He was with ABB-Scandia, Randers, Denmark,from 1987 to 1988. He became an Assistant Pro-fessor in 1992 at Aalborg University, in 1996 anAssociate Professor, and in 1998 a Full Professor inpower electronics and drives. Today he is also Deanof the Faculty of Engineering Science and Medicine.

In 2000, he was a Visiting Professor with the University of Padova, Padova,Italy, as well as a part-time Programme Research Leader in wind turbines atthe Research Center Risoe. In 2002, he was a Visiting Professor at CurtinUniversity of Technology, Perth, Australia. He is involved in more than tenresearch projects within the industry. Among them is the Danfoss ProfessorProgramme in Power Electronics and Drives. He is the author or coauthor ofmore than 500 publications in his research fields including Control in PowerElectronics (New York: Academic, 2002). He is an Associate Editor for theJournal of Power Electronics and Elteknik. He has been very involved inDanish Research policy in the last ten years. His research interests are in powerelectronics, static power converters, ac drives, switched reluctance drives,modeling, characterization of power semiconductor devices and simulation,wind turbines, and green power inverters.

Dr. Blaabjerg received the 1995 Angelos Award for his contribution inmodulation technique and control of electric drives, the Annual Teacher Prizefrom Aalborg University, in 1995, the Outstanding Young Power ElectronicsEngineer Award from the IEEE Power Electronics Society in 1998, five IEEEPrize paper awards during the last five years, the C. Y. O’Connor fellow-ship from Perth, Australia in 2002, the Statoil-Prize for his contributions inpower electronics in 2003, and the Grundfos-prize for his contributions inpower electronics and drives in 2004. He is an Associate Editor of the IEEETRANSACTIONS ON INDUSTRY APPLICATIONS and the IEEE TRANSACTIONS ON

POWER ELECTRONICS. He is a member of the Danish Academy of TechnicalScience, the European Power Electronics and Drives Association, and theIEEE Industry Applications Society Industrial Drives Committee. He is also amember of the Industry Power Converter Committee and the Power ElectronicsDevices and Components Committee, IEEE Industry Application Society.

D. Mahinda Vilathgamuwa (S’90–M’93–SM’99)received the B.Sc.degree in electrical engineeringfrom the University of Moratuwa, Sri Lank, in 1985and the Ph.D. degree in electrical engineering fromCambridge University, Cambridge, UK, in 1993.

He joined the School of Electrical and ElectronicEngineering, Nanyang Technological University,Singapore, in 1993 as a Lecturer and he is now anAssociate Professor. He has published more than80 research papers in refereed journals and confer-ences. His research interests are power electronic

converters, electrical drives, and power quality.Dr Vilathgamuwa was the co-chairman of Power Electronics and Drives Sys-

tems Conference in 2005 (PEDS’05).

Poh Chiang Loh (S’01–M’04) received the B.Eng(with honors) and M.Eng degrees from the Na-tional University of Singapore in 1998 and 2000,respectively, and the Ph.D. degree from MonashUniversity, Victoria, Australia, in 2002, all in elec-trical engineering.

During the Summer of 2001, he was a VisitingScholar with the Wisconsin Electric Machineand Power Electronics Consortium, Universityof Wisconsin, Madison, where he worked on thesynchronized implementation of cascaded multilevel

inverters, and reduced common mode carrier-based and hysteresis controlstrategies for multilevel inverters. From 2002 to 2003, he was a Project Engineerwith the Defence Science and Technology Agency, Singapore, managing majordefence infrastructure projects and exploring new technology for intelligentdefence applications. Since 2003, he has been an Assistant Professor withNanyang Technological University, Singapore.

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Design and Comparison of High Performance Stationary-Frame Controllers for DVR

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