162 IEEE TRANSACTIONS ON ELECTRON DEVICES, VOL. 41, NO. 2, FEBRUARY 1994

Design and Performance Analysis of InP-Based High-speed and High-Sensitivity

Optoelectronic Integrated Receivers Eugene John and Mukunda B. Das, Senior Member, IEEE

Abstract-A novel transimpedance optoelectronic receiver am- plifier suitable for monolithic integration is proposed and ana- lyzed by exploiting state-of-the-art high-speed MSM photodiodes and HBT’s based on lattice-matched InCaAs/InAIAs heterostruc- tures on InP substrates. The projected performance character- istics of this amplifier indicate a high transimpedance (% 3.6 kn), a large bandwidth (17 GHz), and an excellent optical detection sensitivity (-26.8 dBm) at 17 Gb/s for the standard bit-error-rate of lo-’. The latter coresponds to an input noise spectral density, m, of 2.29 PA/- for the full bandwidth. The bandwidth of the amplifier can be increased to 30 GHz for a reduced transimpedance (0.82 kn) and a lower detection sensitivity, i.e., -21 dBm at 30 Gbls. The amplifier also achieves a detected optical-to-electrical power gain of 21.5 dBm into a 50 S2 load termination. The design utilizes small emitter-area HBT’s for the input cascoded-pair stage, followed by a two-step emitter-follower involving one small and one large emitter-area HBT’s. The design strategy of using small emitter-area HBT’s is matched by a low-capacitance novel series/parallel connected MSM photodiode. This combined approach has yielded this amplifier’s combined high performance characteristics which exceed either achieved or projected performances of any receiver amplifier reported to-date. The paper also discusses the issues concerning IC implementation of the receiver, including the means of realizing a high-value feedback resistor.

I. INTRODUCTION N the design of receiver optoelectronic integrated circuits I (OEIC’s) for optical data links, the low dispersion and high

transparency of optical fibers at wavelengths of 1300 and 1550 nm, respectively, necessitates the use of InP-based alloys, and in particular the lattice-matched Ino.53Gao.47 As material with a band gap of 0.75 eV. Hence the transistor technology must also be based on InP substrates, in order to avoid strained layer epitaxy, which among other things complicates the material growth problems. InGaAs which can be grown lattice-matched to InP substrates, has been shown to be a useful photodetector material which will absorb light up to wavelengths as long as 1650 nm. Many research groups have been successful in designing and fabricating monolithic lightwave receivers using p- i -n or MSM photodiodes as lightwave detectors, and HBT’s or MODFET’S as amplifying transistors, in the preferred

Manuscript received May 13, 1993; revised August 30, 1933. The review of this paper was arranged by Associate Editor P. K. Bhattacharya. This work was supported in part by the National Science Foundation under Grant Number

The authors are with Department of Electrical and Computer Engineering, and Electronic Materials and Processing Research Laboratory, The Pennsyl- vania State University, University Park, PA 16803.

ECS-9202642.

IEEE Log Number 9214400.

wavelength range of 1300-1550 nm, and they have reported impressive results [ 11-[5].

In recent years both the HBT’s and MODFET’s have demonstrated their high bandwidth and low noise capabilities beyond 100 GHz as discrete devices, particularly those based on InP substrates [6]-[12]. On the same substrate material MSM photodetectors involving InGaAs active layers have been demonstrated with a ultra-high bandwidth (> 100 GHz) [13]. By combining this high-speed MSM detector with the state-of-the-art HBT’s or MODFET’s in suitable amplifier con- figurations future high performance optoelectronic receivers are expected to be built. Recently we reported [14] the projected performance of a transimpedance feedback amplifier based on a cascoded pair of HBT’s with (1 pm x 25 pm) emitter area designed for standard 50 R output terminations. It was shown that using HBT’s with intrinsic unity current gain frequency fT of 160 GHz, receivers with effective 3- dB bandwidth of 16 GHz with a transimpedance of M 500 R can be achieved. However the sensitivity of such a receiver is limited due to the thermal noise associated with the low feedback resistance ( R f = 500 R), and also due to the shot noise associated with the rather large base current of the HBT. These HBT’s are required to operate at a high current density and their emitter sizes are appropriately tailored to achieve an effective transconductance ( g m ) of 200 mS so that they are capable of providing a transimpedance gain into a 50 R load. From these results it becomes apparent that the sensitivity of the receiver, with respect to noise, can be improved by reducing the emitter-area of the HBT’s used without changing the operating collector current density (w 3 x lo4 A/cm2). This will necessitate a corresponding increase in the collector load resistance of the cascoded circuit which in tum will require an increased feedback resistance value. Thus, in principle both high transimpedance and low-noise performance can be achieved without sacrificing the bandwidth. For the desired 50 R load termination, the cascoded amplifier stage can be fed into a two-step emitter-follower with a small emitter-area transistor at the input, and a large emitter area transistor at the output. Use of HBT’s with small emitter area for the cascoded amplifier reduces its input capacitance which in turn requires that the photodetector capacitance be extremely low. In order to meet this requirement we have used a novel series/parallel connected MSM photodiode which effectively reduces its capacitance by one-fourth without reducing the responsivity below one-half, for the same surface area.

00 18-9383/94$O4.00 0 1994 IEEE

163 JOHN AND DAS: DESIGN AND PERFORMANCE ANALYSIS OF OPTOELECTRONIC INTEGRATED RECEIVERS

This paper is concemed with the design and analysis of a novel casoded transimpedance amplifier circuit suitable for monolithic integration. This amplifier is characterized by high bandwidth, high transimpedance, and low input noise current. This circuit has been designed by a careful choice of small and large emitter-area HBT’s. However, before proceeding with the circuit design aspects of the optoelectronic receiver, it is essential to have a brief discussion on the basic operational and structural aspects of the MSM photodiodes and HBT’s, as considered below.

11. 1II-V DEVICES FOR RECEIVER DESIGN

A. The MSM Photodiode

High-speed photodetectors fabricated with a technology compatible with that of the amplifying devices, are a key element of high speed receiver systems. General device charac- teristics that should be considered in choosing a photodetector (PD) include quantum efficiency, response speed, and device noise. For high-speed OEIC’s the choice for the PD is either a p-i-n or an MSM photodiode as used in receivers based on both HBT’s [2] and MODFET’s [ 13 [4]. A major advantage of the MSM detector arises due to its low capacitance which can provide high receiver bandwidth. Because of its planar nature the MSM photodiodes can also be conveniently arranged in series and or series/parallel fashion to reduce its capacitance further without any serious reduction of its responsivity, as discussed later in this section. From the point of view of minimizing dark current, the MSM diode is not generally as effective as the reverse biased p-z-n diode. However, dark currents less than lop4 A/cm2 have been demonstrated [ 151 in MSM diodes, and even this amount of dark current is usually not the limiting factor in an optoelectronic receiver where the amplifying HBT’s base current is considerably higher, typically by one or two orders of magnitude.

In the optical absorption layer of MSM photodetectors hole- electron pairs are generated by the photon energy hv, where h is the Planck’s constant, and v is the frequency of the incident light wave of optical power, Popt, at a wavelength of A(Av = c, where c is the velocity of light). A related figure of merit of the photodetector is its responsivity, R,, which is the ratio of the photocurrent to the incident optical power, i.e.,

where A is in pm and q is the quantum efficiency defined as the number of electron-hole pair generated per incident photon, and q is the electronic charge. Typical values of R, reported for p-z-n and MSM photodetectors lie in the range of 0.3 to 0.6 A/W depending on the value of q, the factor determining the surface reflection, and the thickness of the absorption layer. The absorption layer thickness is usually selected from the consideration of the PD’s optical sensitivity and the carrier transit time. For InGaAs PD’s the choice of thickness (d) lies between 0.5 pm to 1.5 pm.

O.5pm 0.5pm U

200 i-AllnAs

500 8, CAIlnAdlnGaAs (S.L.)

0.5 pm IT InGaAs

buffer layer

S.I. InP

Fig. 1. Schematic cross-sectional details of the proposed MSM photodiode.

A cross-sectional structural detail of the proposed MSM PD for integration is shown in Fig. 1. An essential feature of this diode is the incorporation of a surface AlInAs layer to enhance the Schottky barrier height. Another important feature is the inclusion of a superlattice of AlInAs/GaInAs altemating layers intended to prevent hole pileup within the PD’s absorption layer [16]. In the MSM structure we assume that the adjacent metallization fingers belong to the two metal electrodes forming the anode and cathode of the diode and they are separated by a distance L = 0.5 pm with a finger stripe width W = 0.5 pm. We also assume that each finger has a length of 10 pm and there are 40 fingers, which gives an effective active area of 400 pm2. Fig. 2(a) gives the theoretically generated capacitance per unit length of the MSM diode [17]. Using this diagram the total capacitance of the MSM diode, with finger length 10 pm and total number of fingers 40, can be calculated to be 24 fF.

The MSM diode can be arranged as a three-terminal device, having a common central electrode which can be directly con- nected to the input terminal of the transimpedance amplifier, and the other electrodes for biasing. In this arrangement the individual half-diode will contribute a capacitance of 12 fF for a total length of 200 pm. Thus the total capacitance of the diode will be 24 fF. In order to reduce the capacitance further, the diode (with the same area) can be arranged into series elements before placing them in parallel as shown in Fig. 2(b). In this configuration the total effective capacitance of the diode will be 6 fF, a reduction by a factor of four, as expected. However, it can be shown from the equivalent circuit model, as shown in Fig. 2(c), that the effective responsivity of this seriedparallel structure is half of what can be expected from the two-element parallel structure. To the best of our knowledge this is the first time this novel arrangement for the MSM diode has been suggested. This type of MSM diode arrangements can also be used as balanced twin-photodetectors for optical coherent receivers [ 181.

Another important parasitic effect of the MSM diode that needs to be considered is the series resistance associated with its electrode metallization. For practical gold metallization thickness of 0.1 pm, the relevant sheet resistance can be calculated to be approximately 0.2 R per square. Based on this sheet resistance value an approximate series resistance of 7 R and 5 R can be calculated for the parallel and series- parallel MSM structures, respectively. Thus the respective

164 IEEE TRANSACTIONS ON ELECTRON DEVICES, VOL. 41, NO. 2, FEBRUARY 1994

o

A \

0.1 0 . 3 0 . 5 0.7 0 .9 WI(W+L)

(a)

n’lnGaAs n+ -200 i, 5x1d’cm.~

n AllnAs Base

-1 U

lp/4 T I 6fF

AMPLlflER I), I- I LOWER I

I I HALF

(C)

Fig. 2. (a) Theoretically predicted MSM capacitance per unit length versus the metallization finger width (W) to width plus the gap ( V7 + L ) ratio. (b) Topological electrode layout of three-terminal series/parallel connected MSM photodiode and (c) Its equivalent network model.

parasitic RC time constant values can be calculated to be 0.17 ps and 0.03 ps, respectively. As will be seen later that optoelectronic receivers based on state-of-the-art HBT’s and with unity current gain frequency f T = 160 GHz, will provide bandwidth no more than 15 to 30 GHz. In this situation, it can be concluded that the parasitic RC time constant effect, and the carrier transit time effect [19], in properly designed MSM diodes, are of little importance in so far the amplifier’s bandwidth or bit-rate limitation is concemed.

Emitter

1oooAo , =

AllnAs isolation layer

(PD layer)

S.I. InP

Fig. 3. Schematic cross-sectional details of the proposed HBT structure.

B . The HBT: Structures for Integration

Several research groups have recently demonstrated mm- wave HBT’s with measured unity current gain f,-values of 160 GHz [6], 63 GHz [7], 120 GHz [8], and 90 GHz [9]. These are lattice-matched structures with heavily doped p-type InGaAs base layer (= 500 A), and a low-doped n-type InGaAs collector layer (x 0.3 pm) grown on InP substrates. A critical examination of these device structures led us to propose an improved device structure [ 141 as shown schematically in Fig. 3. The structure utilizes a two-step emitter doping in order to achieve a lower emitter base junction capacitance (5 5 x lop7 F/cm2).

In Fig. 3 the collector and emitter designs are shown as n-type InGaAs and AlInAs layers which are suitable for MBE growth. For MOCVD growth they can be replaced by InP layers. The HBT equivalent network model [20]-[22] with reference to this structure is given in Fig. 4(a). The values of the intrinsic transconductance (gmi) and emitter series resistance (reet) in this structure are 300 mS at a collector current of 7.5 mA and 1.6Q2, respectively. The intrinsic base-charging capacitance (c,i z gmiTec) is 278 fF since the emitter-to-collector transit time, T,, = 0.925 ps. The components of T~~ are defined below.

In our proposed reference device, emitter stripe width of 1 pm and length of 25 pm are assumed in contrast to 3.6 pm ~ 3 . 6 pm as employed in [6]. This linear structure reduces the emitter resitance (ree/) to 1.6 R and the base resistance ( rbb t ) to 5 R in contrast to the values of 6 s2 and 122 R respectively reported for the 3.6 pm x 3.6 pm design. Fig. 4(b) shows the simplified equivalent circuit of the reference device, and in this representation the effect of ree/ is included in the emitter feedback factor (1 + gmireet), in determining the reduced effective values of c,, g,, gm and go from their respective intrinsic values. The ratio gm/gr is a measure of the current gain Po, assumed to be 50. The output conductance of the HBT is usually much smaller than gm/lO, and a value of gm/20 is assumed here. The inner collector capacitance (cc) lies directly under the emitter stripe and is determined by the collector-base depletion width, whereas the exterior base-to- collector capacitance cbc lies directly under the base contact which is quite large in area. For the assumed structure c, = 10

JOHN AND DAS: DESIGN AND PERFORMANCE ANALYSIS OF OPTOELECTRONIC INTEGRATED RECEIVERS

6 E

(a)

g, = 200mS

2 R C

10 mS

E , o E

LE = 25 pm,

(b)

SE = 1wm

Fig. 4. (a) Equivalent network representation of the HBT based on device physics and structural details. (b) Modified equivalent network representation of assumed HBT structures operated at collector current density, .J, = 3 x 10' A/cm2 (1 pm x 25 pm emitter-area).

fF and cbc = 35 fF, and a further reduction of cbc could be achieved by recourse to oxygen or proton ion implantation [23]. For a device with emitter of 1 pm x 2.5 pm the equivalent circuit parameters will be approximately scaled by 1/10 for all the components except for the series resistances which will be magnified by a factor of ten.

The emitter-to-collector transit time T,,,, excluding the ef- fects of collector and emitter series resistances for the HBT can be expressed as follows [20] and [21]:

where wb is the base width, D, is the electron diffusion constant, xdc is the collector depletion width, cj, is the emitter-base junction capacitance and vsat is the high field carrier saturation velocity. The reported low intrinsic base and collector transport time (M 0.5 ps), represented by the last three terms of (2), implies that the diffusion constant and the carrier saturation velocity are quite high in these structures [6], i.e., D, M 70 cm2/V.s and usat x 4.7 x lo7 cm/s. In the assumed device structure since T , ~ M 0.925 ps, the contribution due to Cj,/gmi M 0.425 ps, which implies an approximate emitter-base junction depletion width of M 220 A. Although T~~ represents the overall carrier transit time, the unity current gain frequency, (fr), measured directly on a device includes the effects of the total collector-base junction capacitance Cj, = (cc + cbc), as given in the expression

rq Cp = 24 fF

_____

165

(a) (b)

Fig. 5. Schematic diagram of basic HBT transimpedance amplifier circuits with 50 0 load resistor: (a) The common-emitter configuration. (b) The cascoded-pair configuration.

In (3) rcc' is the collector series resistance. In the assumed 1 pm x 25 pm device structure when operated with I, = 7.5 mA, the contribution due to the parasitic term becomes 0.310 ps, yielding fr = 128.74 GHz, which is much lower than the value corresponding to f r = 1/(2wec); when 7ec = 0.925 ps. For practical reasons and as an alternate design choice, we would like to increase T,, to 1.5 ps, and this would allow us to reduce the base doping by half and increase the base width to 800 W. These changes would improve the device current gain performance with a modest increase in the base resistance. For this altemate device structure fT can be calculated to be 88GHz. However, the equivalent network parameter values as given in Fig. 4(b) will also apply to this altemate device structure except for the value of c, which should be approximately 300 fF instead of 185 fF, and 30 fF instead of 18.5 fF, for the device scaled to one-tenth. We will analyze receivers incorporating both these designs of the HBT's.

111. THE TRANSIMPEDANCE AMPLIFIER

A. Design Concepts

A photodetector directly converts optical input power into electrical current, and for high-speed signal processing pur- poses the detected electrical current needs to be amplified and delivered into a 50 i2 system. A low-pass transimpedance or &-amplifier with a large bandwidth is ideally suited for this application with the photodetector placed in parallel across the input. The HBT structure as described in the previous section, with a transconductance of 200 mS, can be used for this ZT- amplifier with a 50 R collector resistance acting as the load as schematically shown in Fig. 5(a). A feedback resistance, Rf M lo&, in this circuit will ensure a modest current gain (A,f M lo), and an approximate transimpedance of 500 0 (ZT = A ~ ~ R L ) . This amplifier's bandwidth is limited because of its high input capacitance arising due to the collector-base capacitance ( Cjc) and the associated Miller effect. To improve this situation a common-emitter/common-base or the so called cascoded amplifier can be used involving two HBT's with transconductance of 200 mS, as shown schematically in Fig.

166 IEEE TRANSACTIONS ON ELECTRON DEVICES, VOL. 41, NO. 2, FEBRUARY 1994

IP 6

m where

Fig. 6. Schematic circuit diagram of a conceptual cascoded Zr-amplifier with reduced emitter-area HBT’s followed by a two-step emitter follower.

5(b). This arrangement will improve the bandwidth compared to that obtainable in the single HBT common-emitter amplifier.

Both the common-emitter, and the cascoded-pair amplifiers will suffer from a high shot noise current at the input due to their large dc base current, (IB NN 150 PA). In order to satisfy the requirement of a 50 R load termination as well as a low base input shot noise current, an amplifier consisting of a cascoded-pair of HBT’s with reduced emitter-size (x 1/10) followed by a two-step emitter follower, involving a reduced emitter-size (x 1/10) HBT followed by a full-size (1/1) HBT, can be used as shown schematically in Fig. 6. In this ZT- amplifier arrangement, use of R, = 500 R also helps to increase R f to 5 k R, thus increasing the transimpedance by ten folds. The bandwidth of this amplifier can be expected to be slightly degraded, compared to that of the straight cascoded- pair circuit shown in Fig. 5(b), due to the input capacitance of the emitter-follower stage which appears across R, (500 R). It should be mentioned that since the input capacitance of the reduced emitter-size HBT is very low (E 18.5 E), an MSM photodiode with low capacitance, i.e., the seriedparallel structure described in the previous section, can be used with this ZT-amplifier in order to achieve a wide bandwidth.

and

Using the parameter values: R, = 50 R, Rf = 500 R, gm = 200 mS, re, = 0.925 ps, cp = 24 fF, Cj, = 45 fF, Po = 50, PL = gm(rTllRf) = 33.33, one obtains A;fo x 7.69, f 3 - d B = 5.63 GHz, and ZTO = 385 R. These results indicate that the simple common-emitter amplifier provides a bandwidth which is only 3.5% of the HBT’s f,-value.

The Cascoded-Pair Amplifrer: An approximate analysis of the cascoded amplifier stage shown in Fig. 5(b) can be carried out by considering the fact that the common-emitter input stage has an effective collector resistance of (rcc, + k) due to the common-base output stage. Including the loading effect of R f at the input, the current gain without feedback will have three poles involving the time constants:

re, = 0.925 ps rC = RCCjc = 2.25 PS.

Since re, and r, are both much smaller than PLr’, the gain function can be characterized by two prominent poles identified by

and

With feedback, involving R f in parallel with C f that satisfies the condition R f C f = r,, the magnitude of the current gain

B . Analyses of Amplifier Performance The Common-Emitter Amplifier: A transimpedance or ZT-

amplifier provides an output voltage in response to an input

output current into the load resistance (Rc), the following simple relationship holds well:

be written as

(1 1) current. Since the output voltage is due to the flow of the 1

Z, x A;R,. (4) where fo = J f l f 2 ( 1 + K R , / R f ) , and Q o = fo/(fl + f2) . Using the larger emitter-area HBT’s (with gm = 200 mS),

and an MSM photodiode with cp = 24 fF as used in the previous case one obtains

In (4) Ai represents the current gain of the various amplify- ing stages as discussed above. In the amplifying circuits where an emitter-follower is used, the &-value given above needs to be multiplied by the gain of the emitter follower which is approximately unity. Including the loading effect to R, at the input in the common-emitter circuit, shown in Fig. 5(a), the

= 3.00 GHz, Qo = 0.481, f 3 - d ~ = 15.5 GHz,

f 2 = 50 GHz, f o = 25.48 GHz, ZTO = 385 0.

current gain with feedback can be expressed, in the following manner: These results indicate a significant improvement of the

amplifier bandwidt compared to that of the simple common-

( 5 ) emitter amplifier. A low value of Qo(< 0.7) implies a gain response with negligible peaking.

JOHN AND DAS: DESIGN AND PERFORMANCE ANALYSIS OF OPTOELECTRONIC INTEGRATED RECEIVERS 167

Ohms

Fig. 7. A detailed schematic circuit diagram of the ZT-amplifier of Fig. 6 suitable for monolithic integration (where RI = R2 F= 500 0). The four MSM PD’s shown in the figure represents the single series/parallel connected 3-terminal MSM diode (as shown in Fig. 2(b)).

Cascoded Amplifier with Reduced Emitter-Area HBTs: The cascoded amplifier stage shown in Fig. 6 utilizes three HBT’s with one-tenth emitter area (gm = 20 mS). For simplicity of analysis of this amplifier, we will assume that the input capacitance at the base of the first emitter-follower is essentially capacitive (x 4.8 fF) which increases the collector time-constant, T,. For low input-capacitance loading in this case, a small-capacitance (cp M 6 fF) MSM photodiode will be used. Including the loading effect of Rf(5k 0) at the input, the current gain without feedback in this case can be characterized by three poles similar to that given in (9) above. However in this case the time constant T’, and T, assume the following numerical values. T’ = T,, + cjc + - + 2~ = 1.765 ps, and T~ = 0.5kQ(4.5 fF + 4.8 fF) = 4.65 ps.

Since T,, and 7, are both much smaller than PAT’, the current gain Ai can be characterized with two poles with fl = 2.7 GHz and fi = 28.52 GHz (see (10)). With the application of feedback, the current gain becomes identical to what is given in (1 1). Further numerical calculations show that in this case we have

(reel s ~ m ) g m

fo = 25.48 GHz, f 3 - d ~ = 14.5 GHz Qo = 0.585, ZTO = 3.85k Q.

In this amplifier, instead of using the modified low-capaci- tance MSM diode, if standard MSM diode with c f = 24 fF was used the bandwidth would have reduced to 9.5 GHz from 14.5 GHz.

These results clearly indicate that by using reduced emitter- area HBT’s in a cascoded amplifier it is possible to increase ZTO by a factor of ten with nearly the same bandwidth compared to that obtained using the large emitter-area HBT’s. A reduction of the input shot noise current by a factor of ten

make this amplifier even more attractive for its high sensitivity considered later in Section IV.

C. Frequency Response of Integrated-Circuit Amplifiers

Fig. 7 represents a possible integrated circuit implementa- tion of the cascoded amplifier considered in section 3 B.Note that the biasing is achieved by +3.7 V and -1.5 V supply sources with the emitter of the input transistor grounded. The two outer electrodes of the MSM photodiode are connected to +3.7 V supply line and its third terminal is connected to the base of the input HBT. The forward bias emitter-base voltage is assumed to be M 1.1 Volt for the InAlAsflnGaAs HBT’s. It should be pointed out that the base current of input HBT is supplied mainly by the feedback resistor ( R f ) . The MSM photodetector consists of seriedparallel quad arrangement as shown in Fig. 2(b). This integrated structure occupies a rela- tively small active area (10 pm x 14 pm) and can be assumed to be uniformly excited by the optical input.

The resistors are all obtainable from the sub-collector doped n-InAlAs layer with sheet resistance adjusted by controlled depth of etching. The feedback resistance Rf = 5 kQ may be the most challenging element in this circuit to be realized. Since the doped n-AlInAs layer is usually formed above a semi-insulating AlInAs buffer layer grown atop the S.I. InP substrate, after proper surface treatment or growth of surface passivating superlattice structures, it should be possible to achieve a high sheet resistance (M 500 R/square) doped n-InAlAs layer with negligible capacitance coupling to the substrate. Using single-stripe layout for Rf , with length-to- width ratio of 10 and contact pad size not exceeding 10 pm2, it should be possible to minimize its end-to-end parasitic capacitance below 1 fF. Parasitic capacitance existing between

168 IEEE TRANSACTIONS ON ELECTRON DEVICES, VOL. 41, NO. 2, FEBRUARY 1994

TABLE I SUMMARY OF PERFORMANCE FOR DIFFERENT TRANSIMPEDANCE AMPLIFIER CIRCUITS

Amplifier Tec Rf f T CP Cf f3-dB‘ ZTO Configuration (PS) (0) GHz) (fF) (fF) (GHz) (KW (Ma)

A. Common Emitter 0.925 500 117.0 24 0 5.63 0.38 6.71

(Fig. 5(a)) 1.5 500 82.2 24 0 4.94 0.38 6.7 1

B. Cascoded-Pair 0.925 500 100.3 24 4.5 15.5 0.38 6.73

(Fig. 5(b)) 1.5 500 73.6 24 4.5 12.2 0.38 6.74

0.925 5K 90.0 6 0.45 14.5 3.85 2.24

(5.63)

(4.871

(0) (14.8)

(0) (10.5)

C. Cascoded-Pair with (0) (18.3) Emitter Follower 24 0.45 9.5

0 (1 1.8) (Fig. 6) 1.5 5K 68.0 6 0.45 11.0 3.85 2.24

D. Integrated Circuit 0.925 5K 90.0 6 0 17.1 3.63 2.29 version of (C) 1 16.2

(Fig. 7) 1.5 5K 68.0 6 0 14.4 3.63 2.33

IK 53.2 6 13 27.4 0.82 5.13 PSPICE Data

+Data given in parentheses for A, B, and C are determined by PSPICE simulation. ‘Values given correspond to the respective maximum bit-rate B = f 3 - d e

(0) (14.8)

0.925 1K 71.0 6 9 30.1 0.82 5.0

1 13.6 1.5

the end pads and the grounded substrate will have negligible impact on Cf. The circuit in Fig. 7 will usually perform in an optimum fashion with Cf = 0 or up to 1 fF, as will be seen later in this section. In the circuit layout shown in Fig. 7, a full-size HBT is used as a “diode” in series with the collector of the output transistor as a means to protect the latter from possible collector-emitter breakdown voltage (VCEO) which could be quite low. For the purpose of bias level shifting a larger emitter area HBT configured as a diode can be used between the emitter of the first emitter-follower (1/10) and the base of the second emitter-follower. This would provide a low impedance (highly capacitive) coupling between the first emitter follower and the output emitter follower stage at high frequencies, thus improving the available bandwidth by as much as 10%. However, for simplicity we have used a device with one-tenth emitter area, since this is not a critical requirement for the operation of the circuit.

The curves showing the dependence of the bandwidth (f3-dB). and transimpedance magnitude ( 2 ~ 0 ) on the feed- back resistance Rf are presented in Fig. 8. The bandwidth data were calculated using PSPICE simulation of the schematic circuit diagram given in Fig. 6. For the PSPICE simulation the full-size and the one-tenth size HBT’s were presented by their respective scaled equivalent networks. In the PSPICE simulation trial-and-error method was used to find the opti- mum values of Cf, and they were found to be rather too low (fractional fF) and therefore, responses with Cf = 0 were considered close to the optimum responses. This finding contrasts with the requirement, Cf = T,Rf, suggested in obtaining (11) for the analytical 2-pole approximation of a 3-pole transfer function. Use of a 2-pole function although is satisfactory for the calculation of the magnitude of the gain, it seriously underestimate the phase changes with frequency,

75, 1-45

$5 1 1.5 2 2.5 3 3.5 4 4.5 5 110

Rf (K-Ohms)

Fig. 8. Dependence of the 3-dB bandwidth and the transimpedance mag- nitude on the feedback resistance for the cascoded amplifier shown in Fig. 6.

and consequently use of Cf satisfying C f = r,Rf invariably results in an overcompensation in actual amplifiers. The results presented in Fig. 8 and Table I (Row C) clearly indicate that there is disagreement between the f3-dB values obtained analytically and by the PSPICE simulation. Higher f 3 4 B

values were predicated by the PSPICE simulation compared to that obtained by analytical means, for the circuit given in Fig. 6. However lower f 3 - d ~ values were predicted by the PSPICE compared to that obtained by analytical means for circuits given in Figs. 5(a) and 5(b) (see Table I, rows A and B). By separately modeling the input admittance of the two-step emitter-follower in Fig. 6, we found that there is a negative conductance which assumes a peak of (-1.4 mS) between 10 and 30 GHz. This obviously increases the effective collector load resistance and thus helps maintain a high transimpedance gain, Ai, R,, at higher frequencies, hence the higher bandwidth.

JOHN AND DAS: DESIGN AND PERFORMANCE ANALYSIS OF OPTOELECTRONIC INTEGRATED RECEIVERS

g 75- c 2 70-

65- W U z 60.. d

a 2 50- I/: z 45- d

g 55-

I69

Rf = 5K

R f = l K

The complete transimpedance amplifier frequency responses based on the IC implementation with Rf-value of 5 kR and 1 kR with different Cf-values are presented in Figs. 9(a) and (b) for HBT's with T~~ = 0.925 ps and 1.5 ps, respectively. It should be noted that for Rf = 1 kR there is substantial gain peaking near the 3 - dB cutoff frequency, whereas for Rf = 5 kR the response shows negligible gain peaking. The shape of the response curves in Figs. 9(a) and (b) with Rf = 5 kR with Gf = 0 or 1 fF are not greatly dissimilar and the bandwidth reduction when Cf = 1 fF is less than 3%. In the integrated circuit version we do not expect more than M 1 fF parasitic shunt capacitance (via the SI substrate) between the two ends of the 5 kR resistor if it can be arranged as a rectangular element with length-two-width ratio of 10 and up to 10 pn2 area of the end pads. There would be no need to physically adjust the capacitance for optimization of the receiver response, since the only capacitance that will appear across Rf (= 5 kR) is the parasitic shunt capacitance. The response curve with Rf = lkR show a high gain peaking, M 25 dB in Fig. 9(a), and M 20 dB in Fig. 9(b) with Cf = 0 fF, and they become considerably reduced (within 3 dB) with Cf = 9 fF in Fig. 9(a), and C, = 13 fF in Fig. 9(b). The latter capacitance values could be realized in the form of an interdigitated capacitor on the S.I. substrate. The results of detailed bandwidth and transimpedance calculations based on PSPICE simulation are presented in Fig. 10 for the same amplifier, assuming HBT's with T,, = 0.925 ps and 1.5 ps. Finally, the results, based on analytical and PSPICE simulations, obtained for various &--amplifier are summarized in Table I for ready reference purposes. In particular, we note that how the bandwidth is improved with the use of modified MSM diode (c, = 6 f'F) compared to that obtained with the standard MSM diode (c, = 24 fF) in the amplifier configuration C.

Iv. RECEIVER SENSITIVITY AND NOISE

A. The Equivalent Input Noise Current The input transistor of the &amplifier, shown in Fig. 7,

has its base current ( I B ) related mean-squared shot noise current, i.e., 2 q I ~ a f , where Af is a small bandwidth. There is also the collector mean-squared shot noise current (Zy1,A.f) uncorrelated [24], [25], [21] to the base shot noise current. The latter can be transferred to the input base terminal using the overall small-signal current gain of the input transistor, i.e., I , q m / ~ C ~ ( 2 , where CT represents the total capacitance that appears at the input terminal. Besides these two shot noise sources we must also add the thermal noise associated with the feedback resistance ( R f ) , and the llf noise of the input transistor. The thermal noise associated with the base resistance (rbt,!) and the emitter resistance ( rppj ) must also be transformed into an equivalent noise current by recognizing the capacitance elements (X chr + c p ) that exist between the extemal base and emitter terminals. Thus the sum of the integrated mean-squared input noise current for a given bit-rate ( B ) , involving Personick's integrals [26] can be expressed as

h

E 80 c

$ 75

2

z g 55

5! W 70

d 65

8 60

Q 50

U" I

1 100 10 FREQUENCY (GHz)

!z + 40

35 0 1

(b)

Fig. 9. Frequency response of the ZT-amplifier shown in Fig. 7: (a) For the amplifier based on HBT's with T~~ = 0.925 ps. (b) For the amplifier based on HBT's with T~~ =1.5 ps.

Cl0 +5 1 1.5 2 2.5 3 3.5 4 4.5 5

Rf (K-Ohms)

Fig. 10. Dependance of the 3-dB bandwidth and the transimpedance mag- nitude on the feedback resistance for the receiver amplifier shown in Fig.

IEEE TRANSACTIONS ON ELECTRON DEVICES, VOL. 41, NO. 2, FEBRUARY 1994 170

In (12) I2,13, and If are weighting functions which are dependent only on the input optical pulse shape, and the equalized output pulse shape. For nonretum-to-zero (NRZ) coding format I 2 = 0.562,13 = 0.0868 and If = 0.184 1271. The remaining parameters in (12) are defined as follows:

fc = comer frequency where the base shot-noise current and l/f noise current are identical in magnitude, fo = lower end frequency of the l/f noise spectra assumed lop2 Hz, f+ = gm/%CT, where CT is the total input capacitance at transistor base including the photodiode capacitance, and fex = 1/27r(Tbb’ + T e e / ) C e x , where C,, is the extrinsic capacitance appearing between external base and the grounded emitter, i.e., (ep + Cbc), and any stray or fringe capacitance.

B . The Minimum Detectable Optical Power The responsivity of a photodiode, as given in ( l ) , is de-

termined by its quantum efficiency, v, which usually varies from detector to detector. Therefore, in characterizing an optoelectronic receiver it is appropriate to use the detected optical power, vPOpt, as the reference input explicitly given by the detected photocurrent ( I p ) namely,

1.24 x vPOpt = Ip- Watts (13)

where Ip is in Amperes and X is in micrometer. For an RF-modulated optical signal with modulation index, m, the detected rms photocurrent (ip) can be expressed as

If the number of photo-generated electrons contained in one signal bit is large (> lo4), Gaussian statistics can be used to describe the detection probabilities in receivers involving direct detection by MSM diodes. By choosing the decision threshold to yield equal probability of error in two states (i.e., 0 and l), the probability of making an error or the so called bit error rate (BER) can be expressed as BER

found that Q = 6. For all practical purposes Q is equivalent to the square root of the signal-to-noise ratio, a. The signal-to-noise ratio of the receiver can be expressed as

+3 - - -2- exp(-Q2/2). For the standard BER of lo-’ it can be

h E -20

Y 3

s a

w 4 m - 3 0

uw E: :s -40

ZE $ 0

W L

-50 .l 1 1 0 100

BIT RATE (GWs)

Fig. 1 1 . The minimum detectable optical power or the sensitivity (qFopt) of the receiver amplifier shown in Fig. 7, versus the bit-rate (B) for a BER of

o r m = 6. Sensitivity due to various individual noise sources, and that due to the quantum limits are shown for comparison.

power as given in [29]. Thus according to the above definition the sensitivity will be lower by 1.5 dB, compared to the case where the factor fi is ignored.

The receiver sensitivity, qpopt(min), with reference to 1 mW optical power is graphically presented in Fig. 11 for the standard BER of lo-’, for the cascoded ZT-amplifier shown in Fig. 7. The contributions of individual noise sources are also delineated in this graphical representation. It becomes clear from these data that the contributions from the l/f noise and the series resistance (Tbb/ + re,/) and parasitic capacitance C,, are quite negligible. The contribution of the collector shot noise assumes importance only at high bit-rates which is above the band-width or bit-rate capacity of the amplifier as indicated. It should be pointed out that the contribution of the l/f noise to the HBT-based and FET-based 1271 2,- amplifiers is proportional to h ( B ) and B2, respectively. The reason behind this is that in HBT’s the l/f noise current generator physically appears across the input emitter-base junction, whereas that for FET’s appears across the output, i.e., between the drain and source electrodes.

Concerning the optimization of the collector bias current I, of input HBT 1271, it should be noted that an optimum 1, becomes effective only when contributions from the collector shot noise current and the base shot noise current become comparable at high bit rates. In the cascoded amplifier design considered in Fig. 7 this does not occur within the 3-dB bandwidth of the amplifier as indicated by the vertical dotted line in Fig. 11. In other words, the receiver’s sensitivity is determined, to a large measure, by the base shot noise current source and all other noise sources play a minor role. The diagram also shows the limit due to quantum induced shot noise whose impact on the amplifier sensitivity is also seen to be insignificant. This noise is associated with the detected photocurrent, 2qIpB12, which when substituted for if in (16), yields the quantum limit of optical power, required for a given

where s;n is the mean-squared noise current at the input of the amplifier as given in (12). Using (12)-( 15) and assuming a modulation index of 1, one obtains the following expression for the minimum detectable average optical power qpop,(min) or the receiver sensitivity for a given SIN ratio.

vP,pt(min) = (16)

It is to be noted that the factor fi in the above equation is not included in the definition of minimum detectable optical

JOHN AND DAS: DESIGN AND PERFORMANCE ANALYSIS OF OITOELECTRONIC INTEGRATED RECEIVERS 171

(S IN) ratio, namely,

1.24 5’ A N

- vPopt(quantum) = - ( - )4 ,1~2~ .

For ready reference, all the key results obtained in this paper are summarized in Table I, where in particular, one could see the relative performance of various receiver amplifiers

by examining the values of ZTO, f 3 - d B , and dz at B = f3-dB.

V. CONCLUSION A design rational is presented in this paper for the realization

of high-speed optoelectronic IC receiver amplifiers based on MSM photodiodes and HBT’s on InP substrates. It is shown that the cascoded amplifier configuration is capable of providing a high transimpedance, and a moderately high bandwidth. However, the latter is less than 20% of the intrinsic unity current gain frequency of HBT’s used. In this amplifier the optical sensitivity requirement dictates that small emitter- area HBT’s be used for the input cascoded-pair circuit in order to reduce the base current and the related shot noise current appearing at the input. The emitter-area suggested is only 1 pm x 2 .5 ,um for the input amplifier stage, and it may not be an easy task to reduce this emitter area any further. Consequently, any significant improvement of the amplifier performance than that has been reported in this work may be difficult. By using a novel concept of low-capacitance series/parallel connected MSM photodiode, its impact on the receiver bandwidth has been significantly reduced. At the highest transimpedance the projected sensitivity of the re- ceiver, i.e., -26 . 8dBm at 17 Gb/s, represents an excellent performance figure. However, it could be a challenging task to realize this by practical integration of the proposed IC receiver amplifier. The results presented in Fig. 1 1 , clearly delineate the effects of contributions from each individual noise sources for a better appreciation of the receiver’s limitations. It is to be pointed out that the data shown beyond f 3 - d B (vertical- dotted line) have no practical significance because the weight functions I2 and I3 were defined assuming a flat frequency response of the receiver transfer function.

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[I ] W. P. Hong, G. K. Chang, R. Bhat, C. Nguyen, and M. Koza, “Monolithically integrated waveguide-MSM-detector-HEMT amplifier receiver for long wavelength lightwave systems,” IEEE Photon. Techno/. Lett. vol. 2, pp. 156158, 1991.

[2] S. Chandrasekhar, B. C. Johnson, M. Bonnemason, E. Tokumitsu, A. H. Gnauck, A. G . Dentai, C. H. Joyner, J. S. Perino, G. J. Qua, and E. M. Monberg, “An InP/InGaAs pin/HBT monolithic transimpedance photoreceiver,” IEEE Photon. Technol. Lett., vol. 2, pp. SOS-506, 1990.

13) G. K. Chang, W. P. Hong, J. L. Gimlett, R. Bhat, C. K. Nguyen. G. Sasaki, and J. C. Young, “A 3 GHz transimpedance OEIC receiver for 1.3-1.55 pm fiber-optic Systems,” IEEE Photon. Tec.hno1. Lett., vol. 2, pp. 197-199, 1990.

[4] Y. Akahori, Y. Akatsu, A. Kohzen, and J . Yoshida, “IO-Gb/s high- speed monolithically integrated photoreceiver using InGaAs p-i-n PD and planar doped In-AIAshGaAs HEMT’s,” IEEE Phofot7. Techno/. Lett., vol. 4,. pp. 754-756, 1992.

[SI W. S. Lee, D. A. H. Spear, M. J . Agnew, P. J. G. Dawe, and S . W. Bland, “ I .2 Gbitsh fully integrated transimpedance OEIC receiver for 1.3-1 .SS Itm transmission system,” Elec.fron. Tec,ht7o/. Lett., vol. 26, 377-379, 1990.

[6] Y. K. Chen, R. N. Nottenburg, M. B. Panish, R. A. H a ” , and D. A. Humphrey, “Sub-picosecond InPnnGaAs heterojunction Bipolar transistors,” IEEE Electron Dev. Lett., vol. 10, pp. 267-269, 1989.

[7] B. Jalali, R. N. Nottenburg, Y-K. Chen, D. Sivico, D. A. Humphrey, and A. Y. Cho, “High-frequency sub-micrometer AIInAs/InGaAs het- erostructure bipolar transistors,” IEEE Electron Device Lett., vol. 10, pp. 391-392, 1989.

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[9] H. Fukano, H. Nakajima, T. Ishibashi, Y. Takanashi, and M. Fujimoto, “Effect of hotelectron injection on high-frequency characteristics of abrupt In0 5 2 (Ga~-,Al,)o.~~/InGaAs HBT’s,” IEEE Trans. Electron Dev. , vol. 39, pp. 5 G 5 0 6 , 1992.

[ lo] R. N. Nottenburg, Y. K. Chen, M. B. Panish, D. A. Humphery, and R. H a ” , “Hot-electron InGaAsnnP heterostructure bipolar transistors with f~ of 110 GHz,” IEEE Electron Device Lett., vol. 10, no. I , pp. 3CL32, 1989.

[11] P. C. Chao, A. J. Tessmer, K-H. G. Duh, P. Ho, M-Y. Kao, P. M. Smith, J. M. Ballingall, S-M. J. Liu, and A. A. Jabra, “W-Band low- noise InAl As/InGaAs lattice-matched HEMT’s,” IEEE Electron Device Lett., vol. 11, pp. 59-61, 1990.

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[I61 0. Wada, H. Nobuhara, H. Hamaguchi, T. Mikawa, A. Tackeuchi, and T. Fujii, “Very high speed GaInAs metal-semiconductor-metal photodiode incorporating an AIInAs/GaInAs graded superlattice,” Appl. Phys. Letr.,vol. 54, no. 1 , pp. 1 6 1 7 , Jan. 1989.

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[20] R. L. Pritchard, Electrical Characteristics of Transistors. New York: McGraw-Hill, pp. 298400, 1967.

[21] M. B. Das, “High-Frequency Performance Limitations of Millimeter- Wave Heterojunction Bipolar Transistors,” IEEE Trans. Electron Dev. , vol. 35, pp. 604-614, 1990.

[22] M. B. Das, “HBT device physics and models,” Ch. 4, in HEMTs and HBTs: Devices. Fabrication, and Circuits, F. Ali and A. Gupta, Eds. New York: Artech House, pp. 191-251, 1991.

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172

Eugene John received the Bachelor's degree in electronics and telecommunications engineering from the University of Kerala, Trivandmm, India, and the Master's degree in electrical engineering from the University of Texas at El Paso in 1984 and 1990, respectively.

From 1984 to 1998 he worked for KELTRON, India, as Development Engineer. He is currently a Ph.D. candidate in the Department of Electrical Engineering at The Pennsylvania State University.

He is a member of Eta Kappa Nu, Tau Beta Pi, and Phi Kappa Phi.

IEEE TRANSACTIONS ON ELECTRON DEVICES, VOL. 41, NO. 2, FEBRUARY 1994

Mukunda B. Das (S'75-M'62-SM'70) was born in Bagerhat, Bangladesh, on September I , 1931. He received the B.Sc. (hons) and the M.Sc. degrees in Applied Physics from Dhaka University, Dhaka, Bangladesh, in 1953 and 1955, respectively. He received the Diploma of the Imperial College of Science and Technology (DIC), London, UK, and the Ph.D. degree from the University of London, both in 1960.

During 1960-1962, he was a Lecturer of Elec- trical Engineering at the Imperial College. From

1962 to 1964, he served as a Senior Scientitic Officer at the East Regional Laboratories of the council of Scientific and Industrial Research, Dhaka. From 1965 to 1968. he was a Principal Scientific Staff Member and Leader of the MOS Integrated Circuits Group with ASM. Ltd., GEC Hirst Research Center, in Wembley, Middlesex, UK. In 1968, he joined the Pennsylvania Stale University, University Park, PA, as an Associate Professor of Electrical Engineering. He is now Associate Director of Electronic Materials Processing and Research Laboratory at Penn State. He has authored or coauthored more than 90 publications in refereed journals, and hold eight US and UK patents concerning the field-effect device design and fabrication. His current research interests include HF limitations of devices. millimeter-wave device design and characterization, and their applications in optoelectronic circuits.

Dr. Das received the 1967 Blumlein-Browne-Willan Premium Award for his published work in the Proc. IEE, London, UK. He is a member of Sigma Xi.