Design and performance of microwave oscillatorsin integrated fin-line technique

Reinhard Knochel

Indexing terms: Microwave integrated circuits. Microwave oscillators

Abstract: Guidelines are outlined for the design of oscillators for various applications, which are realised inintegrated fin-line technique. The characteristic features arc derived and compared to one another. The resultsare then applied to cavity-stabilised oscillators, injection-locked amplifiers, and varactor-tuned oscillators,which have all been realised in fin-line technique. Stabilisation by an integrated transmission cavity is shownto be the best configuration for fin-line. For injection-locked amplifiers the stable locking range has been con-siderably improved by reactance compensation. Finally, it is shown how a varactor diode should be integratedin order to achieve wideband electronic tunability. Performance data of realised (Gunn or Impatt) oscillatorsare given for each case of application.

1 Introduction

Since there is increasing activity in the millimetre-waveregion directed toward the developemnt of low-cost mass-production components such as communication systems,collision-avoidance radar, and environmental-monitoringequipment, several transmission media allowing simple batchproduction have been proposed. The conventional microstripproved to be disadvantageous beyond 30 GHz owing to radi-ation losses spurious coupling and tolerance problems, whichdegrade the performance of microstrip components withincreasing frequency. Hence different planar structures anddielectric waveguides were considered. Among these, inte-grated fin-line seems to be the most promising.1"5 It consistsof a planar dielectric substrate carrying a metallisation,mounted symmetrically in the /f-plane of a standard rec-tangular waveguide. Thus, integrated fin line combines thebenefits of rectangular waveguides and planar circuits. Itprovides low loss and compatibility with solid-state com-ponents, even at high frequencies.

Several papers have recently been published which dealwith theoretical investigations of the fin line.6"13 Theirmain results are explicit expressions for wave impedance,cutoff frequency, wavelength, and effective dielectric con-stant of the fundamental mode, and a simple design theoryfor some types of filters. The state-of-the-art of the fin linetheory is, however, far from allowing computer-aideddesign of more complex circuits. Experimental investi-gations deal with a description of various filter circuits,p-i-n-diode attenuators, single-ended and balanced mixers,and oscillators.14"18 Owing to the lack of an effecientdesign theory these components have, so far, been devel-oped by trial and error. Moreover, fin line oscillators take aspecial place among them: their development appears to beonly in a preliminary state.

It is the aim of this paper to give a thorough investigationof the design and performance of oscillator circuits, whichare compatible with fin line. Special attention is paid to thediode coupling structure, which must be shaped in a pre-scribed manner to obtain high efficiency, and the loaded(2-factor of the circuit. From these investigations guidelineswill be outlined for the design of three classes of oscillatorcircuits: cavity-stabilised oscillators, injection-locked ampli-fiers (i.l.o.), and voltage-tuned oscillators (v.t.o.).

Paper T344 M, received 9th January 1979Mr. Knochel is with the Institut fur Hochfrequenztechnik derTechnischen Universitat, Postfach 3329, D-3300 Braunschweig,West Germany

MICROWA VES, OPTICS AND ACOUSTICS, MA Y 1979, Vol. 3, No. 3

2 Diode-coupling structure

At present, the only solid-state elements providing negativeresistance at mm-wave frequencies are the Gunn and impattdiode. These active devices cannot be coupled directly tothe fin line slot, because the lowest impedance level obtainedwith this kind of waveguide is about 100 H.6 The diodesmust, however, be operated with a load of the order of1012 or less, if maximum power output is to be delivered.For high-power diodes in particular, the impedance levelhas to be low. For that reason, the diode coupling structureof Fig. 1, which has been proposed in References 16 and 17,will not be adequate in many cases, because it suffers fromtoo high an impedance level. Furthermore, there is no seconddegree of freedom in order to satisfy the complex oscillationequation. Hence, a special coupling network is needed tomatch the diode for the desired output power with lowcircuit losses. Four possible structures are shown in Fig. 2,where only the side views of the circuits are outlined. Agood heat-sink for the diode is achieved by mounting it in

metal fin sliding short circuit

diode

Fig. 1 Simple fin-line oscillator

a

1-D

= 5 3

"I f-'-f

Fig. 2 Various diode mounts

a Transmission-cavity mountb Conductor-strip mountc Antenna mountd Another transmission-cavity mount

115

0308-6976/79/030115 + 06 $01-50/0

the ground- or end-plate of the housing waveguide. Eachdiode mount consists of a resonant circuit and an impedancetransforming element.

In the mount of Fig. 2a, which has been proposed inReference 16, the diode is fixed in the end plate of thewaveguide. At a distance /, about half a wavelength fromthe diode, the fin line slot is bridged by a discontinuity.The bridging element acts as a transformer, and the inter-mediate line to the diode forms the resonant loop.

The conductor-strip coupling of Fig. 2b, and the antennacoupling (Fig. 2c) show essentially the same features: Aresonant cavity of length / to the right of the diode, whichis mounted in the ground plate of the waveguide. Impedancetransformation is accomplished through a conductor stripor antenna, respectively.

Fig. 2d depicts a structure where the active device iscoupled by a metallic strip to one end of a resonator oflength /, whereas the output waveguide is coupled to theother end. The short-circuited end plate of the housingwaveguide to the right of the diode is tuned so that itpresents a high impedance, as seen from the terminals ofthe active device. This mount and that of Fig. 2a have incommon that the output power is extracted through theresonant circuit. They can be described by the equivalentcircuit of Fig. 3a. The mounts of Figs. 2b and c may bemodelled by the equivalent circuit of Fig. 3b. In theseequivalent circuits RL denotes the load resistance and — RD

the resistance of the active device at maximum generatedpower. The various nK are the transmission ratios of thecoupling transformers. The resonant cavity is described byits resonant frequency co0, unloaded <2-factor (2o andinternal loss resistance R.

The circuit of Fig. 3a is, of course, more versatile thanthat of Fig. 3b. It allows optimisation of either the circuitefficiency or the loaded Q-factor by adjusting n2, while theactive element is matched to the load by means of nx. Theconfiguration of Fig. 3b, on the other hand, exhibits highefficiency as long as the resonant circuit is of high unloadedQ and has small R. It has, however, one degree of freedomless than the other circuit. This difference in principlebetween the possible structures of Fig. 2 is completed by apractical one: the circuit of Fig. 2a has indeed twoimpedance transformers nx and n2, it is, however, notobvious how to vary the coupling n i , because the diode isdirectly connected to the metal fins. Hence this transformerdoes not provide a real degree of freedom, and a poorcircuit efficiency may result. This could lead to the con-clusion that the circuit of Fig. 2d is the best realisation of a

n,:l

-R

Fig. 3 Equivalent circuits

a Of transmission cavity mountsb Of conductor stirp and antenna mount

fin line oscillator. There are, however, also advantages ofthe structures of Figs. 2b and 2c. Both circuits can easily betuned mechanically by varying the position of the short-circuit plunger.

metallisation Teflon foil choke bias

serrations

heat sinkprinted board

Fig. 4 Oscillator cross-section

3 Experimental performance of the diode mountsThe experimental investigations have been conducted at15 GHz to enable network analyser measurements, butscaling the circuits to higher frequencies is straightforward.We have used commercially available packaged activedevices. Fig. 4 shows the cross-section of the basic oscillatorstructure. The diode is placed in a copper heat sink, andthreaded to the housing of the planar printed circuit, whichis split in two parts along the iT-plane of a milled-in standardrectangular waveguide. The dielectric substrate of the printedboard is cut out where the hat of the diode enters. Theelectrical connection between the metallisation of the planarcircuit and the diode is provided by a thin gold-platedribbon of flexible metal adjacent to the hat of the device,or simply by soldering the hat to the metallisation of thecircuit.

Biasing the diode is accomplished through the metallis-ation of the printed board. This is insulated from the metalhousing by a thin Teflon foil. Alternatively, stripline filtersmay be used at a place, where they do not influence thecoupling of the active device.

To protect the circuit against radiation loss through thelongitudinally split housing, the choke construction ofReference 1 has been used. To this end the walls of themilled-in waveguide are chosen to be a quarter wavelengththick in the dielectric medium. In addition, the choke isserrated in short segments according to Reference 19,which prevents the flow of currents in a longitudinal direc-tion. Otherwise, resonances are supported which lead toincreased circuit power losses of 1 dB and more. This loss iseven more pronounced in waveguide sections with a standingwave.

The experimental investigation includes the couplingstructures of Figs. 2b—d. The resonant cavity of eachconfiguration is implemented such that the full height ofthe rectangular waveguide is used and no metal fin projectsinto the inside in this section. It has been found that thisgives the highest unloaded Q of the cavity and thus the bestcircuit efficiency of the oscillator. A short-circuit plungerhas sometimes been used as an end plate of the housingwaveguide, but this short circuit may be integrated on theprinted board in the final construction. The integrated

116 MICROWA VES, OPTICS AND ACOUSTICS, MA Y 1979, Vol. 3, No. 3

short-circuit is formed by defining a metal pattern on theprinted board, which divides the terminating section ofwaveguide in two; both sections are below cutoff and arethus inductive and the actual short-circuit plane lies some-what behind the mechanical one.

Employing a Gunn diode as an active device, all themounts show almost the same electrical features in that therated power can be obtained from the oscillator by properlychoosing the physical dimensions of the coupling structures.The tuning characteristic of an oscillator with the mount ofFig. 2b is shown in Fig. 5 as an example. The curves arevalid for an integrated short circuit which has subsequentlybeen moved away from the diode mount. The oscillationfrequency can be tuned over a relative bandwidth of 15%,with an output power variation of less than 1 dB over 10%bandwidth. The external Q of this circuit has been deter-mined by a locking experiment to be about Qex — 150. Thetuning performance of such oscillators is comparable withthe standard post-coupled waveguide Gunn oscillators.Allowing a metal fin to penetrate into the resonant cavityexpands the tuning range of this oscillator. At the sametime, however, efficiency and frequency stability are alsoreduced, due to a degradation of the loaded Q.

The antenna mount (Fig. 2c) leads to a somewhat morenarrowband performance. In order to bias the diode, a thinwire was used to connect the diode coupling structure andthe bias filter. The wire, however, turned out to influencethe impedance level at the active device, and had to beproperly positioned. Hence, the antenna coupling structureis more suitable for unbiased semiconductor devices. Themount of Fig. 2d requires a more complicated tuning pro-cedure, although it allows a greater degree of freedom inchoosing low impedance levels.

The mounts of Figs. 2b and 2d have also been operatedwith an impatt diode. It is well known that a crucial pointfor this kind of oscillator is the build up of bias oscillations,especially if there is shunt capacitance at the bias port ofthe diode. In the case of fin line oscillators, this shuntcapacitance, which is due to the input capacitance of ther.f. choke, is higher than for the coaxial chokes which arenormally used. For that reason, special care had to be takento damp out any bias oscillations. To this end standardtechniques have been applied.20

4 Integrated-cavity stabilized oscillators

In many cases, a better frequency stability is required thanmay be obtained from single tuned oscillators. To this endthe oscillator structures just descibed may be coupled tohigh-(> cavities, which are integrated on the printed boardtogether with the diode mount. The structures realised areoutlined in Fig. 6. We have to distinguish between threedifferent types of stabilisation,21 which employ a reaction(a), reflection (b), or transmission (c) cavity.

The reaction cavity is formed by a resonant slot acting asa bandstop filter (Fig. la). Natural frequency, couplingcoefficient and unloaded 0-factor depend on the geometri-

damping resistor

damping resistor

—"f^ll- ' ^ W2 I

Fig. 6 Various stabilised oscillators

a With reaction cavityb With reflection cavityc With transmission cavity

~TTr li

16 0

15 0 -

£ 14-0

1302 4 6 8 10relative short circuit plane, mm

14

Fig. 5 Performance of the conductor strip mount

MICROWA VES, OPTICS AND ACOUSTICS, MA Y1979, Vol. 3, No. 3

044

043

0 42

0-41

0762

.0-785

003 007 011 015

Fig. 7 Reaction cavity

a Circuit patternb Physical dimensions

a — broad dimension of housing waveguide•̂o = free space wavelength

117

cal dimensions. The unloaded Q increases with increasingslot height h. The resonant frequency is essentially given bythe slot length /. It depends weakly on h in a manner whichis depicted in Fig. 7b. These results have been foundexperimentally. The coupling coefficient of the reactioncavity turns out to be rather tight in practice, because itdepends mainly on the slot height h but only weakly on itsdistance c from the fin line slot. The parameter h cannot,however, be reduced sufficiently to decrease the coupling,because this would also appreciably decrease the unloadedQ. For a constant fin line slot height s/\0 = 0-05 andconstant c/X0 = 0-1, the coupling coefficient (which is theresistance at resonance, normalised to the wave impedanceof the line) dropped by 300% when h/X0 was changed from0-15 to 0-04. At the same time the unloaded Q decreasedfrom Qo = 1500 to about 700. When h/X0 was held con-stant and c/\0 changed from 0-1 to 0-05, this increased thecoupling coefficient by 25%, and affected the unloaded Qonly weakly.

The reflection cavity (Fig. 6b) is mounted in shunt withthe diode port, whereas the transmission cavity (Fig. 6c) isconnected in series with the load. Both cavities are essen-tially formed by the same planar structure (resonant slot oflength /) which is coupled by inductive strips to the outputwaveguides. The coupling coefficients and natural fre-quencies have again been determined experimentally.

As is well known, the intermediate line between a reflec-tion or a transmission cavity and the diode mount mustcontain a damping resistor in order to ensure a single-modeoperation, when the natural frequency of the cavity is varied.As a consequence, an additional reflection cavity means noimprovement of the electrical performance of a fin lineoscillator, as can be seen by comparing the circuits ofFigs. 2b and 6b. Hence, an oscillator with an integrated finline cavity of the reflection type is of no practicalimportance.

Furthermore, it has to be noted that stabilisation of anoscillator by an integrated cavity is advantageous only forthe diode mounts of Figs. 2b and 2c. In these cases theadditional cavity provides a means for an independent choiceof the loaded ^-factor of the oscillator assembly. Thisadditional degree of freedom is, however, an inherent featureof the diode mount of Fig. 2d. It is hence not necessary touse cavity stabilisation in connection with the latter mount.

Integrated-cavity stabilised oscillators with Gunnelements have hence been constructed only with the diodemount of Fig. 2b. The load impedance as seen from thediode has been measured with a network analyser. It isshown for a reaction-cavity-stabilised oscillator in Fig. 8(solid line). The loop in the locus curve denotes the reson-ance effect of the stabilising cavity. Single-mode operationcan be seen to be achieved, because the impedance locuscurve of the active device (approximately sketched by thebroken line) intersects the load impedance locus curve onlyonce. The impedance locus curve of the transmission-cavity-stabilised oscillator looks approximately the same, but canbe shaped with one more degree of freedom, because inputand output coupling are sperately adjustable. Both reaction-and transmission-cavity-stabilised oscillators could beimproved in f.m. noise performance by about 20 dB with asacrifice of 6dB in power. An absolute value of f.m. noise,which can be achieved, is 0-1 Hz/\/Hz effective frequencydeviation at a modulation frequency of 100 kHz. Compar-able results for oscillators which are stabilised by a high-er TE01n-mode circular waveguide cavity are 30 dBreduction in f.m. noise at the expense of typically 1 to 2 dB

118

in power. This is of course due to the high unloaded Q ofthese cavities, which are more than two to three times anorder of magnitude larger.

As has already been explained, the coupling of the reac-tion cavity cannot be made much looser, in order to increasethe power output. Hence this kind of stabilisation is appro-priate only if a comparatively extreme noise reduction isrequired. The transmission cavity, however, can be adoptedwith much more flexibility. In addition, it exhibits thefeature of noise reduction at high modulation frequencies22

contrary to the reaction cavity. Therefore, it should be usedexclusively for integrated-cavity stabilisation of oscillators.

F ig. 8 Impedance locus curve of reation-cavity-stabilised oscillator

5 Broadband locking circuit

Since we can get high frequency stability only at the sacri-fice of a significant portion of the output power, there isneed for an amplifying module. The basic oscillator structurein the injection-locked mode may serve for this purpose.But, in general, the locking range of the circuit will not belarge enough, if the same module is to be used in differentfrequency channels. It is hence desirable to increase thelocking range of the oscillator. This can be achieved byusing reactance-compensation techniques, as has beenproposed in Reference 23. The method is to lower the(^-factor of the passive circuitry by adding another (thecomplementary) resonance circuit. Equivalent circuit andfin line realisations are shown in Figs. 9a and 9b, respectively.If the frequency of the locking signal deviates from thefree-running frequency, one resonant loop stores capacitiveand the other inductive energy, which can nearly cancel oneanother provided that the coupling of the circuits is correct.Analysing this compensation circuit, one finds that there isan optimum coupling with respect to a maximum lockingrange, which depends on the magnitude of the injectedpower or on the locking gain.24 The compensation loop canbe designed in order to obtain a linear phase relationshipbetween the input and the output signal (constant timedelay), or constant output amplitude (good a.m. com-pression), or a compromise between these features as well.

MICROWA VES, OPTICS AND A COUSTICS, MAY 19 79, Vol. 3, No. 3

Such a broadband locking circuit has been implementedas depicted in Fig. 9b. The diode-structure of Fig. 2b hasbeen used. The compensating resonator is provided by acapacitive stub in parallel with the output waveguide. It istuned together with the length of the intermediate linebetween the stub and the diode coupling network. Thecoupling between the various loops has been adjusted bymonitoring the load impedance as seen from the diode portwith a network analyser. The locking range has beenoptimised for locking gains of 20 dB and 10 dB, respectively.It was broadened by a factor of 3-5 at 20 dB and doubled atlOdB, which is roughly in accordance with the theoreticalresults of Reference 24. Absolute values are 1 -7% relativebandwidth at 20 dB and 4-5% at 10 dB for a Gunn oscillatorwith a centre frequency of 14-5 GHz. These results are verysimilar to what can be obtained with oscillators in rectangu-lar waveguide techniques. The amplitude and phase charac-teristics of the compensated injection-locked oscillator areshown in Fig. 10 for a power gain of 20 dB.

-RD

Fig. 9 Broadband-locking circuita Equivalent circuitb Circuit pattern

150

100

$ 5 0

aa

-50

-100

phase

V*U 145frequency.GHz

22

20

18 m

Ca

16 °»

U

12

Fig. 10 Performance of broadband-locking circuit

6 Varactor-tuned oscillator (v.t.o.)

A severe disadvantage of the integrated circuits describedhere is the lack of an inexpensive possibility of mechan-ically tuning the frequency. Therefore, it is very importantto provide electrical tuning. Then active frequency stabilis-ation can be easily accomplished employing well-knowntechniques such as automatic phase control.

Electrical tuning capability is most easily introduced bycoupling a varactor diode to the basic oscillator structure.

In a final application (especially at millimetre wavelengths)diode chips will be used because of their small parasiticsand their low price. We have, however, used a varactordiode in an l.i.d. package owing to its easy mechanicalhandling. It is mounted in a separate coupling structurebetween the active device and the waveguide load, at alength / away from the diode mount. The equivalent circuitis sketched in Fig. 1 la and the fin line slot pattern inFig. lib. We have used the diode mount of Fig. 2b. Thelength / has been optimised experimentally with respect to alarge tuning range. The degree of coupling of the varactordiode (i.e. the power loss in this diode) is adjusted bychoosing the physical dimensions of the varactor couplingstrip. The l.i.d. package is soldered directly to this stripwith an In-Ag alloy. The position of the varactor packagemust be adjusted carefully, because it has a strong influenceon the performance of the oscillator.

L-—____________

Fig. 11 Varactor-tuned oscillator

a Equivalent circuitb Circuit pattern

300r

200

O

3 100

frequency

f=U-48GHz

0 10 20 30varactor voltage.V

Fig. 12 Performance of varactor-tuned oscillator

16

\U

MICROWA VES, OPTICS AND ACOUSTICS, MA Y 1979, Vol. 3, No. 3 119

The active device (Gunn diode) and the varactor diodehave to be biased separately. For that reason, stripline filtersare used, as is depicted in Fig. lib. The design of thesefilters is simple, because the electromagnetic fields both instripline and in fin line are found to be decoupled to a highdegree in the arrangement shown. A typical example for atuning characteristic of the oscillator structure of Fig. libis shown in Fig. 12. It corresponds to a case where thepower dissipated in the varactor diode is approximately0-5 dB, which means the output power of the oscillatordegrades by this value, when the varactor diode is coupledand the free-running frequency readjusted. The tuningrange is 260 MHz when the varactor bias varies between 0and 34 V, with a power reduction of less than 1 dB over thetuning range. Again, the results are comparable to the fam-iliar two-post v.t.o. structure in rectangular waveguide.However, as a difference, the integrated structure gives thepossibility to use non encapsulated varactor diodes withlow parasitics.

7 Conclusions

Various oscillator circuits have been developed in integratedfin line technique. There are several structures for couplingthe active device to the passive circuitry. These diodemounts have been investigated with respect to circuitefficiency and obtainable loaded Q-foctor. A comparisonhas shown that an active device should be coupled to thewaveguide by a planar pattern which resembles the post-coupling structure familiar from conventional rectangularwaveguide (Fig. 2b), by an antenna coupling (Fig. 2c), orthrough a transmission cavity, where care has to be takenthat the coupling at both the diode and the load port canbe adjusted separately (Fig. 2d). To arrive at high circuitefficiency together with a high-loaded Q-foctor, the un-loaded <2-factor of the frequency determining resonatorshould be made as high as possible. That means that nometallic fin should penetrate into the housing waveguideacross the resonant length. The performance of the fin lineoscillator circuits is then similar to that of conventionalrectangular-waveguide oscillators.

The coupling structure utilising a transmission cavitygives considerable freedom since the oscillator loaded Q,which is a measure of frequency stability, can be tradedwith output power. The other coupling structures do notshow this feature. Frequency stability (loaded oscillator Q)may, however, be increased for these patterns, too, bycoupling them to an external integrated cavity. The possiblefeatures of such an assembly have been discussed.

The fin line oscillators can, of course, be operated asinjection-locked amplifiers. Reactance compensation hasbeen used to increase the locking range. This could berealised by integrating an additional capacitive stub on theprinted board at a special distance from the diode mount.Thus, the locking range has been increased by a factor of3-5 at 20 dB locking gain and doubled at lOdB gain,respectively.

Finally, electronic tunability has been provided bycoupling a varactor diode to the integrated oscillator. Astructure is shown, which allows one to adjust the powerloss in the varactor to vary the tuning range. Varactortuning is possible with the diode mounts recommendedabove. The experimental performance of a varactor-tunedGunn oscillator with medium tuning range and low varactorloss has been shown as an example.

8 Acknowledgement

The author is indebted to the Deutsche Forschungsgemein-schaft for financial support.

9 References

1 MEIER, P.J.: 'Two new integrated-circuit media with specialadvantages at millimeter wavelengths'. IEEE-MTT Symposiumdigest, Arlington Hights, 111., USA, 1972, pp. 221-223

2 MEIER, P.J.: 'Equivalent relative permittivity and unloadedQ-factor of integrated finline', Electron. Lett., 1973, 9,pp. 162-163

3 MEIER, P.J.: 'Integrated fin-line millimeter components'.IEEE-MTT G Symposium digest, Atlanta, Georgia, USA, 1974,pp. 195-197

4 MEIER, P.J.: 'Integrated fin-line millimeter components', IEEETrans. 1974, MTT-22, pp.1209-1216

5 MEIER, P.J.: 'Integrated fin-line, a versatile and provenmillimeter transmission line', Micro wave J. 1976, 19, pp. 24-25

6 HOFMANN, H.: 'Fin-line dispersion', Electron. Lett. 1976, 12,pp. 428-429

7 SAAD, A.M.K., and BEGEMANN, G.: 'Electrical performance offin-lines of various configurations', IEE J. Microwaves, Opt. &Acoust. 1977, l,pp. 81-88

8 HOFMANN, H.: 'Calculation of quasi-planar lines for mm-waveapplication'. IEEE MTT-Symposium digest, San Diego,California, USA, 1977, pp. 381-383

9 HOFMANN, H.: 'Dispersion of planar waveguides for millimeter-wave application', Arch. Elektron. & Uebertragungstech., 197731, pp. 40-44

10 HOEFER, W.J.R.: 'Fin-line design made easy'. IEEE-MTTSymposium digest, Ottawa, Canada, 1978, p. 471

11 SAAD, A.M.K., and SCHUNEMANN, K.: 'A simple method foranalyzing fin-line structures'. IEEE MTT Symposium digest,Ottawa, Canada, 1978, pp. 405-407

12 SAAD, A.M.K., and SCHUNEMANN, K.: 'A first order designtheory for integrated fin-line circuits'. Proceedings of the 6thMicrocoll., Budapest, 1978, pp. Ill—7/46.1 —III-7/46.4

13 SAAD, A.M.K., and SCHUNEMANN, K.: 'Field description forgeneralized (multi-slot) fin-line structures'. Proceedings of the8th European Microwave Conference, Paris, 1978, pp. 101-105

14 MEIER, P.J.: 'New developements with integrated fin-line andrelated printed millimeter circuits'. IEEE-MTT Symposiumdigest, 1975, pp. 143-145

15 MEIER, P.J.: 'Planar multiport millimeter integrated circuits'.IEEE-MTT Symposium digest, San Diego, California, 1977,pp. 385-387

16 HOFMANN, H., MEINEL, H., and ADELSECK, B.: 'New inte-grated mm-wave components using fin-lines'. IEEE-MTT Sym-posium digest, Ottawa, Canada, 1978, pp. 21-23

17 COHEN, L.D., and MEIER, P.J.: 'Advances in E-plane printedmillimeter-wave circuits'. IEEE-MTT Symposium digest, Ottawa,Canada, 1978, pp. 27-29

18 BEGEMANN, G.: 'An X-band balanced fin-line mixer"IEEE-MTT Symposium digest, Ottawa, Canada, 1978,pp. 24-26

19 TOMIYASU, K., and BOLUS, J.J.: 'Characteristics of a newserrated choke', IRE Trans. 1956, MTT-4, pp. 33-36

20 BRACKETT, C.A.: 'The elimination of tuning-induced burnoutand bias-circuit oscillations in IMPATT oscillators', Bell Syst.Tech. J., 1973, 52, pp. 271-306

21 KNOCHEL, R., SCHUNEMANN, K., and BUCHS, J.D.: 'Theoryand performance of cavity-stabilised microwave oscillators',IEEJ. Microwaves, Opt. & Acous. 1977, 1, pp. 143-165

22 SCHIEK, B., and SCHUNEMANN, K.,: 'Noise of negative resist-ance oscillators at high modulation frequencies', IEEE Trans..1972,MTT-20, pp. 635-641

23 AITCHISON, C.S., DAVIS, R., and GIBSON, P.J.: 'A simplediode parametric amplifier design for use at S, C, and X-band',ibid., 1967, MTT-15, pp. 22-31

24 KNOCHEL, R., and SCHUNEMANN, K.: 'Reactance compen-sation techniques for injection locked amplifiers', Arch. Elektron.& Uebertragungstech., 1979, 33, pp. 49-56

120 MICROWA VES, OPTICS AND A COUSTICS. MA Y 1979, Vol. 3, No. 3