1076 H. J. Ryoo et al.: Design of High Voltage Capacitor Charger with Improved Efficiency, Power Density and Reliability

1070-9878/13/$25.00 © 2013 IEEE

Design of High Voltage Capacitor Charger with Improved Efficiency, Power Density and Reliability

H. J. Ryoo, S. R. Jang, Y. S. Jin, J. S. Kim, Y. B. Kim Korea Electrotechnology Research Institute

Electric Propulsion Research Center Changwon, Korea

S. H. Ahn, J. W. Gong

University of Science & Technology Department of Energy Conversion Technology

Daejeon, Korea

B. H. Lee Agency for Defense Development

Department of Energy Conversion Technology Daejeon, Korea

and D. H. Kim

College of Catholic SangJi Department of Electric railway

Andong, Korea

ABSTRACT This paper describes the design of a 48 kJ/s high-voltage capacitor charging power supply (CCPS), focusing on its efficiency, power density, and reliability. On the basis of a series-parallel resonant converter (SPRC) that provides high efficiency and high power density owing to its soft-switching, the design of the CCPS is explained in detail, including its input filter, resonant tank parameters, high-voltage transformer and rectifier, as well as its protection circuit. By using two resonances per switching cycle, which provides a trapezoidal instead of a sinusoidal waveform of the resonant current, the proposed CCPS can take advantage of the lower conduction loss and reduced switching loss by improving the crest factor and allowing a higher value of the snubber capacitor, respectively. In addition, the compact design of an input filter without bulky components such as a DC reactor and an electrolytic capacitor allows for high power density, a high power factor, and low cost. In addition, the control loops for the voltage and current were optimized with a fast response time in order to compensate for the low frequency ripple of the input voltage, which results from the reduced filter component. Experiments on the developed charger were carried out with both resistor and capacitor loads in order to measure not only its efficiency and power factor with respect to the output power but also its charging time, in order to estimate the average charging current. The experimental results obtained with a resistor load showed a maximum efficiency of 96% and a power factor of 0.96 for a full-load condition. For the measured charging time of a 4 mF capacitor, with 9.68 s for 10 kV charging, the average charging current was estimated as 4.13 A. Moreover, to verify the reliability of the developed CCPS, a variety of tests, including opening and shorting of the output terminal as well as misfiring of the discharge switch during the charging operation, were performed with a 200 kJ pulsed power system. Finally, it was experimentally confirmed that the developed CCPS shows high performance in terms of efficiency (96 %), power factor (0.96), and reliability with a high power density (820 W/L).

Index Terms – HVDC converters, resonant power conversion, pulsed power systems.

Manuscript received on 26 September 2012, in final form 10 February 2013.

IEEE Transactions on Dielectrics and Electrical Insulation Vol. 20, No. 4; August 2013 1077

1 INTRODUCTION

SINCE the field of pulsed power applications has been growing, the importance of a capacitor charging power supply (CCPS) for a pulsed power system based on capacitive energy storage with a closing switch cannot be overemphasized, from the view point of efficiency, compactness, and reliability. Accordingly, a considerable number of converter topologies and control methods have been proposed [1-17]. In order to implement capacitor charger, the current mode controlled or current source converters can be suitable candidates. Although, the current mode controlled soft-switching converter has the advantages such as low ripple current, easy filter design and reduced switching loss [18, 27, 28], it is not so easy to protect from short circuit condition from high-voltage output. Due to above reason, the resonant current source converter topologies are still dominantly used for high-voltage power supply and high-voltage capacitor charging application [1-17, 20-26]. A series resonant converter (SRC) that operates in a discontinuous conduction mode (DCM) is a widely used topology for CCPS owing to its many advantages, which include its pure current source characteristic, soft-switching for both the turn-on and turn-off transitions, and small component counts [1-6]. In addition, the switching frequency control that is less than half of the resonant frequency allows for a wide load range. The system’s drawbacks, on the other hand, include audible switching noise at low frequency operation and a relatively high conduction loss attributed to the crest factor of the resonant current. Although an SRC operating in a continuous conduction mode (CCM) can reduce the conduction loss as compared to a DCM, the inversely proportional relationship between the switching frequency and the output current limits the load range, particularly for a no-load condition. In order to improve the performance of a CCPS, many types of converter topologies have been proposed depending on different specifications and applications. The phase shifted pulse width modulation (PWM) control technique based on a full-bridge resonant converter, which provides a constant switching frequency and a low crest factor, has been introduced for a high-repetitive solid-state modulator [7, 8]. By using the above resonance region with control of not the switching frequency but with phase delay, the introduced scheme shows low conduction as well as low switching loss. Moreover, a constant switching frequency allows for an easy filter design and a wide load range, even in the case of a no-load operation. Other studies that focus not only on the control method but also on the structure of the resonant tank have reported approaches such as a parallel resonant converter (PRC) [9-11], a series-parallel resonant converter (SPRC, also called LCC) [12-14], and an LCL resonant converter [15-17]. On the basis of a survey of these studies, the characteristics of each topology were investigated, including a voltage boost-up function obtained through the use of a parallel capacitor and the effect of parasitic capacitance due to a large transformer ratio. In addition, a comparison of the design specifications was summarized in terms of the semiconductor device, maximum switching frequency, and

power density [9]. On the basis of the discussion in this reference, it seems reasonable to conclude that a suitable converter topology should be determined not just by its rated power and efficiency but also by the system requirements, such as power density, reliability, power factor, and cost. For example, the power factor correction circuit is important for specific applications that require a continuous charging operation to generate high-repetition pulsed power, even though it increases the volume and cost [18, 19]. A relatively high-power and low-repetition system, on the other hand, is usually designed with an LC input filter to achieve a low cost and high power density [1]. There are also many other reported approaches, which have focused on the cost [20], power density [9], and reliability [2].

In this paper, an enhanced scheme for an SPRC is proposed for a CCPS, using the specifications listed in Table 1. A photograph of the device is shown in Figure 1. The main concepts and distinctive features of the proposed scheme are briefly introduced in Section 2. A detailed discussion of the analysis, design, and implementation of the developed CCPS, which focuses on it efficiency, power density, and reliability, is presented in Section 3. In addition, a PSpice simulation and the experimental results obtained with a resistor as well as a capacitor load are presented to validate the design. And the results of reliability testing are presented, as well.

Figure 1. Photograph of developed CCPS.

Table 1. Specifications of the Developed CCPS.

Numerical specifications of the developed CCPS

Input voltage 380 Vac ± 10%

Maximum charging voltage, Vcharging 12 kV Maximum charging current, Icharging 4A

Continuous output power, PContinuous 30 kW

Instantaneous output power, P Instantaneous

48 kW (5min)

Maximum efficiency, ηmax 96%

Maximum power factor, PFmax 0.96

Dimensions

Width: 435 mm Depth: 400 mm Height: 210 mm

Power density 820 W/L

Additional functions and characteristics of the developed CCPS

Protections

Over-current Over-temperature

Over-voltage

Reliability against malfunction

Output short during charging Output open during charging Load short during charging

1078 H. J. Ryoo et al.: Design of High Voltage Capacitor Charger with Improved Efficiency, Power Density and Reliability

2 DISTINCTIVE CHARACTERISTICS OF THE PROPOSED CCPS

The distinctive characteristics of the proposed CCPS are as follows.

A. Enhanced series-parallel resonant topology-based design:

- A series-parallel resonant converter can take advantages of a series and a parallel resonant converter, current source characteristics, and voltage boost-up function.

- In addition to the general merit of the CCM with the above resonance operation, the zero voltage (ZV) turn-on of the inverter switches, the relatively high value of the lossless snubber capacitor connected with each inverter switch help to dramatically reduce the turn-off loss.

- Two resonant frequencies give the resonant current waveform a trapezoidal shape, which allows for lower conduction loss.

- A wide load range is achieved by adapting two switching frequencies, of a few tens of kilohertz and a few hundreds of hertz, for voltage regulation and the charger’s on/off function, respectively.

- A parallel resonant capacitor is implemented by using capacitors that are connected with high-voltage rectifier diodes, which are required for voltage balancing.

B. Compact input filter design with high-power factor:

- The input filter is designed without using low-frequency bulky components, including a DC reactor and an electrolytic capacitor, and provides high-power density up to 820W/L.

- The optimized design of a three-phase rectifier with capacitive filter improves the distortion power factor and provides a high power factor up to 0.96, without using an additional power factor correction circuit.

C. A simple and reliable protection circuit for inverter switches:

- By sensing the current through the lossless snubber capacitor, a novel protection circuit is created that can handle abnormal conditions, such as hard-switching, leg-short, and over-current.

- A distinctive current-sensing method that employs a small ferrite core can be implemented at the PCB at a low cost.

D. Reliable operation in the face of unintended malfunctions:

- The reliability of the developed CCPS was verified in a test of intentional malfunctions, which included output shorting and opening output even during charging.

3 DESIGN OF THE PROPOSED CCPS 3.1 STRUCTURE OF THE PROPOSED CCPS

The overall scheme of the proposed CCPS, as shown in Figure 2, can be mainly classified into the following six parts.

A. Input filter and Rectifier

One design point that must be noted is that the input filter and rectifier are made up of only a three-phase diode bridge (DBridge) and a filter capacitor (CLink). As compared with a general DC power supply, the CCPS is not required to consider the input and output voltage ripple in great detail, because the load is a large capacitor. Consequently, the proposed CCPS is designed without bulky components, including a DC reactor and an electrolytic capacitor, as well as additional circuits to protect against an inrush current, in order to achieve high-power density. Furthermore, a direct current supply from the rectifier to the inverter, which does not have a large capacitive filter, helps to improve the distortion power factor. A small film capacitor (CLink) is used only for supplying and absorbing the high-frequency current from the resonant inverter.

Figure 2. Overall scheme of the proposed CCPS.

IEEE Transactions on Dielectrics and Electrical Insulation Vol. 20, No. 4; August 2013 1079

B. SPRC-based full-bridge resonant inverter

The topology of the proposed CCPS is basically designed as an SPRC resonant converter composed of a series resonant inductor (Lr) and a capacitor (Cr) with a parallel capacitor (Cp). The parallel capacitor (Cp), however, is not shown in Figure 2 because capacitors (Cp1-1–Cp4-6) that are connected in parallel with the series of stacked high-voltage diodes (Dhv1-1–Dhv4-6) are designed not only for voltage balancing but also to play the role of a parallel capacitor. When the high-voltage diode stack commutates from a reverse to a forward bias, the transformer’s secondary winding current flows in order to discharge the voltage across the parallel connected capacitor. At this transition interval, the series combination of (Cpn-1–Cpn-6) affects the resonant frequency in the same way as a parallel capacitor (Cp). That is, the proposed topology can have two resonant frequencies during one switching cycle, and this allows the resonant current waveform to assume a trapezoidal shape, for a low crest factor. In addition to the relatively low conduction loss, the turn-off switching loss can be reduced by increasing the value of the lossless snubber capacitors (C1–C4), because the energy stored in the resonant tank is sufficient for charging and discharging the voltage across the switches (S1–S4) during the switch-off and switch-on transitions, respectively [7, 8]. Therefore, the proposed CCPS can achieve a ZV turn-on with reduced conduction and turn-off losses.

C. Gate driver (GD) for ZVS

Depending on the current value at the switch turn-off time, the required delay interval for the ZV turn-on of the other switch pair varies owing to the difference in the energy needed for charging and discharging the lossless snubber capacitors. In order to achieve ZV turn-on in the entire range of load conditions, a novel gate drive circuit (ZD for ZVS) with collector diodes (Dc1–Dc4) is suggested, which provides a flexible time delay by sensing the collector-emitter voltage.

D. High-voltage transformer, rectifier, and sensing circuit

Owing to its voltage boosting property, the SPRC topology facilitates the design of a high-voltage transformer by reducing the turn ratio, as well as helps to prevent distortion from arising owing to the effect of parasitic capacitance [9]. As depicted in Figure 2, two secondary windings (N2, N3) are connected with the voltage doubled rectifiers, which consist of a series of stacked diodes (Dhv1-1–Dhv4-6), voltage balancing capacitors (Cp1-1–Cp4-6), and filter capacitors (Cf1–Cf4). In addition, a high-voltage sensing circuit is designed, which uses resistors (Rs1–Rs8) and a capacitor (Cs1–Cs8) divider [4].

E. Switch protection circuit against abnormal operation

In order to protect the inverter switches (S1–S4) against abnormal operations, including hard-switching and leg short, a simple and reliable protection circuit is proposed that uses transformers (TX2, TX3) and a comparator for current sensing and comparing, respectively. Depending on the operational principle of a full-bridge circuit with a lossless snubber capacitor, it is clear that the value of the current through each leg’s snubber capacitor is the same but the polarity is opposite when the converter operates without problems. On the basis of this basic concept, the design positions two transformers on

the current path by means of a small ferrite core. Owing to the aforementioned characteristic of the snubber capacitor current, the sum of the voltage (Vprotection) on each secondary winding is definitely zero, except during abnormal operations. The voltage and current waveforms associated with the protection circuit are explained in the following parameter design with simulation Section.

F. Controller

As shown in Figure 2, the PI controllers, which have separate designs, one for the output voltage and the other for the output current, are connected through diodes (Dor1, Dor2). This design is accompanied by the limitation of a single output, with one output controlling the other [4]. Usually, an output voltage control loop works with a maximum charging current operation for capacitor charging applications. In addition, the controller PI gain is discreetly designed in order to compensate for the low frequency ripple due to the small input filter. Depending on the value of the control signal (Vc), the frequency of the switching signals for a full-bridge circuit are modulated with a 50% duty cycle and transferred through the AND gate. For reliable operation of the CCPS, the block signal (Vblock) generated from the protections include the over-current, over-voltage, over-temperature, and Vprotection cut-off of the gating signals (G1–G4) to the drive circuit.

3.2 PARAMETER DESIGN WITH SIMULATION

The detailed design of the parameters, including the resonant tank, snubber capacitor, and transformer turn ratio, was carried out using the basic formula of a resonant converter and a PSpice simulation. On the basis of the specifications of the proposed CCPS given in Table 1, the design and simulation parameters are summarized in Table 2. An analysis of the SPRC converter and the design methodology of the resonant tank parameters, which include the series resonant inductor (Lr), series resonant capacitor (Cr), and parallel resonant capacitor (Cp), are reported in [21-25] and are based on the assumption of a sinusoidal resonant current waveform.

Table 2. Summary of Design and Simulation Parameters

Design and simulation parameters of the proposed CCPS

Resonant inductance, Lr 33 µH

Resonant capacitance, Cr 16 µF

Fundamental resonant frequency, fo 7 kHz

Fundamental characteristic impedance, Zo 1.44 Ω

Capacitors for voltage balancing and parallel resonant,

Cp1-1–Cp4.6

0.1 µF

Equivalent parallel capacitance, Cp 1.224 µF

Resonant frequency, including Cp, fop 26 kHz

Characteristic impedance, including Cp, Zop 5.396 Ω

High-voltage transformer (TX1) turns ratio, N1:N2:N3 14:60:60

Lossless snubber capacitors, C1–C4 0.3 µF

Input filter capacitor, CLink 80 µF

Output filter capacitors, Cf1–Cf4 14 µF

High-voltage sensing resistors, Rs1–Rs8 2.5 MΩ

High-voltage sensing capacitors, Cs1–Cs8 1 nF

Current sensing transformer (TX2, TX3) turns ratio,

NS1-1(NS2-1):NSN-2(NS2-2)

1:30

1080 H. J. Ryoo et al.: Design of High Voltage Capacitor Charger with Improved Efficiency, Power Density and Reliability

As mentioned in our description of the system’s distinctive characteristics, however, a trapezoidal resonant current is proposed for high-efficiency operation. It is clear that the improvement of the crest factor allows for a lower conduction loss. In addition, the higher value of the current at the turn-off transition provides sufficient energy for the charging and discharging of the snubber capacitors, which are used for reducing the turn-off switching loss. On the basis of the basic equation of the resonant current, the design concept of the proposed CCPS is described as follows.

From the simple circuit that is composed of a voltage source and a series resonant tank (Lr, Cr), the expression of the resonant current can be derived in terms of the induced voltage (Vd), the characteristic impedance (Zo), and the angular resonant frequency (ωo), as shown in the equation below.

00 0 0 0 0

( )( ) ( ) cos ( ) sin ( )r

r r

d cL L

o

V V ti t I t t t t t

Z

(1)

where 0t denotes the initial time of S1 and S2 conducting, and

0 0

12 ,

r r

fL C

0

r

r

LZ

C

Equation (1) implies that Zo and ωo, which determine the peak value and frequency, are the main factors for shaping the resonant current. That is, if the converter operates with two resonant frequencies during one switching cycle, the shape of the current can be transformed from sinusoidal to trapezoidal. Therefore, the proposed SPRC topology-based CCPS chooses two resonant frequencies by using a parallel capacitor (Cp). Besides the fundamental resonant frequency produced by the resonance between Lr and Cr, the series combination of Cr and Cp provides high-frequency resonance during the Cp charging interval. Accordingly, a trapezoidal resonant current can be obtained by increasing the current with a high-frequency resonance and maintaining it with a low-frequency resonance. The equation for the resonant frequency, including Cp (fop), can be easily derived, as shown below.

1

2 /( )op

r r p r p

fL C C C C

(2)

From the overall structure of the proposed CCPS, the value of Cp can be calculated by Cpn-n, which requires charging and discharging for the forward and reverse bias of the high-voltage rectifier diodes. That is, it works not only for voltage balancing but also as a parallel resonant capacitor. Equation (3) shows the calculation of the equivalent parallel capacitance.

2

21 2 3 4

1

( )p p p p p

NC C C C C

N

(3)

where Cpn is the series of connections from Cpn-1–Cpn-6. In equation (3), it should be noted that the value of Cp, which

has the function of voltage boosting, depends on the transformer’s turn ratio. Therefore, it is necessary to consider both parameters together in order to satisfy a given specification. Using the aforementioned design concepts, a PSpice simulation was performed in order to obtain the optimal design of the proposed CCPS. Figures 3 and 4 shows the results of the PSpice simulation for a rated operation using the

design parameters summarized in Table 2. The waveforms in the bottom graph shown in Figure 3 depict the resonant current waveform with a switching signal of S1. Owing to the trapezoidal current waveform, the rms value is estimated at approximately 131 Arms, even in the case of a 48 kW operation.

From the average and instantaneous loss of the insulated gate bipolar transistor (IGBT) depicted in the top graph in Figure 3, 150 W of switch loss is observed on account of the ZV turn-on and with reduced turn-off and conduction loss. In order to more clearly determine the losses of the IGBT, the voltage and current waveforms of switching during one switching period were charted, as shown in Figure 4. The gradual decrease and increase in the collector-emitter voltage (VCE) at the turn on/off switching transitions reflect the discharging and charging of the lossless snubber capacitor. Although the gate drive signal has a small delay time during discharging of snubber capacitor, the ZV turn on was achieved as shown in Figure 3. The waveforms at the switch turn-off transitions elucidate that the slow slope of VCE helps to reduce the turn-off loss by decreasing the cross-section area of the switch current and voltage. That is, the higher value of the snubber capacitor provides a lower turn-off loss. There is a constraint for the increase in capacitance, however, because the charged energy on the lossless snubber capacitor must be discharged prior to switch turn-on for ZV switching. To increase the energy stored in the resonant tank that is used for charging and discharging the lossless snubber capacitor, a relatively low value of the fundamental resonant frequency (fo) is adopted in order to maintain the resonant current.

Figure 4. Simulation waveforms of switch turn on/off transitions. (Top:Collector-emitter voltage (green), current through anti-parallel diode (red), andsnubber capacitor current (brown). Bottom: Resonant current (brown) withswitching signal (blue).)

Figure 3. Simulation waveforms for 42 kW rated operation (Top: Average andinstantaneous IGBT loss (brown). Bottom: Switching signal (blue), resonantcurrent (brown) with rms value (red).)

IEEE Transactions on Dielectrics and Electrical Insulation Vol. 20, No. 4; August 2013 1081

The trapezoidal current waveform facilitates the increase in the snubber capacitance up to 300 nF, which dramatically reduces the instantaneous turn-off loss by up to 1.2 kW. In addition, the GD for the ZVS provides flexible dead time, depending on the value of VCE, because the required time for anti-parallel diode conduction changes with respect to the load condition. As can be seen in the waveform at the bottom of Figure 4, a small time delay of the switching signal can be observed before the ZV condition. The simulation results, including the overall losses and estimated efficiency are summarized in Table 3. The losses for each semiconductor switch include not only the conduction loss due to the saturation voltage or forward voltage drop but also the switching losses at the turn-on and turn-off transitions. From the energy losses of the IGBT, it is clear that the 5 mJ of conduction energy loss per single switching period is dominant, as compared to the turn-on and turn-off energy losses of 0 mJ and 0.8 mJ, respectively. And the losses on the resonant tank including resonant inductor and capacitor are estimated from the rms value of current and the magnetic flux in the ferrite core [26, 27]. In addition, the winding losses of the transformer are calculated with consideration of both the skin effect and the proximity effect [26]. Finally, 94.6% of the overall efficiency is estimated from the 2.6 kW of total loss for 48 kW of rated operation.

The proposed switch protection circuit was verified in a simulation, as shown in Figure 5. In contrast with the balanced voltage across the primary winding of two current sensing transformers that have the same magnitude with an opposite polarity in normal operation, the imbalanced voltages were measured in the case of abnormal operation, which included hard-switching of the switch and leg short. In case of hard-switching and leg short, relatively high value of current through sensing transformer is observed due to the discharging of the stored energy in snubber, instantaneously. Accordingly, the voltage difference due to the unbalanced current activates the switch protection circuit by comparator logic as shown in Figure 2.

4 EXPERIMENT RESULTS

4.1 EXPERIMENTAL RESULTS WITH RESISTOR LOAD

Figure 7. Measured efficiency and power factor of developed CCPS.

Figure 6. Experimental waveforms for 48 kW of rated operation. (Resonantcurrent (blue, 100 A/div.), Collector-emitter voltage (purple, 200 V/div.),Output voltage (red, 2 kV/div.), 10 µs/div.)

Resonant current

Collector-emitter voltage Output voltage

Figure 5. Simulation waveforms of the proposed switch protection circuit.(Top: normal operation. Middle: hard-switching operation. Bottom: leg short. Left: current through TX2 and TX3. Right: voltage across TX2 and TX3.)

Table 3. Summary of PSpice Simulation Results.

Simulation conditions: Vd = 480 V, fs = 26 kHz, Rload = 3 kΩ, Pout = 48 kW

Peak value of resonant current, Ires,peak 184 A

RMS value of resonant current, Ires,rms 131 A

Peak value of secondary winding current, Isec,peak 22 A

RMS value of secondary winding current, Isec,rms 15 A

Crest factor, CF 1.4

Average loss of each IGBT, PIGBT,loss,avg 150 W

Estimated energy losses, Eon, Eoff, Econ 0, 0.8, 5 mJ

Estimated loss of resonant capacitor, PCr 10 W

Estimated loss of resonant inductor, PLr 30 W

Average loss of each high-voltage diode, Pdiode,loss,avg 10 W

Estimated primary winding conduction loss, Ppri 850 W

Estimated secondary winding conduction loss, Psec 500 W

Estimated total loss, Ptotal,loss 2.6 kW

Estimated overall efficiency, ηEstimated 94.6%

1082 H. J. Ryoo et al.: Design of High Voltage Capacitor Charger with Improved Efficiency, Power Density and Reliability

Various experiments using the developed CCPS were performed with a resistor load in order to verify the design described in Section 3. From the experimental results that were obtained, the feasibility of voltage and current control was confirmed. In addition, the waveforms, efficiency, and power factor of the developed CCPS were measured with respect to the output power. The developed CCPS was tested with a 3 kΩ resistor for 48 kW (12 kV, 4 A) of output power operation, instantaneously. The waveforms of the resonant current, collector-emitter voltage, and output voltage depicted in Figure 6 prove the soft switching of the IGBT and the two resonances per switching cycle, as explained in Section 3. Accordingly, the maximum efficiency and power factor were measured as being 96%, and 0.96, respectively.

4.2 EXPERIMENTAL RESULTS OF CAPACITOR CHARGING AND RELIABILITY TESTS

The developed CCPS was tested for a system with 200 kJ of pulsed power. The experimental circuit used for capacitor charging as well as for reliability testing is shown in Figure 8. Mechanical switches (SWcharging, SWshort, SWDump) and thyristors (Thy1, Thy2) were installed in order to demonstrate tests of malfunctioning, which included an open or short charger output terminal and the misfiring of thyristors during the charging operation.

Figure 11. Waveforms when Thy1 misfires during two module chargingoperations. (Top: Capacitor voltage 1 (purple, 5 kV/div.). Middle-upper:Capacitor voltage 2 (green, 5 kV/div.). Middle-lower: Capacitor charger outputvoltage (red, 5 kV/div.). Bottom: Charging current (blue, 10 A/div.), 5 s/div.)

(a) Waveforms when charger output is shorted during charging operation.

(b) Waveforms when charger output is open during charging operation.

(c) Waveforms when Thy1 misfires during charging operation.

Figure 10. Reliability tests waveforms of the developed CCPS. (Top: Capacitor voltage 1 (purple, 2 kV/div.). Middle: Capacitor charger output voltage (red, 2 kV/div.). Bottom: Charging current (blue, 10 A/div.))

Figure 9. 200 kJ normal charging waveform. (Top: Capacitor voltages 1&2(purple, 2 kV/div.). Middle: Capacitor charger output voltage (red, 2 kV/div.).Bottom: Charging current (blue, 10 A/div.), 2 s/div.)

Figure 8. Experimental circuit for capacitor charging and reliability test.

IEEE Transactions on Dielectrics and Electrical Insulation Vol. 20, No. 4; August 2013 1083

Figure 9 shows the normal charging waveforms of a 4 mF (Cstorage1 = Cstorage2 =2 mF) capacitor up to 10 kV. For 9.68 s of measured charging time, the average charging current was calculated as 4.13 A, which is a value greater than the given specifications. In addition, the results of the reliability tests are depicted graphically in Figure 10.

The charger output terminal short and opening during charging a condition were demonstrated by closing SWshort and opening SWcharging, and the results are depicted in Figures 10 (a) and (b), respectively. From the measured waveforms at a charger output short for a charging condition of approximately 6 kV, the current source characteristic of the proposed scheme was experimentally confirmed, even though a gigantic instantaneous current (90 A) was observed owing to the stored energy on the output filter capacitor.

The opening of the charger output test was also performed by opening SWcharging at a charging condition of around 6 kV. However, the results show a waveform similar to that of normal charging, owing to the arc on the mechanical switch. Only a small increase in the charger output voltage was measured, as depicted in Figure 10 (b).

The experiment of misfiring the thyristor (Thy1) during a charging condition was performed in order to ensure the reliability of the developed CCPS against the surge current from the stored energy in Cstorage1. The ballast resistor (Rballast) limits the peak value of the surge current through diodes (Dhv1-1–Dhv4-6), which are located in the high-voltage rectifier. 10 Ω of Rballast protects the CCPS with a relatively small charging loss, as compared to the charging power. After the misfiring of Thy1, the developed CCPS recharges the storage capacitor up to the reference voltage of 10kV.

In addition to the results shown in Figure 10, which were measured using a single module consisting of Cstorage1 and Thy1, an experiment with the misfiring of one module was performed when the developed CCPS charged two or more capacitors in parallel, as depicted in Figure 9. From the results of the Thy1 misfiring during a Cstorage1 and Cstorage2 charging operation, as shown in Figure 11, it is clear that the developed CCPS fulfills the charging operation up to the reference voltage, independent of the load condition. Finally, the reliability of the developed CCPS was verified in a variety of tests of malfunctioning.

5 CONCLUSIONS

A compact and high-efficiency CCPS for reliable pulsed power applications was developed. The compact SPRC-based design of the high-voltage capacitor charger was explained, which included its input rectifier with a small capacitive filter; its modified resonant tank structure, particularly in terms of the high-voltage power supply; its useful gate drive circuit with a flexible dead time; and a novel protection circuit for the inverter switches. The resonant tank design, which gives the resonant current a trapezoidal shape, which in turn allows for a relatively low conduction loss and turn-off switching loss, was explained. On the basis of a basic analysis of the proposed circuit, a detailed selection of parameters and verification, including the estimation of losses as well as the overall efficiency, were accomplished using PSpice simulation.

Finally, the proposed CCPS was developed with 820 W/L of high-power density. The experimental results obtained with a resistor load showed a maximum efficiency of 96% and a power factor of 0.96. Furthermore, the feasibility of controlling the output voltage by compensating for the low frequency ripple of the input

voltage was experimentally verified. In addition, various tests not only during normal capacitor charging but also during malfunctioning operation during charging were also performed. The results confirm that the developed CCPS can be effectively applied to a capacitive energy storage-based pulsed power application that requires compactness and high reliability.

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Hong-Je Ryoo received the B.S., M.S., and Ph.D. degrees in electrical engineering from SungKyunkwan University, Seoul, Korea in 1991, 1995, and 2001, respectively. From 2004 to 2005, he was with WEMPEC at the University of Wisconsin-Madison, as a Visiting Scholar for his postdoctoral study. Since 1996, he has been with the Korea Electrotechnology Research Institute in Changwon, Korea. He is currently a Principal Research Engineer in the Electric Propulsion Research Division and a Leader of the Pulsed Power World Class Laboratory at that institute. Also, he has been an

Professor in the Department of Energy Conversion Technology, University of Science & Technology, Deajeon, Korea, since 2005. His current research interests include pulsed power systems and their applications, as well as high power and high voltage conversions. Dr. Ryoo is a member of the Korean Institute of Power Electronics (KIPE), and the Korean Institute of Electrical Engineers (KIEE).

Sung-Roc Jang was born in Daegu, Korea, in 1983. He received the B.S. degree from Kyungpook National University, Daegu, Korea, in 2008, and the M.S. and Ph.D. degrees in electronic engineering from the University of Science & Technology (UST), Deajeon, Korea, in 2011. His current research interests include high-voltage resonant converters and solid-state pulsed power modulators and their industrial applications. Yun-Sik Jin received the B.S. and M.S. degrees in uclear engineering from Seoul National University, Seoul, Korea, in 1986 and 1990, respectively, and the Ph.D. degree in electrical engineering from Nagasaki University, Nagasaki, Japan, in 1999. His doctoral dissertation dealt with the formation of carbon nitride thin film by hybrid process of plasma and ion beam. Since September 1990, he has been with Korea Electrotechnology Research Institute (KERI),

Changwon, Korea, where he is working on high-voltage engineering and plasma application. His research interests include high-power lasers, pulsed power technology, plasma applications, and radiation physics.

Jongsoo Kim received the B.S. degree in electrical engineering from the Seoul National University, Republic of Korea, in 1982, M.S. degree from the Kyungnam University in 1991, and the Ph.D. degree from the Kyungnam University in 1999. Since 1982, he has been with the Korea Electro-technology Research Institute as a researcher, and senior researcher and principal researcher of electric propulsion research center. His specialized research

area includes microprocessor application for industrial apparatus, and power converters based on power electronics.

Young-Bae Kim received the B.S. degree in electric engineering from Busan Industrial College, Busan, Korea, in 1978. He is currently working in the Electric propulsion Center, Korea Electrotechnology Research Institute (KERI), Changwon, Korea. His research interests include ozone generation, high voltage switch, and pulse power system for electromagnetic launcher.

Suk-Ho Ahn (S’11) received the B.S. degree in electrical engineering from Incheon National University, Incheon, Republic of Korea, in 2009 and is currently pursuing both his M.S. and Ph.D. degrees at the University of Science & Technology (UST) in Daejeon, Republic of Korea. His research interests include the soft switched resonant converter applications and battery charger systems.

Ji-woong Gong received the B.S. in electrical engineering from Chonnam National University, Gwangju, Republic of Korea, in 2012 and is currently pursuing the M.S. degree at the University of Science & Technology (UST) in Daejeon, Republic of Korea. His research interests include the soft switched resonant converter applications and High Voltage Pulsed Power Supply Systems.

Byungha Lee received the B.S. and M.S. degrees in electrical engineering from Chungnam National University, Korea, in 1994 and 1996, respectively, and is currently pursuing the Ph.D. degree in electrical engineering at Chungnam National University. Since 1996, he has been working as a Senior Researcher at Agency for Defense Development. His research interests are military pulsed power systems and electromagnetic launch technologies.

Duk-Heon Kim was born in Daegu, Korea, in 1964. He received the B.S., M.S., and Ph.D. degrees in electrical engineering from SungKyunkwan University in Seoul, Korea in 1990, 1992, and 2002, respectively. Since 1994, he has been an associate professor in the department of electric railway, college of catholic sangji, Korea. His research interests mainly focused on power system & applications. He is a member of the Korean Institute of Power Electronics (KIPE), and the Korean Institute of Electrical Engineers (KIEE).