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Cover image: A view inside a 60-W equivalet

LED bulb from 3M. Photo by Josh Jones

4632

CONTENTS

08 The energy efficiency scam

10 Driving LEDs with SiC MOSFETs Engineers increasingly see the logic of

silicon-carbide MOSFETs for handling a lot of electrical power, but new SiC devices also make sense for high-voltage switching.

18 Better converter efficiency with eGaN FETs

Gallium-nitride FET technology has improved markedly in the past five years. Power converters based on GaN FETs are living up to the promise of low power losses at an economical price.

24 Powering LEDs for illumination ICs that drive LEDs handle the complexities of illumination that include management of short circuits and squelching electrical noise.

28 Vehicle designs get comfortable with LEDs

The trend for the past decade in the automotive industry has been toward vehicular lighting that’s all solid-state. Incandescent headlights are the last hold-out, and they are on their way to the junk yard of history.

32 Efficiency in power conversion circuits A quick review shows the sources of inefficiencies in both linear and switching power supplies.

40 High interest in low-energy communications Advances in the Internet of Things depend on emerging wireless technologies that use little energy and fit in cramped quarters.

44 Teardown: 60-W-equivalent LED bulbs Surprise: A look inside five LED bulbs designed to replace 60-W incandescents reveals design regimes ranging from dead simple to startlingly sophisticated.

62 White Paper: Smart power for smart systems

64 White Paper: Digital inrush current controller

reference design

73 UPDATES: Power quality analyzer checks three-phase lines

— and doubles as BS detector for false energy efficiency claims

74 UPDATES: Getting pure sine waves from electronic inverters

10

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The energy efficiency scamLELAND TESCHLER

Executive Editor

DAN Carnovale sounds extremely pissed off. He says he has three gizmos on

his work bench right now that are all marketed as devices that can be hooked onto power lines to save energy. And, he claims, “All of them are garbage.”

Carnovale manages the Eaton Corp. Power Systems Experience Center in Pittsburgh. Bogus energy efficiency claims are one of his pet peeves. He says he sees a lot of equipment sold on the premise of saving energy that simply doesn’t work. Worse, he’s convinced many of the people putting these devices on the market know full well that their equipment is worthless.

“One airport I’m familiar with installed 50 of these things at about $4,000 a piece. By the time everything was installed, they’d spent about $1 million. But they didn’t save any energy,” he relates. Worse, the victims in situations like this one are often so embarrassed, he says, they typically bury the evidence rather than make a stink about being scammed.

Eaton’s PSEC produced a video depicting how a sales pitch for energy efficiency add-ins often goes: The video depicts a faux salesperson switching the add-on device into and out of ac lines powering an induction motor, which is typically spinning without driving any load. An ammeter is hooked into the motor circuit. With the energy gizmo in place, the meter reads out a current level half of that when the gizmo is not in the circuit. “That’s a 50% savings!” gushes the salesperson.

What’s not shown in the demo is that the kilowatts dissipated in the motor are the same in either case. The gizmo is actually a capacitor that adjusts the power factor phase relationship of the motor voltage and current. Induction motors running unloaded are close to a worst-case situation for power factor.

Indeed, one reason people frequently get bilked by false energy efficiency schemes is a lack of knowledge about the role power factor plays on ac lines. Any cost savings from power factor correction comes from avoiding a tacked-on fee from the utility, not from saving energy. So gizmos like the one described in Eaton’s video

would only save money if the utility added a penalty to the bill for excessive power factor. Often, says Carnovale, the facilities buying these quack energy efficiency remedies aren’t even on a rate schedule that penalizes them for excessive power factor.

“One hospital approached us for an opinion because they were getting ready to spend a lot of money on capacitors. It turned out the main reason for doing so seemed to be the passion of the salesperson selling the devices. There was no money to be saved because the hospital didn’t pay any power factor penalty,” he says. “It is clear to me a lot of the people selling these things are so excited about selling them that they are blind to the fact that their product really doesn’t work.”

But the days of questionable energy efficiency devices may be numbered. The IEEE has formed a working group that is defining tests for determining the effectiveness of add-on energy efficiency gear. IEEE P1889 aims to provide instructions for measuring electrical quantities that reveal how retrofitted energy saving devices (ESD) actually perform. This standard will focus on monitoring the power absorbed or generated with and without the ESD connected. There will be detailed protocols describing step-by-step the testing circuits, the type and accuracy of needed instrumentation, and a special emphasis on sources of measurement errors.

The result could be less money wasted on phoney energy savers, but fewer humorous war stories in the energy efficiency trenches. “The marketing material for one of these things said the inventor was inspired by God. It is comical but also scary to see what these guys come up with,” remarks Carnovale.


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Driving LEDs with SiC MOSFETsADAM BARKLEYSiC Power Device Application EngineerWolfspeed, a Cree Company

VIPINDAS PALA Research ScientistWolfspeed, a Cree Company

Engineers increasingly see the logic of silicon-carbide

MOSFETs for handling a lot of electrical power, but new

SiC devices also make sense for high-voltage switching.

T he benefits of silicon-carbide power MOSFETs are becoming widely understood among designers of power electronics. Compared to

IGBTs, SiC MOSFETs have a forward characteristic with no knee, which results in higher efficiencies when operating at a low fraction of full power. They also experience five to ten times less switching losses with no current tail during turn-off. Additionally, SiC MOSFETs have an internal body diode with low reverse recovery. These advantages have made SiC MOSFETs a means of realizing higher efficiencies in applications like industrial power supplies, PV inverters and auxiliary power supplies.

Most engineers understand the benefits of SiC MOSFETs compared to IGBTs but are less clear about their superiority over super-junction silicon MOSFETs for devices in the 600-to-900-V class. SiC can enable simpler

hard-switched and soft-switched topologies, reduced component counts and simplified over-voltage protection.

A specific case illustrates how SiC MOSFETs can make possible super-efficient and more energy dense power supplies. Researchers devised a single-stage flyback-based LED driver designed around a 900-V, 65 mΩ SiC MOSFET that is part of a recently developed family of devices called C3M. This single-stage flyback topology is widely used in low-power LED drivers (less than 100 W) as often found in tube LED lighting and residential LED lighting. For high-power LED drive applications over 100 W, two-stage topologies, like the boost PFC + LLC half-bridge topology, are preferred because they deliver a more attractive price-performance trade-off when using silicon switches.

The drawback of single-stage topologies for high-power LED drivers is that the switches see higher voltage

A 220-W, two-stage LED driver employing silicon MOSFETs (above) is about 40% bigger than an equivalent unit based on 900-V SiC MOSFETs (left). And the SiC version is more efficient.

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An important feature of 900-V SiC MOSFETs is a low temperature coefficient of on-resistance. On-resistance only increases by 1.4 times between 25 and 150° C, mainly because the MOS channel mobility, which limits on-resistance in SiC MOSFETs, rises at higher temperatures. Data for a similarly rated 650-V silicon super-junction MOSFET is also shown for comparison.

The C3M generation of SiC MOSFETs has good transconductance thanks to judicious engineering of the MOSFET top-cell structure. The devices hit full turn-on at VGS = 15 V at 25° C. For similar reasons, the device fully turns on even at 12 V at 150° C. In power supply applications, this eases drive requirements for the MOSFETs compared to previous generations of SiC MOSFETs that required 18-to-20-V gate bias for full turn-on.

and current stresses compared to two-stage topologies. These stresses are exacerbated when wide input and output voltage ranges are required. As a consequence of over-stress when using 650-V-rated switches, single-stage topologies have lower efficiency, narrower operating voltage range, more expensive EMI and surge-protection components, and high output current ripple (flicker).

Extra voltage headroom over 650-V devices makes 900-V silicon MOSFETs potential candidates for the single-stage topology. But 900-V super-junction devices exhibit excessive conduction and switching losses compared to C3M SiC MOSFETs. By combining the required blocking voltage with lower conduction loss, switching loss and device capacitances, C3M 900-V SiC MOSFETs let the single-stage flyback supply operate at higher power density and higher efficiency than the two-stage 650-V topology based on silicon.

The single-stage LED driver employing SiC MOSFETs takes up 40%

less volume and weighs 60% less than existing technology. It is also more efficient than the 650-V silicon two-stage topology and a 900-V silicon MOSFET single-stage topology.

It is interesting to compare a typical high-performance 220-W LED driver using silicon super-junction MOSFETs in a two-stage topology and SiC MOSFETs in a single-stage flyback topology. Both drivers have similar qualities in terms of power, input voltage range, efficiency, THD and power factor. Single-stage topology using an SiC MOSFET is also capable of meeting EMI Class B requirements and surge requirements to 4 kV, L-L while delivering acceptable output current ripple.

The single-stage design also costs 15% less, mainly through reductions in component count and magnetics. As a result, the cross-over point between single-stage and two-stage topologies rises from 75 to 100 W for silicon LED drivers, to 250 to 300 W for SiC MOSFET drivers.



In a typical UIS event, the avalanche voltage at high current conditions can be as high as 1.4 kV. The high avalanche voltage observed during UIS conditions comes from the high internal device temperatures (close to 600° C) reached during the UIS event. The typical avalanche energy to failure depends on the current at which the UIS event occurs. Notably, SiC devices still function after such an extreme event.

DRIVING LEDs(A

)



Typical blocking qualities of C3M 900-V, 65 mΩ MOSFETs at 25 and 150°C. The typical avalanche voltage is 1.1 kV at 25°C and 1.13 kV 150°C. Improvements in material quality and optimization of device structure have reduced internal fields in the transistor, which has resulted in extremely stable behavior at high temperature reverse bias (HTRB) conditions.

A comparison of switching losses in the C3M 900-V, 65 mΩ MOSFET chip in a TO-247 three-lead package and in a seven-lead D2PAK surface mount package with a Kelvin source connection. The chip has a 5-times-lower switching loss in a 7L-D2PAK compared to the TO-247, primarily because of the Kelvin source connection, which reduces the gate de-biasing that can arise because of the voltage drop across the parasitic source inductance in the package.

IMPROVED SiC DEVICESThis compact supply benefits from design improvements in the recently developed SiC MOSFETs dubbed C3M. Compared to earlier SiC MOSFETs, the C3M family of devices has a smaller cell pitch, optimized cell structure and optimized doping levels. The median specific on-resistance of the 900-V SiC MOSFET is 2.3 mΩ·cm2, a 42% reduction over the previous generation.

An important feature of 900-V SiC MOSFETs is the low temperature coefficient of on-resistance. On-resistance only rises by 1.4 times between 25 and 150° C. The major reason for this phenomenon is that the MOS channel mobility, which limits on-resistance in SiC MOSFETs, rises at higher temperatures. This effect partially compensates for the rise in drift resistance at high temperatures, thereby reducing the overall temperature coefficient of the device.

The C3M generation of SiC MOSFETs also has improved transconductance, realized by engineering the MOSFET top-cell structure. The device hits full turn-on at VGS = 15 V at 25° C. Transconductance is even better at 150° C, due to the improvement in channel mobility, so the device fully turns on even at 12 V. In power supply applications, this eases drive requirements compared to previous generations of SiC MOSFETs that needed 18-to-20-V gate bias for full turn-on.

The typical avalanche voltage is 1,100 V at 25° C and 1,130 V at 150° C. Improvements in material quality and optimization of device structure have reduced internal fields in the transistor, which has resulted in extremely stable behavior at high temperature reverse bias (HTRB) conditions.

When stressed at accelerated voltages close to breakdown (> 1 kV) at a junction temperature greater than 150° C, no device failures have been observed even after nine million device-hours. This indicates that C3M devices

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DRIVING LEDs

don’t see the typical failure modes, like gate oxide rupture, observed in earlier-generation SiC MOSFETs under accelerated HTRB conditions.

C3M 900-V MOSFETs also show extremely stable avalanche behavior. They’ve been fully qualified for avalanche current withstand capability under unclamped inductive switching (UIS). The avalanche voltage at high currents can be as high as 1.4 kV. The high avalanche voltage observed during UIS arises from the high internal device temperatures (close to 600° C) reached during the UIS event. It is impressive that SiC devices still function after such an extreme event.

Aside from driving lower semiconductor costs, low specific on-resistance of the C3M SiC MOSFET technology can lead to improved dynamic performance for two reasons. First, the smaller chip size reduces internal device capacitances. Consequently, there’s a reduction in switching losses for the same absolute on-resistance. Second, a smaller chip size fits in a smaller form-factor package for the same on-resistance, so the parasitic impedances outside the chip can be minimized.

In hard-switched applications, the turn-on and turn-off energy losses for the C3M SiC MOSFET are influenced by the package to a large degree. Consider the switching losses of the C3M 900-V, 65 mΩ MOSFET chip in a TO-247 three-lead package versus those of a seven-lead D2PAK surface mount package with a Kelvin source connection. In one case, the chip has a 5-times-lower switching loss in a 7L-D2PAK compared to the TO-247. This is primarily because of the Kelvin source connection in the 7L-D2PAK, which reduces the gate de-biasing that can arise from the voltage drop across the parasitic source inductance in the package.

C3M 900-V MOSFETs also have a rugged body diode. Improvements in material and process quality have resulted in the elimination of body diode degradation in SiC MOSFETs. Body diode reverse recovery for the C3M MOSFET is also extremely low. A low recovery body diode lets the device work in half-bridge topologies and in other applications where third-quadrant performance is important.

The specific on-resistance of C3M MOSFETs is approximately 30 times smaller than that of conventional 900-V silicon MOSFETs and four times smaller than the state-of-the-art, 650-V, super-junction MOSFETs. The low on-resistance gives the 900-V SiC MOSFET a big advantage over competing technologies in terms of switching performance.

In addition, it is noteworthy that at a junction temperature of 150° C, the on-resistance of silicon and GaN devices rise by a factor of 2 to 2.4 times, whereas the on-resistance of the SiC MOSFET increases only by

An LED driver based on SiC MOSFETs delivers a higher efficiency than both the 650-V, silicon-based, two-stage topology and a 900-V silicon MOSFET single-stage topology.

Body diode reverse recovery for the C3M MOSFET is extremely low, as is visible in the graph. A low recovery body diode lets the device be used in half-bridge topologies and other applications where third-quadrant performance is important.

(ISD

)

15DESIGN WORLD — EE Network 11 • 2015 powerelectronictips.com



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a factor of 1.4. In hard-switched topologies, switching losses are determined by on-off transition times of the switch, which depends on the gate drive. The figure of merit (FOM) that determines the transition time for a certain gate drive current is RDS,ON •QG. C3M SiC MOSFETs have 13 times lower gate-charge than 900-V silicon MOSFETs and 1.9 times lower than best-in-class, 650-V silicon MOSFETs.

In practice, gate drive speeds are limited by external gate resistors, which are needed to control oscillations that arise because

of parasitic impedances in the circuit. The extra voltage headroom afforded by the 900-V C3M SiC MOSFETs over 650-V silicon MOSFETs can be an advantage for power supply designers with regard to allowable voltage overshoots. The fundamental limit to switching loss, in the limit of infinitesimally fast gate transitions, is the energy stored in the output capacitance of the switch, EOSS. In terms of RDS,ON•EOSS FOM, C3M MOSFETs are comparable to 650-V silicon MOSFETs and are about four times better than 900-V silicon MOSFETs.

In half-bridge type topologies, the reverse recovery of the body diode is a big concern—and here the C3M SiC MOSFET completely outshines super-junction devices. The RDS,ON•QRR FOM for C3M MOSFETs is about 100 times better than that of 900-V super-junction MOSFETs and about 50 times better than the best-in-class, 650-V, super-junction device. In short, C3M body diodes are actually usable and, in many cases, eliminate the need for a separate anit-parallel diode.

In soft-switched or resonant topologies, the overlap switching

Comparing 220-W LED drivers650V Silicon Based 2-Stage Topology

C3M 900V SiC Based Single-Stage Topology

Input voltage Range 120-277V AC 120-277V AC

Output Voltage Range 150-210V DC 150-210V DC

Max Output Current 1.45 A 1.45 A

Peak Efficiency 93.5 % 94.4 %

Input THD < 20% < 20%

Output Current Ripple >0.95 >0.95

Output Current Ripple ±5 % ±10 %

Size 220×52×30 mm 140×50×30 mm

Weight 2.7 lbs / 1.3 kg 1.1 lbs / 0.5 kg

Relative cost 1 0.85

SiC MOSFET LED driver Silicon MOSFET LED driver

A single-stage flyback LED driver that uses 900-V SiC MOSFETs is much simpler than a conventional two-stage LED driver employing 650-V silicon MOSFETs.


DRIVING LEDs

Comparison of 650-V and 900-V SiC, Si and GaN FETs Technology RDSON,SP

(mΩ•cm2)RDS,ON•QG (mΩ•μC)

RDS,ON•EOSS (mΩ•mJ)

RDS,ON•QOSS (mΩ•μC)

RDS,ON•Qrr

(mΩ•mC)Avalanche Capable

900-V C3M SiC MOSFET 2.3 2.0 0.5 4.5 3.5 Yes

900-V Si Super-junction MOSFET A 75 26.5 2.1 31.2 1279.2 Yes

650-V Si Super-junction MOSFET A 10 3.8 0.5 25.1 300.5 Yes

650-V Si Super-junction MOSFET B Not Available 5.4 0.5 7.3 238.8 Yes

650-V Si Super-junction MOSFET C 24 11.3 0.9 18.6 473.9 Yes

600-V GaN HEMT Cascode A Not Available 1.1 0.6 5.8 3.3 No

losses are eliminated by use of a resonant tank circuit. Here, EOSS is not an important metric. Switching frequencies in these topologies are determined by the minimum transition time (dead-time), which is limited by the time needed for the drain current to charge or discharge the output capacitance of the switch. This is determined by the stored charge in the switch or QOSS. The RDS,ON•QOSS FOM for C3M MOSFETs is seven times better than for 900-V silicon MOSFETs and 1.6 times better than for 650-V silicon MOSFETs. This indicates that in resonant and soft-switched applications, C3M SiC MOSFETs will bring a substantial improvement to power supply efficiencies and will enable higher switching frequencies than competing technologies.

All in all, 900-V C3M SiC MOSFETs are suited to half-bridge and bi-polar current topologies because of their low recovery body diode, and they offer excellent overvoltage protection thanks to their avalanche ruggedness. The 900-V SiC MOSFETs make it feasible to use a single-stage flyback topology for over 100-W LED drivers and enable a 40% reduction in size at less cost compared to a two stage boost PFC+ LLC half-bridge design.

REFERENCESJ.W. Palmour, L. Cheng, V. Pala, et al. “Silicon Carbide power MOSFETs: Breakthrough performance from 900 V up to 15 kV.” Int. Symp. Power Semiconductor Devices and ICs, 2014, pp. 79-82.

Infineon Technologies. “New 900V class for superjuction devices.” Application Note, May 2008.

R. Singh. “Reliability and performance limitations in SiC power devices.”Microelectronics Reliability, vol. 46, 2006, pp.713-730.

A. Gaito, G. Ardita, L. Abbatelli and S. Primosole. “Impact of the MOSFET parasitic capacitances on the performances of the LLC topologies at light load.” PCIM Europe, 2013, pp. 1468-1475.

B. Hull, S. Allen, D. Gajewski, V. Pala, J. Richmond, S-H. Ryu, M. O’Loughlin, E. Van Brunt, L. Cheng, A. Burk, J. Casady, D. Grider and J. Palmour. “Reliability and stability of SiC power mosfets and next- generation SiC MOSFETs.” IEEE Workshop on Wide Bandgap Power Devices and Applications (WiPDA), 2014, pp. 139-142.

M. Treu, E. Vecino, M. Pippan, O. Haberlen, G. Curatola, G. Deboy, M. Kutschak and U. Kirchner. “The role of silicon, silicon carbide and gallium nitride in power electronics.” IEDM, 2012, pp. 147-150.

Each of the figures of merit shown in the table is the product of on-resistance at 25° C and a capacitance-related term. The specific on-resistance of C3M MOSFETs is about 30 times smaller than that of ordinary 900-V silicon MOSFETs and four times smaller than state-of-the-art, 650-V, super-junction MOSFETs. This leads to a large advantage for the 900-V SiC MOSFET over competing technologies in terms of switching performance.



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Better converter efficiency with eGaN FETsALEX LIDOW & DAVID REUSCH Efficient Power Conversion Corp.

Gallium-nitride FET technology

has improved markedly in the

past five years. Power converters

based on GaN FETs are living

up to the promise of low power

losses at an economical price.

A plot of source-drain current flowing through the FET “body diode” as a function of the source-to-drain voltage at both room temperature and 125° C shows the eGaN FET has a significantly higher forward drop than the Si MOSFET.

Enhancement-mode gallium-nitride (eGaN) FETs can switch at speeds ten times faster than the best available silicon transistors. This speed advantage is the reason GaN

transistors can boost power conversion efficiency and enable new, exciting applications that are simply beyond the reach of the venerable silicon power MOSFET.No wonder, then, that GaN power semiconductors are showing up in an expanding array of efficient power conversion applications. The technology is rapidly developing and product experience in the field is expanding.

GaN devices, grown on a low-cost silicon substrate, include high electron mobility transistors (HEMT) that significantly outperform the aging silicon MOSFET. Material benefits of GaN include a higher critical electric field strength, which allows the device terminals to be pulled closer together, and higher electron mobility, allowing the electrons to move with less friction between these shortened terminal connections. The improved electrical and thermal performance qualities of chip-scale-packaged eGaN FETs and ICs enable new levels of in-circuit performance, raising the bar for power conversion efficiency and economics.

eGaN FETs operate a lot like silicon power MOSFETs, but they have a few fundamental differences that require attention during circuit designs. The most important difference between them is that eGaN gate voltage is limited to a maximum of 6 V. To maximize eGaN FET performance, it’s best to drive the devices between 4 to 5 V. The lower maximum gate voltage makes it advisable to employ gate drive circuitry that can regulate the voltage to ensure safe operation and to minimize the parasitic inductance of the gate drive loop. In collaboration with Texas Instruments, a series of drivers have been developed to bring simplicity and reliability to meeting the challenges of driving eGaN FETs. This family of drivers allows designers to easily adopt eGaN FETs for use in most applications.

Beyond the introduction of GaN-specific gate drivers, the reliability of eGaN FETs has significantly improved since their introduction in 2009. Looking at gate reliability, there are several mechanisms that can contribute to failure in the field. A common means to measure reliability is to apply High Temperature Gate Bias (HTGB) stress at high gate voltage. These failure mechanisms include dielectric failure, gate sidewall rupture


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It’s best to drive eGaN FETs between 4 and 5 V, as illustrated on a plot of on-resistance versus gate voltage for various temperatures. Gate drive at this level keeps the FET operating with a minimum on-resistance.

As GaN matures, the devices are becoming robust. The mean time to failure for an eGaN FET with a 7-V gate bias, well beyond the normal safe operating range, is around 317 years at 150° C.

and a rise in off-state drain leakage resulting from gate stress.

To determine the voltage acceleration of HTGB failure, technicians conducted a matrix of tests between 6 and 7 V, all at 150° C. Note that this voltage range is outside the safe operating range of less than 6 V for eGaN FETs. The time-to-failure is determined from periodic parametric monitoring of the parts every two minutes. The failure criterion is a rise in the off-state drain or gate leakage beyond data sheet limits.

DEAD-TIME AND PARASITICSAs designers continue to boost frequency in power conversion, dead-time management becomes a critical factor in getting high efficiency and good regulation. For eGaN FETs with a higher forward

“diode” drop, this is especially true. The “body diode” mechanism is both better and worse than that of

a silicon MOSFET. An examination of the drain-source current flowing through the “body diode” as a function of the source-to-drain voltage at both room temperature and 125° C shows the eGaN FET has a significantly higher forward drop than the Si MOSFET. In low-voltage applications where the diode conduction period is large, an external Schottky diode has been shown to significantly improve GaN performance.

An offsetting advantage of the eGaN FET “body diode” mechanism, especially at higher frequencies, is that there is no recovered charge, Qrr. The eGaN FET turns off immediately when the voltage is removed from drain-to-source. In a silicon MOSFET, it takes several nanoseconds for all the charge to sweep out of the device, and the reverse recovery charge is a major high-frequency loss component.

Parasitic inductances make converters less efficient and generate unwanted voltage stresses. As GaN power devices continue to switch at higher speeds, the reduction of parasitic inductances must keep pace or designers will have to tradeoff switching speed for lower voltage stresses, sacrificing performance for reliable operation.

Traditionally, the major source of parasitics in a power device has been the device package. A conventional packaging technique for a vertical power MOSFET involves connecting the drain to the printed circuit board, generally with a large pad. The drain pad has a die-attach material that is used to connect the drain pad to the drain of the vertical MOSFET die. On the opposite side of the MOSFET are die-attach materials for the source and gate connections. Lastly, the source and gate connections have clips to allow connection to the circuit

PCB layout for optimal power loop with GaN transistors: top view, top view of inner layer 1, and side view. The gate drive loop, shown in red, and the high-frequency loop, shown in yellow, only interact directly next to the eGaN FET.


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BETTER CONVERTER EFFICIENCY

board. Each of these packaging steps adds resistance, inductance, size, thermal impedance and cost to the power device.

An advantage of high-voltage lateral eGaN FET transistors is that all the electrical connections reside on the same side of the die, eliminating the need for complex, high-parasitic two-sided packaging common to MOSFETs. With eGaN FET chip-scale land grid array (LGA) packages, the eGaN FET mounts directly to the PCB with drain, source and gate connections. This simple packaging technique reduces the resistance, inductance, size, thermal impedance and cost of the power device.

With the lower package parasitics inherent in eGaN FET chip-scale format, the printed circuit board (PCB) layout can become the limiting factor in converter performance. To minimize the common- source inductance added by PCB layout, the gate driver loop and high-frequency power loop should be located where they have little interaction.

Keeping these loops away from each other minimizes the common source inductance to the ultra-low internal eGaN FET package inductance. To reduce the high-frequency power-loop inductance contributed by the PCB, we’ve developed an optimal layout that uses the first inner layer as a high-frequency power-loop return path. Located between the two eGaN FETs is a series of vias, used to connect the top

layer to the inner layer return path, arranged to match the land grid array fingers of the synchronous rectifier (SR). This return path sits directly underneath the top layer’s power loop path, minimizing the physical loop size and providing magnetic field self-cancellation.

A side view of the optimal layout illustrates the concept of creating a low-profile magnetic field self-canceling loop in a multilayer PCB structure. Additional source vias are used on the lower side of the bottom transistor (SR) to further reduce low-frequency resistance and thermal resistance.

PARALLELING eGaN FETs The concept of paralleling devices is simple; the designer can use multiple smaller devices that appear and operate as a single, larger device. The on-resistance drops and the capacitances rise in proportion to the number of devices paralleled. In practice, as parasitic imbalance rises between the parallel devices, their ability to parallel worsens, limiting current-handling capability. Researchers have shown that parasitics must be both minimized and balanced for ultra-fast eGaN FETs operating in parallel to ensure good dynamic current sharing.

We present two designs in the interest of evaluating traditional paralleling layouts and proposing an improved parallel layout. In the first design, four GaN transistors sit in close proximity to operate as a “single”

power device, with a single high-frequency power loop. The drawbacks of this layout are larger parasitic inductances and greater parasitic imbalance between the parallel devices, leading to current sharing and thermal issues.

Exploded views of packages for Si MOSFETs (top) and eGaN FET chip-scale Land Grid Array (LGA) packages (bottom) illustrate that all the electrical connections for an eGaN FET sit on the same side of the die, allowing for the elimination of complex, high-parasitic, two-sided packaging necessary for MOSFETs.

How the performance of 12 Vin POL converters based on GaN transistors have evolved in the last five years.


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22 DESIGN WORLD — EE Network 11 • 2015 powerelectronictips.com1-800-269-6426 www.PioneerMagnetics.com

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At left, the four GaN transistors sit in close proximity to operate as a “single” power device, with a single high-frequency power loop. The result is larger parasitic inductances and greater parasitic imbalance between the parallel devices, leading to issues with current sharing and thermal performance. The design at right uses four distributed power loops, located symmetrically around a single gate driver. The design will provide the lowest overall parasitics for each device pair and most importantly, provide the best balancing of the parasitic elements, ensuring proper parallel operation.

The second design uses four distributed power loops, located symmetrically around a single gate driver. The design will provide the lowest overall parasitics for each device pair and most importantly, provide the best balancing of the parasitic elements, ensuring proper parallel operation.

A thermal imbalance is evident in the conventional paralleling design, where a hot spot develops on the devices handling a greater portion of the power as a result of parasitic inductance imbalance. The top switch closest to the input capacitors, T1, has a maximum temperature more than 10° C higher than the top switch furthest away from the input capacitors, T4.

For the proposed distributed power loop design, there is a good



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REFERENCESEfficient Power Conversion Corp. epc-co.com

Texas Instruments “Overview for GaN Solutions” www.ti.com/gan

A. Lidow, J. Strydom, M. de Rooij, D. Reusch. GaN Transistors for Efficient Power Conversion, Second Edition, Wiley, 2014.

D. Reusch and J. Glaser. DC-DC Converter Handbook: A Supplement to GaN Transistors for Efficient Power Conversion, First Edition, Power Conversion Publications, 2015.

A. Lidow, J. Strydom, M. de Rooij, and Y. Ma. GaN Transistors for Efficient Power Conversion, First Edition, Power Conversion Publications, 2012.

M. A. de Rooij. Wireless Power Handbook: A Supplement to GaN Transistors for Efficient Power Conversion, First Edition, Power Conversion Publications, 2015.

M. Briere. “GaN-based power devices offer game-changing potential in power-conversion electronics.” EE Times, 2008.

H. Umeda, Y. Kinoshita, S. Ujita, T. Morita, S. Tamura, M. Ishida and T. Ueda. “Highly Efficient Low-Voltage DC-DC Converter at 2-5 MHz with High Operating Current Using GaN Gate Injection Transistors.” International Exhibition and Conference for Power Electronics, Intelligent Motion, Renewable Energy, and Energy Management (PCIM Europe), pp. 1025-1032, 2014.

thermal balance and negligible difference in temperature between the devices. There is also a good distribution of the heat by distributing the higher-loss top devices on the PCB and not clustering them together.

For an example of the progress seen with GaN transistors in a short period of time, consider the application of eGaN FETs in 12 Vin POL converters. Five years ago, the performance of GaN transistors suffered from an insufficient understanding of the nuances involved with this new technology. New technological breakthroughs, such as monolithic integration, let GaN transistors again raise the bar for high-frequency power conversion performance.


Powering LEDs for illuminationTONY ARMSTRONG Linear Technology

ICs that drive LEDs handle the complexities of

illumination that include management of short circuits

and squelching electrical noise.

I t’s no secret that LED lighting consumes much less energy than alternative forms of illumination and that the cost of LED lights is dropping fast.

Consequently, the U.S. Dept. of Energy says LED lighting could account for 84% of lumen-hour sales in the general illumination market 15 years from now.

But look beneath the market projections and you’ll find that the general LED lighting market resembles the analog IC market insofar as it is fragmented. For example, LED sub-segments consist of replacement lamps, strips and strings, outdoor area, industrial, commercial, residential, consumer portable, entertainment, retail display, off-grid and safety/security.

General LED lighting is gaining traction in both commercial and residential applications. It represents

~57% of the total LED market of $25.9 billion that analysts project in three years. (The other segments are signage, automotive lighting, mobile devices, back lighting in displays, monitors and other.) Though many consumers still see LED lighting as too expensive, its long-term energy-savings and environment-friendliness, and its associated tax reductions, are expected to boost its use in commercial spaces such as parking lots, offices, factory facilities and warehouses. That’s because LED lights are attractive replacements not only for high-pressure sodium lamps, halogen lights and incandescent bulbs, but also for CFL and fluorescent lights in some areas.

Lighting generally represents 25 to 40% of total energy use in commercial buildings, so it is no surprise that commercial/industrial applications are leading the transition to LEDs. The long hours of high intensity light these applications require shorten the economic payback of LEDs. And the long life of LED fixtures dramatically reduces the replacement costs of lighting. These costs can be significant in such applications as high bay lighting where bulb replacement may entail renting a scissor or boom lift.

The primary driver behind the high growth of LED lighting is its dramatic reduction in power consumption over traditional light sources. LEDs marketed as incandescent equivalents give the same level of light (in lumens) as the bulbs they replace while consuming less than 20% of the electrical power. Other LED advantages include a lifetime that is orders of magnitude longer than incandescent bulbs. The ability to dim LEDs using existing triac dimmers is also a major benefit, especially in residential lighting.

LEDs turn on instantly and don’t experience the warm-up period associated with CFLs, nor are LEDs sensitive to power cycling like their CFL counterparts. Additionally, LED lighting fixtures do not contain any toxic materials, whereas CFLs contain small amounts of toxic mercury. Finally, LEDs enable new low-profile form factors that other technologies can not.

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EE_LTC_Nov_Vs4.indd 24 11/11/15 3:36 PM

POWERING ILLUMINATION


LT3795 Boost LED driver with input current limit and spread spectrum modulation.

While consumers can be fickle, there is clearly a point at which they will pay for an LED 60-W incandescent bulb replacement. Cree recently lowered the price of its 60-W LED equivalent to $7.97 in the U.S. This LED bulb is designed to last 25,000 hours and consumes 14% of the electrical energy consumed by its incandescent counterpart. Cree claims that by replacing a home’s five most frequently used incandescent bulbs with their LED equivalents, consumers can shave an average of about $60 annually from their electric bills.

Although LED replacement fixtures look relatively simple, they place tough requirements on the LED driver ICs. LEDs require a well-regulated constant-current source to deliver a constant level of light output, and powering them from an ac source entails special design techniques. Clearly there’s a great opportunity for makers of LED driver ICs in this growing market.

WHAT LEDs WANTLEDs are not “heaters” like incandescent bulbs, which electrically behave as resistors.

Because LEDs are diodes, they need drivers that provide a tightly regulated current and voltage. LED-based light sources typically employ a string of medium-power LEDs rather than one or two higher-power devices. For example, LED bulbs meant to replace 60-W incandescent bulbs usually contain a dozen or more LEDs in series. The use of LED strings makes drivers more susceptible to open or short circuits.

Furthermore, high temperatures degrade an LED’s useful light output, so thermal management is a big consideration in both the driver circuit and its housing. It can be tough to protect LEDs from thermal overstress. Often the LED sits in a small fixture with little opportunity for heat sinking. So most heat must be dissipated by conduction. This is also why an LED driver circuit that operates with high conversion efficiency is a great help since it produces less heat. State-of-the-art LED drivers now sport low-to-mid-90% efficiencies to help ensure good thermal design.

Of course, an LED only emits light when it conducts, so it needs a regulated dc voltage and current. When working


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Short-circuit current for the LT3795. The top trace shows the current flowing through the LED string. The middle trace is the current through the P-MOS FET. The bottom trace represents a transient short between rails. The LT3795 turns off the P-FET, which rises to a 12-A peak in just 500 nsec, thereby protecting the LED string from an over-current condition. Without this fast response, current could go as high as 50 A.

Comparing LEDs, CFLs and incandescent light sourcesProperty/Source LEDs CFLs IncandescentsEfficacy (lumens/Watt) 80 to 180

Future >20040 to 70 10 to 15

Watts consumed (60W bulb equiv.)

8-10 13-15 60

Lifespan (hours) >25K 2K to 10K 1K to 2K

Driver power DC AC Offline AC

Triac dimmable? Yes No Yes

Instant turn-on? Yes No Yes

Power factor 0.5 without PFC>0.90 with PFC

0.5 1

Sensitive to power cycling? No Yes Yes

Contains mercury? No Yes No

Failure modes None Yes, may catch on fire, smoke, or emit an odor

Some

Cost of 60-W (or equiv.) bulb $8 $3 $1

from ac mains, LEDs need ac-to-dc conversion. Regardless of the voltage source, catastrophic events can harm the LED driver; so it is helpful if the driver ICs have protection mechanisms built in.

Over-voltages, over-currents and many other factors can degrade LED operation. Because an LED is current-driven, supplying it with the right amount of current will let it attain its rated light output. This might seem like a simple task for a single LED; however, it’s more challenging when many LEDs are strung in series.

One example of an LED driver IC optimized for handling complexities that arise in LED strings is the LT3795, employing a boost topology. It also provides several other noteworthy operating features. The first is called spread-spectrum frequency modulation, a technique that dithers the system clock in a way that lowers the radiated noise. The LT3795, like all LED drivers, is basically a switching power supply whose switching frequency lies in the same band as AM radio. Thus, in the absence of spread-spectrum frequency modulation, it emits RF noise at the power supply switching frequency and its harmonics. Modulating the switching frequency using spread-spectrum techniques reduces the radiated energy peaks such that they don’t interfere with other electronics.


POWERING ILLUMINATION


The LT3797 configured as a triple-boost LED driver. Each of its three channels are boost controllers. A P-MOS FET and sense resistor are configured above each LED string. As with the LT3795, when a short circuit arises, the chip can quickly disconnect the LED string and protect it from damage.

The second feature is short-circuit protection, not easily implemented in a boost converter. Short circuit protection comes in handy when the LED string lies some distance away from the driver electronics. This scenario could arise in automotive wiring, where LED tail lights might be at the end of a wiring harness. A boost converter provides short-circuit protection through the addition of a disconnect FET and sense resistor in series with and above the LED string. The P-channel MOSFET and resistor serves as a way to monitor the current flowing through it and the LEDs in the string. (It can also serve as a means of dimming the LED string.)

In the event of a short circuit, the current flowing through the disconnect FET would rise quickly. This current can potentially harm the LEDs in the string. To stop this from happening, the driver IC senses the voltage rising across the sense resistor. The driver chip must then turn off the disconnect FET quickly. This is not easy; it must take place in less than one microsecond. The LT3795 is one of the few chips that can do it.

Another LED driver IC is the LT3797, a multi-topology triple-output LED driver. It has an integrated rail-to-rail current-sense amplifier with an output voltage range of 0 to 100 V. Each of its three channels can be configured for a buck, boost or SEPIC mode of operation, and each output can be operated autonomously from one another.

This IC incorporates other protection features, including short-circuit protection when a channel is operated in boost mode. Also implemented is open LED protection, which is afforded by a resistor network on the disconnect FET, which the chip monitors. If the voltage rises above a certain point on these sensing resistors, current through the LEDs has ceased to flow, signaling an open circuit.

The LT3797 is particularly suited to situations that demand LED dimming. Buck LED drivers give the highest PWM dimming ratios. But buck regulators need relatively high input voltages that might not be available in scenarios such as vehicular electrical systems. The electrical system voltage must first be boosted with a preregulator. The boosted output voltage can then be applied as input to buck-mode LED drivers. To get both the necessary voltage boost and LED drive, the LT3797 can be configured with one of its channels as the boost preregulator feeding the other two channels, which both act as buck-mode LED drivers.

REFERENCESLinear Technology linear.com


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Vehicle designs get comfortable with LEDsJOSHUA ISRAELSOHNJAS Technical Media

Grade-A-performingsystems,asdefinedbyISO7637-1,mustbefullyfunctionalwithsupplyvoltagesaslowas5Vtoaccommodatethecrankingpulsedepictedhere.

Thetrendforthepastdecadeintheautomotiveindustryhasbeentoward

vehicularlightingthat’sallsolid-state.Incandescentheadlightsarethe

lasthold-out,andtheyareontheirwaytothejunkyardofhistory.

LEDs bring a number of well-chronicled advantages over traditional tungsten-filament incandescent light sources. Their improved luminous efficacy

(lumens per watt) and energy savings are particularly helpful for hybrid-electric vehicles (HEVs) and electric vehicles (EVs) that place a high value on fuel economy. Vehicles built with traditional internal-combustion or diesel drivetrains are somewhat less sensitive to energy savings but still use LED lighting as both safety enhancements and differentiating elements of design style.

Incandescent bulbs are approximately 97% efficient —as heaters. Today’s white LEDs offer luminous efficacy about 5 times that of tungsten-filament bulbs, meaning they generate substantially less heat for a given light

output. So the light emitters are not only small, they also operate at a lower temperature. Ordinary thermal management methods for electronics let designers pack multiple LED elements closely together, without generating temperatures high enough to reduce operating lifetime.

Solid-state lights will operate for tens of thousands of hours. Additionally, repeated on-off cycles—common in several vehicular applications—won’t degrade LED operating life. LEDs also stand up better to shock and vibration than incandescent bulbs. Long life and ruggedness both play into the trend toward more reliable vehicles that need less maintenance. The flip side is that vehicle owners have fewer components they can service with just a typical home tool kit.

For example, the changing of a headlight on some vehicle designs entails partial disassembly of front body components. LED headlights that last the life of the car would eliminate such hassles.

Finally, LEDs bring important safety advantages when used as brake lights and center high-mount stop lights. University of Michigan Transportation Research Institute studies show the rapid turn-on time of LEDs lets following drivers brake faster by 170 to 200 msec (14 to 16 ft. at 55 mph) in good lighting. The advantage rises to 300 msec (24 ft at 55 mph) with bright sunlight or high-intensity reflections on the brake light surface.

All-LED automobiles are only just now starting to emerge. Earliest applications were typically for bright red LEDs and included tail, brake and marker lights. Next came backup lights and license-plate illuminators, which could tolerate the harsh color temperature and marginal color quality that characterized early-generation white LEDs. White LEDs now have better light quality and cost less, allowing uses in instrument clusters, cabin interiors, and, more recently, daytime running lights.

EE_ONsemi_Nov_Vs4.indd 28 11/11/15 3:41 PM

Simple, Easy Solutions

Follow us on:www.monolithicpower.com© 2015 Monolithic Power Systems, Inc. Patents Protected. All rights reserved.

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Average use, power and energy for automotive lighting devices by position

Headlights are the final frontier for LEDs in automotive lighting. This application lasts because it constitutes the highest power and most complex lighting in a vehicle. It is also the lighting position subject to the most stringent federal regulation.

SMART LED BALLASTSTungsten-filament bulbs model as resistors with positive temperature coefficients of resistance. They are inherently stable when operating from power sources that don’t exceed a certain voltage level. By contrast, LEDs are current-operated devices—ideally, electrons in/photons out. So they require an electronic interface to serve, at minimum, as a regulated current source. Consequently, LEDs can’t work directly from a vehicle electric power bus.

That means there’s a power supply between the LED and the vehicle electric power bus. The power supply constitutes an additional bit of complexity compared to ordinary incandescent lights. However, the additional complexity brings with it advances in power-management

ICs as well as new safety, performance, and convenience functions and enhancements. Indeed, LED headlights will offer features that are simply impractical with traditional lighting technologies.

One example is in vehicles carrying stop-start technology. Turning off the engine at stop lights can add 4 to 8% to fuel mileage in passenger vehicles simply by eliminating engine idle time. But consider that the light produced by incandescent bulbs follows their applied voltage to the 3.4 power. So tungsten-filament headlights can dim by 30% during a stop interval as the voltage on the power bus dips, assuming a fully charged battery in good condition. High starter motor current during cranking can also dim incandescent headlights by as much as 92% with a cranking pulse conforming to ISO7637-1.

In contrast, LED ballasts designed with buck-boost power regulators can track out the battery voltage and maintain constant light output during stop-start cycles.

Intelligent ballasts for LEDs can

provide diagnostic information not available in tungsten-filament lighting systems. If equipped with a temperature sensor thermally coupled to the LEDs, a smart driver can compensate for temperature effects. All LEDs have a negative temperature coefficient—light intensity drops as temperature rises and the change is non-linear. For example, an LED’s light output can fall by as much as 30% as its junction temperature rises from 25 to 150° C.

Ideally, ballasts for automotive LEDs should be able to drive lights for any spot on the vehicle. The ability to provide a programmable output current handles this need. It lets the same IC drive LED strings of varying lengths and output flux targets. This approach also lets automotive designers take a platform-wide approach to lighting drive design. The precise needs of individual vehicles then get met with model-specific software. Such designs also let automakers exploit advances in emitter design and manufacturing processes without the overhead costs of redesigning driver hardware.

Lighting Function

Average Usage Power/Lamp [W] Engergy Use [kl/mi]

[sec/mi] Mode [hrs/yr] Tungsten LED Tungsten LED

DRL 112.5 D 382 22.9 11.4 5.2 2.6

Low Beam 94.2 N 97.3 56.2 30.3 10.6 5.7

High Beam 9.5 N 9.8 63.9 34.4 1.2 0.7

Parking/Position 103.7 N 107.1 7.4 1.7 1.5 0.4

Turn Signal 10 24.9 53.6 13.8 1.1 0.3

Side Marker 103.7 N 107.1 9.6 3.4 2.0 0.7

Brake 18.2 80.7 26.5 5.6 1.0 0.2

Tail 103.7 N 107.1 7.2 1.4 1.5 0.3

CHMSL 18.2 80.7 17.7 3 0.6 0.1

Backup 0.9 3.8 17.7 5.2 0.0 0.0

Licence Plate 103.7 N 107.1 4.8 0.5 1.0 0.1

Modes: D: Daytime driving only Daytime Total: 7.9 3.2

N: Nighttime driving only Nighttime Total: 20.5 8.4

Data Source: University of Michigan, Transportation Research Institute


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Efficiency in power conversion circuitsSTEVE ROBERTSRecom

A quick review shows the sources of inefficiencies in

both linear and switching power supplies.

Most engineers know that one reason switch-mode power supplies have become widely used is their high energy efficiency. But many

engineers don’t know power supply technology well enough to explain the sources of switch-mode efficiency and inefficiency. So it is useful to review how ordinary linear supplies dissipate energy and why their energy efficiency is so low. It can also be helpful to understand where energy leaks out of switch-mode designs. Different components have different sources of energy loss. And different switch-mode designs lose energy in different ways.

First, consider typical regulators for linear power supplies. They usually consist of a transistor in series between the input and output voltages. This pass transistor is the regulating element, effectively functioning as a variable resistor. It limits the current flowing from input to output. A resistor divider delivers a voltage at the input of an error amplifier equal to that of a reference voltage. The error amplifier produces an output driving the pass transistor. The action is such that the voltage difference between the inputs of the error amplifier is always zero. In other words, the error amplifier responds to changes such as a higher load or a rising or falling input voltage.

The efficiency, η, of a linear regulator is defined by the ratio of the delivered output power POUT to the power consumption PIN:

η = POUT

PIN

From Ohm’s law, Pout = VoutX Iout; Pin = Vin

X Iin; Iin = Iout + IQ; where IQ is the quiescent current of the linear regulator when there is no load. The equation can be rewritten:

η = (VOUT IOUT)

VIN (IOUT+IQ)

Now consider an example of a typical 5-V, three-pin voltage regulator with an input voltage of 10 Vdc, output current of 1 A and a quiescent current of 5 mA. The efficiency calculation is then:

η = 5V x 1A

10V x 1.005A = 0.49

Thus, the overall efficiency is 49%. Note the power dissipation in the converter exceeds the 5 W delivered to the load. If the input voltage is lowered to the minimum of 7 Vdc, the efficiency rises to 70%, but this is the maximum practical efficiency as the regulator needs about 2 V of headroom for proper regulation.

It is immediately apparent from the efficiency equations that the efficiency of this type of regulator depends directly on the input voltage and load. And the energy efficiency is not constant. This also means the voltage regulator must be equipped with a heat sink large enough to allow safe operation under the worst-case conditions of maximum input voltage and maximum output current.

Now consider switching regulators. In contrast to linear regulators, which dump excess power as heat to limit the output voltage, switching regulators exploit the energy-storing properties of inductive and capacitive components to transfer power in discrete energy packets. The packets of energy are stored either in the magnetic field of an inductor or in the electric field of a capacitor. The switching controller ensures that only the energy actually required by the load is transferred in each packet. That’s why this topology is energy efficient.

The most common way to transfer the energy from input to output in a switching regulator is PWM (pulse width modulation), where a variable width pulse with a fixed time interval modulates the amount of energy

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Power IC Solutions for Today’sChanging Automotive Environment

Allegro MicroSystems offers a comprehensive portfolio of automotive-grade products which provide necessary power rails for LCD panels, bias power, LEDbacklights, MCUs, MPUs, GPUs, memory and interface power for infotainment systems.

Allegro’s solutions are extremely robust, handling wide ambient temperature ranges and input/output operating conditions. Design focus is applied to faultmode survival and recovery. It produces industry-leading packaging for enhanced thermal performance.

Allegro’s strong presence with automotive-qualified design, fabrication, assembly, and test locations adds to the high degree of reliability and performancequality.

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Power Management Solutions for Infotainment

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transferred from input to output. The duty ratio of the PWM, δ, is the ratio of on-time ton (the time during which energy is drawn from the source) to the period T (the inverse of the switching frequency ƒOSC):

δ = tON 1

T ƒOSC

, where T =

tON

T

For many switching regulators, the regulated output voltage is directly proportional to the duty cycle of the PWM. The control loop uses the “large-signal” duty cycle to control the power switching element. In contrast, the linear regulator uses the “small-signal” servo loop to limit the current through the pass transistor.

One reason PWM control is much more efficient than linear control is because the main losses occur during each change-of-state of the switch rather than continuously. FETs that are full-on or full-off dissipate little power.

Compared to linear regulators, the process of determining the efficiency of switching regulators is much more complicated. The linear regulator has easily predetermined dc losses; the largest dissipation takes place in the pass transistor. A switching regulator, however, has not only dc losses but also ac losses that arise in the switches and in the components for energy storage.

For example, the total loss of a switch is made up of not just the loss in the on and off states, but also the losses in the transition from switching on and off. In the case of a transformer, the total loss is calculated from the sum of ac (core), ac (winding) and dc (winding) losses. The losses in the core of a transformer are caused mainly by the interaction between the magnetic flux and the core material (hysteresis losses, eddy current losses). The winding losses result

mainly from the material of the transformer winding (ohmic losses, skin effects). Either way, the net effect is a rise of temperature in the transformer.

To calculate the efficiency of a dc/dc converter, the losses of each part of the conversion cycle must be found by averaging the losses over the whole range of the PWM duty cycle. The losses in the magnetic, inductive and capacitive components can be controlled and minimized to realize a high conversion efficiency. Typically only about 4% of the input power is lost and converted into heat.

Non-isolated converters are generally more efficient than their isolated counterparts because fewer parts are involved in the power conversion; non-isolated converters don’t use transformers, so there are no transformer losses. Yet despite a higher degree of complexity, isolated dc/dc converter efficiencies of more than 85% can be realized, depending on the power rating.

One of the major causes of efficiency loss in switch-mode circuits are the output diodes. If the output current is 1 A and the forward voltage drop across the diode is 0.6 V, then 600 mW will be lost in the diode alone. Thus, high output current dc/dc converters often use FETs with synchronous switching to reduce rectification losses.

It may be surprising to learn that lower-power converters generally have lower efficiencies than higher-power converters, especially considering the higher I²R losses that arise at higher output currents. However, the internal power consumption of the switching controllers, shunt regulators and optocouplers (the “housekeeping” consumption) plays a significant role. If the total housekeeping demand is 1 W, then a 10-W converter can not have an efficiency exceeding 90%. But the maximum possible efficiency of a 100-W converter would be 99%.

Housekeeping losses also explain why all dc/dc converters have 0% efficiency under no-load conditions, as the converters still consume power but deliver no output

Typical regulators for linear power supplies usually consist of a transistor in series between the input and output voltages. This pass transistor is the regulating element, effectively functioning as a variable resistor. It limits the current flowing from input to output. A resistor divider delivers a voltage at the input of an error amplifier equal to that of a reference voltage. The error amplifier produces an output driving the pass transistor.

EFFICIENCY IN POWER CONVERSION CIRCUITS


Acopian_EE_ad.indd 35 11/11/15 5:29 PM

Parasitic elements associated with a transformer include interwinding coupling capacitances CWA and CWB, primary and secondary winding capacitance Cs and Cp (usually insignificant except with high-frequency designs), the magnetizing inductance of the core LM, and the leakage inductances LLP and LLS. Leakage inductances are especially troublesome, reducing efficiency and generating radiated EMI.

Switching supplies may incorporate snubber circuits to absorb some of the energy in voltage spikes generated during switching action, reducing the overvoltage stress on the switch and diode. However, a snubber cannot eliminate the power loss caused by the spikes. The snubber network resistors dissipate the power otherwise dissipated in the switch or rectifier diode.

Critical components, such as switching and rectifying elements, magnetic components and filter capacitors, all affect both the switching frequency and also the overall efficiency of the converter. Inductor losses depend greatly on the choice of core material and have operational losses arising from I2R dissipation in the winding and coupling capacitances between the turns.

power. FETs consume more power when switching than in a steady on or off state. This is because their internal gate capacitance must be charged and discharged to switch the output. Peak gate currents of 2 A or more are not unusual. A dc/dc converter running with no load will still be switching the FETs hundreds of thousands of times per second, so it is not unusual for a dc/dc converter to still run warm without any load.

FACTORING IN PARASITICSThe level of energy efficiency among switch-mode converter topologies varies. One reason is that the components they use are non ideal. Textbook descriptions of converter topologies assume ideal components and ignore the parasitic effects. It is, however, a fact of life that inductors have capacitive and resistive elements and vice versa. The choices of components used in a switching supply therefore have a large influence on its performance. Critical components, such as switching and rectifying

elements, magnetic components and filter capacitors, all affect both the switching frequency and also the overall efficiency of the converter.

In particular, semiconductor switches have many non-ideal properties. FETs place high peak current demands on the driving circuit, especially the current needed to charge and discharge the parasitic Miller capacitance between gate and drain. Diodes have a parallel equivalent capacitance that slows their switching speed and, of course, the internal forward voltage drop. Inductor losses depend greatly on the choice of core material and have operational losses arising from I²R dissipation in the winding and coupling capacitances between the turns. Capacitors have parasitic effects such as equivalent series resistance (ESR) and equivalent series inductance (ESL). All these effects depend on frequency, so an inductor can behave as a capacitor at high frequencies, just as a capacitor can behave as an inductor.

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One of the biggest causes of efficiency loss in any converter is the power dissipation in the output diodes. A big leap forward in efficiency improvement has been the development of synchronous rectification. Here FETs serve as rectifying elements and are switched on during the forward part of the cycle and off during the reverse part of the cycle. The disadvantage of these devices is that they must be actively driven, so there are additional timing and drive circuits required.

One major cause of inefficiency in switch-mode circuits is the power loss associated with the output diodes. An alternative would be to replace the diode with an FET switched on with an out-of-phase signal to the PWM signal. The RDS,ON of a FET is low and does not have the forward voltage drop of a diode. The increase in efficiency can be significant under full-load conditions as the power normally dissipated in the catch diode can be reduced by as much as a factor of four in a typical medium-power 15-W synchronous converter. However, at low load (<10% full load), the synchronous design can actually be less efficient than the asynchronous design partly because of additional losses in the low-side FET switching circuit, which also dissipates power charging and discharging the low-side FET gate capacitance. Another reason is that in an asynchronous design, the inductor current is blocked from flowing backward by the diode, but in the synchronous design, both positive and negative inductor currents can flow. Any negative current represents an additional power loss that the asynchronous circuit does not see.



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The efficiency of a switched converter can be different for different input voltages and for different loads. It is normally expressed as the ratio of output power to input power. At zero load, the efficiency is always zero. A specification in percent is common, but it can also be given as a normalized number (≤ 1). Normally, the data is provided under several conditions, such as nominal input voltage and full load.

The advantage of a flyback transformer design is that the output voltage multiplication can be extremely high with short duty cycles. The disadvantage is that the transformer’s air-gapped core should not saturate though there is an average positive dc current flowing through the transformer. So efficiency can be lost if it has a large magnetic hysteresis. Also, eddy current losses in the windings can be a problem due to high peak currents.

the core, and leakage inductances for both the primary and secondary.

These transformer parasitic effects strongly influence the converter performance. Coupling capacitance causes common-mode EMC problems. Core saturation caused by magnetizing inductance limits the transformer current. Leakage inductances are especially troublesome, reducing efficiency and generating radiated EMI.

Leakage inductances are also responsible for the voltage spikes that arise whenever the current changes rapidly in the windings. Such overvoltages stress the primary switch and secondary diodes, so they must either be sized to withstand the peak voltage or fitted with a parallel snubber network to dissipate the energy in the spikes.

However, the energy in the spikes and the power that the snubber has to absorb constitute an energy loss that diminishes the efficiency of the converter. The energy in the spikes and the power that the snubber must absorb can be calculated according to:

12

LLEAK I2LEAKE =

12

LLEAK I2LEAK ƒP =

A snubber cannot eliminate the power loss caused by the spikes. The power that would otherwise be dissipated in the switch or rectifier diode is now dissipated by snubber network resistors instead.

Besides the spikes caused by the parasitic leakage inductance, any coupled reactive system will also exhibit resonant frequencies. Most transformer-based designs try to either reduce these parasitic elements to a minimum or choose operating frequencies where resonance is not an issue. However, a quasi-resonant or resonant converter design deliberately encourages resonance by increasing the winding inductance or by adding additional inductors because controlling this resonance can facilitate an efficient converter design.

As has been mentioned before, one big source of efficiency loss in any converter is the power dissipation in the output diodes. Low forward-voltage-drop Schottky diodes can sometimes serve as an alternative for low-power converters, but they are expensive when sized to cope with higher currents. Even so, the forward drop is around 200 mV, so the power loss can still be significant.

A big leap forward in efficiency improvement has been the development of synchronous rectification.

In a typical circuit with diode rectification, one diode acts as a rectifier and another is a freewheeling diode.

Transformers have similar issues. The disadvantage of using a transformer is that the energy transfer from primary winding to secondary winding involves additional losses. So while a buck regulator can reach 97% conversion efficiency, transformer-based converters struggle to exceed 90%.

PARASITIC ELEMENTS Parasitic effects in transformers include interwinding coupling capacitances for both the primary and secondary windings, a magnetizing inductance of

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Both diodes are alternately loaded with approximately the same current. The losses from the forward voltage drop in the diodes is just the voltage drop times the diode current. With a typical forward voltage of 0.5 V, a relative power dissipation of 0.5 W per amp can be assumed. A 3.3 V/10 A output converter would therefore have a voltage conversion loss of 15% without considering any other conversion losses. The power dissipated in the diode would be 5 W, so the diode would probably have to be heat-sink-mounted to have any useful operating temperature range.

Fortunately, FETs can be used as rectifying elements by switching them

on during the forward part of the cycle and turning them off during the reverse part of the cycle. Their advantage as fast switches with low on-resistance makes them suitable as rectifiers.

The disadvantage of FETs is that they must be actively driven, so there are additional timing and drive circuits required. Synchronous rectifiers need to sense the internal voltages to correctly turn on and off the two FETs synchronously with the output waveform, hence the name of this topology.

In comparison, diodes are passive devices that need no extra circuitry to function, but the low on-resistance of FETs of about 10 mΩ more than offsets

the disadvantage of the more complex circuitry for high output current converters.

Finally, a word about calculating efficiency: The efficiency of a voltage conversion is given by the ratio of output power to input power. At zero load, the efficiency is always zero. A specification in percent is common, but it can also be given as a normalized number (≤ 1). Normally, the data is provided under several conditions, such as nominal input voltage and full load.

REFERENCESRecom Electronic recom-power.com

The advantage of various switching topologies becomes evident in a simple listing of efficiencies. The combination of a buck converter and linear converter may be used to overcome the fact that the PWM regulator feedback circuit requires a minimum output ripple to regulate properly, as the regulation is typically cycle-by-cycle.

Typical converter efficiencies

Linear regulator 49%

Buck converter >97%

Buck converter + linear converter 76%

Resonant-mode converter >95%


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High interest in low-energy communications

Advances in the Internet of Things

depend on emerging wireless

technologies that use little energy

and fit in cramped quarters.

I t looks as though the success of wearable technology such as smart watches and fitness bands could hinge on low-energy versions of wireless communication technology. The low-

energy technology getting the most attention is Bluetooth 4.1 low energy (BLE) specification, which is being marketed as Bluetooth Smart. This is a less power-hungry version of Bluetooth and was first adopted in 2010.

One reason BLE reduces energy consumption is that it eliminates issues related to continuous battery use and constant pairing and re-pairing, which were problems in the earlier versions. Also, BLE devices use less power because they stay in idle mode except when transferring critical data. To keep down energy consumption in devices with small batteries, BLE-equipped notebooks, smartphones, tablets and other consumer electronic devices act as primary devices to send and receive data from secondary BLE devices, such as heart rate monitors, smart watches or any other wearable products.

BLE also makes possible the use of what’s called beacon technology. The industry analyst firm ABI Research says over the last twelve months, the use of iBeacons/BLE beacons really took off among retailers. The firm predicts BLE beacon shipments over the next five years could climb to a point where 60 million BLE units get sold in 2019. Another analyst firm, IHS, calls 2015 the year indoor positioning takes off. Though indoor positioning is possible with WiFi, Bluetooth beacons offer better accuracy. With Bluetooth beacons, you can navigate not just to a store, but also to an aisle or even a specific product.

Bluetooth Smart semiconductors shipped briskly in 2014. Analyst estimates put the global BLE and Smart Ready market at $5,572 million and shipments of as much as 2.7 billion units by 2020, up from just 49 million units in 2013. Expectations are for a compound annual growth rate of 9.38% between 2014 and 2020. Analysts also see BLE taking the lead in annual wireless sensor network (WSN) shipments. The analyst firm ON World projects that by 2018, IEEE 802.15.4, BLE and WiFi will make up more than 80% of the WSN unit shipments.

Hardware vendors are responding to the need for reduced power consumption with innovative ideas on several fronts. In BLE ICs, recent developments include the IS1870 and IS1871 BLE

Development facilities for the Microchip BM70 BLE module come in the form of a daughter board containing the chip.

EE_TDK_Nov_Vs4.indd 40 11/11/15 4:26 PM

HIGH INTEREST IN LOW-ENERGY


The Microchip BM70 BLE module block diagram. Module options include RF regulatory certifications, or non-certified (unshielded/antenna-less) for smaller and more remote antenna designs that will undergo end-product emission certifications.

IoT devices implementing wireless charging might get juice through a pick-up coil like this unit from TDK-Lambda. The modules are just 1.0 mm thick. Three small Rx coils target wearables; their dimensions are as small as 22 × 12 mm and the coils are designed for a maximum output of 2 W.

RF ICs, and the BM70 module from Microchip Technology. The power profile of these devices minimizes current consumption, and form factors are as small as 4 × 4 mm for the RF ICs and 15 × 12 mm for the module. The module options include RF regulatory

certifications, or non-certified (unshielded/antenna-less) for smaller and more remote antenna designs that will undergo end-product emission certifications.

To handle beacon applications, the module supports stand-alone “hostless” operation—in other words, it boots from a Flash memory so there’s no need for an external host processor to download code or run a driver. Microchip’s BLE devices include an integrated, certified Bluetooth 4.2 firmware stack that facilitates communications at up to 2.5 times faster data transfer speeds and connection security, with government-grade (FIPS-based) secure connection support.

On another front, power supply maker TDK-Lambda Americas has devised a number of components with low-power IoT and BLE applications in mind. These devices include RF components and modules based on surface acoustic wave (SAW), bulk acoustic wave (BAW) and SESUB (semiconductor embedded in substrate) technology, tiny and flat multilayer inductors, a variety of advanced sensor technologies, and miniaturized ultra-flat coils for wireless charging.

Consider one particular BLE module. With its miniature footprint of just 3.5 × 3.5 mm and slim insertion height of 1.0 mm, the SESUB-PAN-D14580 is billed as the world’s smallest BLE module. It is nearly 65% smaller than modules that employ discrete components and consumes about a quarter of the power needed by classic Bluetooth devices. It works from 3-V supply voltage and consumes just 5.0 mA when transmitting, 5.4 mA when receiving and 0.8 µA in standby mode.

Based on TDK SESUB integration technology, the module incorporates a DA14580 Bluetooth 4.1 chip from Dialog Semiconductor. Its substrate layers optimally route all the I/Os to a BGA on the module’s bottom surface, to better let designers access the integrated Bluetooth IC. The module makes available all terminals of the discrete chip. The quartz oscillator, capacitors and various other peripheral components mount on the same substrate. The miniature size and low current consumption lets the Bluetooth module handle battery-powered wearable applications where small size, light weight and low power consumption are essential. Mass production started in July 2015.

It isn’t just communication chips that are getting attention. Power supply makers are designing low-power components specifically for wearable devices such as smart watches. An example is the TDK µdc-dc converters, which have a relatively small 2.9 × 2.3-mm footprint with an insertion height of 1 mm. The integrated power modules occupy 65% of the space taken up by conventional discrete components. They have a 92% power efficiency and under light loads, the modules go into a power-save mode using pulse frequency modulation with a typical quiescent current of 24 µA.

It also looks as though a lot of low-power communication electronics will get juice through some kind of wireless charging scheme. A complicating factor is that there is no such thing as a wireless charging standard. Several technologies are in use,


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HIGH INTEREST IN LOW-ENERGY

TDK µdc-dc converters fit in a 2.9 × 2.3-mm footprint with an insertion height of 1 mm. The integrated power modules occupy 65% of the space taken up by conventional discrete components.

including magnetic inductance, magnetic resonance, radio wave charging, microwave charging and laser beam charging.

Magnetic inductance uses an electromagnetic field to transfer power. In magnetic resonance charging, wireless energy transmission between transmitter and receiver is done through tank circuits tuned to same frequency. Radio wave charging uses radio waves to transmit power to receivers at distances ranging as far as 30 ft.

Currently, about 90% of wireless charging takes place through magnetic inductance. Two organizations are trying to develop harmonized standards for wireless charging. The Wireless Power Consortium (WPC) is involved in creating standards specifically for magnetic inductance. The other organization is a recent merger of the Power Matters Alliance (PMA) and the Alliance for Wireless Power

(A4WP)—a new name for the merged organizations has yet to be determined. One goal of the hook-up is to promote interoperability among the various wireless charging schemes.

Wireless charging takes place in what can only be described as cramped quarters; in most wearables, there is little room for charging components such as inductive pick-up coils. But the level of signal strength is proportional to the number of coil turns and to the third power of loop radius. So manufacturers have developed special inductive coils optimized for both small spaces and maximizing signal levels.

For example, Tx coils from TDK handle WPC low-power (≤ 5 W) specifications that include single primary coils with magnetic alignment (specifications A1 and A9) and without magnets (specifications A10 and A11). A linear array with three coils is also available for chargers that allow free positioning with an array. All Tx coils use WPC-approved ferrite sheets. Extremely thin flexible sheets are available.

TDK Rx coils come in thicknesses from 0.50 to 1.12 mm as a way of meeting the varying requirements for wireless charging applications. All Rx coils are designed with magnetic attractor materials to support magnetic alignment. The lineup also includes Rx coils with a combined antenna for near field communications (NFC). Rx modules include an Rx coil with attractor and control unit. The modules feature a thin maximum thickness of just 1.0 mm. Three small Rx coils target wearables; their dimensions are as small as 22 × 12 mm and the coils are designed for a maximum output of 2 W.

REFERENCESMicrochip Technology microchip.com

The SESUB-PAN-D14580 is billed as the world’s smallest BLE module. It is nearly 65% smaller than modules that employ discrete components and consumes about a quarter of the power needed by classic Bluetooth devices.

TDK-Lambdatdk-lambda.com



Hirose_EE_ad.indd 43 11/11/15 5:31 PM

Teardown: 60-W-equivalent LED bulbsLELAND TESCHLERExecutive Editor

Surprise: A look inside five LED bulbs designed to replace 60-W

incandescents reveals design regimes ranging from dead simple to

startlingly sophisticated.

T he average consumer might think that when it comes to light bulbs, one is about the same as another. This view might have been accurate

back when every light socket contained an incandescent lamp. It is certainly not true for the LED bulbs designed as incandescent replacements.

We came to this conclusion after tearing down five LED bulbs marketed as equivalents for 60-W incandescent bulbs. The five bulbs we chose all got high marks from Consumer Reports Magazine. But that’s where the commonality stopped. When we got inside, we found vastly different approaches in construction techniques, thermal management and electronics design.

We start with a bulb called the E27 A19 LED from Home EVER Inc. in Las Vegas. The mechanics of the bulb

and its electronics are dead simple. The two-sided circuit board seems to have been reflow soldered. Two wires connect the board to a metal plate holding 30 LEDs. Two more wires go to the light socket conductors. All four of these wires look as though they were hand-soldered.

The bulb is built around a 2-in.-high heat sink that weighs 2 oz and looks to be a metal casting. The base of the lamp contains a plastic housing that holds the ac/dc converter. The electrical connections to the lamp socket are at one end of the housing. The other end attaches to the heat sink with two small screws.

Additional attachments to the heat sink are a frosted polycarbonate bulb that encloses the LEDs and a 2-in.-diameter metal plate containing the LEDs. The plastic bulb apparently snap-fits into the heat sink while the

The plastic enclosure for the Home EVER ac/dc converter slid out of the heat sink bottom. The converter board (right) sits in the plastic enclosure.


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EE_Teardown_Nov_Vs5.indd 44 11/12/15 12:18 PM

LED plate attaches with three screws. There’s a couple spots of compound for thermal conduction applied between the LED plate and the heat sink.

The ac/dc converter design is straightforward. The only non-SMD components are two big capacitors, a surge resistor on the input and a transformer. Connections from the board to the screw-base and to the LED board are through discrete wires, but the connection to the bulb foot contact was done by machine. The electrical connection to the metal screw threads, though, is simply a length of bare wire squeezed between the plastic housing and the inside surface of the screw threads.

The electronics on the ac/dc converter are bare bones. The diode bridge on the input is four discrete diodes. There is a single IC on the board. It is a buck topology supply designed to provide a constant current and is made by Bright Power Semiconductor (BPS) in China. The chip, dubbed BP2812, incorporates a 600-V MOSFET. The spec sheet lists the chip operating current at 200 µA.

The “typical application circuit” listed on the BP2812 spec sheet comes extremely close to the actual circuit we found on the LED’s circuit board. Seven resistors go into simple networks that handle the Vcc voltage, sensing the buck inductor’s peak current, and regulating the input voltage to the IC. Five capacitors handle chores of ac line filtering, an ac by-pass for the Vcc pin and line-sense pins, and the buck topology. An in-line fuse cuts the power to the whole circuit in the event of too-high current draw.

Judging by graphics on the BPS web site, it looks as though BPS may have assembled the board itself. There are images there of example boards for a few other LED applications that look remarkably similar to this one.

It should be noted that the effect of temperature on LED operation doesn’t seem to be factored into the ac/dc converter. LEDs put out less light as their temperature rises. That’s generally not a problem for small temperature changes. The eye’s sensitivity to light is logarithmic, and the eye is not particularly sensitive to small changes in luminosity. It’s not unusual to see a 10% drop in LED lumen output as junction temperature rises from room temperature to 150° C.

But LED current can also be reduced at higher temperatures as a way of lessening the need for heat sinking. That said, there is no temperature sensing that we could see in the Home EVER bulb’s ac/dc converter. And there is no circuitry for dimming.

But all in all, the LED bulb probably works well in uses that don’t need a dimmable light.

The chip powering the Home EVER LED bulb is basically a constant current source powering an onboard MOSFET. The reference circuit from chip-maker Bright Power Semiconductor is close to what we found on the PCB.

The Home EVER bulb’s heat sink and plastic base holding the ac/dc converter with the metal screw threads removed. Here, the base foot connection is still wired to the converter.


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A view of the Home EVER PCB reveals the four diodes making up the rectifying bridge and the BP2812 chip (bottom). The other side of the board (top) holds the energy handling components and the fusistor on the input.

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The potting material surrounding the Osram bulb’s ac/dc converter board and the plastic case from which it was extracted.

A view of the Osram LED bulb with its plastic globe cut off revealing the pentagon-shaped tower holding the LEDs. The wires from the ac/dc converter board can be seen soldered to the top plate.

The SSL21082AT reference circuit seems to be close to what we found on the Osram PCB. The chip has an input for a NTC resistor, but we didn’t spot one on either the PCB or the metal plates to which the LEDs mount.

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OSRAMOsram Sylvania’s 60-W equivalent LED bulb is notable in that it has a relatively small, two-piece heat sink. One piece is a 1-in.-high pentagon-shaped tower that doubles as a backing for six LED boards, five oriented in a pentagon shape with the sixth sitting atop the pentagon tower. The other is a 0.75-in.-long cylindrical cast heat sink that apparently snap-fits to the upper part of the plastic dome housing the LEDs. The cylindrical cast heat sink and tower together weigh 1.3 oz.

The base of the unit is a one-piece plastic housing that holds the ac/dc converter circuit board. Two wires connect it to the pentagon-shaped tower holding 18 LEDs, three on each face. The connections between the boards appear to have been reflow soldered. But the discrete wires between the circuit board and the LED assembly appear to have been hand soldered. Similarly, connections to the bulb base are discrete wires with one squeezed between the metal screw threads, the other a machine assembled to the bulb foot.

For reasons that are not completely clear, the designers of the Osram bulb chose to pot the ac/dc converter board. The relatively small heat sink in this board, compared to other designs we’ve seen, might indicate the potting is meant to improve thermal dissipation, though potting material doesn’t completely fill the empty space between the electronic components and the external shell. The potting did, however, complicate the process of deciphering the circuit.

The main board for the Osram LED bulb is two sided. It contains two ICs, one a diode bridge for the ac input, the other an SSL21082AT driver IC from NXP Semiconductors. Features implemented on the NXP chip include dimming, over-temperature protection and LED over-temperature control, output short protection, and a restart mode in the event of a brown out. This IC has an integrated internal HV switch and work as a boundary conduction mode (BCM) buck converter.

BCM is a quasi-resonant technique used to enhance energy efficiency. The fundamental idea of BCM is that the inductor current starts from zero in each switching period. When the power transistor of the boost converter is turned on for a fixed time, the peak inductor current is proportional to the input voltage. The current waveform is triangular; so the average value in each switching period is proportional to the input voltage.

Energy stores in the inductor while the switch is on. The inductor current is zero when the MOSFET is on. The amplitude of the current build-up in the inductor is proportional to the voltage drop over the inductor and the time that the MOSFET switch is on. When the MOSFET is switched off, the energy in the inductor releases toward the output. The LED current depends on the peak current through the inductor and on the dimmer angle. A new cycle starts once the inductor current is zero.

The main heat sink for the Osram LED bulb is a cylindrical casting, shown here in four pieces after removal from the body of the bulb. The metal screw threads attach to the plastic case holding the ac/dc converter board, visible here.


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Once the potting material was removed from the Osram bulb’s PCB, the SSL21082AT driver IC from NXP Semiconductors became visible on the PCB. The other IC on the board is a bridge rectifier. Energy handling capacitors and inductors are mounted to the other side of the board.

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3MThe 3M LED has a distinctive look thanks to the 2-in.-high white cylindrical column visible under its semitransparent plastic dome. The column is just a metal heat sink; it apparently has nothing to do with the dispersion of light.

The LEDs sit on a flex circuit board attached to another 2-in.-high heat sink that also forms a support for the base of the bulb. A plastic sleeve goes on the bottom of the heat sink to hold the metal screw threads and support the foot contact at the bottom of the base. The heat sink

The 3M LED bulb with the plastic globe removed. The white column is a heat sink and has little effect on the light output. The LEDs are positioned around the rim of the plastic bulb in the metal heat sink.


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plus column together weigh in at 2.4 oz.The flex circuit board holding the

LEDs also holds the ac/dc driver circuit. It is a CL8800 from Microchip Technology. The reference design consists of the CL8800, six resistors and a bridge rectifier (a Fairchild device). Two to four additional components are optional for various levels of transient protection. Microchip’s reference design is quite close to what we found in the 3M bulb.

The driver circuit divides a string of 25 LEDs into two sets of five, one set of four, and one set of six. We’re not sure why 3M divided the number of string LEDs

this way. Their orientation, however, is interesting. They sit on a ledge formed by the heat sink and are oriented straight up. The transparent carbonate globe fits onto the same ledge, so the LED light output is actually up into the edge of the plastic globe itself, rather than shining through the globe from the inside of the shell.

The LED driver circuitry is quite simple and laid out on the flex circuit without any potting compound to get in the way. According to the Microchip data sheet, six linear current regulators sink current at each tap and are sequentially turned on and off in a manner tracking the input

sine wave voltage. The chip minimizes the voltage across each regulator when conducting, providing high efficiency.

The output current at each tap is individually set by a resistor. An RC network, consisting of a resistor and three capacitors in parallel, on the input of the bridge rectifier provides phase dimming. Two other components handle transient protection on the connection to the ac line. In all, there are 13 discrete components on the flex circuit that make up the transient protection, phase dimming, and set the currents in the LED strings.

The 3M bulb base consists of a plastic sleeve around the heat sink to which the metal threads and foot contact mount. The electrical connections are on the flex circuit holding the LEDs and ac/dc converter. Visible here are the contact that bends over the side of the plastic sleeve to make contact with the metal screw threads, and a second contact that touches a post on the foot contact (right).

The reference circuit for the Microchip CL8800 is close to the circuit found on the 3M LED bulb, though the 3M bulb includes an additional RC network (not shown here) for phase dimming.

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Aclose-upoftheflexcircuiton the 3M LED bulb that holdsboththeac/dcconvertercircuitryandtheLEDs

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FEIT ELECTRIC CO.The bulb from Feit Electric had the oddest orientation for LEDs of any we examined. The 17⁄8-in.-diameter plate onto which the 36 LEDs mount is partly hidden in the assembled bulb by a circular plastic piece with a 1-in.-diameter hole in the middle. This piece mounts over top the LED plate. So, a look at the assembled bulb provides a view of the plastic piece and just five LEDs visible on the center of the plate below the hole in its middle.

We are at a loss as to why Feit installed the plastic piece over top most of its LEDs. The piece blocks most of the light they emit. (We have no way of quantifying the amount of light getting through the plastic. But informal tests here indicate little of it penetrates.) So the vast majority of the emitted lumens come from the five LEDs in the center of the plate.

The rest of the bulb’s mechanical design is less mystifying. The LED plate mounts to the top of a hefty 3.8-oz cast metal heat sink with three screws. The heat sink serves as the main body of the bulb. The ac/dc converter circuitry fits in a plastic cylinder that slides into the base of the heat sink and attaches to it with two screws.

The electronics is potted into the plastic cylinder that serves as its housing. The potting material is extensive, filling the cylinder. It also doubles as a structural element supporting the screw base of the bulb and the contact foot. The circuit board holding the electronics is two-sided and extends back nearly to the foot of the bulb base. The negative lead to the board is held to the metal screw threads by the potting material. Two wires run from the board to the LED board and seem to be hand soldered there. The board itself is reflow soldered.

The potting material obscured some of the details on the PCB, but on the board are two power MOSFETs, a diode bridge chip, five large caps, transformer, and at least 22 discrete components comprised of resistors, small caps and diodes. The input bridge rectifier seems to be protected with a fusistor.

The main chip is an SSL2103T LED driver from NXP Semiconductors. The SSL2103 is basically a flyback converter that operates in combination with a phase cut dimmer circuit directly from the rectified mains. It implements dimming through integrated circuitry that optimizes the dimming curve. Drive outputs are available for resistive bleeder switching.

Potting material on the Feit bulb’s PCB, visible here at the base of the heat sink, doubled as a structural element holding the foot contact in place.

The Feit ac/dc converter circuit was close to the reference circuit the NXP Semiconductors provides for its SSL2103 converter.

Three screws held the LED plate to the heat sink on the Feit LED bulb. The reverse side of the LED plate, visible here, had thermal compound applied between the heat sink and LED plate surfaces.

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Though potting material obscures some of the connection details, the circuit seems to be close to that of NXP reference designs for the chip. The mains voltage is rectified, buffered and filtered in the input section and connected to the primary winding of a transformer. The transferred energy is stored in a capacitor and filtered before driving the LED chain.

The circuit board also includes two power MOSFETs. One seems to be part of a dimming circuit that divides and filters the mains rectified voltage to provide an input for the generation of the dimming curve. A bleeder drive output from the NXP chip drives the MOSFET to switch bleeder resistors that are involved in a timer for the dimming function. The other MOSFET is the main switch for the flyback transformer.

There is also a buffer circuit consisting of two capacitors and an inductor. The circuit stores energy to ensure the converter can transfer power continuously to the LED chain despite any mains power fluctuations. It also filters ripple current generated by the converter to keep down any mains-conducted emissions.

Finally, another portion of the circuit consists of a capacitor, a rectifier diode, a peak-current-limiting resistor and a protection zener diode and is used to generate an external VCC supply for the IC.

The Feit LED bulb positioned a plastic disk over all but five of its 36 LEDs. We’re not sure why.


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Withthepottingmaterialmilledoff,thePCBfortheFeitLEDbulbrevealedadiodebridgeICandSSL2103TLEDdriverfromNXPSemiconductorsononeside,largeenergystorageelementsandpowerMOSFETsontheother.


59DESIGN WORLD — EE Network 11 • 2015 designworldonline.com

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60-W TEARDOWN The Philips LED bulb had no heat sink other than

the two-sided plate holding the LEDs. One reason: temperature compensation. The NTC resistor is visible on this shot of the LED plate.

The diode bridge and npn power transistor is visible on one side of the Philips LED bulb PCB. The other side holds the energy storage components and the two unidentified ICs providing temperature compensation, dimming and power conversion.


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PHILIPS LIGHTING CO.One noteworthy point about the Philips bulb pertains to heat sinking. The other bulbs we examined had metal heat sinks ranging in weight from 1.3 to 3.8 oz. The Philips bulb manages to handle thermal issues without any extra heat sinking. The only component that spreads heat is the 2.5-in.-diameter disk onto which the 26 LEDs mount, 13 to a side. Moreover, you might expect that designers would stagger the LEDs on the disk such that they wouldn’t mount directly opposite each other—this mounting arrangement would also help spread heat. But the LEDs on either side of the disk sit directly opposite each other. It appears that LED heat just wasn’t an issue in this design.

One of the reasons why is the presence of a negative temperature coefficient (NTC) thermistor on the LED board. But it proved to be impossible to trace out the temperature compensation network exactly because the driver PCB has three layers, one hidden. Further complicating the analysis of the circuit is the fact that two six-pin ICs seem to handle the ac-dc conversion and neither is marked with either a manufacturer logo or part number.

Because the main ICs can’t be identified, we can only hypothesize about how the LED driver works. The presence of a transformer, two large capacitors and an npn power transistor (from STMicroelectronics) on the PCB would seem to indicate that the converter has a flyback design. Our guess is that the temperature compensation

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The foot contact attaches to the PCB on the Philips bulb with wiring, visible here. The contact with the metal screw threads is via a wire that squeezes between the threads and plastic enclosure.

There isn’t much to an LED bulb when it can be built with no heat sink. The Philips bulb basically consists of the PCB and LED plate along with the snap-together plastic case which also supports the contact foot.

network is in the biasing of the switch providing current to the LEDs from the flyback transformer. Two transistors seem to handle the LED current. In all, we counted 32 small discrete components made up of resistors, diodes and capacitors. Rounding out the board components were a bridge rectifier chip and three other power capacitors.

It turns out that the mechanical design of an LED bulb that contains no heat sink can be quite simple (and some might call it elegant). The Philips bulb is basically a plastic enclosure that encases

the LED plate and driver PCB while also supporting the metal screw threads and contact foot.

The form factor differs from that of other bulbs because of the two-sided LED plate. The Philips bulb isn’t so much a bulb as a disk. Rather than encasing the LEDs in a transparent globe-like enclosure, the Philips device presents a flat profile with plastic encasing the two-sided LED plate. The enclosure seems to just snap together over top the LED plate and driver PCB.

And because the Philips bulb contains no heat sink, it is quite light weight. But its

disk-like outline might look a little weird to consumers accustomed to screwing things that are shaped like spheres into light sockets. And it beams most of its light out on the two sides defined by the orientation of the LED plates. It relies on diffusion by the plastic enclosure for lighting in other directions.



Smart power for smart systemsGERALD HOVDESTADDirector COTS EngineeringMember of VITA Committee 46.11, 62 & 65 Behlman Electronics

The role of a co-resident system manager with dual-redundant VITA 46.11 chassis managers, courtesy of VITA, 2015.

T he need for smart power sources able to complement smart systems is becoming increasingly important. In the past, system designers would

decide on components needed to accomplish a required function or mission. Once all the components were known, the system power requirement was determined. Designers often tried to squeeze a power supply into whatever corner of the system remained vacant. The task of accommodating that supply with its requisite needs for cooling was usually an afterthought. Performance monitoring and power supply optimization were barely considered if at all. Simple fault monitoring was sometimes possible, but detailed analysis was beyond the scope of most system designs.

Today, system specifications are generally written to emphasize intelligent platform management/intelligent system management. These concepts are mentioned in the recently ANSI-approved VITA (for VMEbus International Trade Association) 46.11 document. The specifications make

a strong case for carefully monitoring and controlling all aspects of the system to significantly improve performance and to drastically reduce downtime and maintenance costs.

Intelligent power supplies are a vital part of this effort. Behlman has recently released IQ versions of its VPX VITA 62 power supplies to provide reliable, flexible and intelligent power for the new systems built to VPX computer bus standards.

Behlman VPX supplies have been used in many systems for several years. Some of these systems require thousands of watts of power. The systems are continuing to perform reliably; however, they use only basic fault and temperature monitoring.

The new VPXtra-IQ series takes the concept of system intelligence to a whole new level. The system manager can now monitor input voltage and current, as well as output voltages and currents. In addition, temperatures are measured at several locations throughout the power supply. While this information is useful in determining if there is a power supply problem, it is also

invaluable to the system manager for total system performance monitoring.

A great deal of system information can be derived through judicious interpretation of the data available from the power supply. For example, an increase in input power could mean many different things, ranging from a normal response to increased workload, changes in local or ambient temperature, power supply problems, or other system anomalies. Input power changes without output changes might indicate a problem with the power supply itself if other monitored parameters, such as temperature, remain unchanged.

WHITE PAPER


EE_Belman_Nov_Vs4.indd 62 11/12/15 12:21 PM

This 6U Behlman VPXtra 1000CD-IQ power supply provides 1 kW (12 Vdc and auxiliary 3.3 Vdc). A 6U VPXtra 1000CM-IQ power supply provides

700 W (12 Vdc, 5 Vdc, and auxiliary 3.3, -12 and 12 Vdc).

An Intelligent System Manager can take all the available information being fed to it and make well-informed decisions about system performance, then alter system operations as necessary. For example, airflow might be increased, or processing workloads modified, depending on the interpretation of monitored parameters. Monitored temperatures could reveal overheating, or perhaps problems with a specific FRU (field-replaceable unit) or other load. The ability to rapidly communicate with all system components, power supplies, chassis, FRUs and so on, as described in VITA 46.11, yields a robust, flexible system.

Besides providing detailed power supply and system performance information, Behlman VPXtra IQ series

supplies give the people managing the system great flexibility in adjusting internal power supply limits and warnings. These parameters are normally fixed in older

“dumb” supplies. However, in the spirit of VITA 46.11 flexibility, Behlman’s smart interface allows the user to adjust warning and limit thresholds for many functions, such as over and under voltage, over current and over temperature.

One point to note in existing VPX specifications is the absence of a clear definition of programmable power supply functions. As discussed in the system management specification VITA 46.11, Behlman has incorporated the well-defined functions in the PMbus specification as a starting point for power supply management. It has been

suggested that a new specification dedicated to power supply management be written, possibly as a subset of VITA 46.11 or VITA 62. The PMbus commands could be used and then modified or added to as the new system configurations evolve.

As the current need for smart systems with more computing power, less weight and smaller size accelerates, it is only a matter of time before more and more VITA VPX parameters must be agreed upon and provided by manufacturers. The new Behlman VPXtra-IQ 1000CD-IQ 1-kW and Behlman VPXtra-IQ 1000CM-IQ 700-W power supplies take today’s concept of system intelligence to a whole new level, and set the stage for even smarter power for the future.

REFERENCESBehlman Electronicsbehlman.com

VITA vita.com

Power & Energy EfficiencyPOWERELECTRONICTIPS.COM


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WHITE PAPER Power & Energy EfficiencyPOWERELECTRONICTIPS.COM


Digital inrush current controller reference design

Figure 1. Functional block diagram of the digital inrush controller

IXYS digital power control technology can work with the capabilities of Zilog’s 8-bit

Z8F3281 microcontroller, a member of the Z8 Encore! XP F6482 Series of MCUs, to control inrush current in ac-dc rectifiers or ac-dc converters. The objective of this reference design is twofold—to highlight the advantages of digital control that overcome many of the shortcomings of current technology, and to enhance interest in digital control of high-power converters, potentially stimulating the development of next generation converters.

Digital control allows for distinctive solutions to control inrush

current in a typical ac-dc rectifier with a capacitive load by limiting the capacitor pre-charge current to a predetermined value at each half sine-wave cycle. This capacitor charge is spread over a number of cycles until the capacitor charges to a peak value of ac voltage source. The capacitor charges according to a time-dependent pulse train.

Pulses are designed to apply a substantially equal voltage increment to the capacitor as a means of maintaining a peak charging current of approximately the same value at each cycle. The number of cycles depends on capacitor value and charge current. A capacitor value

is selected based on the desired amplitude of ripples. The charge current is a function of the number of pulses and their timing position with respect to the rectified sine wave.

This reference design features programmable overload protection and a “Power Good” status signal. It is not sensitive to power outages, brownouts or ambient temperature variations. This reference design can operate within an input voltage range of 80 to 240 Vac and load current up to 3 A. The entire operating process and essential values are fully

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WHITE PAPER Power & Energy EfficiencyPOWERELECTRONICTIPS.COM


Figure 2. Digital inrush control timing

Figure 3. Capacitor C pre-charging during first two cycles—Legend (not to scale): Blue—power line voltage; Red—voltage on capacitor C with respect to common ground; Yellow—driver to Sw1; Green—capacitor current

programmable. The controller may be programmed to handle 50 or 60 Hz, or any other line input frequency.

This digital inrush current controller reference design is valuable for high-power loads with tens of amps of current. It lets users optimize performance, maximize efficiency across the load range, and reduce design time. IXYS power components handle the pre-charge of load capacitors at these values while limiting inrush current to controlled values.

The digital inrush current controller uses the following components: power components, including a diode bridge, inductor and bulk capacitor; a solid-state switch (Sw1) to commutate capacitor pre-charging current; a second solid-state switch (Sw2) to connect/disconnect the load; and digital control based on Zilog’s Z8F3281 MCU.

Digital inrush control aims to charge up the bulk capacitor in substantially equal increments. This takes place by providing control pulses to Sw1, as in Figure 1, resulting in a voltage applied to the bulk capacitor in equal increments. It is possible to apply this charge on a cycle-by-cycle basis considering a cycle is half of a sine wave of line voltage. For example, we can assign N cycles for the inrush control operation, then split

the normalized amplitude of the sine wave cycle into N segments with 1/N increments, as in Figure 2.

During cycle 1, Sw1 is on (conducting) from time t1 to T, as in Figure 2. The voltage across the capacitor rises proportionally to a normalized value 1/N. During this period, the charging current follows LC-resonant behavior, as in Figure 3 (green line). The current rises until the capacitor voltage reaches the input voltage, excluding voltage dropouts. The current continues its resonant behavior as long as Sw1 is on. There is no further oscillation because the

input voltage drops below the voltage on the capacitor, switching Sw1 to the off state (not conducting). The capacitor remains pre-charged to the voltage proportional to 1/N.

In cycle 2, capacitor C is pre-charged by another voltage increment 1/N, in a process similar to cycle 1. The process continues until N cycles have elapsed at which point capacitor C has charged to a voltage proportional to the peak voltage of the input power line.

A simplified timing diagram for inrush control, Figure 2, shows the voltage increment for each cycle is defined by the



number of cycles (N). The capacitor’s charging current is proportional to the voltage increment, 1/N. Thus the number of cycles (N) is the variable controlling peak inrush current.

Another variable to control inrush current is LC time constant. The value of capacitor C depends on the desired ripple value. With the value of C selected, the designer can reduce peak inrush current by boosting the amount of inductance (L). If there are physical limits to the L value, the number of cycles (N) should be used to set the required peak current. Turn-on time for switch Sw1 should be defined for each active cycle.

Assuming delay from zero-crossing (point 0 in Figure 2) to the earliest turn-on of Sw1 happens at T_off, the time to keep Sw1 on is T_on, and the duration of the complete cycle is T.

T_on for each occurrence i is defined as a geometrical transform:

Tπ/2TON(i) = asin (i/N), where i = 1 ... N

where i = 1 … N. The period T is measured by the MCU at initialization. Values T_on are determined by Eq. 1 and stored in memory. Values for T_off are derived by firmware according to:

T_off = T – T_on

Figure 4 illustrates how T_on values are defined when N is 16 for 120 V at 60 Hz. In Figure 4, the blue line is rectified power line voltage. The magenta line represents actual T_on time value in microseconds for each cycle, and the yellow line indicates the T_on pulse position relative to rectified power line voltage. The red line is a once-per-cycle signal from a time base provided by the internal clock. The counter first counts until the T_off value, represented by the green line. When the counter reaches T_off value, it initiates the T_on pulse (yellow line) which continues until the counter reaches the T_on value (magenta line).

Figure 5 displays the timing position and amplitude of the capacitor current (green spikes) with respect to T_on pulses. The inrush controller produces a single current pulse every cycle because the input voltage drops below the capacitor voltage, and the input power line is isolated from the rest of the circuitry through the diode bridge once the capacitor charge is complete. The inductor discharges into the capacitor, and Sw1 is turned off (not conducting) at the end of the cycle.

Figure 4. T_on timing generation—Legend (not to scale): Blue—rectified power line voltage; Red—full cycle period timing counter; Yellow—driver to Sw1; Green—time off to Sw1; Magenta—time on to Sw1; White—period T

Figure 5. Capacitor C pre-charging—Legend (not to scale): Blue—rectified power line voltage; Red—voltage on capacitor C with respect to common ground; Yellow—driver to Sw1; Green—capacitor current

WHITE PAPER


The reference design also includes a load on/off switch, Sw2, overload protection, and a “Power Good” output to display the status of IXYS power components. Sw2 activation is programmable and is enabled after the pre-charge of capacitor C finishes. In the reference design, Sw2 activates at the zero-crossing on the next cycle after the capacitor is pre-charged.

In the simulation depicted in Figure 5, Sw2 activates at time stamp 0.066 msec, when the capacitor current shows up as negative, because the current is sourced from the capacitor. Sw2 switch activation can be programmed to any time stamp point. The overload protection feature protects a device from damage in case of current overload. The overload threshold is programmable and is set to 3.5 A in this reference design. If the comparator detects an overload, the MCU disconnects the load by turning Sw2 and Sw1 switches off.

Overload protection can be programmed for two modes of operation: 1. To immediately shut down the device and wait for user input; 2. To let the device restart after the short circuit is removed, and restart for a predetermined number of repeated short circuit occurrences. In the second mode, the delay between restarts and the number of restarts is programmable. In this reference design, the delay time is set to 1.5 sec and the number of restarts is set to four.

“Power Good” status activation is programmable. For this reference design, activation is delayed by two cycles after the capacitor pre-charge is complete. The “Power Good” status is not set if an overload is detected.

HARDWARE IMPLEMENTATION Figure 6 (a) and (b) depicts the MCU module and main power board. The MCU module is implemented as an add-on device. The module consists of a connector for programming the microcontroller. The MCU module is

powered by 3.3 V for the MCU and 12 V for the gate driver applied to the J4 connector on the main power board.

The main power board is a two-layer surface-mount PCB that provides easy access to test points. A diode bridge and two MOSFETs mount on small heat sinks. Power dissipated on these heat sinks is less than 5 W at a power output of 375 W.

Following a power-on reset and initialization, the MCU analyzes the power line, sets an appropriate timing, and begins pre-charging the bulk capacitor. Actual waveforms taken from a scope at normal operation appear in Figure 8. The inrush current (top blue line) is limited to 10 A. The yellow line shows the signal at the Sw1 gate. After the inrush procedure completes, the gate is set to a high level to let Sw1 continue conducting. One cycle later, the load is connected and load voltage rises from zero to the level on the pre-charged capacitor. After the load is connected, there’s a slight drop in the rectified voltage due to the limited power capability of the isolation transformer.

Figure 6: (a) MCU module and (b) main power board with MCU module

Figure 7. Digital inrush controller setup

A B





Inrush current measurements were taken with the controller reprogrammed to 16 cycles instead of the original 8 cycles to highlight the performance with more inrush control cycles at the same power conditions. As a result, the inrush current dropped two times, as shown in Figure 9.

PERFORMANCE DURING OVERLOADS Normal operation starts with pre-charging the bulk capacitor. A scope snapshot of the start up appears in Figure 10 with a horizontal scale of 1 sec. The yellow line indicates that the gate driver state is wide because of the low graphic resolution for 12 cycles of pre-charge. After the inrush control sequence completes, the load is connected, indicated by the red line going up. The “Power Good” status (blue line) goes up as well.

Continuous overload is applied at 2.3 sec. The load is disconnected and the “Power Good” status goes low. Attempts to restart the device take place at 1.5-sec intervals. The inrush sequence again commences, and the load is connected. However, the overload is sensed right away, the load is disconnected, and the “Power Good” status continues to stay low. There’s another attempt to restart after 1.5 sec and the load stays connected for about one second while there is no overload.

We can estimate the efficiency of the digital inrush controller considering circuit elements after the diode rectifier all the way to the load. At a load power of 375 W, the power loss is 2.1 W, which equates to an efficiency of 99.47%.

This reference design can be configured for different parameters such as input voltage and frequency range, overload threshold, overload recovery time, number of overload events before shutdown, time position for “Power Good” status, and time position to turn on the load. The firmware that controls the operations of this reference design was developed using a Zilog Development Studio II (ZDS II – Z8 Encore! version 5.2.0). The source

Figure 8. Scope snapshot of digital inrush current control—Legend: Blue—power line current (10 A/div); Red—load voltage (50 V/div); Green—rectified input voltage (50 V/div); Yellow—Sw1 commutation signal

Figure 9. Scope snapshot of inrush pulses with N increased to 16—Legend: Blue—power line current (10 A/div); Red—load voltage (50 V/div); Green—rectified input voltage (50 V/div); Yellow—driver to Sw1

Figure 10. Scope snapshot to show performance during overload and restart—Legend: Blue—“Power Good” status; Red—load voltage (50 V/div); Yellow—driver to Sw1

WHITE PAPER



code for this application, IXRD1001-SC01.zip is available free for download from the Zilog website. Here are the programs included in the source code:

Main.c file—To limit the inrush current upon system power up, this program uses look-up tables (LUTs) containing values based on characterizations from a simulation model to charge the buffer capacitors, which connect to the load after the capacitor charging completes. If there’s a fatal over-current event, the comparator will shut down the system and restart the inrush current control for a user-defined amount of time. After the allowed restart attempts, the system remains off.

The program implements a comparator and three timers to generate the current-limiting pulse trains. The pulse trains consist of on and off times based on the LUT values. Effectively, this constitutes a pulse-width-modulated signal. Each of these on/off pulses is applied to the input switch of the inrush current limit circuit to limit the current of the circuit. Three LUTs contain values that are characterized for the capacitors to be charged at 220 V/60 Hz, 120 V/60 Hz and 220 V/50 Hz. The main purpose of the comparators and timers is to create the current-limiting pulse train so the pulses are aligned to a certain position of the rectified power input sine wave. A comparator and a timer, which together measure the zero-crossing points of the input power sine wave, synthesize this pulse train alignment. The pulse train is then synchronized with the zero crossings and is located toward the falling slope of the rectified sine wave.

Upon power up, the first user-definable sine-wave cycles are ignored to let the system stabilize. The next two sine-wave periods are measured using the comparator’s falling edge and

multichannel timer and stored as variable T_period. However, this variable is not implemented; instead, a macro containing the ideal value for either 50 or 60 Hz is used. The macro can be changed so the measured values can be implemented, if required.

The next four input power sine cycles are used to determine the zero-crossing points by measuring the time elapsing from the sine’s falling slope to the sine’s rising slope at a 0.7 V comparator interrupt threshold. A timer called timer1 measures the time between the two comparator interrupt events. When the measurement completes, the zero crossing is determined to be half of the measured time.

After the above information is collected, timer1 and a second timer called timer2 are configured in triggered one-shot mode. Timer2 is loaded with a quantity equal to the power-input sine period value minus the inrushCurrentOffTime value from the LUT, minus the measured zero-cross time and a zero-cross compensation value. The result of this operation becomes the on-time.

The zero-cross compensation value is used to fine-tune the pulse train alignment with the power input sine wave. Timer2 triggers upon the comparator’s rising edge interrupt. When timer2 expires, the gate of the input switch MOSFET is enabled and timer1 triggers. When timer1 expires, the input switch MOSFET is disabled again.

This process repeats until all the values from the LUT have been used. To this point, only the capacitors have charged, not the actual load. The actual load actives once the inrush current procedure for the capacitors has finished and one input power sine cycle has elapsed. After the load connects, a “Power Good” signal is generated after a user-defined delay.

Figure 11. Schematic diagram of digital inrush controller main board





Main.h file—This file contains the function prototypes, structures and macro definitions. The inrush current limit can be selected for 120 V/60 Hz, 240 V/60 Hz and 220 V/50 Hz systems by commenting and un-commenting macros. Uncommenting the ONEHUNDRED_TWENTY macro selects 120 V/60 Hz; commenting out the ONEHUNDRED_TWENTY and TWOHUNDRED_TWENTY_VOLT_50HZ macros selects 240 V/60 Hz, otherwise 220 V/60 Hz is selected.

Depending on the load capacitor values, the amount and durations of the LUT values may have to change. If so, there’s a formula to calculate the new LUT values which can then be loaded into the LUTs. The INDEX macro must be adjusted to the new LUT values.

The RC_DLY_OFFSET_50HZ macro is used to fine-tune the pulse train to the left on the x-axis of the input power sine wave to realize a more linear charging characteristic.

Initialization.c file—This file contains the peripheral setup for the system clock, operational amplifier, ADC, multichannel timers, general-purpose timers and comparators.

The code for this reference design was created on ZDS II – Z8 Encore! version 5.2.0, which can be downloaded from the Zilog website. To change the code:

1. Install ZDS II – Z8 Encore! version 5.2.0 or newer software.

2. Connect the Opto-isolated USB Smart Cable (debugger) to the computer. Connect the ribbon cable to the J3 connector on the main power board.

3. Download the source code file for the reference design (IXRD1001-SC01) from the Zilog or IXYS website and open this project file in the ZDSII IDE.

4. To change the code operation, refer to the instructions in the main.h header files.

5. Compile the program and download it to the Z8F3281 MCU.

In tests of the reference design, the efficiency of the inrush control path has been measured to be 99.5%. This design presented here is capable of working with input voltages ranging from 80 to 240 V. To work with higher power line voltage, a longer control pulse train needs to be programmed. For instance, raising line voltage from 110 to 220 V required twice as much pre-charging time to have the same peak inrush current.

Overload protection is based on continuous monitoring of dynamic current from the bulk capacitor. In an overload, the current drawn from the capacitor instantly increases and triples the comparator,

Figure 12. Schematic diagram of digital control module

WHITE PAPER



initiating system overload mode. Overload current threshold, the number of overload instances, and the period between overload events are programmable. The option to turn the load on/off facilitates the overload mode of operation by disconnecting the load. “Power Good” status is not available in the event of an overload.

This digital inrush controller is the first digital power device based on Zilog’s Z8 Encore! XP F6482 Series of MCUs. An innovative current measurement algorithm lets this controller have common input and load grounds. The reference design can be used as part of an ac-dc rectifier or can be expanded to higher-level devices such as PFC converters. Digital control can be used to build a user interface that would allow users to change device parameters, gather statistics, add a communication interface, remotely monitor performance or change parameters.

REFERENCESIXRD1001 Digital Inrush Controller Reference Design document zilog.com/docs/referencedesign/IXRD1001.pdf

Source code for Digital Inrush Controller Reference Design zilog.com/docs/referencedesign/IXRD1001-SC01.zip

F6482 Series General-Purpose Flash Microcontroller Product Specification tinyurl.com/oauvfvc

F6482 Series Development Kit User Manual zilog.com/index.php?option=com_doc&Itemid=99

F6482 Series API Programmer’s Reference Manual tinyurl.com/nqj7akg

Figure 13: Main board layout (top and bottom layers)



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currents. An instrument able to do that and much more recently became available from Keysight Technologies. Engineers can use the firm’s IntegraVision PA2203A four-channel power analyzer to observe three-phase power consumption under dynamic conditions on a built-in 12.1-in. capacitive touchscreen.

The instrument provides a numerical read-out of power factor, Var, VA, power draw in watts, and several other factors for three-phase lines. It does so with a 0.05% basic accuracy and 16-bit resolution. Additionally, it lets users visualize transients, in-rush currents and state changes with a 5-M sps digitizer that captures voltage, current and power in real time with 2.5-MHz bandwidth.

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The instrument can also double as a truth-detector when gauging claims of energy efficiency. For example, vendors of “black box” energy efficiency equipment sometimes run demonstrations of their “solutions” on lightly loaded motors. The black box frequently consists of a capacitor effecting a power-factor correction. Switching the black box into the circuit reduces a simple readout of amps dissipated in the load. Examining the input in terms of a phasor diagram, however, can give a graphic depiction that clarifies

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Getting pure sine waves from electronic inverters

Pure sine wave inverters work well for powering sensitive electronic devices, including communication equipment, that need high-quality waveforms with little harmonic distortion. This particular board from Magnetic Design Labs employs a TI TMS320F2812 DSP chip, a low-power device operating from a 1.8-V core at 135 MHz.

Digital signal processors (DSPs) are being used in switching power supplies able to synthesize pure sine waves for use in high-reliability (hi-rel) military applications.

Ruggedized digital inverters were installed in the M1068, a Command Communications variant of the U.S. Army’s M113 family of armored personnel carriers. The original M113 Armored Personnel Carrier was first fielded in Vietnam. To date, an estimated 80,000 M113s, including a long list of variants, have been produced and used by over 50 countries worldwide, making it one of the most widely used armored fighting vehicles of all time.

During this time, the M113 has been continuously updated to meet the demands of the modern battlefield. Since then, the M113 family of vehicles are being upgraded, reconfigured and introduced as entirely new systems, including the M1068.

The M1068 variant is used as a tactical operations center capable of long range communications and includes 4.2-kW auxiliary power unit (APU) mounted on the right front of the vehicle to provide 24-V power.

As part of this project, BAE Systems required two hi-rel, ruggedized 2.5-W pure sine wave inverters per vehicle to convert 24 Vdc power generated by the vehicle-mounted APU into usable multi-kilowatt levels of ac for powering communications devices, lighting, computers and other electronic devices.

Although more expensive, pure sine wave inverters provide cleaner utility-grade power than quasi-sine-wave models. Pure sine wave inverters are helpful when operating sensitive electronic devices, including communications equipment, that require high-quality waveform with little harmonic distortion.

In addition, pure sine wave models have a high surge capacity, which means they are able to exceed their rated wattage for a limited time. This lets vehicle motors start easily, though they can draw many times their rated power during start up.

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explained that the 2.5-kW inverters also had to meet shock and vibration requirements that could only be met by hi-rel inverters.

“The shock requirement for the inverters installed on the M1068 was close to 30 gs, in three directions (vertical, horizontal and transverse) and the vibration spectrum was also demanding,” he said.

The pure sine wave supplies come from Magnetic Design Labs in Santa Ana, Calif. According to Kamran Kazem, MDL VP and chief technology officer, despite the 30-g requirement for the M1068, the military specified that the inverters had to withstand 100 gs. This was well beyond the amount of shock the vehicle would ever realistically experience, based on military tests conducted in the roughest terrain that maxed out at 15 gs.

Kazem added that hi-rel electronic systems and components are no longer the exclusive domain of aerospace and defense. Today, medical, transportation, communications, infrastructure and industrial all have applications where the price of failure is high.

“These power converters might not have to withstand the same extreme conditions as the military, but vibration, shock, humidity and other inherent environmental problems are still factors, so the need for hi-rel and rugged power converters certainly applies to those markets as well.”

DSP is also a key element in a new generation of modular, stackable inverter options designed to provide a range of 1

to 20 kW of dc-ac power through a single, customizable unit. This type of system, available from custom power converter designers like MDL, consists of rack-mounted inverter modules that can stack in a parallel configuration, letting the user add as many inverters as needed to meet the power requirements.

Each unit connects to a communications controller that is responsible for synchronization, load sharing and any external communications. The individual inverters are hot swappable, enabling the addition or replacement of modules on the fly.

“This type of modular design provides project managers with a system that fits their power requirements without having to develop a new unit just for their specific project,” Kazem said. “This eliminates the need for many application-specific designs, and could also enable faster delivery of the power converter at a much more economical cost.”

REFERENCESMagnetic Design Labs magneticdesign.com


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