Family of zero-voltage-switching unregulated isolated step-up DC–DC converters

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Published in IET Power ElectronicsReceived on 4th December 2012Revised on 25th January 2013Accepted on 24th February 2013doi: 10.1049/iet-pel.2012.0714

ISSN 1755-4535

Family of zero-voltage-switching unregulatedisolated step-up DC–DC convertersZhilei Yao1, Lan Xiao2

1School of Electrical Engineering, Yancheng Institute of Technology, Yancheng Jiangsu 224051,

People’s Republic of China2Jiangsu Key Laboratory of New Energy Generation and Power Conversion, Nanjing University of Aeronautics and

Astronautics, Nanjing 210016, People’s Republic of China

E-mail: [email protected]

Abstract: This study presents a family of zero-voltage-switching (ZVS) unregulated isolated step-up DC–DC converters. Allswitches can realise ZVS using the magnetising current of the transformer. The ZVS unregulated isolated step-up DC–DCconverters are deduced from the basic ZVS cells using the magnetising current. The ZVS condition for the switches is given.The operating principle is analysed taking the ZVS push–pull forward (PPF) unregulated DC–DC converter as an example.The ZVS-oriented parameter design and example are presented. Finally, the experimental results from a 1 kW ZVS PPFunregulated DC–DC converter support the theory.

1 Introduction

Recently, many concerns have been raised over fossilfuel-electricity generation, since it pollutes our environmentand depletes energy supply. In contrast, renewable energysources, such as solar energy and fuel cell, have gained alot of attention because they are renewable, environmentalfriendly, and flexible for installation [1–9]. As the outputvoltage of the fuel cell or solar cell is often low andfluctuates with load, such as 30− 60 V, it should beboosted and regulated by a DC–DC converter andconverted to an AC voltage by a DC–AC converter [10].Thus, the front-end DC–DC converter should havelow-input current ripple and can operate at wide-input-voltage range.The current-fed converter, in general, exhibits low

transformer turns ratio, small input-current ripple, and lowdiode voltage rating. However, a clamping or snubbercircuit is usually required for the current-fed converter tolimit the transient voltage caused by transformer leakageinductance. Since the duty ratio must be > 0.5, itsinput-voltage operating range is relatively narrow [11, 12].The voltage-fed converter has low switch voltage rating andsmall output current ripple. However, it has large inputcurrent ripple and high diode voltage rating in wideinput-voltage range application.A two-stage DC–DC converter, for example, a regulated

boost converter cascaded with a zero-voltage-switching(ZVS) unregulated isolated step-up DC–DC converter, is agood solution in wide input-voltage range and lowinput-current ripple applications, as shown in Fig. 1 [13].

862& The Institution of Engineering and Technology 2013

Although the two-stage converter has a little lowerefficiency and larger size compared with the single-stageconverter, the rectifier reliability can be improved and bothconverters in the two-stage converter can be designedoptimally. The boost converter can step up the voltage andis suitable for the wide input-voltage range and lowinput-current ripple applications. In addition, the inputvoltage of the ZVS unregulated isolated step-up DC–DCconverter is constant, so it is easy to select the rectifierdiode. The ZVS unregulated DC–DC converter has thefollowing merits: constant duty ratio, simple structure, highefficiency, high reliability, low electromagnetic interference(EMI) and electric isolation [14–20].A resonant inductor (including the transformer leakage

inductor) is utilised to realise ZVS of the switches, but ZVScannot be achieved at light load [14–18]. In [19] and [20],magnetising current is used to achieve ZVS for theswitches, so the switches can realise ZVS at full range ofload. However, ZVS condition for the switches is notanalysed, and the topologies are only suitable for lowoutput-voltage applications.In order to solve the aforementioned problems, the

ZVS unregulated isolated step-up DC–DC convertersusing the magnetising current are deduced from the basicZVS cells, and the ZVS condition for the switches isgiven. Take the ZVS push–pull forward (PPF) unregulatedDC–DC converter as an example, the operating principleis described. The ZVS-oriented parameter design andexample are presented. The experimental results from a1 kW ZVS PPF unregulated DC–DC converter confirm thetheory.

IET Power Electron., 2013, Vol. 6, Iss. 5, pp. 862–868doi: 10.1049/iet-pel.2012.0714

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2 Derivation of the ZVS unregulated isolatedstep-up DC–DC converters

Generally, soft turn-on in ZVS converters is achieved byturning on the switch, while its body diode is conducting.The body diode can be forced into conduction by using themagnetising current of the transformer to fully discharge theoutput capacitances of the switch. There are two basic ZVScells using the magnetising current, as shown in Fig. 2.

2.1 Two switches connected to one winding of thetransformer [see Fig. 2a]

To realise the ZVS of both switches, the magnetisinginductance must be bilaterally magnetised. As the ZVS cellis symmetric about point c, the input-voltage source can beconnected between points a and b, or a and c. It should benoted that the input-voltage source connected betweenpoints b and c is equivalent to that connected betweenpoints a and c.

2.1.1 Input-voltage source connected between pointsa and b: When two cells of Fig. 2a are connected inparallel, that is, the same point name connected, and twoprimary windings of the transformer combines to oneprimary winding, then the ZVS full-bridge unregulatedDC–DC converter is deduced taking full-wave rectifier inthe secondary winding as an example, as shown in Fig. 3a.The same rectifying method is used in the other deducedcircuits except for Fig. 3d.In addition, voltage-type components, such as capacitors,

can be connected to the ZVS cell to magnetise ordemagnetise the magnetic core. Therefore, there are twocombinations. (1) Two equal capacitors are connectedbetween points a and c and points c and b, respectively.Thus, the ZVS half-bridge unregulated DC–DC convertercan be obtained, as shown in Figs. 3b. (2) One capacitor isconnected between points a and c or b and c. Therefore theZVS asymmetrical half-bridge unregulated DC–DCconverter can be gained, as shown in Fig. 3c.

Fig. 2 Two basic ZVS cells using the magnetising current

a Two switches connected to one winding of the transformerb Each switch connected with each winding of the transformer

Fig. 1 Distributed power generation system


2.1.2 Input-voltage source connected between pointsa and c: A capacitor can be connected between points b andc, so the ZVS active-clamped forward unregulated DC–DCconverter can be deduced, as shown in Fig. 3d. Thecapacitor is used to reset the magnetising current and clampthe voltage spike of the switches caused by the leakageinductance, but it cannot transfer the power to the load.Hence, the rectifying method can only be the half-waverectifier.

2.2 Each switch connected with each winding ofthe transformer [Refer to Fig. 2b]

When the input-voltage source is connected between points aand b in Fig. 2b, the ZVS push–pull unregulated DC–DCconverter can be obtained, as shown in Fig. 3e. In addition,when a capacitor is connected between points c and d inFig. 3e, which can clamp the voltage spike of the switchescaused by the leakage inductance, the ZVS PPF unregulatedDC–DC converter can be gained, as shown in Fig. 3f.Comparisons among different ZVS unregulated isolated

DC–DC converters are shown in Table 1. One point thatneeds to be clarified is that the converters shown in Figs. 3band c are not practical in step-up applications since theprimary voltage of the transformer is just half of the inputvoltage.The rectifying method can be full-wave and full-bridge for

all the deduced ZVS unregulated isolated step-up DC–DCconverters except for Fig. 3d. However, when the DC–DCconverter is as the front stage of a three-phase inverter, theoutput voltage of the DC–DC converter should be about 700V, the full-bridge rectifier should be used. When the DC–DCconverter is as the front stage of a single-phase inverter, theoutput voltage of the DC–DC converter should be about 360V, the full-wave or full-bridge rectifier can be used.

3 Operating principle

It should be noted that the principle of operation and featuresof these ZVS unregulated isolated step-up DC–DC convertersare similar, so the ZVS PPF unregulated DC–DC convertershown in Fig. 3f will be taken as an example and beanalysed in this paper.To simplify the analysis, the following assumptions are

made.

1. All diodes, capacitors, and the transformer are ideal,except for the magnetising inductor of the transformer (Lm).2. The output capacitors of the switches S1 and S2 (CS1 andCS2) are equal to CS.3. The clamping capacitor (C ) is large enough to be treated asa voltage source with value of the input voltage (Vin) andC≫CS.4. The output filter capacitor (Cf) is large enough to betreated as a constant voltage source, and the output filtercapacitor current can be neglected.5. NP1 = NP2 =NP, NS1 =NS2 = NS, and n =NS/NP.

Key waveforms of the ZVS PPF unregulated DC–DCconverter are shown in Fig. 4, where Vgs1, Vgs2, Vds1, Vds2,im, iD1, iD2 represent the gate-drive voltages of S1 and S2,the drain–source voltages of S1 and S2, the magnetisingcurrent of the transformer, and the currents through D1 andD2, respectively. When the diode conducts, the diodecurrent is equal to Io according to assumption (4) and


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Fig. 3 Family of ZVS unregulated isolated step-up DC–DC converters using magnetising current

a Full-bridge circuitb Half-bridge circuitc Asymmetrical half-bridge circuitd Active-clamped forward circuite Push–pull circuitf PPF circuit

Fig. 3, which is a flat waveform. For Fig. 3d, difference of thekey waveforms in Fig. 4 is that iD2 does not exist during [t4,t6]. Duty ratio of S1 and S2 (D) is constant and approximates to0.5. There are six switching modes in a switching period, anda set of corresponding equivalent circuits is given in Fig. 5 toaid in understanding each mode.

3.1 Mode 0 [prior to t0]

Prior to t0, the input source and C provide the load current. S2is on and D2 is conducting, while S1 is off and D1 is


reverse-biased. The voltage Vds1 is equal to 2Vin, and thevoltage across D1 (Vd1) is twice the output voltage (Vo).The current im increases in the negative direction.

3.2 Mode 1 [t0, t1] [Refer to Fig. 5a]

At t0, S2 is turned off with zero voltage thanks to CS2. As thetime is very short during this mode, the current im can betreated as a current source with value of − Im. The currentim discharges CS1 and charges CS2, so Vds1 decreases andVds2 increases linearly. The voltages Vds1 and Vds2 can be

Table 1 Comparisons among different ZVS unregulated DC–DC converters

Full-bridge Half-bridge Asymmetrical half-bridge Active-clamped forward Push–pull PPF

no. of switches 4 2 2 2 2 2voltage stresses of the switches Vin Vin Vin 2Vin 2Vin 2Vinno. of primary winding 1 1 1 1 2 2no. of input capacitors 0 2 1 1 0 1suitable for step-up applications yes no no yes yes yes


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calculated as

Vds1(t) = 2Vin −Im2CS

t − t0( )

(1)

Vds2(t) =Im2CS

t − t0( )

(2)

At t1, Vds1 reduces to zero and Vds2 increases to 2Vin. Thus, thebody diode of S1 (DS1) conducts naturally clamping Vds1 at

Fig. 4 Key waveforms of the ZVS PPF unregulated DC–DCconverter


zero. The transformer secondary voltage is larger than theoutput voltage, so D1 conducts. The duration of mode 1 canbe deduced from (1) or (2).

t01 =4VinCS

Im(3)

During this mode, the rising and falling time of the drain–source voltage of the switch is independent of the load from(3).

3.3 Mode 2 [t1, t2] [see Fig. 5b]

During this mode, Vin is applied to Lm, which makes im decaylinearly in the negative direction. The currents im can beobtained as

im(t) =−Im + Vin t − t1

( )Lm

(4)

In order to achieve zero-voltage turn on of S1, the followingconditions should be satisfied from (3) to (4) and Fig. 4.

(0.5− D)Ts . t01 =4VinCS

Im(5)

where Ts is the switching period. At t2, S1 is turned on withzero voltage if (5) is satisfied.

3.4 Mode 3 [t2, t3] [see Fig. 5c]

During this mode, the input source and C power the loadsimultaneously. The inductor Lm continues to be charged bythe input voltage.

Fig. 5 Equivalent circuits of the switching modes

a Mode 1b Mode 2c Mode 3


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At t3, S1 is turned off with zero voltage thanks to CS1. The
current im reaches the maximum value (Im), which can beestimated from (4) omitting the small duration t12.

Im = VinD

2Lmfs(6)

where fs is switching frequency. [t3, t6] is the secondhalf-switching cycle, which is similar to the firsthalf-switching cycle [t0, t3].

4 ZVS-oriented parameter design andexample

In this section, a simplified design procedure and an exampleto determine ZVS-oriented component values of the ZVSunregulated isolated step-up DC–DC converters are given.

4.1 Specifications

The input voltage is 60 V, whereas the output voltage isspecified to 360 V. The proposed unregulated isolatedstep-up DC–DC converter is designed to provide 1 kW andoperates at 82 kHz.

4.2 Selection of duty ratio

The duty ratio D should approximate to 0.5, but it cannot beequal to 0.5, or else the ZVS of the switches cannot berealised and the input-voltage source may be shorted.Therefore the duty ratio D is selected as the maximumvalue of the control chip SG3525, which is about 0.45.

4.3 Selection of transformer turns ratio

As D approximates to 0.5, the transformer turns ratio can becalculated as

n = Vo

Vin= 360

60= 6 (7)

4.4 Selection of MOSFET

Based on the analysis in Section 3, the root-mean-square(RMS) switch current can be estimated as

Isrms ≃��1

Ts

∫Ts/20

nIo( )2

dt

√= nIo��

2√ = nPo��

2√

Vo

= 6× 1000��2

√ × 360

= 11.8A (8)

where Po is the output power.The voltage stress of the MOSFET (Vds) is

Vds = 2Vin = 120V (9)

International Rectifier IRFP90N20D power MOSFET wasused in the ZVS PPF unregulated DC–DC converter. Thepower MOSFET has a drain-source breakdown voltage of200 V, a maximum continuous drain current of 94 A at thecase temperature TC = 25°C and 66 A at TC = 100°C. Thechosen MOSFET has an output capacitance Coss = 1070 pFat Vds = 25 V. When D and fs are defined, t02 is fixed. The


time t12 cannot be too large, or else it is difficult to realiseZVS for the switches from Fig. 4. Therefore, t01 should belarge enough but should be smaller than t02, and then a 33nF/250 V capacitor is selected to parallel with the switch,which also can suppress the voltage spike of the switchcaused by the leakage inductance. As a result, the capacitorCS is about 34 nF.

4.5 Selection of magnetising inductance

The ZVS condition for the switches can be obtained bysubstituting (6) into (5).

Lm ≤ D(1− 2D)

16CSf2s

(10)

When D, CS and fs are defined, the magnetising inductanceshould be no larger than 12.28 μH from (10). A transformerwas built using an EE ferrite core (EE55/28/21) with asmall air gap to satisfy the ZVS condition for the switches,4 turns for primary winding NP, and 24 turns for secondarywinding NS. The inductance Lm measured across primarywinding is about 9.2 μH. However, the air gap of thetransformer cannot be too large, that is, the inductance Lmcannot be too small, or else the conduction losses of thepower devices will be large.

5 Experimental results

A 1 kW ZVS PPF unregulated DC–DC converter has beenbuilt to support the theory with the following parameters:

1. input voltage Vin: 60 V;2. output voltage Vo: 360 V;3. clamping capacitor C: 30 μF;4. turns ratio NP1:NP2:NS1:NS2: 4:4:24:24, and windingresistance referred to the primary side Rcu = 0.054 Ω;5. magnetising inductor Lm: 9.2 μH;6. filter capacitor Cf: two 330 μF capacitors connected inseries;7. switching frequency fs: 82 kHz;8. duty ratio D: 0.45;9. S1 and S2: IRFP90N20D, and the conduction resistanceRds(on) = 0.023 Ω; and10. D1 and D2: DSEI30-12A.

Fig. 6 shows the experimental results at different load.From Fig. 6, S1 sustains twice the input voltage and D1

sustains twice the output voltage. Before S1 is turned on,the voltage Vds1 is zero, so the switch can be turned on withzero voltage from the red dotted line in Fig. 6. S1 is turnedoff with zero voltage thanks to CS and clamped at 2Vin

because of C. The rising and falling time of Vds1 are thesame at different load, that is, about 0.45 μs, which isindependent of the load. The time t01 is 0.457 μs calculatedby (3) and (6). Therefore the measured results support thetheoretical ZVS interval.As shown by Fig. 6a, the rectifier diode D1 is off with

zero-current at light load. The main reason is that iD1 is thesum of the load current and the output filter capacitorcurrent, and when Cf is fully charged, the capacitor Cf canprovide the load current. As can be seen from Fig. 6b, thecurrent spike of iD1 at turn-off is caused by the reverserecovery of D1 at full load. The difference from the theoryabout iD1 is that the output filter capacitor current is omitted


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Fig. 6 Experimental results

a At 150 Wb At 1000 W

in the theoretical analysis owing to the output voltageconsidered as a constant voltage source.Fig. 7 shows the efficiency chart at different output power.

One point that needs to be clarified is that the efficiency graphincludes the power supply generation of the control circuit.The maximum efficiency is 96.12%, and the efficiency is95% at full load.According to the datasheet of the rectifier diode, the

forward conduction voltage (VF) can be estimated as

VF = 1.65+ 0.0182Io (11)

Therefore the output voltage can be estimated as

Vo = nVin − Rds(on) + Rcu

( )n2Io − VF = 6× 60

− (0.023+ 0.054)× 62[ ]

Io− 1.65+ 0.0182Io( )=358.35−2.79Io ≃ 358.35− 2.79Po

360(12)

Fig. 8 shows the output voltage regulation. The output voltagedecreases linearly with the increase of the output power from(12). From Fig. 8, the output voltage in theoretical analysis islower than that in experimental results at low power. Themain reason is as follows: the capacitor Cf can be fullycharged and can provide the load current at low power, sothe current through Rcu and Rds(on) in experimental results issmaller than that in (12). However, the output voltage intheoretical analysis is higher than that in experimentalresults at high power. The main reason is that the voltage

Fig. 7 Efficiency chart of the proposed converter


across the leakage inductance is omitted in (12). The outputvoltage decreases to 348.8 V at full load in experimentalresults, which is larger than the maximum output voltage ofthe inverter (seen from Fig. 1), that is,

��2

√ × 220 = 311V.Therefore, the proposed unregulated isolated step-up DC–DC converter can provide sufficient input voltage for theinverter.

6 Conclusion

This paper has presented a family of ZVS unregulatedisolated step-up DC–DC converters. Take the ZVS PPFunregulated DC–DC converter as an example, the operatingprinciple has been analysed. The ZVS-oriented parameterdesign and example have been presented in detail. Theexperimental results from a 1 kW ZVS PPF unregulatedDC–DC converter confirm the theoretical analysis. TheZVS unregulated isolated step-up DC–DC converters canbe widely applied to uninterruptible power supply,distributed generation, aeronautical static inverter and powerfactor correction, etc.

7 Acknowledgment

This work was supported in part by the Qing Lan Project, bythe Natural Science Foundation of the Jiangsu HigherEducation Institutions of China under Grant 12KJB470013,by the National High Technology Research andDevelopment Programme of China (863 Programme) underGrant 2011AA11A249, and by the Provincial Scientific andTechnological Innovation and Technology Transfer Project

Fig. 8 Output voltage regulation


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under Grant BY2012021 and by the National Nature ScienceFoundation of China under Grant 51107108.
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Family of zero-voltage-switching unregulated isolated step-up DC–DC converters

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