IEEJ Journal of Industry Applications Vol.4 No.3 pp.116–125 DOI: 10.1541/ieejjia.4.116 Paper Harmonics Reduction Control for the Input Current of Electrolytic Capacitor-less High-Power-Factor Inverter for IPMSM Kodai Abe ∗ Student Member, Hitoshi Haga ∗ Member Kiyoshi Ohishi ∗ Senior Member, Toshio Hiraide ∗ Member (Manuscript received April 4, 2014, revised Dec. 4, 2014) This paper describes an inverter control method for reducing the input current harmonics and vibration in an elec- trolytic capacitor-less inverter for an interior permanent magnet synchronous motor (IPMSM). Typically, the back elec- tromotive force (EMF) on an IPMSM is not sinusoidal and contains harmonics caused by the rotor structure, which leads to harmonic distortion in the motor control system and generation of harmonics in the input current. Moreover, the input current vibration occurs at the source side, which is caused by resonance with the line impedance and the DC-link capacitor. This paper proposes three control methods to reduce the current harmonics distortion and vibration. The first method filters out the harmonics of the feedback d-q axis current using a notch filter. The second method compensates the d-q axis voltage references to reduce the input current harmonics. To prevent the input current vibration, the third method adds a cancellation voltage pulse to the three-phase voltage reference of the inverter. The superior performance of these proposed methods are demonstrated through experimental results. The experimental results confirm that the proposed methods meet guideline EN61000-3-2. Keywords: electrolytic capacitor less inverter, interior permanent magnet synchronous motor, input current harmonic, input current vibration, LC resonance 1. Introduction Recently, an inverter-driven AC motor system has been widely used in many fields, and its applications range from home appliances to industrial applications from the view- point of global environmental problems (1)–(3) . Typically, con- trolling the compressor rotating speed in a residential air- conditioner application with an inverter-driven AC motor system would allow for overall system optimization, which could significantly reduce energy consumption. However, the inverter system requires an improvement in the input power factor and the input current waveform of the AC sources. Therefore, home appliances have a power factor correction (PFC) circuit in the rectifier. The purpose of PFC is to im- prove the input current waveform and to obtain a constant DC-link voltage (4)–(6) . Figure 1 shows a conventional PFC circuit. The conven- tional PFC technique requires the use of high-capacitance electrolytic capacitors, heavyweight reactors, and expensive power switching devices (e.g., IGBTs). The electrolytic ca- pacitors in the inverter circuit occupy a large volume, which prevents miniaturization of the circuit and limits its lifetime. Reactors and switching devices significantly contribute to- ward an increase in power loss, weight of the system, and cost. To further conserve energy and resources of the com- pressor drive system, PFC circuits need to be reduced in size and weight and have high efficiency (7)–(10) . ∗ Nagaoka University of Technology 1603-1, Kamitomioka-machi, Nagaoka, Niigata 940-2188, Japan Fig. 1. Conventional power converter The authors have proposed an electrolytic capacitor less single-phase to three-phase power converter for compressor motor applications (11)–(14) . The proposed system consists of a single-phase diode rectifier, a low-capacitance film capaci- tor at the DC link, a three-phase voltage-source inverter, and an IPMSM. According to previous reports (11) (12) , high-power- factor operation has been achieved under many load condi- tions from a low speed to a high speed. However, there are a large number of current harmonics at the AC-source side; the input current vibration still remains at the source side, which is caused by resonance with the line impedance and a low- capacitance capacitor at the DC link. Therefore, the system could not meet guideline EN61000-3-2. This paper details three control methods for reducing the input current harmonics and vibration of the proposed power converter. First, the cause of the input current harmonics and vibration in the proposed power converter is clarified. To overcome these problems, this paper proposes control meth- ods for reducing the current harmonic distortion. The first method filters out the harmonics of the feedback d-q axis c 2015 The Institute of Electrical Engineers of Japan. 116
Transcript
IEEJ Journal of Industry ApplicationsVol.4 No.3 pp.116–125 DOI: 10.1541/ieejjia.4.116
Paper
Harmonics Reduction Control for the Input Current ofElectrolytic Capacitor-less High-Power-Factor Inverter for IPMSM
Kodai Abe∗ Student Member, Hitoshi Haga∗ Member
Kiyoshi Ohishi∗ Senior Member, Toshio Hiraide∗ Member
(Manuscript received April 4, 2014, revised Dec. 4, 2014)
This paper describes an inverter control method for reducing the input current harmonics and vibration in an elec-trolytic capacitor-less inverter for an interior permanent magnet synchronous motor (IPMSM). Typically, the back elec-tromotive force (EMF) on an IPMSM is not sinusoidal and contains harmonics caused by the rotor structure, whichleads to harmonic distortion in the motor control system and generation of harmonics in the input current. Moreover,the input current vibration occurs at the source side, which is caused by resonance with the line impedance and theDC-link capacitor.
This paper proposes three control methods to reduce the current harmonics distortion and vibration. The first methodfilters out the harmonics of the feedback d-q axis current using a notch filter. The second method compensates the d-qaxis voltage references to reduce the input current harmonics. To prevent the input current vibration, the third methodadds a cancellation voltage pulse to the three-phase voltage reference of the inverter. The superior performance of theseproposed methods are demonstrated through experimental results. The experimental results confirm that the proposedmethods meet guideline EN61000-3-2.
Keywords: electrolytic capacitor less inverter, interior permanent magnet synchronous motor, input current harmonic, input currentvibration, LC resonance
1. Introduction
Recently, an inverter-driven AC motor system has beenwidely used in many fields, and its applications range fromhome appliances to industrial applications from the view-point of global environmental problems (1)–(3). Typically, con-trolling the compressor rotating speed in a residential air-conditioner application with an inverter-driven AC motorsystem would allow for overall system optimization, whichcould significantly reduce energy consumption. However, theinverter system requires an improvement in the input powerfactor and the input current waveform of the AC sources.Therefore, home appliances have a power factor correction(PFC) circuit in the rectifier. The purpose of PFC is to im-prove the input current waveform and to obtain a constantDC-link voltage (4)–(6).
Figure 1 shows a conventional PFC circuit. The conven-tional PFC technique requires the use of high-capacitanceelectrolytic capacitors, heavyweight reactors, and expensivepower switching devices (e.g., IGBTs). The electrolytic ca-pacitors in the inverter circuit occupy a large volume, whichprevents miniaturization of the circuit and limits its lifetime.Reactors and switching devices significantly contribute to-ward an increase in power loss, weight of the system, andcost. To further conserve energy and resources of the com-pressor drive system, PFC circuits need to be reduced in sizeand weight and have high efficiency (7)–(10).∗ Nagaoka University of Technology
The authors have proposed an electrolytic capacitor lesssingle-phase to three-phase power converter for compressormotor applications (11)–(14). The proposed system consists of asingle-phase diode rectifier, a low-capacitance film capaci-tor at the DC link, a three-phase voltage-source inverter, andan IPMSM. According to previous reports (11) (12), high-power-factor operation has been achieved under many load condi-tions from a low speed to a high speed. However, there are alarge number of current harmonics at the AC-source side; theinput current vibration still remains at the source side, whichis caused by resonance with the line impedance and a low-capacitance capacitor at the DC link. Therefore, the systemcould not meet guideline EN61000-3-2.
This paper details three control methods for reducing theinput current harmonics and vibration of the proposed powerconverter. First, the cause of the input current harmonics andvibration in the proposed power converter is clarified. Toovercome these problems, this paper proposes control meth-ods for reducing the current harmonic distortion. The firstmethod filters out the harmonics of the feedback d-q axis
Harmonics Reduction Control for Electrolytic Capacitor-less Inverter（Kodai Abe et al.）
current by using a notch filter to improve the input currentharmonics (13). The second method compensates the d-q axisvoltage references to reduce the input current harmonics. Thed-q axis compensation voltages are obtained by a feed for-ward controller (14). The third method adds a cancellation volt-age pulse to prevent the input current vibration. The cancel-lation voltage pulse is added to the three-phase voltage ref-erence of the inverter. The experimental results confirm thatthe proposed methods improve the input current waveformand maintain high power factor control.
2. Electrolytic Capacitor less Inverter for anIPMSM Drive System
2.1 Conventional Single-Phase PFC to Three-PhasePower Converter Figure 1 shows a conventional single-phase to three-phase power converter that controls the inputpower factor and the output voltage regulation. The inputcurrent waveform is controlled by the boost chopper. Theboost chopper obtains a sinusoidal current using a reactor, anelectrolytic capacitor, and a high-frequency switched powerdevice. In the system, the reactor and capacitor are bulky andexpensive, and the power loss in the reactor is large. The ca-pacitor determines the lifetime-limiting factor of the conven-tional power converter. Moreover, the loss in the boost chop-per is large in conventional systems and requires a noise fil-ter. In a conventional system, source-side single-phase powershould smooth at the DC link through the use of a large-capacitance electrolytic capacitor. In this manner, the invertersupplies constant power to the motor.2.2 High-Power-Factor Operation in an Electrolytic
Capacitor less Power Converter Figure 2 shows the pro-posed power converter (11) (12), which consist of a single-phasediode rectifier, a low-capacitance film capacitor at the DClink, and a three-phase voltage source inverter. The capac-itor is used to absorb the DC-link current ripple due to thePWM of the inverter. The proposed system has few energystorage devices in the power converter. Thus, source-sidesingle-phase power is provided to the IPMSM directly. Theripple power is smoothed by the moment of inertia of theIPMSM; therefore, torque ripples are present in the IPMSM,and the motor speed has some ripples and is controlled aver-agely. The proposed power converter can be usefully appliedto a residential compressor drive system that does not requirespeed response with high precision.
In the proposed system, the inverter controls both the motorspeed and the source-side current waveform to obtain high-power-factor operation. Figure 3 shows the control block di-agram of the inverter-output-power controller. The controlmethod using the inverter output power is applied as the high-power-factor control method (12). The inverter output poweris controlled with the twice-synchronized source voltage fre-quency. The inverter output power reference p∗inv is calcu-lated by subtracting the compensation power of the DC-linkcapacitor from the input power reference p∗in, which is de-termined by multiplying the output of the motor speed con-troller with sin 2ωint generated from the input voltage vin. TheDC-link capacitor power pc is calculated by using vin. Thedifference between p∗inv and pinv is controlled by the inverter-output-power controller, and its output is the q-axis currentreference i∗q. By using the control method, high-power-factor
Fig. 2. Proposed power converter
Fig. 3. Control block diagram of inverter output powercontrol
operation is achieved under many load condition from a lowspeed to a high speed.
3. Input Current Harmonics and Vibration ofProposed Power Converter
In a conventional inverter system with a large electrolyticcapacitor and a PFC circuit, the PFC circuit controls the inputcurrent waveform at the source side. The inverter switchingpattern and motor operation do not influence the source sidebecause of the large electrolytic capacitor.
However, the input current waveform of the proposed elec-trolytic capacitor less power converter is controlled by theinverter and affected by the motor side harmonics because ofthe small DC-link capacitor. Moreover, input current vibra-tion occurs because of resonance with the line impedance andthe DC-link capacitor. This chapter clarifies the influence ofmotor-side harmonics and LC resonance on the input currentharmonics and vibration.3.1 Spatial Harmonics of IPMSM The back EMF
of an IPMSM has spatial harmonics with amplitude and fre-quency proportional to the motor speed because the shapeof the rotor is not a sine wave. Figure 4 shows the backEMF waveform of the tested IPMSM. Figure 5 shows theFFT analysis result of the back EMF. The largest harmoniccontent is the fifth followed by the seventh. The motor-sideharmonics affect the input current waveform.3.2 Influence of Spatial Harmonics for d-q Axis Cur-
rent The spatial harmonics cause disturbances in the d-qaxis current controller. The distortion of the d-q axis currentincreases the harmonics of the input current. This sectionshows the influence that the spatial harmonics of the IPMSMhave on the d-q axis current controller.
The fifth harmonic included in the three-phase current iu5,iv5, and iw5 can be expressed as⎡⎢⎢⎢⎢⎢⎢⎢⎣
iu5
iv5iw5
⎤⎥⎥⎥⎥⎥⎥⎥⎦ = ωeψc
⎡⎢⎢⎢⎢⎢⎢⎢⎢⎣sin 5θe
sin(5(θe − 23π))
sin(5(θe +23π))
⎤⎥⎥⎥⎥⎥⎥⎥⎥⎦ · · · · · · · · · · · · · · · (1)
where, ωe is the rotor electrical angular speed [rad/s], θe isthe motor position [rad], and ψc is the back EMF coefficient.
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The current harmonics in the d-q axis frame can be expressedas
[idh
iqh
]=
√23
[C]
⎡⎢⎢⎢⎢⎢⎢⎢⎣iu5
iv5iw5
⎤⎥⎥⎥⎥⎥⎥⎥⎦
=
√32ωeψc
[sin 6θe
cos 6θe
]· · · · · · · · · · · · · · · · · · (2)
where,
[C]=
⎡⎢⎢⎢⎢⎣ cos θe cos(θe− 23π) cos(θe +
23π)
− sin θe − sin(θe− 23π) − sin(θe+
23π)
⎤⎥⎥⎥⎥⎦ · · · · · · (3)
Hence, the harmonic current appears as the sixth harmonic inthe d-q axis current controller.3.3 Input Current Vibration in the Proposed Power
Converter Figure 6 shows the input current vibration inthe proposed power converter. In the proposed system, theDC-link capacitor has a low capacitance, and the input cur-rent waveform is controlled by the inverter. When there isline impedance at the source side, input current vibration oc-curs there. The vibration is caused by resonance with the lineimpedance and the DC-link capacitor when a single-phase
Fig. 4. Back EMF waveform of tested IPMSM
Fig. 5. Frequency analysis of back EMF
Fig. 6. Input current vibration in the proposed powerconverter
diode rectifier with nonlinear characteristics is turned on. Theturn-on-timing of the single-phase diode rectifier is varied byan amount of the charge of DC-link capacitor. The charge isoccured by the regeneration in the DC-link capacitor becausethe DC-link voltage decrease to almost zero. Then, the am-plitude and generation timing of the input current vibrationis also changed. The current vibration causes current distor-tions and a lower power factor. For example, if the line induc-tance is 0.2 mH, and the capacitance of the DC-link capacitoris 14 μF, the vibration frequency of the input current is ap-proximately 3 kHz. In the proposed system, the input powerfactor is controlled by the inverter. Normally, the switchingfrequency of the inverter is approximately 16 kHz. Therefore,it is difficult to apply conventional dumping feedback controlfor vibration restraint of the input current.
4. Proposed Control Method
4.1 Current Harmonics Reduction Method usingHarmonics Filter Figure 7 shows the control block dia-gram of the first method that uses a notch filter. In the pro-posed power converter, the input current is controlled by theinverter because of a small DC-link capacitor. Therefore,if the voltage references include the harmonics due to spa-tial harmonics of an IPMSM, the input current harmonics in-crease. The first method filters out the harmonics in the d-qaxis voltage references and prevents input current harmonicscaused by the voltage references distortion. The harmonicsnewly cause harmonics at other frequency through PI con-troller as described in section 4.5. The harmonics filter alsoreduces the influence on other frequency. Then, the methodclarifies influence on input current caused by spatial harmon-ics; thereby the second method to be described below can beapplied effectively. In this study, the filter is designed for thefifth harmonic shown in Fig. 5. Hence, the targeted harmonicorder is the sixth in the d-q axis frame; the filter is designedto the sixth order.4.2 Current Harmonics Reduction Method using
Compensation Voltages Figure 8 shows the controlblock diagram of the second method that uses d-q axis com-pensation voltages. The second method reduces the har-monic content caused by the spatial harmonics of the mo-tor. The proposed method compensates the d-q axis voltagereferences to reduce the input current harmonics. The d-qaxis compensation voltages vdcmp and vqcmp are obtained by
Fig. 7. Control block diagram of harmonics filter
Fig. 8. Control block diagram of compensation voltages
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a feed forward controller. The compensation voltages vdcmp
and vqcmp are calculated by using Eq. (4) and Eq. (5).
where, n is the targeted harmonic order number. In this study,the compensation voltages are designed for the fifth harmon-ics shown in Fig. 5. Hence, n is set at 6. φd and φq are thephase differences between the electrical rotor position θe ob-tained from the encoder. Kd and Kq are the gains of the com-pensation voltages. In this study, φd, φq, Kd and Kq are ob-tained through an off-line experimental test.4.3 Current Vibration Reduction Method using Can-
cellation Voltage Figure 9 shows the principles of theproposed current vibration reduction method. Figure 10shows the control block diagram of the third method that usesa cancellation voltage. In the proposed system, the vibrationon the input current iin and the DC-link voltage vdc occurswhen the diode of the rectifier is turned on. The third methodcancels the vibration of vdc by changing the amount of dis-charge in the DC-link capacitor. The amout of discharge ischanged in accordance with the turn-on time of switching de-vices. Hence, the input current vibration is reduced. In orderto prevent the current vibration, the proposed method adds acancellation voltage pulse vcs to the three-phase voltage ref-erence for the inverter as follows:
where A is the amplitude of the cancellation voltage. Thefrequency of the cancellation voltage is equal to the reso-nant frequency fr = 3 kHz. For this reason, sampling fre-quency should be set at higher than five times the frequencyof the resonance in order to decouple the effects by sampling.Moreover, only one cycle of the cancellation voltage is addedto the three-phase voltage reference. In this study, the can-cellation voltage is added to one phase. The phase x is deter-mined by the amplitude of the three-phase reference voltages.
Fig. 9. Principle of current vibration reduction method
Fig. 10. Control block diagram of cancellation voltage
The phase is the middle-voltage phase of the three phases be-cause there is a case in which other phase reference voltagesare saturated by the DC-link ripple voltage. In Eq. (6), sgn(ix)implies that the phase of the cancellation voltage refers to thepolarity of the x-phase motor current ix. When the polar-ity of ix is positive, the signal has a value of −A sin(2π frt).When the polarity of ix is negative, the signal has a value ofA sin(2π frt). Thus, the current vibration is canceled withoutchanging the added phase. In this study, A and the timing ofthe addition are obtained through an offline experimental testso as to be the minimum THD of the input current.4.4 Proposed Control Block Diagram Figure 11
shows the diagram of the entire circuit with the proposedcontrol method using the cancellation voltage in the inverter-output-power controller. Control sensors are installed for thesource side vin, the DC-link voltage vdc, and the motor cur-rents iu and iv. The IPMSM is driven at the voltage refer-ences vre f
u , vre fv , and vre f
w obtained by v∗u, v∗v , v∗w, and vcs. vcs
is obtained by using v∗u, v∗v , v∗w, iu, and iw. The voltage refer-ences v∗u, v∗v , and v∗w are from the output of the d-q axis currentregulator. For the rotor speed control, the error between thespeed reference ω∗m and the calculated speed ωm is the inputto the PI controller; ωm is calculated by differentiating therotor position obtained from the encoder. The output of thespeed PI controller corresponds to the peak value of the inputpower required by the proposed system. The DC-link voltageis increased because the electromotive force increases whenthe motor speed increases. As a result, the input power fac-tor decreases, and the current harmonic components increase.To solve this problem, the d-axis current is provided by theflux-weakening control method, and the d-axis current refer-ence i∗d is constant. The currents id, and iq are obtained bycoordinate conversion from the detected currents iu and iw,respectively. These currents are subtracted from the refer-ence currents i∗d and i∗q, and each current is regulated by thePI controller. The voltage references are generated by thedecoupled control outputs of the PI controllers.
In the proposed control system, the DC-link voltage variessynchronously with the source voltage. Because of the DC-link ripple voltage, the outputs of the PI controllers for d- andq-axis currents are saturated. As a result, the input current iinincludes harmonic distortions. To improve the input currentwaveform, a PI control method is used that considers the volt-age limitations of the inverter for the d- and q-axis voltages.Figure 11 also shows the control block diagram of the PI cur-rent controller with saturation feedback of the d- and q-axisvoltages. When the output of the PI controller is saturated, itis calculated again using the values Δvd and Δvq obtained un-der saturated conditions. The voltage limitation value variessynchronously with the DC-link voltage vdc. Therefore, theproposed PI current controller operating with feedback usingthe values obtained at saturation improves the input currentwaveform.4.5 Design of d-q Axis Current Controller In the
proposed electrolytic capacitor less power converter, the per-formance of the d-q axis current controller influences the in-put current harmonics. This section presents a design methodfor the d-q axis current controller to improve the input currentwaveform. In this study, the d-q axis proportional gains Kdp
and Kqp are designed as follows:
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Fig. 11. System configuration with proposed control methods
where, ωc is the cutoff frequency of the current controller.Kdp is not equal to Kqp because of Ld and Lq. The harmoniccontent of the motor current passes this current controller andbecomes the inverter voltage reference. The harmonic com-ponents of the d-q axis voltage reference v∗dh, v∗qh are calcu-lated by using Eq. (2), Eq. (7) and Eq. (8).
[v∗dhv∗qh
]=
√32ωeψc
[Kdp sin 6θe
Kqp cos 6θe
]
=
√32ωeωcψc
[Ld sin 6θe
Lq cos 6θe
]· · · · · · · · · · · · · (9)
Then, the harmonics in the u-v-w axis frame can be expressedas,⎡⎢⎢⎢⎢⎢⎢⎢⎣v∗uhv∗vhv∗wh
Eq. (10) shows that there are harmonics of the fifth and theseventh order. The seventh-order harmonics are caused by thedifference between Kdp and Kqp. The harmonic content of theinverter output voltage increases with the cutoff frequency ofthe d-q axis current controller ωc. The decoupling controlleralso causes harmonics except for the fifth-order harmonics.Hence, the proposed electrolytic capacitor less power con-verter should design the cutoff frequency of the d-q axis cur-rent controller to have a low frequency.
5. Experimental Results
Table 1 lists the parameters of the tested motor. The ex-perimental results validate the proposed control method. Thesource voltage and frequency are set at 200 Vrms and 50 Hz,respectively. The DC-link capacitance is 14 μF, and the sam-pling frequency is 16 kHz. The line impedance is set at0.2 mH and 0.5Ω. The motor speed is set at 4500 rpm and1500 rpm, and the load torque is set at 1.8 Nm. In this study,in order to apply the cancellation voltage effectively, it is nec-essary to impart a periodicity to the input current vibration.Therefore, the proposed control method is especially effec-tive when the output frequency of the inverter is set as themultiples of the power supply frequency. Thus, the inverteroutput voltages are synchronized to the input voltage and theinput current vibration is the same waveform every period.The bandwidth of the current regulator is set at 1500 rad/s.
Figures 12–15 show the experimental results of the con-ventional control method, which does not use any proposedmethod, and the proposed control method at 4500 rpm. Fig-ure 16 shows the FFT analysis result of the input current at4500 rpm in the proposed power converter.
Figure 12 shows the experimental results of the conven-tional control method. The control method involves onlythe inverter output power controller in Fig. 3, and the d-axis
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current reference is constant. The conventional methodcauses harmonic distortion on voltage references v∗d and v∗qand the input current.
Figure 13 shows the results of the proposed control methodwith the harmonics filter. The harmonics filter is designed toreduce the target sixth-order harmonic component. The firstmethod reduces the harmonic distortion of the voltage refer-ences in Fig. 13. Therefore, the twenty-first harmonic of theinput current due to voltage reference distortion is reducedby 76.0% in Fig. 16. However, the influence of the harmon-ics in the motor side is clear, and the seventeenth and nine-teenth harmonics of the input current are increased, as shownFig. 16. Figure 14 shows the results of the proposed controlmethod with the harmonics filter and the compensation volt-ages. The compensation voltages are designed for the targetsixth-order harmonic to reduce the input current harmonicsdue to the motor-side harmonics. The second method reducesthe input current harmonics in Fig. 14. Figure 16 shows theseventeenth and nineteenth harmonics are reduced by 21.9%and 52.4%, respectively. These control methods meet guide-line EN61000-3-2 in Fig. 16. However, the vibration on theinput current and the DC-link voltage due to the resonancestill occurs. In Fig. 14, the pulsation on DC-link voltageis occurred near 100 V separately from resonance vibration.The pulsation is caused by regeneration because the proposedsystem uses a low-capacitance film capacitor at DC link and
(a) Voltage and current waveform at source side
(b) d-q axis voltage references
Fig. 13. Proposed control method with harmonics filterat 4500 rpm
DC-link voltage decrease to almost zero. Figure 15 shows theresults of the proposed control method with the harmonicsfilter, the compensation voltages, and the cancellation volt-age. In the experimental results, the middle voltage phaseis a v-phase, and the cancellation voltage is added to the v-phase. The third method prevents current vibration by usingthe cancellation voltage pulse in comparison of Fig. 14(a) andFig. 15(a). Although the proposed power converter has motorcurrent ripple, the motor speed can be controlled averagely.
Table 2 summarizes the input power factor and THD of theinput current waveform at 4500 rpm. When the harmonicsfilter is applied, the twenty-first harmonic is reduced, but theseventeenth and nineteenth harmonics are increased as shownFig. 16. Therefore, the power factor and THD with harmon-ics filter are worse than those with the conventional method.By combining the three proposed methods, the power factorand THD of the input current are improved effectively.
Figures 17–20 show the experimental results of the con-ventional control method and the proposed control methodat 1500 rpm. Figure 21 shows the FFT analysis result of theinput current at 1500 rpm.
Figures 17–19 show the experimental results of the con-ventional control method, the proposed method with the har-monics filter and the proposed method with the harmonicsfilter and the compensation voltages. In Fig. 17, the input cur-rent harmonics at 1500 rpm are small because the back EMFis small in a low speed and the influence caused by spatialharmonics decreases. Therefore, the effect of the harmon-ics filter and the compensation voltages decrease as shownin Figs. 18, 19, and the input current using only the conven-tional method meet guideline EN61000-3-2 in Fig. 21. Fig-ure 20 shows the experimental results of the proposed methodwith the harmonics filter, the compensation voltages and the
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(a) Voltage and current waveform at source side
(b) d-q axis voltage references
(c) DC-link voltage
Fig. 14. Proposed control method with harmonics filterand compensation voltages at 4500 rpm
Table 2. Input power factor and THD of input current at4500 rpm
P.F. [%] THD [%]Conventional method 98.06 18.02
Proposed method with filter 98.01 18.79Proposed method with filter, vdcmp and vqcmp 98.47 16.65
Proposed method with filter, vdcmp, vqcmp and vcs 98.51 16.56
cancellation voltage. The input current vibration due to res-onance do not relate to spatial harmonics and still remain.Hence, the effect of the third method does not change. Thecancellation voltage pulse prevents the current vibration inFigs. 20, 21. In the experimental results, the middle voltagephase is a w-phase, and the cancellation voltage is added tothe w-phase.
Table 3 summarizes the input power factor and THD of theinput current waveform at 1500 rpm. There is small improve-ment in the input power factor and the THD with the harmon-ics filter and the compensation voltages because the influenceof spatial harmonics decrease in a low speed. Since there isthe input current vibration due to resonance, the cancellationvoltage improve the input power factor and THD.
(a) Voltage and current waveform at source side
(b) DC-link voltage
(c) cancellation voltage
(d) motor current
Fig. 15. Proposed control method with harmonics fil-ter, compensation voltages and cancellation voltage at4500 rpm
Table 3. Input power factor and THD of input current at1500 rpm
P.F. [%] THD [%]Conventional method 91.51 30.38
Proposed method with filter 91.52 30.36Proposed method with filter, vdcmp and vqcmp 91.52 30.35
Proposed method with filter, vdcmp, vqcmp and vcs 91.57 28.50
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Fig. 16. FFT analysis result of the input current at 4500 rpm
(a) Voltage and current waveform at source side
(b) d-q axis voltage references
Fig. 17. Conventional control method at 1500 rpm
6. Conclusion
This paper proposed a new PFC method using an inverter-driven IPMSM. The proposed circuit realizes the high powerfactor of the single-phase diode rectifier by using three-phaseinverter control and a load-side IPMSM. The proposed sys-tem consists of a diode rectifier, a three-phase inverter, anda small film capacitor at the DC-link. The DC-link capaci-tance is 14 μF (0.33 J/kVA). It consists of several energy stor-age elements, and the single-phase power ripple is smoothedby the moment of inertia of the load motor. In addition,this paper proposed three control methods to reduce the in-put current harmonics and vibration of the proposed powerconverter. The first control method filters out the harmon-ics from d-q axis current feedback by using a notch filter.Thus, the input current harmonics caused by the influence of
(a) Voltage and current waveform at source side
(b) d-q axis voltage references
Fig. 18. Proposed control method with harmonics filterat 1500 rpm
voltage reference harmonics are reduced. The second con-trol method compensates the d-q axis voltage references toreduce the input current harmonics due to the spatial harmon-ics of the motor. These two control methods meet guidelineEN61000-3-2. Moreover, the third control method improvesthe input current vibration due to the resonance with the lineimpedsance and the DC-link capacitor. Future work lies inthe examination of calculating the cancellation voltage on-line.
The input power factor and THD of the input currentare improved by the proposed control method. The maxi-mum power factor obtained by using the proposed methodis 98.51% and the THD is 16.56%. The effectiveness of theproposed method has been verified through experiments incomparison with the conventional method.
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Harmonics Reduction Control for Electrolytic Capacitor-less Inverter（Kodai Abe et al.）
(a) Voltage and current waveform at source side
(b) d-q axis voltage references
(c) DC-link voltage
Fig. 19. Proposed control method with harmonics filterand compensation voltages at 1500 rpm
References
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(a) Voltage and current waveform at source side
(b) DC-link voltage
(c) cancellation voltage
(d) motor current
Fig. 20. Proposed control method with harmonics fil-ter, compensation voltages and cancellation voltage at1500 rpm
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124 IEEJ Journal IA, Vol.4, No.3, 2015
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Fig. 21. FFT analysis result of the input current at 1500 rpm
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Method for Electrolytic Capacitor-less Inverter for IPMSM Drive System”,IEEE Int. Symp. Ind. Electron. (ISIE) (2013)
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Kodai Abe (Student Member) received the B.S. degree in ProductionSystems Engineering from the Akita National Col-lege of Technology, Akita, Japan in 2013. Now heis a candidate of the M.S. degree in Electrical, Elec-tronics and Information Engineering from NagaokaUniversity of Technology, Nagaoka, Japan. His re-search interests include power electronics.
Hitoshi Haga (Member) received B.S., M.S. and D.Eng. degrees inenergy and environmental science from the NagaokaUniversity of Technology, Nagaoka, Japan, in 1999,2001, and 2004, respectively. From 2004 to 2007, hewas a Researcher with Daikin Industries, Ltd., Osaka,Japan. From 2007 to 2010, he was an Assistant Pro-fessor with the Sendsai National College of Technol-ogy, Sendai, Japan. Since 2010, he has been with theDepartment of Electrical Engineering, Nagaoka Uni-versity of Technology. His research interests include
power electronics.
Kiyoshi Ohishi (Senior Member) received the B.S., M.S., and Ph.D.degrees in electrical engineering from Keio Univer-sity, Yokohama, Japan, in 1981, 1983, and 1986, re-spectively. From 1986 to 1993, he was an AssociateProfessor with Osaka Institute of Technology, Osaka,Japan. From 1993 to 2003, he was an Associate Pro-fessor with Nagaoka University of Technology, Ni-igata, Japan. Since August 2003, he has been a Pro-fessor at the same university. He is an administrationcommittee member of the IEEE Industrial Electronics
Society, the Institute of Electrical Engineers of Japan (IEEJ), the Japan Soci-ety of Mechanical Engineers (JSME), the Society of Instrument and ControlEngineers (SICE), and the Robotics Society of Japan (RSJ).
Toshio Hiraide (Member) received the B.S. and M.S. degree in elec-trical engineering from Nagaoka University of Tech-nology, Nagaoka, Japan, in 2011 and 2013, respec-tively. He is currently with Sanyo Denki Co., Ltd.,Ueda, Japan. His research interests include motordrive and power electronics.
