High Step-Up Converter Based on Charge Pump and Boost Converter

2484 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 5, MAY 2012

High Step-Up Converter Based on Charge Pumpand Boost Converter

K. I. Hwu, Member, IEEE, and Y. T. Yau, Student Member, IEEE

Abstract—In this paper, a novel step-up converter is presented,where the charge pump concept, combined with the traditionalboost converter structure, is utilized. Although two inductors areused in such a converter, the difference in value between thetwo inductors affects the regulation performance of this converterslightly. Furthermore, the behavior of this converter is similar tothe traditional boost converter, and hence, the control of this con-verter can be realized easily. Above all, the energy stored in thetwo inductors, connected in series with the energy stored in thecharge pump capacitor and the input voltage, is released to theload during the demagnetization period. In this paper, the basicoperating principles of the proposed converter are presented alongwith some experimental results to demonstrate the effectiveness ofthis converter.

Index Terms—Boost converter, charge pump, demagnetizationperiod, high step-up converter.

I. INTRODUCTION

A S generally recognized, step-up converters have beenwidely used in many applications, such as battery-

powering device, uninterruptible power supply (UPS), photo-voltaic (PV) system, etc., requiring some circuits transferringlow voltages to high voltages used as input voltages for dc–acconverters.

Up to now, there have been many researches on how to getcircuits with high voltage conversion ratios, based on severalconverters with output voltages connected in series [1], [2]or the coupling inductor concept [3]–[8] or on the chargepump concept [9]–[18] or even on the last two concepts com-bined [19]–[24]. The step-up converters mentioned before havesome demerits. For example, in [4]–[9], [11], [15]–[17], [20],and [23], there are many complicated circuits presented, wheretwo switches or more and mass passive components cause con-version efficiency to be degraded. Only low-power applicationsare suitable in [3], [9], [11], [14]–[18], and [24]. Some switchesare floating in [4], [9], [10], [13]–[15], and [17], thereby causingadditional isolated gate driving circuits to be needed, and hence,making systems complex. The nonlinear relationships between

Manuscript received July 31, 2011; revised October 2, 2011; acceptedOctober 25, 2011. Date of current version February 27, 2012. This work wassupported by the National Science Council under Grant NSC 100-2628-E-027-004. Recommended for publication by Associate Editor Y. C. Liang.

The authors are with the Department of Electrical Engineering, Na-tional Taipei University of Technology, Taipei 10608, Taiwan (e-mail:[email protected]; [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2011.2175010

Fig. 1. Proposed step-up converter.

the input and output voltages exist due to high voltage conver-sion ratios in [3], [4], [11], [16], [17], and [19]–[21], therebycausing the system to be difficult to control.

According to the aforementioned, in this paper, a step-up con-verter with a high voltage conversion ratio is presented, whosetwo inductors are magnetized simultaneously, and, together withthe input voltage and the energy stored in the charge pump ca-pacitor, pump energy into the output. Such a circuit is simple,combining the charge pump concept with the traditional boostconverter structure. Although there actually exists the differencein value between the two inductors in this circuit topology, theproposed converter possesses a good and robust performance ofregulation, with the behavior similar to that of the traditionalboost converter and hence with easy control.

II. PROPOSED CONVERTER STRUCTURE

Fig. 1 shows the proposed step-up converter that contains twoinductors L1 and L2 , two diodes D1 and D2 , charge pump ca-pacitor Ce , one output capacitor Co , two MOSFET switches Q1and Q2 with two body diodes Db1 and Db2 connected in parallel,respectively, and one output resistor RL . The gate driving signalcreated from the digital control effort is employed to drive Q1that is used as a main switch, and the gate driving signal com-plementary to that for Q1 is utilized to drive Q2 that is used asa synchronous rectifier.

It is noted that the reason why the proposed circuit uses asynchronous rectifier is described next. As generally acknowl-edged, if the converter operates in discontinuous conductionmode (DCM) with the output voltage regulated using the outputdiode, the less the load is, the less the duty cycle. Hence, thecorresponding control is not so easy because a too small dutycycle is sensitive to the noise. For the convenience of controland analysis, a synchronous rectifier is used herein instead of theoutput diode, thereby causing that the duty cycle is not changedtoo much all over the load range. It is noted that the main switch

0885-8993/$26.00 © 2011 IEEE

HWU AND YAU: HIGH STEP-UP CONVERTER BASED ON CHARGE PUMP AND BOOST CONVERTER 2485

and the synchronous rectifier are driven only by one half-bridgegate driver, such as HIP2101, and this is very easy to implement.

Furthermore, one comparison is made between the convertershown in [25, Fig. 9] and the proposed converter. In the circuitshown in [25, Fig. 2], two inductors are magnetized simulta-neously, and the input voltage, together with the energy storedin two inductors, passes the energy to the output, whereas inthe proposed circuit, two inductors are also magnetized simul-taneously, but the input voltage, together with the energy storedin the charge pump capacitor and two inductors, passes the en-ergy to the output. Therefore, the former has a smaller voltageconversion ratio, corresponding to 2D/(1−D), than the latter,corresponding to 2/(1−D). It is noted that in [25], there is nodiscussion about what will happen if the value of L1 is not iden-tical to the value of L2 . If two inductances are different, then inthe condition of the same turn-ON time and the same voltageapplied, two inductors get magnetized. The moment the switchis turned OFF, the small inductance has a larger current than thebig inductance. And hence, at this instant, two inductors con-nected in series will conflict with the Kirchhoff’s current law(KCL).

On the other hand, the other comparison is made between theconverter shown in [26, Fig. 2] and the proposed converter. Inthe circuit shown in [26, Fig. 2], during the turn-ON period, twoinductors and one charge pump capacitor are magnetized andcharged simultaneously, and the input voltage, together with theenergy stored in two capacitors, pumps the energy into the out-put, whereas during the turn-OFF period, the energy stored intwo inductors and one charge pump capacitor connected in se-ries is released to these two capacitors. In the proposed circuit,two inductors are also magnetized simultaneously during theturn-ON period, but the input voltage, together with the energystored in two inductors and one charge pump capacitor, passesthe energy to the output during the turn-OFF period. That is,the voltage boosting in the former is based on the input volt-age connected in series with two series capacitors, whereas thevoltage boosting in the latter is based on the input voltage con-nected in series with two series inductors and one charge pumpcapacitor. Therefore, the former has a voltage conversion ratioof (3+D)/(1−D) that is larger than the voltage conversion ratioof 2/(1−D) for the latter. However, the former has more compo-nents than the latter, and the difference in number of componentsbetween the two is four.

III. BASIC OPERATING PRINCIPLES

The following description focuses on discussing the basicoperating principles of the proposed converter. The values ofL1 and L2 will affect the operating behavior of this converterremarkably. And hence, there are three cases to be describedas follows, with the assumption that the voltages across all theswitches and diodes are zero during the turn-ON period, Ce islarge enough to keep the voltage across Ce , ve is constant atthe input voltage vi , and the blanking times between the twoswitches are zero. It is noted that this converter always operatesin the continuous conduction mode (CCM). Hence, the currents

Fig. 2. Key waveforms relevant to case 1.

flowing through two inductors can be positive or negative, butthe corresponding average currents must be zero or positive.

Moreover, there are some symbols to be given as follows:1) the input voltage and current are signified by vi and ii ; 2) theoutput voltage is expressed by vo ; 3) the voltage and current inCe are represented by ie and ve ; 4) the currents flowing throughL1 and L2 are denoted by i1 and i2 ; 5) the currents in Q1 , Q2 ,D1 , and D2 are indicated by ids1 , ids2 , id1 , and id2 , respectively;and 6) the gate driving signals for Q1 and Q2 are signified byvgs1 and vgs2 , respectively.

A. Case 1

This is an ideal case under the assumption that the value of L1is equal to that of L2 . Fig. 2 shows the key waveforms relevantto this case, where Ts is the switching period. For analysisconvenience, let

L = L1 + L2 (1)

and

i1 = i2 = i. (2)

1) State 1 (t0∼t1): As shown in Fig. 3, Q1 is turned ON,Q2 is turned OFF, and D1 and D2 are forward biased. Since thevoltage across Ce is equal to vi , L1 and L2 are to be magnetized.During this state, the output energy required is supplied fromCo , and Ce is charged. Therefore, the corresponding differentialequations are

⎧⎪⎪⎪⎨

⎪⎪⎪⎩

Ldi

dt= 2vi

Codvo

dt= − vo

RLii = 2i + ie .

(3)


Fig. 3. Current flow in state 1 of case 1.


2) State 2 (t1∼t0+Ts): As depicted in Fig. 4, Q1 is turnedOFF, Q2 is turned ON, and D1 and D2 are reverse biased. Atthis moment, vi plus L1 and L2 releases energy to the load,thereby causing L1 and L2 to be demagnetized. Besides, Ce isdischarged. Therefore, the corresponding differential equationsare

⎧⎪⎪⎪⎨

⎪⎪⎪⎩

Ld i

dt= 2vi − vo

Codvo

dt= i − vo

RLii = i.

(4)

Prior to obtaining the average equations from (3) and (4), thereis a symbol 〈y〉 that is used to represent the average value of avariable y, where y indicates the voltage or current as follows:

〈y〉 =1Ts

∫ Ts

0y dτ . (5)

According to (3)–(5), the averaged equations can be obtainedto be

⎧⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎩

Ld 〈i〉dt

= 2 〈vi〉 − (1 − d) 〈vo〉

Cod 〈vo〉

dt= (1 − d) 〈i〉 − 〈vo〉

RL〈ii〉 = (1 + d) 〈i〉 + d 〈ie〉

(6)

where d is a variable denoting the duty cycle of the pulsewidthmodulation (PWM) control signal for Q1 .

Based on the ampere–second balance, 〈ie〉 can be expressedas a function of 〈i〉 to be

〈ie〉 =1 − d

d〈i〉 (7)

and hence, by substituting (7) into (6), (6) can be rewritten as⎧⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎩

Ld 〈i〉dt

= 2 〈vi〉 − (1 − d) 〈vo〉

Cod 〈vo〉

dt= (1 − d) 〈i〉 − 〈vo〉

RL〈ii〉 = 2 〈i〉 .

(8)

Prior to obtaining the small-signal ac model from (8), theperturbation and linearization of (8) are indispensable. First ofall, 〈y〉 is represented by the corresponding dc quiescent valueY plus the superimposed small ac variation y, along with theassumption that ac variation is small in magnitude compared tothe dc quiescent value. Let

⎧⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎩

〈vi〉 = Vi + vi

〈vo〉 = Vo + vo

〈ii〉 = Ii + ii

〈i〉 = I + i

d = D + d

with

⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

|vi | � Vi

|vo | � Vo∣∣∣ii

∣∣∣ � Ii

∣∣∣i

∣∣∣ � I

∣∣∣d

∣∣∣ � D.

(9)

Next, by substituting (9) into (8), the following equations areobtained:

⎧⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎩

Ld(I + i)

dt= 2(Vi + vi) − (1 − D + d)(Vo + vo)

Cod(Vo + vo)

dt= (1 − D − d)(I + i) − (Vo + vo)

RL

Ii + ii = 2(I + i).

(10)

Consequently, the dc quiescent equations from (10) can beobtained to be

⎧⎪⎪⎪⎨

⎪⎪⎪⎩

0 = 2Vi − (1 − D)Vo

0 = I(1 − D) − Vo

RL

Ii = 2I

(11)

and hence, the corresponding voltage conversion ratio of thisconverter from (11) can be obtained to be

Vo

Vi=

21 − D

. (12)

On the other hand, with the second-order ac terms neglected,the small-signal ac equations can be obtained to be

⎧⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎩

Ld i

dt= (3 + D)vi + (Vo + Vi)d − (1 − D)vo

Cod vo

dt= (1 − D)i − dI − vo

RL

ii = 2i.

(13)

And hence, the resulting small-signal ac model of the pro-posed high step-up converter with L1 equal to L2 is shown inFig. 5 according to (13), where T1 and T2 are the ideal trans-formers with the turns ratios of 1:2 and (1−D):1, respectively.Hence, by taking the Laplace transform of (13), the relationship


Fig. 5. Small-signal ac model for the proposed high step-up converter withL1 equal to L2 .

between vo(s), vi(s), and d(s) can be expressed to be

vo(s) = Goi(s)vi(s) + God(s)d(s) (14)

where

Goi(s) =vo(s)vi(s)

∣∣∣∣d(s)=0

=2/(1 − D)

1 + s(L/RL (1 − D)2) + s2(LCo/(1 − D)2)

(15)

God(s) =vo(s)

d(s)

∣∣∣∣∣v i (s)=0

=(Vo/1 − D) [1 − s(LI/Vo(1 − D))]

1 + s(L/RL (1 − D)2) + s2(LCo/(1 − D)2)

(16)

where Goi(s) is the input-to-output transfer function and God (s)is the control-to-output transfer function.

B. Case 2

This is an actual condition under the assumption that the valueof L1 is larger than that of L2 and this converter operates at ratedload. Fig. 6 shows the key waveforms pertaining to this case.In this case, as shown in Fig. 7, the duty cycle D multiplied bythe switching period Ts is defined as the turn-ON time of theswitch Q1 , tON , and Ts minus tON is defined as the turn-OFFtime of the switch Q1 , tOFF , corresponding to (1−D)Ts . If thevalue of L1 is identical to the value of L2 , then iL 1 is equal toiL 2 for any time. However, if the value of L1 is not identical tothe value of L2 , say, the value of L1 is larger than L2 in this case,then as soon as the switch is turned OFF, iL 1 is smaller than iL 2 ,and hence, the corresponding time interval ATs gets started. Atthis instant, L2 goes to demagnetization immediately due to thevoltage across L2 being Vi minus Vo . At the same time, the KCLat the point X shown in Fig. 1 must be obeyed, and hence, D2is forced to turn ON, thereby causing L1 to be still magnetized.The moment iL 1 is equal to iL 2 , this time interval goes to theend. Hence, there are three operating states in this case, to bediscussed as follows.

1) State 1 (t0∼t1): As shown in Fig. 8, Q1 is turned ON,Q2 is turned OFF, and D1 and D2 are forward biased. Since thevoltage across Ce is equal to vi , L1 and L2 are to be magnetized.During this state, the output energy required is supplied fromCo , and Ce is charged. Besides, i1 is smaller than i2 due to L1

Fig. 6. Key waveforms pertaining to case 2.

Fig. 7. Classification of operating states for case 2.




larger than L2 .

⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

L1di1dt

= vi

L2di2dt

= vi

Codvo

dt= − vo

RL

ii = i1 + i2 + ie .

(17)

2) State 2 (t1∼t2): As shown in Fig. 9, Q1 is turned OFF, Q2is turned ON, and D1 is reverse biased. During this state, i1 issmaller than i2 , thereby causing D2 to be forward biased. Hence,L1 to be still magnetized but L2 to be demagnetized. Besides,Ce is discharged. As soon as i1 is equal to i2 , the operation goesto state 3.

⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

L1di1dt

= vi

L2di2dt

= vi − vo

Codvo

dt= i2 −

vo

RL

ii = i2 .

(18)

3) State 3 (t2∼t0+Ts): As shown in Fig. 10, Q1 is still turnedOFF, Q2 is still turned ON, D1 is still reverse biased, and D2is reverse biased. At this moment, the input voltage, togetherwith the energy stored in L1 and L2 , pumps energy into the load,thereby causing L1 and L2 to be demagnetized. Besides, Ce isstill discharged.

⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

L1di1dt

=L1

L1 + L2(2vi − vo)

L2di2dt

=L2

L1 + L2(2vi − vo)

Codvo

dt= i1 −

vo

RL

ii = i1 .

(19)

For analysis convenience, let

M1 =L1

L1 + L2and M2 =

L2

L1 + L2. (20)


Therefore, the equations can be expressed to be⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

L1di1dt

= M1(2vi − vo)

L2di2dt

= M2(2vi − vo)

Codvo

dt= i1 −

vo

RL

ii = i1 .

(21)

According to (5) and (17)–(21), the averaged equations canbe obtained to be⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

L1d 〈i1〉

dt= d 〈vi〉 + a 〈vi〉 + M1(1 − d − a)(2 〈vi〉 − 〈vo〉)

L2d 〈i2〉

dt= d 〈vi〉 + a( 〈vi〉 − 〈vo〉 )

+M2(1 − d − a)(2 〈vi〉 − 〈vo〉)

Cod 〈vo〉

dt= d

(

−〈vo〉RL

)

+ a

(

〈i2〉 −〈vo〉RL

)

+(1 − d − a)(

〈i1〉 −〈vo〉RL

)

〈ii〉 = d(〈i1〉 + 〈i2〉 + 〈ie〉) + a 〈i2〉 + (1 − d − a) 〈i1〉(22)

where a is a variable with the corresponding dc quiescent valueof A.

Based on the ampere–second balance, 〈ie〉 can be expressedas a function of 〈i1〉 and 〈i2〉 to be

〈ie〉 =a 〈i1〉 + (1 − d − a) 〈i2〉

d(23)

and hence, by substituting (23) into (22), (22) can be rewrittenas

⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

L1d 〈i1〉

dt= (d + a + 2M1 − 2M1d − 2M1a) 〈vi〉

+(−M1 + M1d + M1a) 〈vo〉

L2d 〈i2〉

dt= (d + a + 2M2 − 2M2d − 2M2a) 〈vi〉

+(−M2 + M2d + M2a − a) 〈vo〉

Cod 〈vo〉

dt= −〈vo〉

R+ (1 − d − a) 〈i1〉 + a 〈i2〉

〈ii〉 = 〈i1〉 + 〈i2〉 .

(24)

After this, vi , vo , ii , i1 , i2 , d, and a will be represented bythe corresponding dc quiescent values plus small ac variationsthat are much smaller in magnitude than the corresponding dc


quiescent values:

⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

〈vi〉 = Vi + vi

〈vo〉 = Vo + vo

〈ii〉 = Ii + ii

〈i1〉 = I1 + i1

〈i2〉 = I2 + i2

d = D + d

a = A + a

with

⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

|vi | � Vi

|vo | � Vo∣∣∣ii

∣∣∣ � Ii

∣∣∣i1

∣∣∣ � I1

∣∣∣i2

∣∣∣ � I2

∣∣∣d

∣∣∣ � D

|a| � A.

(25)

Sequentially, by substituting (25) into (24), (24) can be rewrit-ten to be⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

L1d(I1 + i1)

dt=[(D +d)+ (A + a)+2M1 − 2M1(D + d)

−2M1(A + a)](Vi + vi) + [−M1 + M1

×(D + d) + M1(A + a)](Vo + vo)

L2d(I2 + i2)

dt=[(D+d) + (A + a) +2M2 − 2M2(D + d)

−2M2(A + a)](Vi + vi) + [−M2 + M2

×(D + d) + M2(A+a)−(A + a)](Vo +vo)

Cod(Vo + vo)

dt= − (Vo + vo)

RL+ [1 − (D + d) − (A + a)]

×(I1 + i1) + (A + a)(I2 + i2)

(Ii + ii) = (I1 + i1) + (I2 + i2).(26)

Based on (26), the small-signal ac equation can be obtainedto be⎧⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎨

⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎪⎩

L1di1dt

= (D + A + 2M1 − 2M1D − 2M1A)vi + (−M1

+M1D + M1A)vo + (Vi − 2M1Vi + M1Vo)a+(Vi − 2M1Vi + M1Vo)d

L2di2dt

= (D + A + 2M2 − 2M2D − 2M2A)vi + (−M2

+M2D + M2A − A)vo + (Vi − 2M2Vi + M2Vo

−Vo)a + (Vi − 3M2Vi + M2Vo)d

Codvo

dt= − vo

RL+ (1 − D − A)i1 + A i2 + (−I1 + I2)

×a − I1 d

ii = i1 + i2(27)

and hence, according to (27), the corresponding small-signal acmodel can be obtained as in Fig. 11.

If the values of both L1 and L2 are identical, Fig. 11 canbe retrieved to Fig. 5 under the corresponding condition thatM1 = M2 = 0.5, L = L1 + L2 , A = a = 0, I = I1 = I2 , andi = i1 = i2 .

C. Case 3

This is an actual condition under the assumption that the valueof L1 is smaller than that of L2 and this converter operates at

Fig. 11. Small-signal ac model for the proposed high step-up converter withL1 larger than L2 .



100% of the rated load. Since the behavior of the converter inthis case is similar to that in case 2, only the corresponding basicoperating principles are described herein.

1) State 1: As shown in Fig. 12, Q1 is turned ON, Q2 is turnedOFF, and D1 and D2 are forward biased. Since the voltage acrossCe is equal to vi , L1 and L2 are to be magnetized. During thisstate, the output energy required is supplied from Co and Ce ischarged. Besides, i1 is larger than i2 due to L1 smaller than L2 .

2) State 2: As shown in Fig. 13, Q1 is turned OFF, Q2 is turnedON, and D2 is reverse biased. During this state, i1 is larger thani2 , thereby causing D1 to be forward biased, and hence, L2 tobe still magnetized but L1 to be demagnetized. Besides, Ce isdischarged. As soon as i2 is equal to i1 , the operation goes tostate 3.

3) State 3: As shown in Fig. 14, Q1 is still turned OFF, Q2is still turned ON, D2 is still reverse biased, and D1 is reversebiased. At this moment, the input voltage, together with theenergy stored in L1 and L2 , pumps energy into the load, thuscausing L1 and L2 to be demagnetized. Aside from this, Ce isstill discharged.



D. Voltages Conversion Ratios for Three Cases

In the following, the voltage conversion ratio correspondingto each case is obtained based on the voltage–second balance,to be described as follows.

1) Case 1 with L1 = L2: The voltage conversion ratio in thiscase can be obtained from the following equation:

ViD + 0.5(2Vi − Vo)(1 − D) = 0, for L1 and L2 . (28)

Hence, the resulting voltage conversion ratio is 2/(1−D) thatis identical to (12).

2) Case 2 with L1 > L2: The voltage conversion ratio in thiscase can be obtained from the following equations:

ViD + ViA +L1(2Vi − Vo)

L1 + L2(1 − D − A) = 0, for L1

(29)

ViD + (Vi − Vo)A +L2(2Vi − Vo)

L1 + L2(1 − D − A) = 0,

for L2 . (30)

Hence, by adding (29) and (30) and doing some calculations,the corresponding voltage conversion ratio is 2/(1−D) that isalso the same as (12).

3) Case 3 with L1 < L2: The voltage conversion ratio canbe obtained from the following equations:

ViD + (Vi − Vo)A +L1(2Vi − Vo)

L1 + L2(1 − D − A) = 0,

for L1 (31)

ViD + ViA +L2(2Vi − Vo)

L1 + L2(1 − D − A) = 0, forL2 .

(32)

Hence, by adding (31) and (32) and doing some calculation, theaccompanying voltage conversion ratio is 2/(1−D) that is stillidentical to (12).

IV. CONTROL METHOD APPLIED

Fig. 15 shows the proposed overall system configuration forthe proposed step-up converter. The one-comparator counter-based PWM control strategy without any A/D converter (ADC)based on the field-programmable gate array (FPGA) [27] is uti-lized, and the parameters of the PI controller are obtained at

Fig. 15. Overall system configuration.

rated load. The output voltage information after the voltage di-vider is obtained via the comparator COMP, and then, sent toFPGA having a system clock of 100 MHz, so as to create thedesired PWM control signals vgs1 and vgs2 to drive the MOS-FET switches Q1 and Q2 after the gate drivers, respectively.Furthermore, one half-bridge gate driver can be used to driveQ1 and Q2 .

V. KEY PARAMETER CONSIDERATIONS

Before this section is discussed, there are some requirementsgiven as follows: 1) the rated dc input voltage is from 10 to 16 Vwith 12 V rated; 2) the dc output voltage is 60 V; 3) the rateddc output current Io-rated is 1 A; 4) the switching frequency fsis 195 kHz, i.e., the switching period Ts is 5.1 μs; 5) one 330μF Rubycon capacitor is selected for Co ; 6) the product nameof Q1 and Q2 is IRF3710ZS; 7) the product name of the half-bridge gate driver is HIP2101; 8) the product name of D1 andD2 is STPS15H100CB; and 9) the product name of FPGA isEP1C3T100.

The key parameter design contains design of two inductorsL1 and L2 and one charge pump capacitor Ce , to be described asfollows. Before this topic is discussed, there are some assump-tions to be given: 1) the values of L1 and L2 are identical and areset to L with the current slew rate di/dt being 0.6 A/μs duringthe magnetization period; and 2) the converter operates in therated condition with the voltage sag ΔV on Ce being 0.15% ofthe voltage across itself during the discharge period. Aside fromthis, the component stresses are also included herein.

A. Design of Two Inductors

The values of two inductors can be figured out according tothe following equation:

L =Vi

(di/dt). (33)

Therefore, the value of L can be calculated to be 24 μH un-der the rated input voltage. As generally recognized, in prac-tice, an inductor, made of the ferrite core with air gap, hasa tolerance of about 20%, that is, if this inductor is made


TABLE ICOMPONENT STRESSES

TABLE IICOMPONENT STRESS VALUES

of the ferrite core with air gap, the corresponding induc-tance will be increased or reduced by a factor of about 20%.Hence, for observation convenience, the values of two in-ductors with 20% tolerance of the calculated value are setto 19.2 or 28.8 μH. Finally, four T106-18 toroidal cores,made by micrometals along with twisted copper conductorof φ0.1 × 140, are used to obtain these four inductances:20 μH with 17 turns, 20 μH with 17 turns, 24.1 μH with 19turns, and 29.4 μH with 21 turns.

B. Design of Charge Pump Capacitor

The value of the charge pump capacitor can be worked outbased on the following equation:

Ce =ΔV

∫ (1−D )Ts

0 ie dt. (34)

Therefore, the value of Ce can be calculated to be 270.3 μF.Finally, one 270 μF OSCON capacitor is chosen for Ce .

C. Component Stresses

In this section, the steady-state stresses on the componentsof the proposed converter except the body diodes of MOSFETswitches are tabulated in Table I, on the assumption that thevoltage across any MOSFET or diode during the turn-ON periodis negligible and the voltage across Ce is Vi . In addition, thefollowing component stresses are obtained under the conditionthat the value of L1 is identical to the value of L2 and theconverter operates in the rated condition. Hence, Table II givesthe stress value for each component in the proposed circuit.

In Table I, the symbols from 1) to 7) have the followingmeanings.

1) Vo .2) 0.5Vo .3) Max{Vi, 0.5(Vo − 2Vi)}.4) Vi .5) Io-rated

Vo

Vi+ DVi

Ts

L .

6) 0.5Io-ratedVo

Vi+ 0.5DVi

Ts

L .

7) 0.5Io-ratedVo

Vi+ 0.5DVi

Ts

L − Io-rated .

Fig. 16. Measured waveforms from top to bottom iL 1 , iL 2 , Δve , and vg s 1with L1 equal to L2 under rated input voltage at (a) no load, (b) half load, and(c) rated load.

VI. EXPERIMENTAL RESULTS

At rated load, Figs. 16–18 show the currents in L1 and L2 ,iL 1 and iL 2 , the ac portion of the voltage on Ce , Δve , and thegate driving signal of Q1 , vgs1 , for L1 equal to L2 , L1 smallerthan L2 , and L1 larger than L2 , respectively, with no load, halfload, and rated load considered. In Fig. 16, iL 1 almost overlapswith iL 2 ; in Fig. 17, iL 1 is larger than iL 2 ; and in Fig. 18, iL 1is smaller than iL 2 . Above all, Figs. 16(a)–18(a) show that thisconverter still operates in CCM at no load because of positiveand negative currents flowing through two inductors, therebycausing the duty cycle to be changed slightly for any load, andhence, this converter to be easy to control. Besides, iL 1 andiL 2 are synchronized with each other, and the difference inaverage value between iL 1 and iL 2 is not too much even if thedifference in value between L1 and L2 is up to 40%, implyingthat the proposed converter possesses robustness to some extent,that is, this converter has the capability of enduring componenttolerance. As for the oscillation on Ce , it occurs due to L1


Fig. 17. Measured waveforms from top to bottom iL 1 , iL 2 , Δve , and vg s 1with L1 smaller than L2 under rated input voltage at (a) no load, (b) half load,and (c) rated load.

and L2 resonating with the junction capacitances of D1 andD2 . On the other hand, Figs. 19 and 20 show load regulationsunder identical and different inductances at three different inputvoltages, respectively. It can be seen that the percentages of loadregulations are all within 3% in Figs. 19 and 20. Aside fromthis, Figs. 21 and 22 display the curves of efficiency versusload current under identical and different inductances at threedifferent input voltages, respectively. It can also be seen that forthese two cases, the values of efficiencies in the rated conditionare about 90%.

Figs. 23 and 24 show the load transient responses to furtherdemonstrate the performance of the proposed converter. Sincethe controller parameter tuning method is widely used in theindustry, there are three steps to online tune the parameters ofthe voltage controller, to be described as following:

1) Step 1: The proportional gain kp is tuned from zero to thevalue that makes the output voltage very close to about

Fig. 18. Measured waveforms from top to bottom iL 1 , iL 2 , Δve , and vg s 1with L1 larger than L2 under rated input voltage at (a) no load, (b) half load,and (c) rated load.

80% of the prescribed output voltage. And eventually, kp

is set at 1.0375.2) Step 2: After this, the integral gain ki is tuned from zero to

the value that makes the output voltage very close to theprescribed output voltage but somewhat oscillate. Then,ki will be reduced to some value without oscillation. Andfinally, ki is chosen as 0.01.

3) Step 3: From this time onward, the differential gain kd istuned from zero to the value that accelerates the dynamicresponse but somewhat oscillate. Then, kd will be reducedto some value without oscillation. And eventually, kd isselected to be 2.

Figs. 23 and 24 show the load transient responses due to theload change from 50% to 100% of the rated load and from 100%to 50% of the rated load, respectively. Both the correspondingrecovery times are about 10 ms, and both the resulting voltagedroops are about 2 V.


Fig. 19. Curves of load regulation with L1 equal to L2 under three differentinput voltages.

Fig. 20. Curves of load regulation with L1 larger than L2 under three differentinput voltages.

Fig. 21. Curves of efficiency versus load current with L1 equal to L2 underthree different input voltages.

Fig. 22. Curves of efficiency versus load current with L1 larger than L2 .

From the aforementioned, it can be seen that the proposedtopology has good operating performances both in the steadystate and in the transient.

Fig. 23. Load transient response due to load change from 50% to 100% of therated load.

Fig. 24. Load transient response due to load change from 100% to 50% of therated load.

VII. CONCLUSION

A novel high step-up converter is presented herein. There aresome features in this converter, as summarized as follows:

1) The voltage boosting concept is based on the energy storedin the two inductors, together with the input voltage andthe energy stored in the charge pump capacitor, which isreleased to the load during the demagnetization period.

2) Easy control of this converter can be achieved since thisconverter is always operated in CCM.

3) A good performance of load regulation can be obtainedeven though the difference in value between L1 and L2 isup to 40%.

However, there are three main drawbacks in the proposedconverter, to be described as follows. First of all, at start-up,the surge current will occur. Second, the proposed converteris suitable for the low-current application, but not suitable forthe high-current application. This is because a large chargingcurrent will flow through the charge pump capacitor in the high-current application, due to the voltage drop across this capacitorbeing too large. Third, the main switch has to endure the out-put voltage during the turn-OFF period. These drawbacks willbe improved in the future study. Furthermore, the basic op-erating principles of the proposed converter working withoutsynchronous rectification will also be taken into account in thefuture.


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K. I. Hwu (M’06) was born in Taichung, Taiwan, onAugust 24, 1965. He received the B.S. and Ph.D. de-grees in electrical engineering from National TsingHua University, Hsinchu, Taiwan, in 1995 and 2001,respectively.

From 2001 to 2002, he was the Team Leader of theVoltage-Regulated Module (VRM) at AcBel Com-pany. From 2002 to 2004, he was a Researcher at theEnergy & Resources Laboratories, Industrial Tech-nology Research Institute. He is currently an Asso-ciate Professor at the Institute of Electrical Engineer-

ing, National Taipei University of Technology, Taipei, Taiwan, where he was theChairman of the Center for Power Electronics Technology from 2005 to 2006.His current research interests include power electronics, converter topology, anddigital control.

Dr. Hwu has been a member of the Program Committee of the IEEE Ap-plied Power Electronics Conference and Exposition since 2005. He has alsobeen a member of the Technical Review Committee of the Bureau of Standards,Metrology, and Inspection since 2005. Since 2008, he has been an associatemember of The Institute of Engineering and Technology (IET).

Y. T. Yau (S’08) was born in Tainan, Taiwan, onNovember 23, 1980. He received the B.S. and M.S.degrees in electrical engineering from Tamkang Uni-versity, Tamsui, Taiwan, in 2002 and 2004, respec-tively. He is currently working toward the Ph.D.degree at the Institute of Electrical Engineering,National Taipei University of Technology, Taipei,Taiwan.

In 2002, he was with Acbel Company for sixmonths. He is currently a Researcher with the In-dustrial Technology Research Institute, Hsinchu,

Taiwan. His current research interests include power electronics, convertertopology, and digital control.

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High Step-Up Converter Based on Charge Pump and Boost Converter

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