IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. EMC-21, NO. 3, AUGUST 1979
How Switches Produce Electrical NoiseE. KEITH HOWELL, SENIOR MEMBER, IEEE
Abstract-This tutorial paper describes the fundamental mechanismby which mechanical switches produce electrical "noise," and theparameters which determine the complex waveforms, frequency com-ponents, and amplitudes produced. Understanding the fundamentalsremoves some of the mystery often associated with this ubiquitoussource of noise and transients, and can assist in development of effec-tive methods of preventing resultant malfunctions and damage inelectronic equipment. The primary focus is upon switches operatinglow-current inductive loads on 120-V residential power lines, and showspeak amplitudes up to several kilovolts and frequencies up to severalhundred megahertz produced on the supply line.
I. SWITCHES AND NOISES WITCHES produce transient electrical "noise," the un-
desirable effects of which range from trifling annoyance tocatastrophic malfunction and failure. The noise sensitivity ofAM radio has contributed to the growth of FM radio, whichis but one illustration of the significance of annoyance. Theeffects of more serious malfunctions are well recognized. Thecontribution of switch noise to irreversible damage of insula-tion, dielectrics, and semiconductors is gaining recognition asthe result of greater use of electronic equipment.
Switches are ubiquitous and come in all sorts of solid-stateand mechanical varieties. Solid-state switches are now well un-derstood, but mechanical switches are not. The "noise" theyproduce is so called because it is 1) undesired, and 2) very dif-ficult to define.
Attempts at definition of mechanical switch noise have metwith considerable problems because of limitations in under-standing of controlling factors and reasons for the variabilitiesencountered in measurements. Mathematical circuit transientanalysis, based on the ideal (nonexistent) switch, is welldeveloped and quite useful when applied properly to real-world switching situations.
II. THE IDEAL SWITCH
In the elementary circuit of Fig. 1, let the switch S1 beideally perfect and assign unit values, V, = 1 V, R1 = 1 Q.When S, closes, the current i instantaneously changes fromzero to 1 A, and the load voltage v changes from zero to 1 V.
This step-function change in i and v would then have theclassical Fourier transform in which the amplitude is inverselyrelated to frequency. This 1/f continuum also appears when S,opens and i and v make a step-function change to zero.
If that were all there was to it, definition of switch noisewould be simple. However, the circuit of Fig. 1 exists only on
Manuscript received March 8, 1978; revised March 23, 1979.The author is with the Circuit Protective Devices Department,
General Electric Company, 41 Woodford Ave., Plainville, CT 06062.(203)-747-76 10.
a-~~ C ---0sI
VS RI
Fig. 1. Elementary switching circuit.
5 L L55 LR
Fig. 2. Inductive-load switching circuit.
paper. In a true step function, di/dt and dv/dt are both in-finitely large, now consider that:
L di/dt i = Cdv/dt.
The inevitable presence of inductance and capacitance makethe true step ftunction impossible and adds some interestingfeatures.
If a finite inductance L is added to the circuit, as in Fig. 2,the current will build up to the V1/R1 value in the classicalexponential manner with a time constant of L1/R1 whenswitch S, closes. However, when S, opens (ideally) the cur-rent would abruptly drop to zero, and the L1 di/dt voltagedeveloped across L1 would be infinitely large. Since no insu-lation system could withstand that voltage, the real circuitmust be different.
The addition of a capacitance C1, as in Fig. 3, creates amore realistic situation. Now when S, opens, the storedinductive energy (Li2/2) transfers to the capacitance (CV2/2)and oscillates back anid forth until all the energy is consumedin losses. The maximum voltage can then be deternined byequating energies, assuming negligible losses. Hence, V max -
-i NL /C1 = -i Z1 where Z1 is the characteristic resonantimpedance ofL1 and Cl. This circuit will oscillate at a resonantfrequency fi - 1/2ir/L1C1 or lower, depending on losses.(The waveform of v is shown in Fig. 6 for the condition whereR1 is small in comparison with Z1.)
III. THE REAL SWITCH
While the circuit of Fig. 3 describes the load side of S5reasonably well, we must also consider what happens on thesource (line) side of the switch. If S succeeds in interruptingload current, it must interrupt source current, resulting in
some transient disturbance. A convenient and elementary re-
presentation of the source (and line) impedance is shown inFig.4.
Typically, R2, L2, and C2 are much smaller than R1, L1,and C1, and thus will ring at a much higher frequency f2.
When S1 opens, current i is interrupted and the maximumvoltage v2 max = i Z2 + V,. Since the first peak of v2 is posi-tive, and v1 is negative, the peak voltage across the switchcontacts (the vector sum of v, and v2) can be greater thaneither.
Thus far we have seen that the successful opening of theswitch inevitably produces transient disturbances on both linevoltage and load voltage, typically of a damped oscillatorynature with predictable amplitudes and frequencies deter-mined by both lumped and distributed circuit parameters.Now, let us consider the nature of the switch itself.
IV. SWITCH CLOSURE
The most prevalent kind of switch employs solid metalliccontacts operating in air at roughly standard atmosphericconditions. Switch closure is typically a rapid series of clo-sures and openings resulting from bouncing, sliding, rocking,and surface contaminants. The initial closure must charge loadcapacitance C1 (Fig. 4) through some impedance Z2, whichhas a capacitive component C2. Thus the initial current may
rise very rapidly to a high peak value then decay to the normalload current. This decay is most probably a damped oscillatorywave with a frequency approximately f3 = 1 /27rVL ~-. Sincethe first area of contact may be extremely small, the rapidlyrising current density could be high enough to melt or vaporizemetal, resulting in contact welding, or vapor pressure reopeningthe contacts, and arcing. The resonance f3 is typically of muchhigher frequency than the bounce rate, hence may be repeatedwith each closure. However, the voltage on C1 may not decayappreciably (at f1) during open intervals of the bounce,thereby reducing amplitudes in subsequent closures.
V. SWITCH OPENING
The opening process of our typical switch also involves an
initial series of preliminary openings and closures as the resultof rocking or sliding actions. External shock and vibration can
also cause reclosure, as can contact sticking forces being brokenand vibrating contact members. A much more subtle mechanismresponsible for multiple opening operations is the electrostaticforce of attraction between the contacts at close spacing. Theforce between parallel plates is given by
F= 4.515 X 10-10 r V2 AIx2 grams force,Er relative dielectric constant (=1 for air),V voltage across contact, volts,A area of contact, in same units asx2,x separation of contacts (gap).
For 2.5-mm-diameter flat contacts separated 0.006 mm, theforce of attraction is 9 g at 200 V, and is 20 g at 300 V. Con-sidering that the opening force on the contacts is often coupledthrough a resilient (spring) member, and is being countered byinertia, the electrostatic attraction becomes a significant re-closing mechanism. Evidence of this action is shown later inexperimental results, and has been confirmed in separate ex-periments not included in this paper. Creep-action switches,such as certain thermostats, are notably subject to electro-static forces on both opening and closing actions, modified bysticking (weld) forces.
Since load current is presumed to be well established at thetime of opening, the current density in the last area of contactmay be high enough to cause metal vaporization. When thisoccurs, a metallic-vapor arc is established, and the arc imped-ance and voltage drop is suddenly inserted in the circuit. Thiscreates a shock excitation, primarily of f3. Subsequent re-closure in this opening series will reestablish load current.
If load current is below vaporization level, opening of thecontacts initiates the ringing frequencies f1 and f2. As notedearlier, f2 is typically much higher than fl, and Z2 is muchlower than Z1, hence v1 max is generally much higher than v2max. Considering that the mechanical separation of the con-tacts may be quite slow in comparison with f1 and f2, itwould not be surprising to find that the voltage across the con-tacts could exceed the breakdown voltage for a short air gap.
Breakdown of a spark gap occurs (simplistically) when thevoltage gradient times the electron mean-free-path lengthequals the ionization potential of the gas. For very short gaps,however, ion recombinations at the metal surfaces reduceavalanche probability. Thus the minimum breakdown voltagefor a gap occurs when the gap spacing equals some criticallength for the existing gas density. For most contact shapes,the fact that breakdown voltage increases at spacings less thanthat critical length is meaningless because that length occurs atsome point along the edges of the contacts. From the Paschencurves relating breakdown voltage to gap spacing, for air atstandard atmospheric conditions, we find that the minimumbreakdown voltage is about 350 V, and this occurs for a gapspacing of 0.006 mm. Experience shows that this minimumbreakdown is influenced by other factors, such as electrostatic-force conditions, temperature, residual ionization and metallicions, and the presence of very-high-frequency voltages, whichtend to reduce the observed breakdown voltage.
When switch contacts open, one of two possible conditionsmay occur. If current is high enough, a hot-cathode arc may becreated at opening, which starts with a characteristic voltage
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IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. EMC-21, NO. 3, AUGUST 1979
v2CRO(8. 5 pF)
GROUNDPLANE
Fig. 5. Te
drop on the order of 15 V and then rises as the contacts separ-ate, stretching and cooling the arc. The arc is extinguishedwhen current goes to zero as the result of an ac power supply,or arc drop approaching supply voltage, or arc negative-resistance characteristic, or some combination of these factors.However, the thermionic arc current rarely stops abruptly be-cause the ion plasma and hot-cathode spot will not permitappreciable ringing voltages.
On the other hand, low current will permit switch openingwithout an arc, but with ringing of circuit resonances whichcan exceed the contact-gap breakdown voltage, with resultsakin to the old spark-gap transmitters.
It has been observed that it is the second case (low current)that produces the most serious "noise" disturbances which can
result in malfunctions and/or damage to electrical and elec-tronic equipment.
VI. EXPERIMENTAL VERIFICATION
In one particular instance, switching of a small timer motoron 120 V, 60 Hz, was causing serious malfunctions of an elec-tronic device operating from the same power line. Attempts toobserve the "noise" on the line with an oscilloscope resultedonly in "hash," i.e., not readily interpreted waveforms. There-fore, an experiment was set up to study the switching phenom-ena with this motor. The test circuit, Fig. 5, employed a regu-lated and well filtered adjustable dc power supply to permit re-
peatable observations over the full range of currents at whichthe switch could open in the normal ac operation.
The normally closed contacts of a relay (G.E. type CR120-HCA1J02) were used as the switch, and a solid-state pulsingcircuit of adjustable height and pulse width was used to openthe contacts and trigger the oscilloscope, thereby permittingobservation of the complete opening process. Contact openingspeed and distance (gap length) were varied by control of thepulse. Armature spring-mass resonance produced repeatabletiming, and resulted in the opening gap dimension followingthe crest of a sine wave for weak excitation pulses.
Line impedance was represented simply by a 5-,uH air-core inductor L2, having a total of about 14-pF stray and dis-tributed shunt capacitance C2. With the oscilloscope probeconnected for observing v2, the total shunt capacitance was
22.5 pF. In this condition, the line impedance Z2 = =
471 Q2, and self-resonant frequency f2 = Z2/2m7L = 15 MHz.This impedance was deliberately chosen to get the resonant
; ; 1l- V R CRO_RF
st circuit.
frequency within the writing range of the storage oscilloscopebeing used.
The timer motor, measured on a bridge at 1 kHz, hadan inductance of 6.80 H and a dc resistance of 1448 Q.Measurements at 120 V, 60 Hz showed a total impedanceof 5480 Q at 61 degrees, drawing 22-mA rms (31-mA peak).This impedance breaks down to a resistance of 1448 Q inseries with inductance of 13.54 H, which is paralled by 22kQ, representing core losses. The shunt capacitance of themotor coil was found to be 80 pF by square-wave analysis.Using the 1 kHz inductance of 6.8 H and the peak currentof 31 mA, the peak voltage resulting from transfer of allthe inductive stored energy to the shunt capacitance is V1 =0.03lV.8/80 X 10 12 = 9000 V. The impedance Z1 = 300kQ, and the ringing frequency fi = l/21rjiTC= 6.8 kHz;hence, the voltage could rise to 9 kV in 37 ,us. This conditionwill, obviously, result in breakdown of the contact gap. Inorder to reduce the impedance, frequency, and voltage, aceramic-disk capacitor of 4730 pF was added in parallel withthe motor terminals for part of the testing. Using a totalcapacitance C1 of 4800 pF and inductance L1 of 6.8 H, theimpedance Z1 = 37.6 kQ2 and the resonant frequency f1 =880 Hz (1.13-ms period). A rather simple RF detector wasused to show, qualitatively, the existence of substantial voltagedrop in a short length of hookup wire, typically used in elec-tronic equipment.
The waveform of v1, in Fig. 6, was taken with a supply of2 V and 1.38 mA. The peak voltage observed is 52 V, and theperiod is 1.17 ms, which converts to a frequency of 855 Hz.The calculated peak voltage is V1 = I Z1 = 52.0 V. The closeagreement between observed and calculated values confirmsthe principles and justifies the use of the inductance valuemeasured at 1 kHz. This waveform also illustrates the success-ful opening of the contacts with no preliminary openings andwith no arcing. Reclosure of the contacts is observed 2.5 msafter opening, with a series of bounce openings and closures.The drive on the relay coil was set for minimum consistentopening, which accounts for the relatively short open interval.The contact voltage at initial reclosure implies the effect ofelectrostatic attractive force on the point of closure, as de-scribed previously, and will be apparent also in subsequentwaveforms.
Increasing the dc supply to 16 V produced 1 1 mA in themotor coil and the waveform of v1 shown in Fig. 7. The cal-
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HOWELL: HOW SWITCHES PRODUCE ELECTRICAL NOISE
Fig. 6. Load voltage transient with ideal switch opening. IO = 1.38 mA.Scale: 20 V/Div.; 0.5 ms/Div.
Fig. 8. Load voltage with reversing and nonreversing breakdowns ofswitch. Io = 16.6 mA. Scale: 200 V/Div.; 0.5 ms/Div.
Fig. 7. Load voltage with one breakdown of switch. Io = 11 mA. Scale:200 V/Div.; 0.5 ms/Div.
culated peak voltage is 415 V for this case, but the waveformshows a breakdown of the contact gap at about 320 V. Thisbreakdown voltage is in reasonable agreement with the predic-tion. The breakdown not only discharged C1, but also reversedits charge by 80 V. Subsequent inductive energy transfer re-charged the capacitor to 200 V peak, indicating relatively littleenergy loss in the breakdown event.
The waveform of v1 in Fig. 8 was taken for a supply of 24V, 16.6 mA, giving a calculated peak V1 of 623 V. Two break-downs are shown in this photograph, both occurring in the300-320 V range, as expected. However, the first event re-sulted in a reversal of voltage on C1, whereas the second eventappeared to be only a partial discharge. An expanded scaleview of these two events, in Fig. 9 (actually taken of anotheroccurrence), shows the same discrepancy in discharge of C1.An explanation is in order.
Breakdown of the contact gap suddenly connects thecharged C1 to the line. Since L2 and C2 are much smaller thanL1 and C1, the resulting circuit is basically the resonant circuitof C1 and L2, having a characteristic impedance Z3 and reso-nant frequency f3. In this experiment, (L2 = 5 pH, C1 = 4800pF) the impedance Z3 was 32.3 Q and the frequency f3 was1.03 MHz. Therefore, the discharge of C1 through breakdownof the contact gap should be oscillatory.
The waveforms of v1 in Figs. 10-13 confirm this. In Fig.10, corresponding to the first breakdown event of Fig. 9, it isseen that the discharge of C1 consisted of a single half cycle ofthe f3 ringing frequency. The oscillation was terminated bycommutation of the gap arc at the first current zero, whichoccurred at the reverse voltage crest, therefore leaving C1charged to this reverse voltage.
Peak arc current is calculated from Vp/Z3 at about 8 A, and
Fig. 10. Expansion of first (reversing) breakdown in Fig. 8, showingarc commutation at first current zero. IO = 16.6 mA. Scale: 200V/Div.; 0.5 jus/Div.
Fig. 11. Expansion of second (nonreversing) breakdown in Fig. 8,showing arc commutation at second current zero. Io = 16.6 mA.Scale: 200 V/Div.; 0.5 us/Div.
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IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. EMC-21, NO. 3, AUGUST 1979
Fig. 12. Load voltage, showing arc commutation at third currentzero. Io = 16.6 mA. Scale: 200 V/Div.; 0.5 Ms/Div.
the di/dt at commutation (2lrTf3I) about 52 A/ps. This may
seem remarkable to those who are accustomed to thyristorcommutation characteristics, but it becomes astounding even
when the dv/dt following commutation (shown later) is
considered.Fig. 11, corresponding to the second event in Fig. 9, shows
that the gap arc failed to commutate on the first current zero
but did clear on the second zero, thereby leaving the voltageon C1 at a reduced value of the same polarity prior to break-
down. This explains the apparent partial discharge observed in
Figs. 8 and 9 in the second event.It can be inferred that arc commutation could occur at any
current zero in the oscillatory discharge, and the waveforms in
Figs. 12 and 13 confirm this surmise. From the observed
damping in Fig. 13, it is apparent that the loss of stored energy
is relatively small in each half-cycle of this discharge. This
situation indicates that the gaseous discharge is in the form of
a low-voltage thermionic arc.
When the gap is conducting, the line voltage V2 is essentiallythe same as the load voltage v1. Therefore, commutation of
the arc leaves the line impedance Z2 charged to v1 at that in-
stant, hence V2 must then decay, according to Z2, in an oscil-
latory manner. Since f2 is a much higher frequency than f1,
this oscillatory decay takes place before v1 can change apprec-
iably. The voltage across the contacts at commutation has a
dv/dt of 27rf2 V1 and a maximum value (neglecting losses) ap-
proaching 2u1, where v1 is taken at the instant of commuta-
tion.The waveform of line voltage V2 is shown in Fig. 14 for a
breakdown event corresponding to the second event of Figs. 8
and 9, and to Fig. 11. This shows V1 at commutation of about
150 V which leads to a dv/dt of 14 000 V/,us and a peak voltageacross the gap of 300 V. The waveform also shows a discon-
Fig. 14. Line voltage, with arc commutation at second current zero,showing line resonance. IO = 16.6 mA. Scale: 200 V/Div.; 0.5 Ms/Div.
tinuity, not well recorded, at the reverse voltage crest, indi-cating an attempt at commutation. The voltage at that crestwas about 180 V, hence maximum gap voltage followingcommutation would have been 360 V, which is high enoughto reestablish the arc by voltage breakdown of the gap. Thiseffect can explain failure to commutate at the first currentzero, and possibly also at the second, but it does not ex-plain Figs. 12 and 13 where commutation did not occur atpeak voltages below 100 V. A reasonable conjecture for thelater cases involves the thermionic emmission from the hotcathode spots, and the balance of arc energy heating thosespots with energy loss into the surrounding metal of the con-tacts.
Thus arc commutation is easiest at the first current zeroprovided that 2 V1 is less then breakdown voltage, because thecontact which had served as the anode must next become thecathode and no hot spot had been produced on it during thefirst half-cycle. Commutation at the second current zero is,likewise, less likely because the temperature of the first cath-ode hot spot may still produce sufficient electron emission torestore the arc. Probability of commutation should improvethen, as the oscillating current decays, reducing cathodeenergy input and allowing time for heat flow into the adjacentmetal. It will be later shown, however, that the breakdown arccan be translated to a low-current cold-cathode glow dischargeunder proper conditions.
The waveform of v1 in Fig. 15 was taken with a dc supplyof 50 V, 34.5 mA, representing the peak current in the clockmotor in normal operation at 120 V, 60 Hz. This photo showsthe breakdown voltage starting at 320 V and rising to 660 V
as the contact gap increases. Commutation of the arc appearsto have occurred at the second zero crossing in the firstfour breakdowns, then shifted to the third current zero inthe fifth and sixth events because of the higher voltages. Thepeak current in the last arc must have been about 20 A.
The same conditions were used, with expanded timescale in Fig. 16, showing the waveform of v1 in the upper(A) trace. In this case, the first breakdown occurred at 230 V,possibly indicating a reclosure resulting from electrostaticforce. The first arc commutated at the first current zero,the next four at the second zero, and the last at the thirdzero.
The lower (B) trace of Fig. 16 shows the output of thediode detector VRF indicating the RF voltage developed acrossa 5-cm length of AWG 22 wire. This represents an inductance
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HOWELL: HOW SWITCHES PRODUCE ELECTRICAL NOISE
Fig. 15. Load voltage at rated peak current. Io = 34.5 mA. Scale:200 V/Div.; 0.5 ms/Div.
Fig. 16. Load voltage (A) and RF detector voltage (B) at rated peakcurrent. Io = 34.5 mA. Scale: (A) 200 V/Div.; (B) 10 V/Div.;100 Ms/Div.
of about 50 nH, and was used because of suspected RF-cou-pling problems on a similar wire in the electronic device re-
sponsible for this investigation. As expected, the output of thedetector occurs only at breakdown of the gap, but the 10-Vmagnitude was larger than expected. On further examinationof the circuit, however, a series-resonant circuit is found in C2and the wiring inductance which is estimated to be about 0.1pH. Breakdown of the gap applies V1 to this circuit, which hasa resonant frequency of about 135 MHz, and about half thatpeak voltage would appear at the detector input. However, thedetector capacitance is larger than C2 and the diode efficiencyis not very high at that frequency; therefore, the 10 V observedis much lower than would be present without the detector.The triangular waveform on the RF detector was obtained byuse of small forward bias (restoring) current, not shown in thecircuit diagram, and not always used, as evident in later wave-
forms.The observations thus far have apparently established the
basic mechanisms involved in switch production of electricalnoise, but the circuit was padded by additional capacitors. Infurther study, the discrete capacitor C1 was removed, leavingonly probe and distributed shunt capacitance as C1.
In Fig. 17, the upper (A) waveform shows v1 and the lower(B) trace shows VR F for a dc supply of 10 V, 6.9 mA. For thefirst time, a preliminary opening is observed, probably theresult of contact wipe since the open time is estimated to be0.5 ps, the voltage is only 80 V, and the succeeding closed in-terval is relatively long. The RF detector shows that this pre-
liminary opening was actually two openings in quick succes-
sion. Variations in breakdown voltage from 180 to 320 V are
partly explained by the chopped display not recording theactual peak voltage, and partly by the probable persistance of
Fig. 17. Load voltage (A) and RF detector voltage (B) at 20-percentcurrent and with C1 removed. Io = 6.9 mA. Scale: (A) 200 V/Div.;(B) 10 V/Div.; 5s0 s/Div.
line (v2) ringing voltage which increases contact voltage overthe observed v1 voltage. An interesting observation here is thatthe RF detector voltage reaches 17 V, even though the motorcurrent and shunt capacitance are lower than used in Fig. 16.The v1 waveform shows what others have given the colorfulname "showering arc."
The preliminary opening in Fig. 18 is much more pro-nounced, perhaps the result of higher dc supply (22 V, 15.2mA) producing faster rise times and higher energy discharges.In the v1 waveform (A), the relaxation oscillations are reason-ably consistent for about 370 Ms, then abruptly cease in astable cold-cathode discharge at 300 V drop. This indicatesthat the current- and time-dependent negative resistance char-acteristic of the glow discharge, in combination with the lossesand resistances in the circuit, results in a net positive resistanceand stable operation. During the relaxations, the RF detectoroutput reached 22 V, but decayed to zero during the glow dis-charge. Which leads to a very emphatic conclusion:
The intrinsic noise of a gaseous discharge is insignifi-cant in comparison with the "noise" produced by theabrupt switching functions of a gaseous relaxationoscillator.
A short segment of the v, relaxation waveform is expandedin Fig. 19. The relaxation frequency f4, is seen to range from380 to 760 kHz. Breakdown voltages range from 270 to 350 V.The presence of a small, 5 MHz ring on the v1 waveform showsthat we are approaching the point where the simple, lumped-constants circuit is no longer an accurate representation.
At 50-V 24.5-mA dc supply, the tendency toward a stableglow discharge was much greater, as shown in Figs. 20 and 21.During relaxation oscillations, the RF detector reached 30 V.Fig. 22 shows a segment of relaxation oscillations with fre-quency f4 ranging from 0.9 to 2 MHz.
With the same dc supply condition, the drive on the relaycoil was increased to produce a faster separation of the con-tacts. This produced a rising breakdown voltage as shown inthe v1 (upper) waveform of Fig. 23. The highest breakdownvoltage clearly recorded here is 1020 V and the highest RF-detector output is 32 V. Other observations showed break-down voltages up to 1.4 kV under similar conditions. Thedecrease in RF-detector output as breakdown voltage increasedis contrary to reason, but is simply caused by the inability ofthis detector to follow relatively low duty cycle, very high-
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IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. EMC-21, NO. 3, AUGUST 1979
Fig. 18. Load voltage (A) and RF detector voltage (B), with C1removed, at 44-percent current, showing brief glow dischargeperiod. IO = 15.2 mA. Scale: (A) 200 V/Div.; (B) 20 V/Div.; 100,s/Div.
Fig. 19. Expanded load voltage of Fig. 18 showing relaxation rate.
Io = 15.2 mA. Scale: 200 V/Div.; 1 Ms/Div.
Fig. 20. Load voltage (A) and RF dector voltage (B), with C1 re-
Fig. 22. Load voltage relaxations of Fig. 21, expanded. IO = 24.5 mA.Scale: 200 V/Div.; I ps/Div.
Fig. 23. Load voltage (A) and RF detector voltage (B) with highervelocity switch opening, C1 removed. IO = 24.5 mA. Scale: (A)200 V/Div.; (B) 20 V/Div.; 200 ,s/Div.
frequency ringing waveforms. The detector output does,however, verify that the observed breakdowns are taking placein the contact gap rather than in the clock-motor coil itself.
VII. SUPPRESSION
In specific situations, however, certain corrective actionscan be taken. The most effective technique is one that pre-vents the generation of noise at the switch. Fig. 24 shows thewaveform of v1 (upper trace) and VRF (lower trace) for thesame test conditions used in Fig. 23, but with a metal oxidevaristor, GE-MOV type V130LAIO, connected directly acrossthe switch contacts. The varistor completely eliminated therelaxation oscillations by holding the voltage below the break-down level while dissipating the stored energy in the-system.The varistor did not, and could not, eliminate the initialcontact reclosures resulting from electrostatic forces and/orwiping or rocking action of the contacts. A suitable capacitorconnected in parallel with the varistor and the contacts can re-duce the rate of rise of voltage, thereby eliminating electro-static reclosure and reducing voltage excursions from mechani-cal reclosures. However, when the contacts close, this capaci-tor is discharged in a very low-impedance path with high peakcurrents that can create explosive reopening of the contactsand possible reclosure on molten metal with resultant welding.Connecting the capacitor in parallel with the load allows thesource impedance to limit contact current, but also permitsringing of this impedance. A series RC network can then beconnected across the varistor and contacts to dissipate storedenergy in the line without oscillatory ringing upon opening.
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HOWELL: HOW SWITCHES PRODUCE ELECTRICAL NOISE
Fig. 24. Load voltage (A) and RF detector voltage (B) with 130-Vvaristor across contacts. Io = 24.5 mA. Scale: (A) 200 V/Div.;(B) 10 V/Div.; 200 ,us/Div.
VIII. CHARACTERISTICS OF SWITCH NOISE
The term "noise" applied to any unwanted electrical signalor disturbance, but generally implies a random distribution ofamplitude and frequency parameters, randomly distributed intime. The "noise" produced by switches fits broadly in thisdescription. This investigation has shown, however, that switch"noise" is not completely random and can be described to a
certain extent.1) Noise is produced by the abrupt transitions between con-
ducting and nonconducting states of the metallic contacts andof gaseous discharges between those contacts.
2) Comparatively insignificant noise is produced by contin-uous hot- or cold-cathode gaseous discharges.
3) The load-circuit relaxation oscillation is basically a saw-
tooth wave which can sweep across the frequency (f4) spec-
trum from 107 to 104 Hz with each switch operation. Thepeak-to-peak amplitude may sweep from about 100 V toseveral kilovolts within each switch operation, and is generallyinversely related to frequency f4. The duration of this relaxa-tion oscillation may range from 0.1 ms to several milliseconds.Specific load-circuit impedance, instantaneous operating con-
dition at time of switching, and specific switch characteristicsare the controlling parameters.
4) Each breakdown of the gap or reclosure of the switchproduces on the line a ringing wave at frequency f3 as voltageon the load capacitance is applied to the line. Duration of thering may be as short as one half-cycle, and may extend tomany cycles. Amplitude of first-peak voltage may range toseveral kilovolts, but probability of being less than 300 V isvery low. In each switch operation, amplitude may sweep from300 V to several kilovolts. The frequency f3 is generally fixedby specific line and load-circuit impedances, and does not vary
appreciably in operation. Limited observation places the range
off3 from 1 to 100 MHz.5) At each commutation, or attempted commutation, of
the gap discharge, and to some degree at its initiation, a ringingmay occur on the line at its specific self-resonant frequency f2,which may range from 10 to 1000 MHz. The first-peak ampli-tude of f2 will vary within each operation, over an expectedrange to several hundred volts, with highest probability on theorder of 100 V, measured at the switch. This amplitude willdecrease with line distance away from the switch according topropagation characteristic of the line. Branch circuits, wiringdevices, and loads can introduce additional resonances, reflec-
tions, and absorptions, resulting in a complex frequency spec-trum unique to each installation.
6) In summary, switch-produced noise on the ac power linehas two primary frequency components f2 and f3 unique toeach installation, which appear as damped oscillations, at a rep-etition frequency f4 which varies within each switching opera-tion over a range characteristic of the installation and instan-taneous supply voltage and current at time of switching. Amp-litudes are also dependent on the specific installation andelectrical conditions at switching.
With such a wide range of frequencies and amplitudes beinggenerated, it is no wonder that switch-produced noise (tran-sients) can break down insulation, destroy electronic compo-nents, interfere with radio and TV, and cause malfunctions ofmore serious nature in other electronic equipment. It is alsoapparent that the noise produced in two similar, but notidentical, installations of the same basic equipment (switchesand loads) can have significantly different effects. Similarly,no single installation can be considered as having effects typi-cal or representative of others, nor as "worst case," especiallyas far as the frequencies f2 and f3 are concerned.
IX. EFFECTS OF SWITCH NOISE
The voltage gradients appearing in insulation systems arewell known to be frequency and rise-time dependent, particu-larly where discontinuities in dielectric constant are present,and where wire inductance and capacitance self-resonanceoccurs. Thus the integrity of a specific insulation system mustbe examined over a broad range of frequencies in order toprove its ability to survive in various installations. Similarly,the voltage distributions in electronic equipment must be ex-amined over a wide frequency range to ensure adequate com-ponent protection at the high voltages likely to be encountered.
Malfunctions in electronic equipment, as the result ofswitching noise, are extremely difficult to analyze, predict,and eliminate because of the wide ranges of frequencies, amp-litudes, wave shapes, and durations involved. When regarded asa receiver, most electronic equipment will show a responsesensitivity spectrum having a multitude of peaks and valleys,most of which are caused by stray resonances and couplingsthat change with lead lengths and wire dress, and are oftendependent on signal amplitude. Unfortunately, the responsesalso change when attempts are made to find them by directmeasurements. If the noise frequency componentsf2,f3, orf4should coincide with a peak in the sensitivity spectrum, a mal-function response could occur. Conversely, no malfunctionwould be expected in a device that does not respond to thenoise frequencies, excluding effects purely caused by highvoltages. Since both the generated frequencies and the re-sponse frequencies are heavily dependent on specific wiringparameters, a wide disparity of responses is to be expected fora given class of devices in a given class of environments.
The large amplitudes at high frequencies found in switchingnoise complicate laboratory measurements and analysis. Con-sider, for example, a line voltage ringing waveform of 100 Vamplitude at 5 MHz (f3 case) and an electronic circuit whichcan respond to a signal of 10 mV and having an impedance of
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300 2. A capacitance between line and circuit of only 0.01 pFis sufficient to produce that 10-mV signal.
On the other hand, tests made on the circuit at lower voltagecontinuous-wave RF might well show no response. The ringingwaveforms constitute a modulation which can also affect cir-cuit performance in a totally different mannei than a CW testwould show.
X. NOISE TESTING
Attempts have been made to evaluate the noise response ofelectronic equipment by various methods, including:
1) CW and moaulated RF at discrete frequencies;2) CW RF swept over a range of frequencies;3) repetitive switching operations of devices "known"
to have caused problems, such as relays, timers, seriesmotors, etc.;
Obviously, all of these test methods are significant if a mal-function occurs, but the absence of malfunction in any one orall of these test carries no defined statistical assurance of noiseimmunity in "field" application, nor in another implementa-tion of the same type of test, nor in other devices of the same
class tested. No single test waveforn, however complex, canprovide even passable correlation with the "real world."
As a practical matter, it is rarely possible to eliminateswitching noise at its source. The electrical and electronicequipment must be made to withstand such noise withoutdamage or serious malfunction. A tremendous amount of workhas been done to this end, yet more needs to be done. There isyet no statistical quantitative description of the switchingnoise on low-voltage (120/240 V) power systems. There is yetno definitive way to measure such noise. There is yet no testmethod, equipment, or procedure by which the performanceof a device in service can be reasonably predicted. This paperqualitatively describes switching noise and some of the mech-anisms by which it is produced, and provides some indicationof the sort of amplitudes, frequencies, waveforms and dura-tions inherent in such noise. It is suggested that a study of avariety of switches (including motor brushes), loads, sources,lines, and wiring impedance, could lead to a statistical de-scription of noise characteristics to be expected in varioussituations, and thence to a test method and criteria of statisti-cal significance.
REFERENCES[1 W. C. Kotheimer and L. L. Mankoff, "Electromagnetic inter-
ference and solid state protective relays," IEEE Trans. PowerApp. Syst., vol. PAS-96, pp. 1311-1317, July/Aug. 1977.
