A Resonant Bidirectional DC-DC Converter for Aerospace Applications

Alireza Safaee, Student Member, IEEE, Alireza Bakhshai, Senior Member, IEEE, Praveen Jain, Fellow, IEEE Queen’s Centre for Energy and Power Electronics Research (ePOWER)

Queen’s University, Kingston, Ontario, Canada

Abstract — A wide-range input-output bidirectional dc-dc power converter suitable for power rating of several kilowatts is introduced in this paper. This converter provides a high frequency operation with zero voltage switching (ZVS) for all the switches, with no auxiliary circuitry or increased component stresses. The transformer leakage inductance is used as the resonant inductor. The proposed control algorithm includes a combination of variable frequency control and pulse width modulation methods. The control provides the minimum current for any active power level, with no compromise in zero voltage switching, through precise selection of the control variables of frequency, phase shift and duty cycle. The experimental results verify the feasibility and performance of the proposed topology.

I. INTRODUCTION Various types of dc-dc converters are available

nowadays, each with its specific advantages and limitations. Some converters are only capable of providing output voltages lower than input voltage (step-down) while others increase the voltage level (step-up) and some can do both. Another distinction is the direction of power flow. A unidirectional converter delivers current to raise its output voltage to a desirable level, but it cannot absorb current to inhibit the output voltage to be higher than set point level. Bidirectional converters can transfer energy in both directions between two dc sources while the voltage polarity and level at either end is within the operational range.

Another important distinction factor is full presence or lack of isolation between the input and output circuits. Non-isolated converters are generally used where the voltage needs to be stepped up or down, while isolated converters transfer energy between two circuits with galvanic isolation. Several topologies have been investigated for low power [1, 3, 8] and high power applications [4-12].

This paper proposes a wide-range input-output bidirectional dc-dc power converter. The proposed topology is a combination of a voltage-fed full bridge converter with a modified three-level resonant converter in order to gain a

bidirectional system. This converter provides a high frequency operation with zero voltage switching for all the switches, with no auxiliary circuitry or increased component stresses. The proposed control algorithm includes a combination of variable frequency control and pulse width modulation methods. The control variables of frequency, phase shift and duty cycle are selected precisely to have minimum currents for any active power level while zero voltage switching is maintained.

II. THE PROPOSED TOPOLOGY The proposed bidirectional converter topology is

illustrated in Fig. 1. For the power levels beyond 1000W, full bridge is the standard choice due to its better switch capability utilization. There is a voltage-fed full bridge block at the low voltage side to produce a nearly square wave voltage. This is a voltage-fed converter with no bulky dc inductor. There is a series resonant tank connected to its ac lines to shape the current waveform. The inductor of this tank is the leakage inductance of the transformer to reduce the size of the system toward a compact design. The selected switching frequency is always greater than the tank resonant frequency. On the other side of the tank there is a high frequency transformer for voltage level adjusting and galvanic isolation.

The secondary winding of the transformer reaches to ac lines of a three-level converter block. There are several reasons for selecting a three-level voltage-fed converter for the high voltage side: (1) the optimal operation of the circuit is based on having three level of voltage for VEF (see Figs. 1 and 4). A conventional half bridge leg is not able to provide such a waveform; (2) the circuit should be able to work with two-wire or three-wire voltage buses at high voltage end. A conventional full bridge block does not provide any center point for three-wire bus systems. In a three-level converter the midpoint of the splitting capacitor branch is available and can be connected to the zero voltage level of the three-wire bus (see the optional connection of the mid-point in Fig. 1). This is a highly demanded requirement in more electric

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airplanes, because this arrangement allows the operation of the same system in two or three wire bus architectures.

Any of the step-up and step down modes of operation can be divided into 18 intervals during one switching cycle. Theses intervals are illustrated in Fig. 2 and 3.

Fig. 1: Simplified block diagram of the proposed topology.

Fig. 2: Step-up operation intervals, waveforms and semiconductors status.

Fig. 3: Step-down operation intervals, waveforms and semiconductors status.

I

VAB VEF

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III. OPERATION PRINCIPLES

To extract the equations, we transferred VEF to the primary side, it is named VE’F’ and its amplitude, V2m, is scaled down by the turns ratio of the transformer. The amplitude of VAB is V1m. The switching frequency, 1/T, is higher than resonant frequency of the tank and the tank behaves as an inductor. The ideal waveforms of VAB, VE’F’ and I for both the step-up and step-down operation modes are shown in Fig. 4 in which T1, T2 and T are introduced. Appling T1, T2 instead of more familiar variables of duty cycle and phase shift provides better insight for the control as is shown in Fig. 5. In the case of having T1 ahead of T2 the converter is in step-up mode (power flow to high voltage side, HV) and when T1 happens after T2 the converter works in step-down mode (power flow to low voltage side, LV).

(a)

(b)

Fig. 4: Theoretical waveforms of VAB, VE’F’ and I for (a).step-up and (b) step-down modes.

The constraint on T comes from the requirement of switching frequency variation up to 20%, for example 250 to 300 kHz. Also, T1 and T2 should satisfy the condition of T1+T2<T/2. For each power level there are many possibilities of T, T1 and T2 to choose, as shown in Fig. 5. The task of controller is to find the optimum point in which these criteria are satisfied: (1) required power flow is achieved; (2) ZVS operation for both LV and HV sides is obtained; and (3) minimum Imax (or equivalently minimum Irms) is guaranteed.

Fig. 4 shows that to have ZVS for LV side switches I should be negative at rising edge of VAB. Similarly I should be positive at both rising edges of VE’F’ in order to have ZVS for HV side switches. The operating points satisfying both the conditions at shown in Fig.5. The transition from ZVS to

non-ZVS for LV side is when I becomes zero at rising edge of VAB. So this border is when a zero current switching (ZCS) happens. The same is true for HV side when I is zero at rising edges of VE’F’. The analytical formula for the LV-ZCS and HV-ZCS contours of Fig. 6 are calculated as (1) and (2), respectively. These formulas are essential in choosing proper operating points, i.e., selection of T, T1 and T2.

Fig. 5: Variations of P for different T1 and T2

at T=4µsec.

⎥⎦

⎤⎢⎣

⎡−⎟

⎠⎞

⎜⎝⎛ −+−= −

4 sin2

4sinsin1

4

2

11

112

TVVTTTTT

m

m ωωω

(1)

⎥⎦

⎤⎢⎣

⎡⎟⎠⎞

⎜⎝⎛ −−+= −

m

m

VVTTTTT

2

11

12

4

sin24

sinsin14

ωωω

(2)

Fig. 6: Contour of 1600W to find minimum Imax.

Best operating

points

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In Fig. 6 the contour for 1600W is drawn on top of Imax plot. For this power level, the best operating points are the ones with smallest value of Imax. This is achievable when the operating point is selected from the points near the LV-ZCS contour (see Fig. 6). This observation is true for all the power levels. Choosing an operating point on the constant power contour close to LV-ZCS contour minimizes the conduction loss and retains the ZVS for both the LV and HV switches. Selecting such an operating makes a relation between T1 and T2. The condition of being in vicinity of LV-ZCS contour fixes the T1 value.

Fig. 7: Variations of P for different V1m and V2m

at T=3.33µsec.

Fig. 8: Variations of P for different V1m and V2m

at T=4µsec.

Fig. 6 also reveals that using a constant T it is not possible to achieve best operating point defined above for any desired power level. Thus T should be smaller for lower powers (Fig.7). The active power also depends on both V1m and V2m as illustrated in Fig. 8. By increasing the switching

frequency it is possible reduce power level in order to compensate the variation of V1m and V2m as well as variation of power, based on the demand or charging curve of the batteries connected to the LV bus.

IV. EXPERIMENTAL RESULTS

A prototype converter was developed to pass 2kW from 28V (20-32 V) to 270V (240-290 V) and vice versa. The parameters are given in Table 1. The experimental waveforms of VAB, VE’F’ and I for step-up operation for a power of 1kW are shown in Fig. 9. The ZVS of all switches are evident: I < 0 at rising edge of VAB, which mean ZVS in LV side and I > 0 for both rising edges of VE’F’ which is the indicator of ZVS for HV side, as expected.

Table 1: Component Values.

Parameter Value Ls, resonant tank inductance 1.25µH Cs, resonant tank capacitance 386nF

N, transformer turns ratio 3 F, switching frequency 250kHz V1m, amplitude of VAB 20V V2m, amplitude of VE’F’ 240/2/N = 40V

Fig. 9. Experimental waveforms of VAB, VE’F’ and I in step-up mode. (10v/div, 100A/div and1µs/div)

An overall efficiency of 94% is achieved at the conditions of Fig. 9, which is very high considering that LV side has a low voltage and high current. This shows the importance of minimizing the current level in order to decrease the inevitable conduction losses.

V. CONCLUSION

In this paper, a resonant power converter system to transfer energy from a 28V dc bus to another 270V dc bus

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and vice versa has been presented. This topology is able to provide soft switching for all the semiconductors involved in the system. The possibility of a reduced conduction loss with no compromise in ZVS operation or power handling has been expressed. With no bulky dc inductor, this converter can be very compact, with a high power density, which is essential in aerospace applications. Theoretical analysis and evaluation results have been presented to confirm the feasibility and performance of converter system.

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