Electronic Front End for Rogowski Coil Current Transducers with Online Accuracy Self Monitoring

Joris PascalI, Richard BlochI, Stephane Isler2, Lucas Georges3 I ABB Switzerland Ltd, Corporate Research, Baden-Dattwil, Switzerland

2 ABB Secheron Ltd, Geneva, Switzerland 3Ecole Centrale Paris, Paris, France

joris.pascal@ch.abb.com

Abstract-. In order to lower the overall amplitude error of the combination of a medium voltage Rogowski coil current transducer (RCCT) with its associated readout electronics, a specific electronic architecture is presented here. It implements online accuracy monitoring, i.e. an online self calibration, that enhances accuracy performances of the electronics. This new architecture also provides transducer connection diagnostic. As illustrated by the experimental results, this new accuracy drift monitoring function allows one to divide the amplitude error caused by the drifts of electronic components by a factor of 10. Besides, the electronic front end automatically detects the disconnection between the coil and the usually remote electronics. The new Rogowski coil read out electronic front end is implemented with standard low cost CMOS electronics supplied with 5V. However it is compatible with high primary current measurements which correspond to several tens of Volts at the RCCT output.

I. INTRODUCTION

In many medium voltage industrial and power distribution systems, the Rogowski coil current transducers (RCCT), known as non conventional instrument transformers (NCIT), are good candidates to replace the conventional ferromagnetic cored instrument transformers for current measurements. RCCT offer higher dynamic range and better linearity thanks to their non magnetic cores. Besides, they can be manufactured at lower cost and in smaller size. However, they still suffer from low accuracy performances in comparison to conventional ferromagnetic cored current transformers [1]. External electromagnetic field sensitivity and temperature dependency are typical error sources of a Rogowski coil [2]. Therefore, in parallel to transducer improvements, one has to lower as much as possible the error budget of the read out electronics. Indeed, this should lead to a combination transducer-electronics which exhibits accuracy performances comparable to those of conventional instrument transformers, e.g. class 0.5 according to IEC 6 0044-8 [3]. In this paper we present a Rogowski coil read out electronics with self calibration technique and detection of transducer disconnection. The next section illustrates a conventional electronic architecture for interfacing Rogowski coils and exposes the challenging aspects in terms of accuracy and stability of electronic parts. The new approach for accuracy improvement is then described. Finally, the experimental

results obtained on an electronic prototype are given, followed by a conclusion.

II. ROGOWSKI COIL READ OUT ELEClRONICS

A. Conventional approach

In a conventional architecture the read out electronics for Rogowski coils is a differential amplifier loading an analog to digital converter as illustrated in Fig.I. Since the transducer is delivering a voltage proportional to the derivative of the current, its output signal needs to be integrated [4]. This integration can either be performed by an analog set up or in the digital domain. For the sake of simplicity, this integration is not depicted in the different schematics in this paper.

Fig. I. Simplified schematics of a conventional Rogowski coil read out electronics. The necessary integration of V, is not represented here.

The gain is configured according to the rated primary current Ir• For high currents the amplifier stage can be set as an attenuator to limit the voltage at the integrated amplifier inputs to less than the supply voltage also used for the rest of the circuitry. To avoid the use of additional supply voltages, attenuation of more than factor 10 can be necessary. The use of discrete resistors is then required when designing above IMO input impedance (RI•3 on Fig. 1) with commercially available integrated circuits. The RCCT often require high load to operate with maximal stability performances [1]. Unfortunately, the higher the resistor value, the more it is expected to drift with temperature and ageing [5]. This leads to a non negligible effect on the gain stability of the amplifier and therefore on the overall accuracy performances of the

978-1-4673-0342-2112/$31.00 ©2012 IEEE 1037 leIT 2012

RCCT based measuring system. In order to compensate this drift, as well as the one from the ADC gain, we propose a specific self calibrating architecture as described in the next section.

B. Online self calibration architecture

The principle of an online self calibrating instrumental chain is well known and has been implemented in many high accuracy sensor applications [6 ]. This principle allows calibrating without interrupting the measurement, and is therefore called online. In our application, this characteristic is a must since RCCT are used to perform protection functions which need to be triggered with minimal delay [2]. In addition, the reference used for calibration is generated by the device itself which prevents the need of any external reference source and saves maintenance operations. This principle is called self calibration.

A typical online self calibrating instrumental chain is depicted in Fig. 2. The sum of the rated input signal ( the RCCT secondary voltage V s) and the analog to digital converter reference V,ef is performed at the input of the instrumental chain. This sum will then be amplified by the product of GA and GB which corresponds to the amplifier and ADC gains, respectively. The ADC is then comparing the reference signal V,ef with its input signal (V,ert Vs)' If Vs and V,ef are located in different frequency domains it is easy to numerically separate the two signals in order to isolate the component of the output signal which is equal to

GA·GB·V,ev'V,eFGA·GB. This component of the signal is constant when the gains GA and GB are constant. This is valid independently of the stability of the reference voltage V,ef. Indeed, the analog to digital conversion is a comparison, i.e. a ratio, between the ADC input and the ADC reference V,ef. Therefore the drift of GA'GB can be tracked and corrected numerically. The calculated drift of GA'GB is used to correct the component of the signal which carries the RCCT secondary voltage Vs. By this means, the RCCT read out electronics monitors and compensates its own gain drift.

Fig. 2. Online self calibration set up. The same reference voltage as used by the analog to digital converter is injected at the input of the instrumental chain as the calibrating signal.

C. The new self calibrating electronics

Two challenging operations appear when the online self calibration principle described in the previous section has to

be implemented for a RCCT read out electronics similar to the one depicted in Fig. 1: first, the summation of the reference signal and the RCCT output signal V s, and second, the modulation of the reference.

The input signal has to be superimposed to the reference at the input of the instrumental chain where the RCCT output voltage Vs settles. The possibly high voltage value of V s makes it difficult to operate a physical sum of two signals at this node of the circuit. Certainly, Vs can reach higher amplitudes than the supply voltage of the integrated circuits, e.g the amplifier. For example a RCCT with a sensitivity of 12� V/A-Hz will generate Vs = 36V when I, = 4kA at 750Hz. In addition to this, we have to modulate V,ef in order to shift the reference to a different frequency domain than the rated signal which is typically [50Hz-750Hz] [3]. Besides, we cannot simply inject the reference at DC as it is done at the ADC input. Actually we have to separate the reference from any offsets that could be added by the amplifier or by the ADC. This modulation operation needs to be performed without adding any error on V,ef since V,ef will be used for drift correction.

The proposed solution is depicted in Fig. 3.

Data Ol ADC

Fig. 3. New online self calibration architecture applied to a RCCT read out electronic front end. Instead of being injected at the high voltage input nodes where V, settles, the reference voltage, also used by the analog to digital converter, is injected between the low voltage nodes P and S (R,<Rt).

In order to sum the reference signal V,ef and the RCCT output signal Vs without having access to the possibly high voltage nodes, we place two voltage reference sources directly at the amplifier summing points S. In this manner we access to low voltage nodes since the attenuation of the voltage is set by the ratio R2/R1<1.

Fig. 3. illustrates the sum of the reference signal modulated by a square carrier ± V ref and the input signal V s' In practice we have to implement the modulation of V,ef without introducing any drift on the amplitude value which is to remain precisely equal to V,ef' In Fig. 4, the voltage source symbol ± V,ef used in Fig. 3 is depicted in the way it is actually implemented. Two capacitors are alternatively connected between the nodes P and S or between V ref and ground. This connections and disconnections are performed with analog switches at the frequency fH• It allows one to maintain a constant voltage V ref across the capacitors. By

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operating the switches fast enough the capacitors do not have enough time to discharge through the resistor RJ. the coil and the resistor R3• A second switching scheme performed at lower frequency fL inverts the voltage across the capacitors. In this way the capacitors will be alternatively charged to Vref or to -Vref. This is necessary to provide an AC square voltage source between the nodes P and S.

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Fig. 4. Detail of the square voltage source with amplitude V ref and frequency fl. The high frequency fH switching is used to compensate the natural discharge of the capacitors through RI• the coil and R3. The low frequency fl switching inverts the V ref voltage sign across the capacitors.

Finally one square voltage source is introduced at both summing points S of the amplifier as depicted in Fig. 3. The two V ref voltage sources are in opposite phases in order to create a differential reference signal to be transmitted to the ADC.

D. Self diagnostic function

The calibration scheme presented in the previous section introduces a reference current which flows through the RCCT winding since the two Vref voltage sources are in opposite phases. It therefore creates an easy way to detect if the transducer is actually connected to its electronic front end input or not. In fact the Rogowski coil is a rather low resistive device of about few tens of Ohms [4]. As a consequence, the amplitude of the measured square reference signal in normal conditions will be drastically higher than the one with an open circuit, i.e. a disconnected transducer. Besides calibration, this additional feature is of interest for ensuring a diagnostic of the RCCT cable, which can be for example accidentally cut or mounted with a defect connector.

III. EXPERIMENTAL RESULTS

A. Test conditions and experimental test bench

The test conditions and relevant design details are summarized in Table I.

TABLE I DESIGN DETAILS AND TEST CONDITIONS OF THE TESTED PROTOTYPE

RCCT secondary voltage Vs O.5V

Rated primary current frequency f, 50HZ

Input resistors RI) IMO Feedback resistors R3,4 180kO

Switched capacitors CI) IOOnF

Reference voltage Vref 2. 5V

Switching frequency fH I. 4KHz

Modulated reference frequency CLK V"f fl 1.25Hz

ADC output data rate fs 4KHZ

In order to evaluate experimentally the efficiency of the proposed self calibrating front end we have introduced in our prototype some variations of component values. One of the most critical parameters in the instrumental chain depicted in Fig. 3 is the input resistors values which will set the gain. Their high ohmic value makes them sensitive to ageing and temperature [5]. For the test purpose we have therefore increased RJ between 0% and 4% from its initial value. Each modification has been analyzed on the digital output signal as depicted in Fig. 5. A reference measurement has been performed with a calibrated acquisition card (NI PXI 446 2). In order to maintain the measured signal at a stable value, and to avoid using a test set up generating primary current Ir in the kA range, the RCCT has been replaced for the test purpose by an arbitrary waveform generator with differential output. It is important to note that the use of a waveform generator instead of a RCCT has no influence on the performances evaluation of our system. As a matter of fact our architecture is aimed at correcting only errors generated by the electronics and not by the possible RCCT sensitivity variations. The gain drift correction is applied in the software domain. In the present prototype this software has been written on a PC using Lab VIEW. In future development it will be implemented on an embedded target. The ADC clocks and the control signal which generates the V ref square voltage source at the frequency fL are also generated by the software. In this way, we ensure a synchronous demodulation of the ADC output signal as it is detailed in the next section. Finally, the residual error Er is computed and serves as conclusion about the efficiency of the method.

Fig. 5. Experimental test bench. The signal processing of the measured data is performed on a PC and the residual error £, on the delivered signal Vco=ted is calculated after comparison with the measurement of V, performed by a calibrated acquisition card (NJ PXI 4462).

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B. Time domain results analysis

A time domain measurement with the parameters described in Table 1, and according to the test set up of Fig. 5 is depicted in Fig. 6 .

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Fig. 6. Time domain analysis. The output of the ADC V ADC provided by our device under test is reconstructed in software to deliver a corrected value V oorrec�d to the final user. Vcorrected does not contain any more the low frequency reference that was injected by the hardware and its amplitude has been corrected as a gain drift has occurred in the instrumental chain.

The output of the device under test is labeled with V ADC

and corresponds to our electronic front end ADC output. We clearly see that V ADC contains two kinds of information: the rated sinusoidal signal at 50Hz and the square modulated reference signal at 1.25Hz. These two signals have been superimposed by our hardware set up ( see Fig. 3). By this means, both signals have been transferred from the RCCT output (V 5) to the ADC output through the same gain chain. The signal V ADC is then processed in software in order to separate it into two signals, the reference signal V refmod and the demodulated output signal V demod. The signal V refmod is obtained by a low pass FIR digital filtering with a 151 notch located at the rated frequency fr in order to optimally remove the rated signal. The demodulated signal V demod is obtained by synchronously adding to or subtracting from V ADC the value Vref. The delivered signal is then proportional to the input signal V5• In order to correct V demod by the appropriate gain value taking into account the possible gain error, the algorithm has to extract the gain value from a known signal, in our case Vref. The amplitude measured on the Vrefmod signal allows one to calculate the gain of the electronic front end. The reference voltage Vref is constant and equal to 2.5V. As can be seen in Fig. 6 the gain is approximately 0.45/2.5=0.18.

This calculation takes into account every possible variation of electronics components that could affect the gain value ( temperature and ageing drifts of the resistors and ADC gain). In our experiment we have modified the resistors value to introduce a gain drift and calculated the accuracy of the gain drift correction provided by our self calibration. The results are given in Fig. 7 and show a reduction of factor 10 of the gain drift. Actually in our differential set up, as we modified only one input resistor, a drift of 4% on a resistor introduces a gain variation of 2%. As shown in Fig. 7, after correction, our residual gain error Er is less than 0.2%.

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Fig. 7. Residual error calculation. An improvement by a factor of approximately 10 is achieved. A 4% drift on a single input resistor which introduces 2% gain drift is corrected down to less than 0.2%.

IV. CONCLUSION

The new read out electronic front end for RCCT presented here offers transducer connection diagnostic and electronics accuracy drift monitoring. The gain drift introduced by the electronics through ageing or temperature dependency is lowered by a factor 10. Despite low voltage electronic power supply, e.g. 5V, the new front end is compatible with transducers signal output which can reach few tens of volts. By lowering the error budget of the electronics, this new front end, opens the way to the design of Rogowski coil based electronic current transformers with high accuracy, i.e. comparable to the one offered by conventional ferromagnetic cored transformers.

REFERENCES

[1) W.F. Ray, c.R. Hewson, "High performance Rogowski current transducers" in Proc. IEEE Industry Applications Conference, pp. 3083-3090. 2000

(2) IEEE Guide for the Application of Rogowski Coils Used for Protective Relaying Purposes, Feb. 22 2008.

(3) Instrument Traniformer-Part 8: Electronic Current Transformers, IEC 60044-8 first edition, JuL 2002.

(4) E. Abdi-Jalebi, R. McMahon, "High-Performance Low-Cost Rogowski Transducers and Accompanying Circuitry", IEEE Transactions on instrumentation and measurement, vol. 56, no. 3, June 2007.

(5) R. W. Kuehl "Stability of thin film resistors - Prediction and differences base on time dependent Arrhenius law." Microelectronics Reliability, vol. 49 pp. 51-58,2009.

[61 M. Pastre, M. Kayal, and H Blanchard, "A Hall Sensor Analog Front End for Current Measurement With Continuous Gain Calibration" IEEE Sensors Journal, vol. 7, no. 5, May 2007.

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