SCIENCE

Interference between braided coaxial cablesA.H. Badr, B.Sc, M.Eng., Ph.D., Prof. F.A. Benson, D.Eng., Ph.D., C.Eng., F.I.E.E., and J.E. Sitch, B.A.,

M.Eng., Ph.D.

Indexing terms: Noise and interference, Coaxial cables

Abstract: The concept of transfer (coupling) impedance is applied in the determination of the mutualcoupling between braided coaxial cables in free space or over a metallic ground plane. Good agreementis found between experimental and calculated results using measured values of the transfer impedance.

1 Introduction

Modern developments in the miniaturisation of electronicequipment have made possible the concentration of complexsystems within confined spaces, as in aircraft. With the con-centration have come many problems of mutual interference,and a considerable part of this interference can arise owingto leakage of signals from the coaxial cables used to connecttogether the units of the systems installed. Various methodsof immunising against such interference are possible. Eachsuch method, however, usually requires additional circuitcomplexity or an increase in system size and weight. Hence,their use should first be justified on the basis of known, or.accurately estimated, levels of interference. Cables withhomogeneous tubular outer conductors have negligible leakagethrough their shields if the frequency is sufficiently high(associated with skin effect [1]); but braided screens, owingto their interwoven nature and the gaps in the braid, exhibita marked worsening of the shielding properties at high fre-quencies compared to those of solid tubes. Two additionalprocesses are involved in the transfer of electromagneticenergy through the braid of a coaxial cable; a form ofmagnetic coupling due to the inductive nature of the braidand an electric coupling due to mutual capacitances betweenthe cable and its environment. The magnetic coupling iseffected through the transfer impedance ZT of the braid andit is this coupling which has mainly been investigated inthe present work. Trie transfer impedance of the screen isdefined as the ratio of the voltage generated outside theshield to the current flowing on the inside, and it largelydetermines the shielding against interference currents; thelower the value of transfer impedance, the better the screeningproperties of the cable. The transfer impedance has beencalculated for coaxial cables with solid screens [1] but anytheoretical analysis [2—5] attempting quantitative resultson the transfer impedance of braided cables is extremelydifficult and prone to errors owing to the physically complexconstruction of the screen. However, it is possible to measurethe transfer impedance using a triaxial tester. It was thereforedecided to determine the values of the transfer impedancefor the cables under investigation experimentally, and touse these results in the theoretical analysis for calculatingthe coupling factor. The transfer impedance measurementswere performed by using a standard triple-coaxial apparatus[6] in the frequency range from 100 kHz to 150 MHz. Thisapparatus is based on the principle of measuring leakageby collecting the leakage energy in a coaxial system com-pletely surrounding the leakage source. In the frequencyrange of 100 MHz to lGHz the measurements of ZT werecarried out using a triaxial device designed [7] for thatpurpose, and similar to that used by Ditscheid [8]. The

Paper 1368A, first received 25th November 1980 and in revised form25th March 1981The authors are with the Department of Electronic & ElectricalEngineering, University of Sheffield, Mappin St., Sheffield Si 3JD,England

triaxial device used for the measurements is similar in principleto the triple-coaxial apparatus used for the low-frequencymeasurements, the main difference being that it uses properlyterminated outer and inner coaxial circuits. Also, the velocityof propagation in the outer system is matched to that in theinner by incorporating within the outer coaxial line a dielectricmaterial having the same dielectric constant as the cabledielectric.

The transfer impedance is a property of the cable screen,but the leakage from any installed cable is not only a functionof this transfer impedance; it is influenced by the externalcircuit within which the leakage signals flow. A knowledgeof the transfer impedance will define the leaked-signal voltageapplied to the external circuit, but the current flowing andthe power drawn from the leakage source will depend onthe load impedance which the external circuit presents. Thisexternal circuit may be sharply resonant at the workingfrequency, in which case the leakage may be serious, or itmay be damped so as to present a low-(? resonance wellaway from the working frequency, in which case the effectswill probably be acceptable.

Once the leakage voltage is known from a determinationof the transfer impedance, the cable current and the lengthof the cable, the next problem is to determine the equivalentcircuit corresponding to the external system. This is the mostdifficult part of making quantitative estimates of screeningperformances. Much of the early work on this problem wascarried out at the Bell Laboratories with special referenceto the crosstalk between coaxial telephone lines [9—11].Formulas for the crosstalk between parallel cables placedeffectively in free space were published by Schelkunoffand Odarenko [9]. Experimental studies confirmed theaccuracy of their predictions. The case of coupling betweenscreened lines over a common metallic ground plane hasbeen considered by Mohr [12,13], but the use of a lumped-circuit model limited his work to frequencies below about65 MHz. For any particular system this method cannot beused at frequencies for which the cable lengths are appreciablefractions of a wavelength. However, an attempt has beenmade in the present work to obtain a general expressionfor calculating the coupling between two braided coaxialcables in free space or over a metallic ground plane withbraids short-circuited at both ends, using the measured valuesof the transfer impedance of the cables under investigationand using the braiding factor according to Blackband forcalculating the attenuation of the braided cables and thetertiary circuit between the coupled lines.

2 Theoretical expressions

2.1 Coupling between coaxial cables in free spaceThe general case of far-end coupling between two coaxiallines of length / with the tertiary short-circuited at eachend (the tertiary circuit consisting of the two outer shellsof the coaxial conductors) is investigated as follows:

Fig. 1 shows the two transmission lines (1) and (2), both

IEEPROC, Vol. 128, Pt. A, No. 5, JULY 1981 0143-702X/81/050347 + 07 $01.50/0 347

lines being terminated in their characteristic impedanceZo i and Z 0 2 , and their propagation constants per unit lengthare yx and y2, respectively. The tertiary circuit (3) betweenthe two transmission lines is short-circuited at both endsand has a characteristic impedance and propagationconstant of Z0 3 and y3, respectively. The voltage impressedon the disturbing coaxial circuit (1) induces currents andvoltages in the intermediate circuit (3), which now actsas a disturbing circuit for the second coaxial circuit (2),thus causing coupling. If the disturbing voltage E is applied

a!

Z32"T

01

x cJx ^ d y

l

Fig. 1 Coupling between two coaxial lines

to the left end of line (1), and the induced voltage Vf ismeasured at the right end of line (2), the ratio Vf/E is definedas the far-end coupling. From Fig. 1, at any point at a distancex from the sending end, the elemental voltage on the outersurface of the outer coaxial conductor, will be iZx3e~lx xdx,where Z13 is the surface transfer impedance per unit lengthof cable (1) and / is the current delivered by the source.Each differential element iZX3e~lxXdx of this voltage dropwill produce a current iKL in the tertiary circuit determinedby the impedance Z' and Z" of the tertiary as seen in thetwo directions from this point

, -7i*

Z'+Z"dx

where

Z' = Z0 3 tanh y3x

Z" = Z0 3 tanh73(/— x)

0)

(2)

(3)

From transmission-line theory, the current at any point xon a line in terms of the current at the sending end is:

= h cosh yx — Vs/Zo sinh yx (4)

In Fig. 1 at any other point at a distance y from the sendingend, the tertiary current i3(y) due to the voltage iZi3e~Jxxdxis obtained as follows:

(/) for y > xV8 = IKLZ' and Is = iKL, by substituting in eqn. 4

' 300 = IKL cosh 73 (y -x)-iKLZ"/Z03 sinh y3(y -x)

(5)

From eqns. 1—3 using eqn. 5:

'a 00 =' 03

cosh y3(l— y)

tanh y3x cosh 73 (/ — x) + sinh 73 (/ — x)(6)

From this the transfer admittance A(x, y) between thesetwo points is

1

cosh 73 (I —y)

A(x,y)

tanh 73* cosh 73(/ — x) + sinh y3(l~x)

(7)

(«) for y < x=

IKLZ' a nd I8 = IKL> by substituting in eqn. 4:

hiy) = IKL cosh 73 (x -y)—iKLZ'/Z03 sinh y3(x —y)

(8)

A(x,y)

Similarly foiy <x, this transfer admittance is obtained as:

1

y<x ^03

cosh y3y

sinh y3x + cosh 73x tanh 73 (/ —x)(9)

dx

coaxial (1) L_v zy3'*O

coaxial (2)

—e,/2 e , / 2 ^

groundplane

Fig. 2 Coaxial coupling schematic diagrams

348 IEEPROC, Vol. 128, Pt. A, No. 5, JULY 1981

The tertiary current z3 (x) will be

i3(x) =

= \*iZae->*yA(pc,y) dyy<x

+ dy (10)

The first integral has the following form:

/Z13 cosh 73 (/—JC)

2Z03 sinh 73/

1

[73 — 7i

and the second integral:

/Z13 cosh 73 JC

2Z03 sinh 73/

1

73 ~7i

7i +73

(ID

1

7i +73

(12)

Adding eqns. 11 and 12, and collecting the common terms,z3 (x) is given by

iZ 13

2Z03 si

1

+73{cosh73(/— x)

— e~7>' cosh 73X + e~7lX sinh 73/}

{cosh73 ~7i

— e~7lZ cosh 73*}

sinh73/

(13)

This tertiary current /3 (x) will induce an elementary electro-motive force in the second coaxial conductor given by e =/3(x)Z32cfcc. The contribution of this electromotive forceto the voltage across the far end of the second coaxial pairwill be

dVf =2Z Q2

[l

f =I JO

)dx

(14)

(15)

For two identical cables: Z01 =Z02 =Zo,Zu =Z32 =ZTand7i = 72 = 7, hence from eqns. 13 and 15, after performing

the integration, the following expression for the far-endcoupling can be obtained:

r y, _E

(cosh 73/—cosh 7/))

(16)

where Z33 is the series impedance of the tertiary circuitper unit length, Z33 = 73Z03. Or

C=201og10 |F|dB

where Cis the coupling factor in dB.In the above analysis the reaction of the induced currents

on the disturbing line has been neglected, assuming weakcoupling.

2.2 Coup/ing between coaxial cables over a metallic groundplane

In the case of two coaxial cables running parallel over ametallic ground plane, there are two indirect far-end couplingcomponents (indirect coupling means that the energy transfertakes place via an intermediate circuit causing the coupling[9]), one between the two coaxial lines and another betweenthe lines and ground plane. To analyse the coupling in thiscase, the problem can be tackled as follows.

Consider first an elementary section dx of a long singlecoaxial line in free space as indicated in Fig. 2a. If theimpressed current at that point is Ix, an open-circuit voltageequal to ex =IlZi3dx is developed on the outer surface ofthe outer coaxial conductor. The term Z13 represents thesurface transfer impedance (mutual impedance) per unitlength between the inner and outer surface of the outercoaxial conductor. Now suppose that another long coaxialline is placed parallel to the first one as shown by Fig. 2b.The open-circuit voltage ex on length dx of the first coaxialouter conductor will now cause current to flow in the inter-mediate circuit composed of the two outer conductors. Theparameters of this circuit are 73 and Z03. On returning inthe second coaxial outer conductor, this current causes coupl-ing into the second coaxial circuit. It is convenient at thispoint to replace the original impressed voltage ex by theset of EMFs shown in Fig. 2c. The insertion of equal andopposite voltage ei/2 on the outer surface of the disturbedcoaxial outer conductor does not change conditions, butenables the consideration of certain effects separately. Thefirst effect to be considered is that due to the pair of equaland opposite voltages ex/2 in the loop composed of thetwo coaxial outer conductors. These voltages combine toform a 'balanced' voltage ex which tends to drive currentaround the balanced circuit composed of the two outerconductors. For the present, the voltages exl2 which arein the same direction in the outer conductors should notbe considered. The current in the 'balanced' intermediatecircuit of characteristic impedance Z03 and propagationconstant 73 due to the balanced voltage ex is given by eqn.13 in the preceding Section (the intermediate circuit consistsof the outer shells of the coaxial conductors and is short-circuited at both ends). This current flowing along the outercoaxial conductor of the disturbed circuit produces a voltage^2 = h{x)Zyj,dx on the inner surface of this outer conductor,and this voltage in turn causes a current i2o [the subscript oin i2o relates this current to the balanced (odd) mode current]in the disturbed coaxial circuit equal to e2/2Z02, where Z02is the coaxial characteristic impedance. This current willcause a voltage across the far end of the second coaxial cable.When the effects are integrated, the far-end coupling forthis component (odd mode) can be obtained in a similar

IEEPROC, Vol. 128, Pt. A, No. 5, JULY 1981 349

way to eqn. 16 in the preceding Section (for identical cables,ZQI =Z02 =Zo,Zn = Z32 =ZT and7i = y2 = 7).

Z\e

2ZoZ33{y23-y

7 3 ' 273 72 (cosh 73 / — cosh 7/),2\2(73 - 7 ) sinh73/

(17)

where Zo and 7 are the characteristic impedance and thepropagation constant of the coaxial cables, respectively.

Now consider the coupling contribution due to thelongitudinal voltage ex\2 acting along both coaxial outerconductors in parallel. When the two cables are placedsymmetrically above the ground plane, as shown in Fig. Id,the longitudinal voltage sends a current equal to /4 aroundthe circuit composed of the two paralleled outer conductorswith the ground return (the two outer conductors are short-circuited to the ground at both ends).

' 4 =ex/2

Z'+Z"(18)

where Z' and Z" are the impedances of the tertiary circuitas seen in the two directions from any point x on the line.This longditudinal current /4 in eqn. 18 can be found, in asimilar way to eqn. 13, to be:

4Z04ZQl sinh 74/ 17 + 74

1{cosh 74 (/ — x)

74-7

e~yx sinh74/}

{cosh 74 (/ — x) — e yx sinh 74 /

— e cosh 74;t} (19)

where Z 0 4 and 74 are the characteristic impedance and thepropagation constant of the ground/outer-conductor circuit,respectively. Half of this longitudinal current flows on thedisturbed coaxial outer conductor in opposition to thebalanced current i3 flowing there. Following the previousprocedure it can be shown that, in the elementary length,a coupling current i2e = i^ZTdx/4Zo will flow in the dis-turbed coaxial current (the subscript e relates this currentto the even mode current). When the effects are integratedover a length / (similar to eqn. 16) the far-end coupling forthis component is found to be:

Z\e~yX \ y\l 27472 (cosh 7 4 / - c o s h 7 / )

i\-l2 (74 ~ 7 2 ) 2 sinh 74/

(20)

spectrumanalyser

50H termination

ground plane>

aluminum plategenerator

Fig. 3 Experimental setup for coupling measurements

350

where Z44 = Z 0 4 7 4 = series impedance per unit length of thecircuit composed of the outer conductors with ground return.

The total far-end coupling is the vector sum of the abovetwo components:

F = + F, =z_h'yl

2Z0Z33 I 73 - 7

273 72 (cosh 73/ — cosh 7/)

(73 ~ 7 2 ) 2 sinh 73/

8ZnZ 44

747

7 4 - 7 2

274 72 (cosh 74 / — cosh 7/))

(74 - 7 2 ) 2 sinh 74/ j

(21)

Measurement techniques

The experimental setup for the coupling measurements infree space or over a metallic ground plane is shown in Fig. 3.The cables are placed in parallel over a metallic ground planewhich is of course removed when the free space situationis examined. The two coaxial cables are fitted with bulkheadconnectors at each end, which are mounted on aluminiumplates bolted to the ground plane. The signal source is con-nected at the end of one line, and the detector at the farend of the other line, while the two other ends are terminatedin their characteristic impedance Z o .

Each time the coupling is measured, the spectrum analyseris tuned to obtain a suitable signal level. The cable sampleis then disconnected and the attenuator is adjusted to givea signal level identical to the first. The attenuator reading indecibels is a direct measure of the far-end coupling betweenthe two lines.

All the connections to the sample under test are madeusing solid jacketed cables (Andrew Heliax — type FSJ1-50)to avoid any measurable coupling between the connectingcables and the cable under test.

-20r

-40

-60CO^ . -80o

5-100

'3.-120

8-140

x measured- computed

-160105

Fig. 4

10b 10' 10° 10a

frequency, Hz

Coupling against frequency in free space

L = 1 m, D = 1 cm

4 Results and discussion

The calculations and measurements of coupling between twosimilar coaxial cables were carried out on samples of UniradioNo. 43 cables under different conditions:

(i) varying cable-to-cable separation(ii) varying cable-to-ground distance(iii) using different sample lengths.

The results of the transfer-impedance measurements whichwere used in the calculations of the coupling factor for thecable under investigation are shown in Fig. 10.

IEEPROC, Vol. 128, Pt. A, No. 5, JULY 1981

The results of coupling presented here are divided intotwo groups: first, the coupling between two braided coaxialcables in free space (without a metallic ground plane), andsecondly, the coupling between two braided coaxial cablesover a metallic ground plane. All the experiments were carriedout on samples of cables with their PVC protective jacketsremoved. The reason for removing the PVC jacket is tocompare the experimental results with those predicted bytheoretical analysis, since the analysis is based on the assump-tion that the dielectric medium surrounding the cables ishomogeneous.

L\ the first group, Figs. 4 and 5 show the theoretical andexperimental results of the coupling factor against frequency

-20r

105 106 107 108

frequency, Hz

Fig. 5 Coupling against frequency in free space

L = 1 m, D = 2 cm

109

x measured- computed

x x ~ x x

10* 107

frequency, Hz108 109

Fig. 6 Coupling against frequency over ground

L = 1 m, D = 1 cm,H = 6 cm

for two similar samples of Uniradio No. 43 (UR43) with cable-to-cable separations of D = 1 and 2 cm, respectively, thecable length / was 1 m. The theoretical graphs presented assolid lines were obtained by plotting the coupling factorusing eqn. 16, against frequency, and the points are exper-imental results.

In the second group, the theoretical predictions for cablesover a ground plane were obtained from eqn. 21. Figs. 6and 7 show the results of coupling between two similarcoaxial cables of type UR43 against frequency. The cableswere 1 m in length and the cable-to-cable separation was 1 cm.the heights of the cables over the ground plane were 6 cmand 2 cm, respectively. Fig. 8 is similar to Fig. 6, except that

-20

-40

o

en

•Q.

8

x measured- computed

-16010s 10° 10'

frequency, Hz10°

Fig. 8 Coupling against frequency over ground

L = 2 m,D = I cm, H = 6 cm

the lengths of the coupled lines were 2 m instead of 1 m.Fig. 9 is similar to Fig. 7, except that the cable-to-cableseparation was changed from 1 cm to 6 cm.

The experimental results can be discussed in the light ofthe theoretical calculations in each of the two previous groups.These results can be divided into two frequency ranges

(a) low-frequency range, where the length of the coupledlines is much smaller than quarter wavelength (i .e . /= 30MHzat /=X/10 for a length equal to 1 m of the intermediatecircuit)

(b) high-frequency range, for frequencies above 30 MHz.In the low-frequency range, the coupling factor, given byeqn. 16, reduces to F = Zj<l/2ZOZ33 where Z33 is the seriesimpedance of the intermediate circuit. Comparing theexperimental results with the theoretical ones in Fig. 4, theagreement is within 4 dB. The effect of varying cable-to-cableseparation in this low-frequency range can be seen in Figs. 4and 5. In these graphs the coupling factor is decreasing with

107

frequency, Hz

Fig. 7 Coupling against frequency over ground

L = 1 m, D — 1 cm, H = 2 cm

10s 10'frequency, Hz

10° 103

Fig. 9

L = 1 m

Coupling against frequency over ground

D = 6 cm, H = 2 cm

IEEPROC, Vol. 128, Pt. A, No. 5, JULY 1981 351

increasing cable-to-cable separation D, because 1/Z33 isdecreasing according to a logarithmic law with increasing D.

In the high-frequency range, referring again to Fig. 4,considering the coupling factor given by eqn. 16, it can beseen that, owing to the presence of the hyperbolic functionsin that equation, the resulting curve of coupling againstfrequency is a line with sharp peaks and troughs superimposedon a smooth curve, with successive maxima at frequencieswhere the length of the intermediate circuit becomes amultiple of half wavelengths. The smooth curve is givenby the magnitude of the first term of eqn. 16 which is•\Z\12ZOZ^\. In the same Figure, the peak expected at60 MHz is missing. This collapse of the maximum at such afrequency is not a general case, but is only observed forpolythene cable dielectric (velocity ratio K = 0.667) whenthe dielectric of the intermediate circuit is air (velocity ratioK' = 1). If the dielectric of either of them is changed, thecollapse of the maxima can happen at other frequencies,or it may not occur at all in the frequency range of interest.The maxima are caused by tertiary resonances, whereas

w

010

EaM

-110

-21C

•

•

•

-_

-

.

7 '7

77

\

7

7 ^7

7

77

7

77

7

7

7

77

7

7

7

7

7

/r

105 10b 107

frequency, Hz108 10b

Fig. 10 Transfer impedance against frequency for UR43 (wire-braided)

the minima are caused by cancellation of the transfer intoand out of the coaxial cables, due to differing phase velocities.When they coincide, either or both may disappear. As fre-quency increases the losses of both the cables and the tertiarycircuit increase and imply damped resonances, but at thesame time the transfer impedance takes on large values andthis increases the coupling. Comparing the experimentalresults with the theoretical ones in Fig. 4 in the higherfrequency range, the agreement is within 6 to 12 dB, exceptat the peaks where the experimental values are much lessthan the computed ones (coupling back into the disturbingline becomes important). The effect of varying cable-to-cableseparation in the high-frequency range can be seen by compar-ing the results in Figs. 4 and 5. The coupling values are seento decrease away from the peaks with increasing cable-to-cable separation, whereas the peak values for the couplingslightly increase, and reach a limiting value after a certainseparation (D > 2 cm, this has been deduced from thetheoretical calculations). This is because the attenuationof the intermediate circuit decreases with increasing separ-ation between the two lines. At resonant frequencies (150,3 0 0 , . . . MHz), this attenuation plays an important part in

352

determining the level of coupling (see eqn. 16) as the circu-lating current in the resonant tertiary is limited only bylosses (which include coupling into the disturbing and victimlines). At the same time 1/Z33 in eqn. 16 is decreasing withincreasing separation, which implies a decrease in the valuesof coupling factor.

In the second group of results the effect of the presenceof the metallic ground plane on the coupling between thetwo cables has been considered. Eqn. 21 shows that thecoupling component due to the presence of the metallicground plane opposes the component which is presentbetween the two cables in free space (see eqn. 16) so thatthe resulting coupling is less than the value computed ignoringthe presence of the gound. This reduction is due to the factthat part of the return current formerly flowing on the dis-turbed outer conductor now flows in the ground instead.

At low frequencies (J<\/4), the coupling factor givenby eqn. 21 can be reduced to F = Zfy/2ZO(1/Z33 - \\AZ^t

which is similar to the expression given in Reference 18.Comparing the practical and theoretical results in Figs. 6-9in the lower frequency range, the agreement is within 4 dB.A similar conclusion to that in Reference 18 for the frequencyrange can be drawn. To achieve low coupling the cables haveto be far apart and very close to the ground plane (this caseis illustrated in Fig. 9 for D = 2-6 cm and h = 2 cm).

At high frequencies, the problem is more complicatedthan in free space. At frequencies for which / = «X/2, thetwo intermediate circuits (the intermediate circuit formedbetween the two outer conductors of both cables and thatformed between the two outer conductors of the cablesand the ground plane) resonate, producing a sharp increasein the coupling factor, while at the same time the averagemagnitude of coupling, which is given by a smooth curveof magnitude (Z^/2ZOZ33)-(Z^/8ZOZ44), is less thanthat in free space (Zx/2ZOZ33). This can be seen by com-paring Figs. 6 and 7 with Fig. 4. In Fig. 9, the couplingaround the resonant frequencies can be seen to have beenreduced by a considerable amount compared to that in Fig. 7by increasing cable-to-cable separation from 1 cm to 6 cm,whereas at the resonant frequencies the values of couplingare not much affected by increasing the separation of thecables. The effect of doubling the length of the coupledlines (see Figs. 6 and 8) is an increase of the coupling factorby 6 dB at low frequencies, whereas at high frequenciesthe peaks have been shifted to occur at different frequenciescorresponding to the multiple of half wave lengths of theintermediate circuit (i.e. / = 75, 150, . . . MHz for / = 2 m,assuming air dielectric of the intermediate circuit).

5 Conclusion

From the above discussions it can be said that, while reducedcoupling is achieved to a considerable extent at low fre-quencies by placing the cables well apart and very close tothe ground plane, this configuration is not very favourablefor high-frequency performance because of the sharp increasein the coupling factor at the resonant frequencies. However,this condition of laying the cables well apart and near tothe ground so as to reduce the coupling can still be usefulat high frequencies, providing the frequency of operationavoids the resonant frequencies. Furthermore, since thecoupling factor is proportional to the square of the transferimpedance (which is a measure of the screen efficiency ofthe cable), by reducing the transfer impedance (i.e. reducingthe leakage), less coupling can be obtained.

6 Acknowledgements

The authors wish to thank M.M. Rahman, S. Sali and J.

IEEPROC, Vol. 128, Pt. A, No. 5, JULY 1981

Hellinas for helpful suggestions and advice, and J.L. Goldbergof British Insulated Callendars Cables for valuable discussionsand the supply of cable samples.

The authors are also grateful to the UK Ministry of Defencefor support of the work.

7 References

1 SCHELKINOFF, S.A.: The electromagnetic theory of coaxialtransmission lines and cylindrical shields', Bell Syst. Tech. J.,1934,13, pp. 532-578

2 TYNI, M.: 'The transfer impedance of coaxial cables with braidedouter conductors'. EMC symposium, Wroclaw, Poland, Sept. 1976

3 VANCE, E.F., and CHANG, H.: 'Shielding effectiveness of braided-wire shields'. Technical memorandum 16, Stanford ResearchInstitute, Menlo Park, California, Nov. 1971

4 VANCE, E.F.. Shielding effectiveness of braided-wire shields',IEEE Trans., 1975, EMC17, pp. 71 -77

5 PERKOWSKI, Z.: 'Optimization of construction of multiwirescreens for high frequency coaxial cables - Part I, Principles ofthe theory', Rozpr. Electrotech., 1973, 19, pp. 517-560

6 BS 2316: 'Radio frequency cables'7 RAHMAN, M.M., SITCH, J., and BENSON, F.A.: 'Leakage from

coaxial cables', IEEProc. A, 1980, 127, (2), pp. 74-80

8 DITSHEID, H.L.: Theorie und Messung des Kopplungswider-standes'. NTG conference, Berlin, Oct. 1966

9 SCHELKUNOFF, S.A., and ODARENKO, T.M.: 'Crosstalk betweencoaxial transmission lines', Bell Syst. Tech. J, 1937, 26, pp. 144-164

10 BOOTH, R.P., and ODARENKO, T.M.: 'Crosstalk between coaxialconductors in cable', ibid., 1940, 19, pp. 358-384

11 GOULD, K.E.: 'Crosstalk in coaxial cables - analysis based onshort-circuited and open tertiaries', ibid., 1940, 19, pp. 341 —357

12 MOHR, R.J.: 'Coupling between open and shielded wire linesover a ground plane', IEEE Trans., 1967, EMC-9, pp. 34-45

13 MOHR, R.J.: 'Coupling between lines at high frequencies', ibid.,1967, EMC-9, pp. 127-129

14 BLACKBAND, W.T.: 'A theory of electrical conduction in wirebraids'. Technical Report 72153, RAE Farnborough, Oct. 1972

15 BLACKBAND, W.T.: 'Losses in tape braid conductors'. TechnicalReport 72030, RAE Farnborough, March 1972

16 SHENFELD, S.: 'Coupling impedance of cylindrical tubes', IEEETrans., 1972, EMC-14, pp. 10-16

17 HOMANN, E.: 'Geshirmte Kabel mit Optimalen Getlechtschirmen',Nachrichtentech. Z., 1968, 21, pp. 155-161

18 BADR, A.H., BENSON, F.A., and SITCH, J.E.: 'Coupling betweencoaxial cables over a ground plane at low frequency', IEE Proc. A,1980,127, (8), pp. 549-552

IEE North Western Centre Supply Section:Chairman's AddressD.F. Binns, M.Sc, Ph.D., C.Eng., M.I.E.E.

EVOLUTION OF POWER SYSTEM INSULATION

Indexing terms: Insulators and insulation, Power systems and plant

Power system insulation has evolved to take advantage of newmaterials that have become available and to satisfy thechanging needs of the electricity supply industry. From theopening of the Deptford Power Station in 1889 through thesetting up of the UK National Grid system in the 1920s andthe nationalisation of the UK electricity supply industry in1947, we have now reached a crossroads in 1980. Do wefollow the nuclear option, with extra EHV transmissioncapacity, or the CHP alternative with additions only to low-voltage networks?

Whatever happens in the UK, many exciting developmentsare taking place worldwide. Important and relatively newinsulants include polyethylene, SF6, cast resin and siliconefluid. Insulants may be better used together, as in oil-impregnated tape for cables and transformers. However,insulation on the microscopic scale is never made up of onestate of matter alone. Solids, such as polyethylene, containgaseous voids and water as well as particulate contamination.Liquids contain particles and also gas which is normallydissolved but may appear as bubbles; gases contain particles.Breakdown mechanisms for all states of matter have muchin common, indeed breakdown in solids and liquids maybe triggered through inclusions of gas and water. Breakdownprocesses are least well understood in liquids and relativelywell understood in gases. However, over a life of 30 yearsor more, insulation deteriorates and design stresses atworking voltage are low compared with ideal breakdownvalues. This is a function also of transient overvoltages.

Progress in insulation is a measure of the success achievedby research and development, but we have also learned fromour occasional failures. The 26th January 1963 holds memoriesfor UK linesmen and system operations personnel whostruggled to cope with nationwide insulator flashovers infreezing fog. The experience led to washing and greasing ofinsulators in areas of heavy pollution and to the development

IEE PROC, Vol. 128, Pt. A, No. 5, JULY 1981

of better insulator shed designs; it also added to the con-viction that clean air is a necessity. Similarly, the failure oftwo directly buried 132kV cables in July 1962 due tosoil drying out led to a big advance in knowledge of soilthermal behaviour. Selected backfill came to be used, andultimately forced cooling of 400 kV cables to match in onecircuit the power handling capacity of overhead lines. SF6

circuit breakers and metalclad busbar installations have beena great success worldwide. The three-phase busbar in SF6

at 145 kV is standard and 345 kV three-phase busbar systemsare being developed. Of course SF6 may not be the last wordin gases, both for arc quenching and insulation, and this addsan extra impetus to the development of metalclad substations.

Changes in transformer insulation are perhaps overdue. Theneed for less flammable materials has brought in silicone fluidfor small transformers in special applications, e.g. in rail cars,and more recently alternatives such as MIDEL have appeared;these have taken the place of toxic PCBs. The possibilities forSF6 -insulated transformers is being looked at as it could givethe dual benefits of elimination of fire hazard and greatercompatibility with SF6 metalclad substations.

In all work on insulation design, the statistical nature ofthe phenomena must be appreciated. Measurements oninsulation samples, having small stressed electrode areas andinsulation volumes, have to be related to the combined chanceof failure in many large-scale equipments. The law of thedistribution of breakdown voltages or discharge magnitudesshould be known so that extrapolation to the power systemscale can take place. For this the three-component Weibulldistribution is widely taken to describe insulation performancemost adequately.

Abstract 1348A of address delivered at the University of ManchesterInstitute of Science & Technology, 28th October 1980Dr. Binns is with the Department of Electrical Engineering, Universityof Salford, Salford MS 4WT, England

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