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VC 2013 Wiley Periodicals, Inc.

LOW-POWER PHOTONIC CONTROL OF AMICROWAVE RING RESONATOR USINGBULK ILLUMINATION

Mohammad Ali Shirazi-Hosseinidokhtand Mani Hossein-ZadehCenter for High Technology Materials, 1313 Goddard, SE,Albuquerque, NM 87106; Corresponding author:mhz@chtm.unm.edu

Received 22 October 2012

ABSTRACT: We demonstrate the feasibility of bulk illumination

technique for low-power photonic control of RF resonance. Using thistechnique, the transmitted RF power through a microstripline-ring filter

on a junction-less silicon substrate is changed by 11 dB with less than 2mW of interacting optical power. VC 2013 Wiley Periodicals, Inc.

Microwave Opt Technol Lett 55:1594–1599, 2013; View this article

online at wileyonlinelibrary.com. DOI 10.1002/mop.27606

Key words: light-controlled RF devices; RF ring resonator;RF-photonics

1. INTRODUCTION

Photonic control of RF signal propagation is an important topic

that continues to be an active area of research and development

in microwave photonics [1–14]. Photonic control has many

advantages over conventional electrical control. High degree of

electrical isolation between the control signal and the microwave

circuit, immunity to parasitic electromagnetic radiation, high

power handling, overall weight reduction, high-speed control,

and timing precision are among the most important benefits of

photonic control. In particular, electrical isolation between the

control signal and the microwave structure is crucial for design

and fabrication of reconfigurable antennas [15–18] where

the radiation pattern and efficiency are affected by the

presence of control devices and circuits in the vicinity of

antenna pattern [15].

A large variety of techniques, devices, and materials have

been explored for designing photonically controlled switches,

phase shifters, and attenuators. In almost all these approaches,

photonic carrier generation in a semiconductor controls the am-

plitude and the phase of the RF signal propagating on microstrip

or coplanar transmission lines. Free carrier generation in biased

and unbiased junctions as well as junction-less regions have

been used to control the RF field in discontinuities [8, 11], stubs

[9, 13], resonators [6], and terminations [4, 5, 9]. Except few

cases where the photosensitive element is added to a transmis-

sion line fabricated on a low-loss RF substrate [1, 15, 18], in

most proposed structures the RF circuit is fabricated on the pho-

tosensitive semiconductor substrate in order to reduce the com-

plexity of the fabrication process and keep the device mono-

lithic. Although compound semiconductors have also been used

as the structural materials in these devices [3, 10], implementa-

tion of photonically controlled RF devices on silicon substrates

is more attractive for monolithic integration of microwave and

mm-wave devices using well-developed fabrication processes.

Independent of the material and the device structure, in all

these approaches and devices a laser wavelength between 600

and 900 nm is used to maximize optical absorption and carrier

generation. As a result, the optically affected region has been

confined at the surface (due to small optical penetration depth at

these wavelengths). In contrast to the previous work, here we

explore the potential application of bulk illumination at a longer

wavelength (1064 nm) combined with high-Q RF resonance to

maximize the RF-optical field overlap. Moreover, we use a side-

coupled RF ring resonator to confine the RF field and enhance

the interaction of RF field and photogenerated carriers. Note

that although previously certain planar resonant structures were

used for photonic RF control, the confinement of free carrier on

the surface has limited the interaction of the resonant field and

the free carriers. In these cases, the presence of free carriers has

been mainly modifying the electrical properties at the bounda-

ries of the resonator (effectively tailoring the conductor size).

This article describes a photonically controlled RF ring reso-

nator suitable for controlling a variety of silicon-based micro-

wave integrated circuits. By optically controlling the density of

free carriers in the regions of the substrate with large resonant

RF field, we have demonstrated up to 8 dB of transmission loss

variation with only 1.9 mW of interacting optical power from a

commercial fiber pigtailed laser diode operating at 1064 nm. To

our knowledge, this is the largest optical sensitivity of RF trans-

mission ever reported for a passive junction-less photonically

controlled microwave device.

2. PHOTONICALLY CONTROLLED RF RING RESONATOR

2.1. Bulk Versus Surface IlluminationAs photonic free carrier generation controls the RF propagation

in most optically controlled components, strong optical absorp-

tion is the main criteria for choosing the photoconductive mate-

rial and the corresponding wavelength. On the other hand to

reduce the fabrication cost and complexity, preferably only one

type of material is used in the device structure. As a result, low-

loss optical waveguides cannot be easily integrated with the de-

vice to deliver light directly to the sensitive region, and almost

all devices are controlled by top illumination to avoid absorption

before reaching the sensitive region. Silicon is one of the most

common substrates used in phonically controlled RF devices

(mainly because of compatibility with IC fabrication and low

fabrication cost). Figure 1(a) shows the absorption depth (d, the

depth at which the light intensity drops to 36% of its value at

the interface) plotted against wavelength for silicon substrate.

Below 900 lm, d is less than 30 lm, and all the photogenerated

carriers are effectively confined at the surface (interface between

air and silicon). That is why so far the optical control has been

mainly achieved by tailoring the conductive structure on top of

the semiconductor substrate using laser beam.

Here, we examine an alternative approach by choosing a

wavelength with an absorption depth larger than the substrate

1594 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 7, July 2013 DOI 10.1002/mop

thickness to enable bulk illumination (photogeneration) and,

therefore, enhanced interaction between the RF field and the

photogenerated carriers. In addition, we use an RF ring resona-

tor that confines and passively amplifies the RF-field in a rela-

tively small region. Bulk illumination of the volume with maxi-

mum electric field intensity results in a large RF-optical (and,

therefore, RF-free carrier) overlap integral. In our experiment,

we use the 1064 nm wavelength with an absorption depth of

about 1 mm that is two times the thickness of the substrate. So

the photocarriers are generated almost uniformly distributed

across the wafer.

2.2. Experimental SetupThe structure consists of RF ring resonator side coupled to a

microstripline. RF ring resonator has been extensively investi-

gated in the context of planar antennas and frequency-selective

surfaces [19–21]. The ring resonator and the transmission line

are fabricated on a 500-lm silicon substrate (100 orientation)

with a resistivity of about 2000 X cm. A thin layer (�2 lm) of

copper was coated on both sides of the silicon wafer using RF

sputtering (50 nm of chromium was used between copper and

silicon to improve the attachment). Next, we patterned the top

copper layer using photolithography and wet etching.

Figure 1(b) shows the ring resonator side coupled to the

microstripline. The ring resonator has a diameter of 5.3 mm and

a width of 0.43 mm. The microstripline is a 50-X line (width

� 0.43 mm), and two SMA launchers were used to couple RF

power into and out of the microstripline [Fig. 1(b)]. When a

ring resonator is side coupled to microstripline, the degeneracy

between frequencies of the even and odd resonant modes will

be removed because of the asymmetric coupling [19, 20]. As a

result, two dips appear near each resonance in the transmission

spectrum of the microstripline. We have used a finite element

microwave modeling software (CST) to calculate the frequencies

and the field distribution for the first two modes of the ring reso-

nator. Figure 2 shows the simulated transmission spectrum (S21)

as well as the electric field distribution for the fundamental and

the second-harmonic mode of the ring resonator. Below 15

GHz, four modes are excited: the odd fundamental mode (M1-

o), even fundamental mode (M1-e), odd second-harmonic mode

(M2-o), and even second-harmonic mode (M2-e).

Using the simulation results, we have identified three differ-

ent locations corresponding to the minimum field-strength for

each one of the RF modes for optical illumination [see Fig.

3(a)]. A small circular aperture on the copper ring is used to

expose the substrate to laser at each one of these locations.

The diameter of aperture is 0.2 mm (about 50% of the ring

width) to minimize its effect on the RF resonance [Fig. 3(b)].

We fabricated three samples with identical rings and coupling

gaps but with different aperture positions (so each ring has

only one aperture). As shown in Figure 3(c), the 1064-nm laser

light from a fiber pigtailed laser diode is fed to a fiber pigtailed

collimator and is vertically coupled to the substrate through the

aperture.

Figure 1 (a) Absorption depth of the silicon plotted against wavelength. (b) A photograph of the ring coupled to a microstripline on a silicon sub-

strate. [Color figure can be viewed in the online issue, which is available at wileyonlinelibrary.com]

Figure 2 Simulated transmission spectrum of the microstripline coupled to the ring resonator. (a) Near the fundamental resonance. (b) Near the sec-

ond-harmonic resonance. The arrows show the even (fe) and odd (fo) modes for each resonance. The insets show the electric field magnitude on a plane

located in the middle of the silicon substrate (250-lm depth). [Color figure can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 7, July 2013 1595

Figure 4 (a) The laser is illuminating Location 3 (the minimum field intensity for even second-harmonic mode). (b) The laser is illuminating Position 1

(the minimum field intensity for even fundamental mode). (c) The laser is illuminating Position 2 (the minimum field intensity for odd fundamental mode)

Figure 3 (a) The location of the aperture on three different ring resonators. (b) Close-up photograph of a ring with an aperture at Position 3. (c) Sche-

matic diagram of the collimator vertically coupling laser light to the silicon substrate. [Color figure can be viewed in the online issue, which is available

at wileyonlinelibrary.com]

1596 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 7, July 2013 DOI 10.1002/mop

2.3. Measurement and ResultsFigure 4 shows the S21 measurement results near fundamental

and second-harmonic resonance in the absence of the laser illu-

mination and with three different levels of incident optical

power (0.43, 1.23, and 2.71 mW). The incident optical powers

are estimated based on the optical output power from the colli-

mator and the aperture area. The resonant frequencies of the odd

and even fundamental modes as well as the even second-har-

monic modes are in good agreement with valued from the S21

measurement. According to the simulation, the M2-o is weakly

coupled to the microstripline (S21 � �4 dB) and that is why it

is hidden in the parasitic background transmission in the meas-

ured spectrum.

In Figure 4(a), the laser is illuminating Location 3 where

the E-field intensity for the M2-e is minimum (according to

simulation). As this location is near the maximum of the M1-

e, this mode is suppressed by photocarrier generated loss. The

resonant dip of the M1-o and the M2-e are also reduced; this

is due to three main reasons: the resonant electric field of

these modes is not completely zero near in the illumination

region, the real location of the minimum field intensity might

be slightly different from the simulation, and some carriers

will diffuse to high-field region. In Figure 4(b), the laser is

illuminating Location 1 where electric field intensity for M1-e

is minimum. As expected, the M1-e is minimally affected,

whereas M1-o is effectively disappeared and M2-e is severely

attenuated (Position 1 has a significant overlap with the field

intensity of M2-e). In Figure 4(c), the laser is illuminating

Position 2 where the maximum of M2-e is located. In this

case, we observe the maximum attenuation for M2-e mode.

These results show that using bulk illumination one can selec-

tively control the loss factor of different modes. Due to the

absence of the M2-o, near 14 GHz the illumination only con-

trols the magnitude of the even second-harmonic with mini-

mal frequency shift. So effectively near the second-harmonic

resonance, the device functions as a single frequency optical

RF switch. Figure 5 shows the RF quality factor (unloaded),

frequency and the transmitted RF power (S21) at fRF ¼ f2,e

� 13,890 MHz (resonant frequency of M2-e) plotted against

the interacting optical power. Note that at a single wavelength

(1064 nm), optical reflection from the silicon–air interface

can be easily canceled by depositing two layers of dielectric

on top of the silicon with almost no effect on the RF proper-

ties. Therefore, to estimate the ultimate performance of the

device, here we have considered the optical power inside the

silicon (or the interacting optical power) instead of the inci-

dent optical power. The unloaded RF quality factor in each

case has been estimated using the S21 spectrum based on the

3-dB linewidth measured from the bottom of each transmis-

sion dip [22].

As expected, the quality factor of the modes degrades due to

loss generated by free carriers. The frequency does not change

when aperture numbers two and three are illuminated, and it

changes only by 1% when aperture number one is illuminated.

Note that in almost all resonant cases studied previously [4–6],

the optically induced attenuation is accompanied with significant

frequency shift. Here, the frequency change is minimal, because

the photocarriers do not increase the RF-length as opposed to

previous resonant structures where the photogenerated carriers at

the surface change the conducting surface (therefore the resona-

tor size) and the resonant frequency. Decoupling the frequency

shift and attenuation is very important, because in narrow-band

Figure 5 Measured values of RF quality factor (a), resonant frequency (b), and S21 at resonance (c) plotted against optical power in the substrate

(interacting optical power). These parameters are only measured for M2-e (f2,e � 13,890 MHz) and V1–V3 correspond to illumination at Positions 1–3.

[Color figure can be viewed in the online issue, which is available at wileyonlinelibrary.com]

TABLE 1 Comparison Between the Performance of the Proposed Devices and Previously Shown Optically Controlled RF Devices

Reference Wavelength (nm) dB/(W/mm2) dB/mW Geometry Frequency (GHz) Material

This work 1064 173 5.5 Ring 13.7 Silicon

[4] 513 1.9 X Stub 6 Silicon

[5] 513 19 X Stub 4 Silicon

[10] X X 4.1 OVC 14 III–V

[8] 870 X 1 Interdigital 6 Silicon

[11] 820 X 0.8 CPW-gap 6 Silicon

[12] Xe arc lamp 85.7 X Waveguide 50 Silicon

[13] 850 X 0.35 Stub 7 Silicon

[18] 808 X 0.09 Si-gap 8.4 Silicon

[9] <1000 X 1.15 Stub 6.9 Silicon

[14] 808 X 0.08 CPW-gap 6.9 Silicon

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 7, July 2013 1597

resonant systems (such as filters) attenuation at certain frequen-

cies is desired.

Table 1 summarizes the some of the results from previous

works on photonically controlled RF circuits. Independent of the

material and the specific design, all devices use optical illumina-

tion to control the RF field through photocarrier generation.

Clearly in terms of maximum RF transmission change per 1

mW of optical power, our device outperforms the previous devi-

ces. All these devices function based on optical surface effects

(k < 900 lm) and some of them use complex structures, junc-

tions (biased and unbiased), and compound semiconductor mate-

rial. The device presented in this work is completely made on a

uniform silicon substrate and does not use any bias voltage or

p–n junction, yet it has the largest optical sensitivity. The

enhanced sensitivity is a result of large overlap between the

photocarrier density and the oscillating RF field in the silicon

substrate.

3. DISCUSSION

The device presented here has many applications in optical

switching of microwave power and the design of optically

reconfigurable RF circuits and antennas. The same structure

can also function at higher frequencies by reducing the ring

resonator diameter or using higher harmonics of the same

ring. We expect better optical sensitivity in smaller rings due

to the larger ratio between the optically attenuated RF field

and the total resonant RF filed. Moreover, by coupling more

rings, several frequencies can be switched simultaneously

resulting in more flexible and versatile RF transmission spec-

trum. Due to low power consumption, one laser can feed sev-

eral resonators using low-cost power splitters and collimators.

Figure 6(a) shows three identical coupled ring resonators com-

prising a three-pole RF band-stop filter (three stop bands

shown in the figure). As shown (gray trace), laser illumination

of all three rings can eliminate the second stopband without

significantly changing the other stopbands. By properly illumi-

nating different locations on each ring, one can tailor the

transmission spectrum of the filter without any physical con-

tact or using any electronic/mechanical element. Figure 6(b)

shows a design where ring resonators with different diameters

are side coupled to a single transmission line. In this case, a

series of optically controlled interleaved stopbands (with dif-

ferent mode spacing) can provide more control over the

desired region (dashed box).

In this work, the ring resonator is side coupled to the

microstripline resulting in bad-stop operation. These configu-

rations and methods can be extended to photonically con-

trolled band-pass filters using capacitive gap coupled (as

opposed to side coupled) ring resonators. Optically controlled

band-pass RF filters with relatively low insertion loss can be

designed by implementing optimized capacitive coupling

methods between rings [23]. Although for the sake of simplic-

ity in this proof-of-concept demonstration, we used a colli-

mated beam for illuminating the substrate, the illumination ef-

ficiency can be improved using a diverging beam that spreads

out right after passing through the aperture resulting in a

larger spatial distribution of the photogenerated carriers.

Moreover, the size of the aperture and its location can still be

optimized.

4. CONCLUSION

We have shown bulk illumination combined with high-Q RF

resonance can significantly reduce the power consumption in

optically controlled RF attenuators. Using side-coupled RF

ring resonator on a silicon substrate and choosing a laser

wavelength that generates free carriers across the substrate (as

opposed to substrate surface), we have been able to show a

record low optical power control of RF transmission near 14

GHz. Our device is fabricated simply by copper deposition on

a high-resistibility silicon substrate, and no junction or com-

pound semiconductor has been used in the structure of our de-

vice. The device design and employed wavelength were lim-

ited by available lasers. Using the same methodology and

only by optimizing the wavelength, substrate thickness and

the RF coupling (between the ring and the transmission line)

higher efficiencies can be obtained. The band-stop frequencies

of this device can be easily tailored by changing the ring

radius.

ACKNOWLEDGMENT

This work was supported in part by Air Force Office of Research

(Grant: FA9550-12-1-0049).

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VC 2013 Wiley Periodicals, Inc.

PERFORMANCE ENHANCEMENT OF ACIRCULARLY POLARIZED PATCHANTENNA FOR RADIO FREQUENCYIDENTIFICATION READERS USING ANELECTROMAGNETIC BAND-GAPGROUND PLANE

Sylvain Pflaum, Philippe Le Thuc, Georges Kossiavas, andRobert StarajLEAT-Universit�e de Nice-Sophia Antipolis-CNRS Batiment Forum,Campus Sophi@Tech, 930 Route des Colles, 06903 SophiaAntipolis, France; Corresponding author: sylvain.pflaum@unice.fr

Received 29 October 2012

ABSTRACT: A miniaturized circularly polarized (CP) patch antennawith wide impedance matching and axial ratio bandwidths and a good

efficiency using an electromagnetic band gap (EBG) ground plane ispresented. CP is obtained with the truncated corners technique applied

on a square patch and size reduction of this radiating element is basedon the slow wave phenomenon created in the EBG structure. Thisantenna is fabricated on a 117 � 117 � 4 mm3 FR-4 substrate (eT ¼4.4 and tan d ¼ 0.22) and dedicated to the European UHF radiofrequency identification band (865–868 MHz). The measured bandwidth,

realized gain, and total efficiency are 14.4%, 3.7 dBi, and 79%,respectively. Experimental and theorical results are presented anddiscussed. VC 2013 Wiley Periodicals, Inc. Microwave Opt Technol Lett

55:1599–1602, 2013; View this article online at wileyonlinelibrary.com.

DOI 10.1002/mop.27642

Key words: microstrip circularly polarized patch antenna; radio

frequency identification; electromagnetic band-gap ground plane

1. INTRODUCTION

Nowadays, UHF radio frequency identification (RFID) technol-

ogy is widespread in many industrial applications such as mer-

chant flow tracking, stock control, and supply chain management

[1]. For the reader part of these RFID systems, circularly polar-

ized (CP) antennas are usually used in order to be able to

receive RF signals that can be emitted from arbitrarily oriented

tag antennas. Microstrip patch antennas are suitable in such

applications, because they are compact, low profile, low cost,

and easy to manufacture. The circular polarization on a single

feed square microstrip antenna can be obtained by cutting two

corners off [2]. However, in many cases, matching and axial ra-

tio bandwidths are narrow band (around 5 and 1%, respectively),

and the antenna size is considered too large (k/2 � k/2). Several

methods have been studied to reduce the microstrip antenna size

such as the use of short circuits as in PIFA antennas, fractal

shapes, or high-permittivity dielectric substrates [3]. However,

using these previous techniques, it is not easy to simultaneously

combine, good matching, circular polarization in the broadside

direction, wide bandwidth [4, 5], and moreover the last tech-

nique commonly involves an increase of production cost [6]. To

overcome this last drawback, some authors propose to use meta-

materials [7] to increase the relative permittivity by etching slots

in the ground plane without any important additional production

cost [8]. In this article, an electromagnetic band gap (EBG)

ground plane (GP) is used to both enhance the performance and

reduce the dimensions of the microstrip antenna.

2. ANTENNA GEOMETRY

Figure 1 shows the truncated corners square microstrip antenna

printed on a FR-4 substrate (thickness of 4 mm, dielectric

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 7, July 2013 1599