1

CHAPTER 1

Introduction

1.1 Background Vehicles are becoming mobile electronic communication systems, part of a wider telematics

network with applications at microwave and millimeter wave frequencies. Many low frequency

antennas below 1 GHz are printed on glass screens in the motor industry to reduce costs, hide the

antennas and protect them from vandalism. Microstrip patches are widely used as cheap,

conformal antennas for a wide variety of higher frequency applications and so there is currently

much interest in printing such antennas on, or within, the glass areas of vehicles for intelligent

transport and telematics systems. References [1], [2] reported on the performance of patch

antennas fixed directly to glass which formed a superstrate. Mounting antennas within the glass

offers the prospect of reducing costs but presents production problems such as thermal distortion

of the glass during processing and feeding the signal to the embedded antenna. In addition it is

not possible to print a solid conductor area on glass if it exceeds a few millimeters across as the

metal area reflects heat and distorts the glass during the shaping/lamination process. In that case

the metal must be meshed.It is not the intention for this seminar to look solely at glass based

applications but to present some interesting general properties of meshed antennas that are not

reported in the literature. Discussion on square antennas where the patch and ground plane were

meshed in various combinations and their effects of the varying line widths and line density on

gain, cross-polarization, resonant frequency and bandwidth. The antennas were fed using coaxial

probes but coplanar feeds can also be used.

1.2 Relevance The topic is related to Antenna designing which is also a part of our curricular. Here, traditional

patch antennas are described, later on which are meshed to form a antenna which is transparent

The transparency of the antenna can be varied but it adds to a significant change in the

parameters of the antenna.

2

1.3 Literature Survey 1.3.1 Dipole Antenna A dipole antenna is a radio antenna that can be made of a simple wire, with a center-fed driven

element. It consists of two metal conductors of rod or wire, oriented parallel and collinear with

each other (in line with each other), with a small space between them. The radio frequency

voltage is applied to the antenna at the center, between the two conductors. These antennas are

the simplest practical antennas from a theoretical point of view. They are used alone as antennas,

notably in traditional "rabbit ears" television antennas, and as the driven element in many other

types of antennas, such as the Yagi. Dipole antennas were invented by German physicist

Heinrich Hertz around 1886 in his pioneering experiments with radio waves. Dipoles have an

radiation pattern, shaped like a toroid (doughnut) symmetrical about the axis of the dipole. The

radiation is maximum at right angles to the dipole, dropping off to zero on the antenna's axis.

The theoretical maximum gain of a Hertzian dipole is 10 log 1.5 or 1.76 dBi. The maximum

theoretical gain of a λ/2-dipole is 10 log 1.64 or 2.15 dBi.

Fig: 1.1 Electric field and magnetic field radiated by the Dipole.

3

Fig:1.2 Radiation Pattern of a Diploe Antenna

1.3.2 Fractal Antenna Benoit Mandelbrot, the pioneer of classifying this geometry, first coined the term ‘fractal’ in

1975 from the Latin word derived from the Latin fractus meaning broken, uneven. Any of

various extremely irregular curves or shape that repeat themselves at any scale on which they are

examined. A fractal antenna is an antenna that uses a fractal, self-similar design to maximize the

length, or increase the perimeter (on inside sections or the outer structure), of material that can

receive or transmit electromagnetic radiation within a given total surface area or volume.

Such fractal antennas are also referred to as multilevel and space filling curves, but the key

aspect lies in their repetition over two or more scale sizes, or "iterations". For this reason, fractal

antennas are very compact, multiband or wideband, and have useful applications in cellular

telephone and microwave communications. A good example of a fractal antenna as a space

filling curve is in the form of a shrunken fractal helix. Here, each line of copper is just a small

fraction of a wavelength. A fractal antenna's response differs markedly from traditional antenna

designs, in that it is capable of operating with good-to-excellent performance at many different

frequencies simultaneously. Normally standard antennas have to be "cut" for the frequency for

which they are to be used—and thus the standard antennas only work well at that frequency. This

makes the fractal antenna an excellent design for wideband and multiband applications. For

example Mandelbrot discusses the basics of fractal theory as applied to the characteristics of a

coastline. The length of a coastline depends on the size of the measuring yardstick. As the

yardstick we use to measure every turn and detail decreases in length, the coastline perimeter

increases exponentially. As the view of a coastline is brought closer, we discover that within the

coastline there lie miniature bays and peninsulas. As we examine the coastline on a rescaled

4

map, we discover that each of the bays and peninsulas contain sub-bays and sub-peninsulas.

There is a self-similar trait observed as we look at the coastline at various resolutions. The

number of microscopic structures begins to approach infinity. In fact, because of the large

number of irregularities, the physical length of a coastline is virtually infinite.

Fig: 1.3 Coast Line

Fig.1.3 represents an imaginary coastline. The grey lines are rulers being used to measure

the length of the coastline (L). These rulers are of the length S. Using the first ruler we see that it

L = 2 * S. When we decrease the length of S the number of times that S is used increases. What

these rulers illustrate is that as the size of the measuring device becomes smaller the accuracy of

the measurements becomes more and more accurate. From this fact we can assume that

eventually we will be able to get an exact measurement of the coastline. This statement is false.

As we decrease the size of the measuring device the length that we have to measure becomes

greater. We can see this by zooming in on the coastline. As we get closer and closer we will

notice that it looks very similar to how it looked from a greater distance away. Only now we are

much closer. This observation shows the self-similarity of the coastline. Therefore as we

decrease the size of the measuring device the length of the coastline will increase without limit,

thus showing us its fractal nature.

1.4 Motivation With the growing number of telematics systems used in cars there is an increased need for the

integration of antennas into the structure. Microstrip patch antennas which are lightweight and

low in cost can be integrated into different parts of the car body. Integrating the patch antenna

into the windscreen poses practical problems:

It is not possible to screen print large solid metal areas (e.g. a patch at 1.5 GHz) because this

would distort the windscreen during the heating process by which it is formed.

5

This problem is overcome by replacing the solid metal areas of the patch by a mesh structure

which gives the additional advantage that the antenna gains a degree of transparency. Meshing

antenna although increases its transparency; it also changes its resonant frequency, Gain and

other parameters severely. Hence there is need to briefly discuss the effects meshing on the

above parameters.

1.5 Scope The Meshed antenna designed can be used in vehicular applications. As the Antenna is

transparent it can be used for some of the more complex applications, which includes embedding

these antennas on the solar panel.

1.6 Organization of Report Chapter 2 starts with the theory of Microstrip Patch antennas, it illustrates what are the

microstrip antennas how do they radiate, the different shapes available for microstrip patch

antennas. Chapter 3 explains different feeding methods for microstrip antennas along with

appropriate illustrations along with their equivalent circuit diagrams. Chapter 4 explains the

designing of microstrip antennas later on the designed antenna will be meshed to form Mesh

Patch Antenna, also we will discuss the effects on Gain, resonant frequency, by changing

meshing parameters such as line spacing, line width etc.

6

CHAPTER 2

Microstrip Antenna

2.1 Introduction Microstrip antennas received considerable attention starting in the 1970s, although

the idea of a microstrip antenna can be traced to 1953 and a patent in 1955. Microstrip antennas,

as shown in Figure 2.1 consist of a very thin (t << λ0, where λ0 is the free-space wavelength)

metallic strip (patch) placed a small fraction of a wavelength (h << λ0, usually 0.003 λ0 ≤ h ≤

0.05 λ0) above a ground plane. The microstrip patch is designed so its pattern maximum is

normal to the patch (broadside radiator). This is accomplished by properly choosing the mode

(field configuration) of excitation beneath the patch. End-fire radiation can also be accomplished

by judicious mode selection. For a rectangular patch, the length L of the element is usually

λ0/3 < L < λ0/2. The strip (patch) and the ground plane are separated by a dielectric sheet

(referred to as the substrate), as shown in Figure 2.1. There are numerous substrates that can be

used for the design of microstrip antennas, and their dielectric constants are usually in the range

of 2.2 ≤ εr ≤ 12. The ones that are most desirable for good antenna performance are thick

substrates whose dielectric constant is in the lower end of the range because they provide better

efficiency, larger bandwidth, loosely bound fields for radiation into space, but at the expense of

larger element size. Thin substrates with higher dielectric constants are desirable for microwave

circuitry because they require tightly bound fields to minimize undesired radiation and coupling,

and lead to smaller element sizes; however, because of their greater losses, they are less efficient

and have relatively smaller bandwidths. Since microstrip antennas are often integrated with other

microwave circuitry, a compromise has to be reached between good antenna performance and

circuit design. Often microstrip antennas are also referred to as patch antennas. The radiating

elements and the feed lines are usually photo-etched on the dielectric substrate. The radiating

patch may be square, rectangular, thin strip (dipole), circular, elliptical, triangular, or any other

configuration.

7

a) Microstrip Antenna

b) Side View of Microstrip Antenna

Fig: 2.1. Microstrip Antenna and its Side view

These and others are illustrated in Figure 2.2. Square, rectangular, dipole (strip), and circular are

the most common because of ease of analysis and fabrication, and their attractive radiation

characteristics, especially low cross-polarization radiation. Microstrip dipoles are attractive

because they inherently possess a large bandwidth and occupy less space, which makes them

attractive for arrays. Linear and circular polarizations can be achieved with either single elements

or arrays of microstrip antennas. Arrays of microstrip elements, with single or multiple feeds,

may also be used to introduce scanning capabilities and achieve greater directivities.

8

Fig: 2.2. Shapes of Microstrip patch Elements

9

CHAPTER 3

Feeding Methods There are many configurations that can be used to feed microstrip antennas. The four most

popular feeding methods are the microstrip line, coaxial probe, aperture coupling, and proximity

coupling. These are displayed in Figure 3.1. One set of equivalent circuits for each one of these

is shown in Figure 3.2. The microstrip feed line is also a conducting strip, usually of much

smaller width compared to the patch. The microstrip-line feed is easy to fabricate, simple to

match by controlling the inset position and rather simple to model. However as the substrate

thickness increases, surface waves and spurious feed radiation increase, which for practical

designs limit the bandwidth (typically 2–5%). Coaxial-line feeds, where the inner conductor of

the coax is attached to the radiation patch while the outer conductor is connected to the ground

plane, are also widely used. The coaxial probe feed is also easy to fabricate and match, and it has

low spurious radiation. However, it also has narrow bandwidth and it is more difficult to model,

especially for thick substrates (h > 0.02 λ0). Both the microstrip feed line and the probe possess

inherent asymmetries which generate higher order modes which produce cross-polarized

radiation. To overcome some of these problems, non-contacting aperture-coupling feeds, as

shown in Figures 3.1 (c ,d) have been introduced.

(a) Microstrip Line feed (b) Probe Feed

10

(c) Aperture Coupled Feed

(d) Proximity Feed Coupled

Fig: 3.1 Typical Feeds for microstrip Antenna

The aperture coupling of Figure 14.3(c) is the most difficult of all four to fabricate and it also has

narrow bandwidth. However, it is somewhat easier to model and has moderate spurious

radiation. The aperture coupling consists of two substrates separated by a ground plane. On the

bottom side of the lower substrate there is a microstrip feed line whose energy is coupled to the

patch through a slot on the ground plane separating the two substrates. This arrangement allows

11

(a) Microstrip Line (b) Probe

(c) Aperture-coupled (d) Proximity Coupled

Fig: 3.2 Equivalent Circuits for Typical Feeds

independent optimization of the feed mechanism and the radiating element. Typically a high

dielectric material is used for the bottom substrate, and thick low dielectric constant material for

the top substrate. The ground plane between the substrates also isolates the feed from the

radiating element and minimizes interference of spurious radiation for pattern formation and

polarization purity. For this design, the substrate electrical parameters, feed line width, and slot

size and position can be used to optimize the design. Typically matching is performed by

controlling the width of the feed line and the length of the slot. The coupling through the slot can

be modelled using the theory of Bethe, which is also used to account for coupling through a

small aperture in a conducting plane. Of the four feeds described here, the proximity coupling

has the largest bandwidth (as high as 13 percent), is somewhat easy to model and has low

spurious radiation. However its fabrication is somewhat more difficult. The length of the feeding

stub and the width-to-line ratio of the patch can be used to control the match .

12

CHAPTER 4 Transmission line Model

The rectangular patch is by far the most widely used configuration. It is very easy to analyze

using both the transmission-line and cavity models, which are most accurate for thin substrates.

The transmission-line model is the easiest of all but it yields the least accurate results and it lacks

the versatility. However, it does shed some physical insight. the cavity model, a rectangular

microstrip antenna can be represented as an array of two radiating narrow apertures (slots), each

of width W and height h, separated by a distance L. Basically the transmission-line model

represents the microstrip antenna by two slots, separated by a low-impedance Zc transmission

line of length L.

(a) Microstrip Line

(b) Electric field Lines

13

(c) Effective Dielectric Constant

Fig: 4.1 Microstrip line, its electric field lines, and effective dielectric constant geometry.

4.1 Fringing Effects Because the dimensions of the patch are finite along the length and width, the fields at the edges

of the patch undergo fringing. This is illustrated along the length in Figures 2.1(a,b) for the two

radiating slots of the microstrip antenna. The same applies along the width. The amount of

fringing is a function of the dimensions of the patch and the height of the substrate. For the

principal E-plane (xy-plane) fringing is a function of the ratio of the length of the patch L to the

height h of the substrate (L/h) and the dielectric constant εr of the substrate. Since for microstrip

antennas L/h>> 1, fringing is reduced; however, it must be taken into account because it

influences the resonant frequency of the antenna. The same applies for the width. For a

microstrip line shown in Figure 4.1(a), typical electric field lines are shown in Figure 4.1(b).

This is a nonhomogeneous line of two dielectrics; typically the substrate and air. As can be seen,

most of the electric field lines reside in the substrate and parts of some lines exist in air. As

W/h >> 1 and εr >> 1, the electric field lines concentrate mostly in the substrate. Fringing in

this case makes the microstrip line look wider electrically compared to its physical dimensions.

Since some of the waves travel in the substrate and some in air, an effective dielectric constant

εreff is introduced to account for fringing and the wave propagation in the line. To introduce the

effective dielectric constant, let us assume that the center conductor of the microstrip line with its

original dimensions and height above the ground plane is embedded into one dielectric, as shown

in Figure 4.1(c). The effective dielectric constant is defined as the dielectric constant of the

14

uniform dielectric material so that the line of Figure 4.1(c) has identical electrical characteristics,

particularly propagation constant, as the actual line of Figure 4.1(a). For a line with air above the

substrate, the effective dielectric constant has values in the range of 1 < εreff < εr. For most

applications where the dielectric constant of the substrate is much greater than unity (εr >> 1),

the value of εr will be closer to the value of the actual dielectric constant εr of the substrate. The

effective dielectric constant is also a function of frequency. As the frequency of operation

increases, most of the electric field lines concentrate in the substrate. Therefore the microstrip

line behaves more like a homogeneous line of one dielectric (only the substrate), and the

effective dielectric constant approaches the value of the dielectric constant of the substrate.

Typical variations, as a function of frequency, of the effective dielectric constant for a microstrip

line with three different substrates are shown in Figure 14.6.

For low frequencies the effective dielectric constant is essentially constant. At intermediate

frequencies its values begin to monotonically increase and eventually approach the values of the

dielectric constant of the substrate. The initial values (at low frequencies) of the effective

dielectric constant are referred to as the static values, and they are given by,

W/h >> 1

ε𝑟𝑒𝑓𝑓 = 𝜀𝑟+12

+ 𝜀𝑟− 12

+ �1 + 12 ℎ𝑤�−0.5

(4.1)

4.2 Effective Length, Resonant Frequency, and Effective Width Because of the fringing effects, electrically the patch of the microstrip antenna looks greater than

its physical dimensions. For the principal E-plane (xy-plane), this is demonstrated in Figure 14.7

where the dimensions of the patch along its length have been extended on each end by a distance

3L, which is a function of the effective dielectric constant εreff and the width-to-height ratio

(W/h). A very popular and practical approximate relation for the normalized extension of the

length is

15

(a) Top View (b) Side View

Fig: 4.2 Physical and effective lengths of Rectangular microstrip antenna

∆𝐿ℎ

= 0.412 �𝜀𝑒𝑓𝑓+0.3� �𝑊ℎ+0.264�

�𝜀𝑒𝑓𝑓−0.3� �𝑊ℎ+0.8� (4.2)

Since the length of the patch has been extended by 3L on ach side, the effective length of the

patch is now (L = λ/2 for dominant TM010 mode with no fringing)

Leff = L + 2∆L (4.3)

For the dominant TM010 mode, the resonant frequency of the microstrip antenna is a function of

its length. Usually it is given by ,

(𝑓𝑟)010 = 12𝐿√𝜀𝑟 �𝜇𝑜𝜀𝑜

= 𝑣𝑜2𝐿 √𝜀𝑟

(4.4)

where 𝑣𝑜 is the speed of light in free space. Since (14-4) does not account for fringing, it must be

modified to include edge effects and should be computed using ,

(𝑓𝑟𝑐)010 = 𝑞 𝑣𝑜2𝐿 √𝜀𝑟

(4.5)

Where ,

𝑞 = (𝑓𝑟𝑐)010(𝑓𝑟)010

(4.6)

The q factor is referred to as the fringe factor (length reduction factor). As the substrate height

increases, fringing also increases and leads to larger separations between the radiating edges and

lower resonant frequencies.

16

4.3 Design Based on the simplified formulation that has been described, a design procedure is outlined

which leads to practical designs of rectangular microstrip antennas. The procedure assumes that

the specified information includes the dielectric constant of the substrate (𝜀𝑟) , the resonant

frequency (𝑓𝑟 ), and the height of the substrate h. The procedure is as follows:

Specify: εr , 𝑓𝑟 (in Hz), and h.

Determine: W,L

Design procedure:

1. For an efficient radiator, a practical width that leads to good radiation efficiencies

Is :

W = 12fr�µ0ε0

� 2εr+1

= 𝑣02fr

� 2εr+1

(4.7)

Where, 𝑣0 is the free space velocity of light.

Determine the effective dielectric constant of the microstrip antenna using, ε𝑟𝑒𝑓𝑓

Once W is found, determine the extension of the length ∆L using

The actual length of the patch can now be determined by solving for L, or

𝐿 = 12𝑓𝑟�𝜀𝑟𝑒𝑓𝑓�𝜇0𝜀0

− 2∆𝐿 (4.8)

Calculate for εreff and ∆L/h using the equations (4.1) and (4.2) respectively.

17

Fig: 4.3 Circular and square mesh patches with their solid equivalents

The microstrip patch antenna is further meshed as shown in the figure. We are concentrating on

the designing of the Square Patch antenna. Square mesh patches have an overall dimension of ‘a’

with square holes of side ‘c’ spaced apart. If is the number of holes in one direction then the

amount of metal in the meshed patch compared to the solid patch is given by 𝑚 = 1 − 𝑛2𝑐2

𝑎2 and

the number of lines per wavelength by λ by λ𝑐+𝑑

. A reference patch, a conventional solid metal

patch and ground plane printed on RT Duroid substrate with εr = 2.33, was used in all cases for

comparing the measured parameters to provide a consistent benchmark.

4.4 Meshed Patch Over the solid ground plane. A series of meshed patches were manufactured with different line widths and line densities to

establish the basic properties of gain, cross-polarization and resonant frequency. The input

impedance is higher for the meshed patch and so the feed point is closer to the centre. The

18

meshed patches were placed over solid ground planes at this stage. The first measurements

examined the effects of changing the line width and the line spacing on gain and cross-

(a) (b)

Fig: 4.4 Current distribution on meshed patch over solid ground plane

Polarization. The measured results for five samples are plotted in Fig. 4.5,where it can be seen

that the gain improves as the line width increases and the spacing decreases, i.e. as the area of

metal increases over the patch. On the other hand thin, widely spaced lines have better cross-

polarization. There is, therefore, a trade-off between gain and cross-polarization for a given

geometry. More work is needed to understand the effect of the meshing parameters on the

bandwidth. In general the bandwidth remained at about 1% for this patch study but variations up

to 0.3% were noted. The resonant frequency reduces as the percentage of metal decreases as

shown in Fig. 4.6, e.g. a meshed patch with side a=65mm, c= 2.5mm , d=0.7mm resonates at

1.37 GHz (52% metal) while the same standard patch antenna unmeshed has a resonance at 1.48

GHz. Hence for a given patch size the resonant frequency goes down as the number of mesh

lines is reduced resulting in a smaller antenna at a given frequency. The relationship was not

linear as the frequency of resonance reduces more quickly when the metal percentage falls below

60% as seen in Fig. 4.6. The effects noted in Figs. 4.5 and 4.6 were investigated further using the

simulation package Momentum. Fig. 4.4 shows the current distribution computed over the

meshed patch in two forms, Fig. 4.4(a) shows the magnitude while Fig. 4.4(b) shows the vector

19

Fig: 4.5 Effect of line width and line spacing on measured normalized gain and cross-polarization. (a) Line spacing (b) line width. ——— gain change - - - - -- - cross-polar level.

20

Fig: 4.6 Measured resonant frequency of mesh square patch as a percentage of metal content

compared to a solid metal patch.

The feed point is clearly visible. Fig. 4.4(a) shows that current is distributed on each of the

vertical lines of the mesh uniformly whereas for a conventional patch the current density is high

only at the edges of the patch. Note also the high current magnitudes at the edges of each mesh

line in Fig. 4.4(a). These excited mesh lines are closely coupled and the resulting radiation

pattern is similar to that measured for a standard patch. The loss in gain noted in Fig. 4.5 is

mainly accounted for by the conductor losses due to the high currents at the edge of each mesh

line. The current vector diagram in Fig. 4.4(b) shows that the currents flowing from the top to the

bottom of the patch flow into the horizontal conductor lines as well at the junctions with the

vertical lines. The consequence of this is that the current paths are longer and hence the meshed

patch radiates at a lower frequency than a standard patch. Therefore thicker mesh lines give rise

to a lower resonant frequency than thin ones.

4.5 Meshed ground plane The ground planes used in this study were about 2.5 times the size of the patch, resulting in some

radiation diffracted to the rear. An experimental study investigated meshing the ground plane in

a similar way to that of the patches, thus creating a more optically transparent antenna. A square

mesh structure was used for the rectangular patches operating in the fundamental mode. It should

21

be noted that using a standard patch over a meshed ground offered no significant benefits.

A number of effects were observed when the ground plane was meshed and combined with a

meshed patch. Meshing the ground plane improved the bandwidth which increased typically

from 0.6% to 1.6% for 25% metallization while the resonant frequency reduced further to 1.21

GHz. Hence the resonant frequency of the standard patch at 1.48 GHz was reduced to 1.21 GHz

for the fully meshed patch, a reduction of 32%.The radiation patterns were most affected as

shown in Fig. 6. The most notable change was in the back radiation which increases inversely

with the density of the mesh. This is because the ground plane effectively leaks radiation through

the mesh, the more holes in the mesh the greater the leakage. The meshing also improves the

cross polarization levels in the forward direction by about 5 dB.

Fig: 4.7 Measured radiation patterns in H plane for rectangular meshed patch with solid and meshed ground planes. Solid ground plane; - - - - Meshed ground plane.

22

CHAPTER 5

Applications and Future Challenges

It is widely used in the vehicular data communication application, also implemented in the places

where you want to hide the antenna.

Autonomous communications systems often involve the use of separate solar cells and antennas,

which necessitate a compromise in the utilization of the limited surface area available. These

separate items may be combined thus saving valuable 'real estate', provided that the antennas and

solar cells are compatible. One method for achieving this is to integrate the two kinds of device

on the same element i.e. solar cells are intimately combined with printed antennas, providing a

new device called SOLANT.

Mesh Patch antennas can also be integrated into a car wind screen.

Future Challenges is to increase the transparency of the antenna while maintaining the radiation

properties of the antenna.

23

CHAPTER 6

Conclusions

This seminar explains the effects of meshing a Patch antenna and the ground plane. The radiation

patterns are not significantly affected by meshing the patch alone, keeping a solid ground plane,

but the gain suffers by up to 3 dB when compared to a standard patch. In general, as the meshing

gets denser, the gain of the antenna also increases. Bandwidth is a function of meshing space,

bandwidth increases with increase in mesh spacing. However, gain of the antenna reduces with

the increase in the mesh spacing. Meshing the patch lowered the resonant frequency by up to

20% and on meshing the ground plane as well as the patch radiation leaks through the mesh

increasing the radiated fields in the reverse direction dependent on the mesh density. The

resonant frequency drops further and a reduction of 32% was measured. The meshed patch offers

a complex trade-off between parameters but gives opportunities for improving the bandwidth

and reducing the cross polarization and the antenna dimensions at the expense of the gain.

24

References

[1] L. Economou and R. J. Langley, “Circular microstrip patch antennas on glass for vehicle

applications,” in Proc. Inst. Elect. Eng. Microwave, Antennas and Propagation, vol. 145,

1998, pp. 416–420.

[2] P. Lowes, S. R. Day, E. Korolkiewicz, and A. Sambel, “Performance of microstrip patch

antenna with electrically thick laminated superstrate,” Electron. Lett., vol. 30, pp. 1903–

1905, 1994.

[3] S. Vaccaro, P. Torres, J.R. Mosig, A. Shah, J.-F. Ziircher, A.K. Skrivervik, F. Gardiol, P.

de Maagt and L. Gerlach, “Integrated Solar Panel Antenna,” in Electronics Letters, vol.

36, 2nd March 2000, pp. 390-391

[4] G. Clasen and R.J. Langley, “Meshed patch antenna integrated into car windscreen”,

Electronics Letters, vol. 36, 27th April 2000, pp. 781-782

[5] Timothy W. Turpin and Reyhan Baktur, “Meshed Patch Antennas Integrated on Solar

Cells” in IEEE Antennas and Wireless Propagation Letters, vol. 8, 2009, pp. 693-696

[6] Constantine A. Balanis, Antenna Theory: Analysis and Design, 3rd Edition; John Wiley

Sons Inc. New York, 2001,pp 811-872