MIMO wireless communication

Study of the Diversity Antenna Array for the MIMO Wireless Communication Systems

by

Choo Chiap Chiau

A thesis submitted to the University of London in partial fulfilment of the requirements for the degree of Doctor of Philosophy

Department of Electronic Engineering Queen Mary, University of London

United Kingdom

April 2006

To my family, girlfriend Ms Wendy Kang and

in memory of my brother Mr Yang Wee Chiau

i

Abstract

Recent developments in wireless communications have shown that by using

multiple antenna elements at both transmitter and the receiver, it is possible to

substantially increase the capacity in a wireless communication system without

increasing the transmission power and bandwidth. This system with multiple antenna

elements at both link-ends is termed the MIMO (Multiple-Input Multiple-Output)

system. Despite considerable research being done on MIMO systems, the design of

realistic diversity antennas on mobile terminals for MIMO systems remains a

challenging issue.

The main challenge in designing two or more antennas on a small mobile

terminal is to achieve a high isolation between the antennas. It is very difficult to

achieve a high isolation with the existing handset antennas, such as PIFA (Planar

Inverted-F Antenna), which induce substantial surface currents on the ground plane.

In this thesis, two approaches are proposed to address this challenge in antenna

design. Firstly, a compact self-balanced antenna which is a folded loop antenna with

a loaded dielectric slab is proposed. The second method is to employ

Electromagnetic Band-Gap (EBG) structures on the ground plane of the antennas to

suppress the surface currents at specific frequencies band (i.e. stopband region). A

new EBG structure with smaller dimensions and wider stopband is developed.

Based on the proposed compact self-balanced antenna, a four-element

diversity antenna array is designed and implemented on a PDA terminal. An

isolation of more than 20dB is achieved between each pair of antennas in the

measurement. This high isolation leads to a diversity gain of 14.22dB at 99%

reliability in an indoor environment. Furthermore, a ray tracing simulator (i.e.

Wireless InSite) is used to assess the spectral efficiency of a 4x4 MIMO system in an

indoor environment with the proposed diversity antenna array as the receivers. The

channel capacity achieved with the proposed diversity antenna array in the 4x4

MIMO system is more than twice the capacity achieved in a SISO system.

In conclusion, a novel solution for the design of multiple antennas on a small

terminal is presented in this thesis.

ii

Acknowledgements

First of all I would like to express my deep and sincere gratitude to my

supervisor, Dr Xiaodong Chen, for his guidance, supervision and stimulating

discussions throughout my PhD study, as well as his constructive comments in the

writing up of this thesis. I would also like to express my warm and sincere thanks to

Professor Clive G. Parini for his valuable suggestions and encouragement during the

course of my PhD study.

My special thanks to Mr John Dupuy for his guidance and assistance during

my measurement works conducted in the Antenna Measurement Laboratory at

Queen Mary University of London.

I want to thank all my colleagues and friends for all their help, advice and

valuable contributions to this thesis. I am especially obliged to Dr Jianxin Zhang, Dr

Yasir Alfadhl, Mr Yue Gao, Miss Marianna Setta and Mr Sunil Sudhakaran.

I would like to give my special thanks to my parents, brothers, sister and Miss

Wendy Mingli Kang for their support, patience and love. Without their

encouragement, motivation and understanding it would have been impossible for me

to complete this work.

The financial support of the Department of Electronic Engineering, Queen

Mary University of London is thankfully acknowledged.

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Table of Contents

Abstract i

Acknowledgments ii

Table of Contents iii

List of Figures vii

List of Tables xvi

Abbreviations xvii

1 Introduction......................................................................................................... 1

1.1 Introduction ....................................................................................................1

1.2 Review of the State-of-Art .............................................................................3

1.3 Motivation ......................................................................................................5

1.4 Organisation of the thesis...............................................................................7

References ............................................................................................................ 8

2 MIMO and Diversity ........................................................................................ 11

2.1 Introduction ..................................................................................................11

2.2 MIMO Systems ............................................................................................11

2.2.1 Channel Capacity ..............................................................................12

2.2.2 Spatial multiplexing ..........................................................................15

2.2.3 Space-time coding.............................................................................16

2.3 Antenna Diversity ........................................................................................17

2.3.1 Diversity Combining Techniques .....................................................19

2.3.1.1 Switched Combining.......................................................... 19

2.3.1.2 Selection Combining.......................................................... 20

2.3.1.3 Equal Gain Combining....................................................... 20

2.3.1.4 Maximum Ratio Combining .............................................. 21

2.3.2 Diversity Gain ...................................................................................21

2.3.3 Correlation ........................................................................................23

2.3.4 Branch Power Ratio and Mean Effective Gain .................................25

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2.3.5 Antenna Diversity Techniques..........................................................26

2.3.5.1 Spatial Diversity................................................................. 27

2.3.5.2 Pattern Diversity................................................................. 28

2.3.5.3 Polarisation Diversity......................................................... 29

2.4 Environmental factor....................................................................................29

2.4.1 Angular density functions in outdoor urban environments...............30

2.4.2 Angular density functions in indoor environments...........................31

2.4.3 Proposed statistical model.................................................................31

2.4.4 Cross Polarisation Power Ratio (XPR) .............................................32

2.4.5 Proposed propagation model.............................................................32

2.5 Summary ......................................................................................................33

References ......................................................................................................... 34

3 Small Antennas on Mobile Terminals ............................................................. 38

3.1 Introduction ..................................................................................................38

3.2 Small Antennas on Mobile Terminals..........................................................38

3.2.1 Design Parameters for Antennas on Small Mobile Terminals..........39

3.3 Review of Small Antennas on Mobile Terminals ........................................39

3.3.1 Monopole ..........................................................................................40

3.3.2 Normal Mode Helical Antenna (NMHA) .........................................41

3.3.3 Meander line antenna ........................................................................43

3.3.4 Inverted-L Antenna (ILA) and Inverted-F Antenna (IFA) ...............44

3.4 New Concept for Mobile Terminal Antenna Design ...................................47

3.5 Summary ......................................................................................................51

References .......................................................................................................... 51

4 Electromagnetic Bandgap (EBG) Structures ................................................. 54

4.1 Introduction ..................................................................................................54

4.2 EBG Background .........................................................................................54

4.2.1 Metallo-dielectric structures .............................................................56

4.2.2 Uniplanar Compact PBG (UC-PBG) Structure ................................57

4.2.3 Square lattice of circles etched on the ground plane.........................58

4.3 Multiperiod EBG structure on the ground plane..........................................60

4.3.1 1-D multi-period EBG structure........................................................61

4.3.2 2-D multi-period EBG Structure.......................................................65

4.3.3 Microstrip patch antenna on the multi-period EBG structure...........67

v

4.4 Multi-period EBG structure in the substrate ................................................77

4.5 Summary ......................................................................................................80

References .......................................................................................................... 81

5 Diversity Antenna Array for MIMO Systems................................................ 84

5.1 Introduction ..................................................................................................84

5.2 Single Folded Loop Antenna .......................................................................84

5.2.1 Single Folded Loop Antenna at 5.2GHz...........................................84

5.2.2 Single Dielectric Loaded Folded Loop Antenna at 5.2GHz .............92

5.3 Four-Element Diversity Antenna Design.....................................................96

5.3.1 Return loss and isolation performances ............................................99

5.3.2 Radiation patterns ...........................................................................101

5.4 Diversity Performance of the Four-element Diversity Antenna Array......107

5.4.1 Correlation and MEG......................................................................107

5.4.2 Diversity Gain .................................................................................109

5.4.3 Effect of cross polar ratio (XPR) ....................................................110

5.5 Channel Capacity .......................................................................................112

5.6 Dielectric Loaded Folded Half-loop Antenna............................................115

5.6.1 Design Concept ...............................................................................115

5.6.2 Return loss performance .................................................................118

5.6.3 Radiation patterns and gain.............................................................118

5.6.4 Effect of the ground plane...............................................................120

5.6.4.1 Effect on the size of the ground plane............................. 120

5.6.4.2 Effect of the antenna’s location on the ground plane...... 122

5.6.5 Four elements of Dielectric Loaded Folded Half-loop Antennas on a PDA.........................................................................................123

5.7 Summary ....................................................................................................126

References ........................................................................................................ 128

6 Conclusions and Future Work....................................................................... 129

6.1 Summary ....................................................................................................129

6.2 Key Contributions ......................................................................................131

6.2.1 Multiperiod EBG Structure .............................................................131

6.2.2 Four-element diversity antenna array on a PDA.............................131

6.2.3 Miniature Dielectric Loaded Folded Half-loop Antenna ................132

6.3 Future work ................................................................................................132

vi

Publications............................................................................................................. 134

Appendix A ............................................................................................................. 139

Appendix B ............................................................................................................. 148

Appendix C ............................................................................................................. 149

vii

List of Figures

Figure 1-1: Experimental MIMO systems at 2.1GHz from Lucent Technologies, (a)

multi-antenna base station with four omnidirectional vertically polarised ¼

dipole antennas at 4λ apart along a line at a height of about 3m, (b) receiver

terminal antennas are bow-tie printed dipoles with alternating 45º polarisation

fitting on the back of a laptop.(Reproduced from [10]) ......................................... 4

Figure 1-2: MIMO wireless routers by (a) Netgear (Model: RangeMax 240) [16], (b)

Linksys (Model: Wireless-G with SRX 400) [17] and (c) Belkin (Model:

Wireless Pre-N Router) [18]. Three monopoles are used on each router and

these designs are not suitable for mobile terminals. .............................................. 5

Figure 2-1: A MIMO system created by two antenna arrays, comprising Tn transmit

elements and Rn receive elements. ...................................................................... 12

Figure 2-2: Comparison of channel capacity for SISO and MIMO systems, assuming

flat Rayleigh fading and zero correlation between all transmission coefficients

in the channel. (Plotted in MATLAB according to the model from I-METRA

[6])........................................................................................................................ 14

Figure 2-3: A 2x2 MIMO system with a spatial multiplexing scheme. The original

message is demultiplexed into two sub-streams (Red and Blue) and transmitted

simultaneously from each transmitting antennas. ................................................ 16

Figure 2-4: A 2x2 MIMO system with a space-time coding scheme. The original

message is transmitted simultaneously from all the antennas at the transmitter

without sub-streaming them................................................................................. 17

Figure 2-5: Diagram showing two versions (Signals 1 and2) of transmitted signals

available from two different channels in a multipath environment. Diversity

technique is used to combine both the different signals. The combined signal

always has the highest signal level compared to the individual signals. ............. 18

Figure 2-6: Diagram showing two signals are combined in a basic diversity receiver.18

Figure 2-7: Diagram showing four types of diversity combining techniques can be

employed at the receive diversity......................................................................... 19

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Figure 2-8: Block diagram of switched combining for N branches/antenna elements

with only one receiver.......................................................................................... 19

Figure 2-9: Block diagram of selection combining for N branches/antenna elements.20

Figure 2-10: Block diagram of equal gain combining for N branches/antenna

elements. .............................................................................................................. 20

Figure 2-11: Block diagram of maximum ratio combining for N branches/antenna

elements. .............................................................................................................. 21

Figure 2-12: Cumulative distribution function of Rayleigh fading signals for a

different number of diversity branches (Plot based on equation 2.6). ................. 22

Figure 2-13: Diagram showing the relation of angular coordinates to Cartesian

coordinates. .......................................................................................................... 24

Figure 2-14: Diagram showing the effect of antenna spacing to correlation

coefficient............................................................................................................. 28

Figure 3-1: Photograph of a handset’s diversity antenna used in Japan for the PDC

system. (Handset model: P501i by NTT DoCoMo) ............................................ 40

Figure 3-2: 1/4-wavelength monopole antenna: (a) practical structure, and (b)

antenna image and current distribution. ............................................................... 41

Figure 3-3: NMHA used in a GSM mobile handset. (Reproduced from [1]) .............. 42

Figure 3-4: Geometrical configuration of a helix. ....................................................... 42

Figure 3-5: Meander printer antenna on a core. (Reproduced from [1]) ..................... 43

Figure 3-6: Dual-band meander line antenna............................................................... 43

Figure 3-7: Inverted-L antenna (ILA). ......................................................................... 44

Figure 3-8: Wire inverted-F antenna (IFA).................................................................. 45

Figure 3-9: Planar inverted-F antenna (PIFA). ............................................................ 45

Figure 3-10: Top view of a dual-band PIFA with an L-shape slot on the planar

element. (Reproduced from [16])......................................................................... 46

Figure 3-11: Photograph of multi-band PIFA in a Sony Ericsson T68i handset.

(Reproduced from [19]) ....................................................................................... 46

ix

Figure 3-12: The folded dipole antenna. ...................................................................... 48

Figure 3-13: (a) A half-wavelength dipole with a coaxial feed line; (b) a half-

wavelength folded dipole with a coaxial feed line............................................... 49

Figure 3-14: Structure of a folded loop antenna constructed from a folded dipole

where L ≈ ½ λ..................................................................................................... 50

Figure 3-15: Folded loop antenna mounted on a ground plane. .................................. 50

Figure 3-16: Surface current distribution on the ground plane of (a) a folded loop

antenna and (b) PIFA at 1.8GHz.The simulated results are obtained from CST

Microwave Studio®.The ground plane sizes are 40mm x 100mm.(c) The

geometry of PIFA is shown for clarity................................................................. 50

Figure 4-1: For frequency in the stopband, the incident wave does not travel through

the media. ............................................................................................................. 55

Figure 4-2: (a) Cross section of a high impedance surface. (b) Top view of the high-

impedance surface, showing a triangular lattice of hexagonal metal plates. ....... 56

Figure 4-3: Capacitance and inductance in the high-impedance surface. .................... 56

Figure 4-4: Effective circuit used to model the surface impedance............................. 57

Figure 4-5: Schematic of (a) UC-PBG structure etched on the ground plane of a

microstrip line, and (b) unit cell of UC-PBG structure........................................ 58

Figure 4-6: Schematic diagram of a microstrip on an EBG ground plane. The EBG

structure is a square lattice of etched circles........................................................ 59

Figure 4-7: Schematic representation of the proposed multiperiod EBG structure by

cascading two single period EBG structures........................................................ 60

Figure 4-8: Photographs of the ground plane of the EBG circuits with periods of

(a) a1 =7.5mm, structure’s dimensions = 104mm x 15mm and (b) a2 = 12mm,

structure’s dimensions = 62.5mm x 15mm. The radius of each lattice circle,

r=3mm.................................................................................................................. 61

Figure 4-9: Simulated and measured S21 parameters for EBG circuits with period of

(a) a1 =12mm, and (b) a2 = 7.5mm....................................................................... 62

x

Figure 4-10: Schematic representation of 2 models of the 1-D multiperiod EBG

structure with 8 lattice cells. Unit lattice (circle) radius=3mm, a1=7.5mm,

a2=12mm. (a) Model A and (b) Model B. ........................................................... 63

Figure 4-11: Simulated S21 curves for (a) Model A and (b) Model B in Figure 4-10. 64

Figure 4-12: Photograph of the ground plane of the proposed 1-D multiperiod EBG

structure. The dimensions of this structure are 86mm x 15mm........................... 64

Figure 4-13: Simulated and measured S21 curves for the proposed 1-D multiperiod

EBG...................................................................................................................... 65

Figure 4-14: Schematic representation of the folded 1-D multiperiod EBG structure. 66

Figure 4-15: Photograph of the ground plane of the folded 1-D multiperiod EBG

structure measured. The dimensions of this structure are 86mm x 27mm. ......... 66

Figure 4-16: Simulated and measured S21 for the folded 1-D multiperiod EBG

structure................................................................................................................ 67

Figure 4-17: Schematic diagram of the microstrip patch antenna designed at a

resonant frequency of 7.4GHz. (L= 6mm, W= 8.3mm, Wo= 0.6mm, W1=

0.7mm, yo=2.517mm) .......................................................................................... 68

Figure 4-18: Cross section of the (a) conventional patch antenna, and (b) patch

antenna on the sandwiched EBG structure. ......................................................... 69

Figure 4-19: Schematics (top view) of the patch antenna on the sandwiched EBG

structure with (a) single period a = 12mm, and (b) multiperiod a1 = 7.5mm, a2 =

12mm. .................................................................................................................. 70

Figure 4-20: Photographs of the (a) single period EBG layer, and (b) multiperiod

EBG layer before it is sandwiched to the ground plane....................................... 70

Figure 4-21: Photograph of the antenna under test in the anechoic chamber. ............. 71

Figure 4-22: Simulated (Red) and measured (Blue) input return (S11) for (a)

conventional patch antenna, (b) patch antenna on a sandwiched single period

EBG ground plane, and (c) patch antenna on a sandwiched multiperiod EBG

ground plane......................................................................................................... 72

xi

Figure 4-23: Simulated (+) and measured () radiation patterns of E-plane co-

polarisation for (a) conventional antenna ,(b) antenna on a sandwiched single

period EBG ground plane, and (c) antenna on a multiperiod EBG ground plane.74

Figure 4-24: Simulated (+) and measured () radiation patterns of H-plane co-

polarisation for the (a) conventional antenna ,(b) antenna on a sandwiched

single period EBG ground plane, and (c) antenna on a multiperiod EBG ground

plane. .................................................................................................................... 75

Figure 4-25: Simulated surface distribution of the electric field at 7.4GHz for the (a)

conventional antenna (b) sandwiched single period EBG antenna, and (c)

sandwiched multiperiod EBG antenna................................................................. 76

Figure 4-26: Simulated S21 parameter for the single period EBG substrate circuit of

period (a) a1 = 10mm, and (b) a2 = 7.5mm. The radius of the unit lattice circle,

r is 3mm. .............................................................................................................. 78

Figure 4-27: Fabrication process of the multiperiod EBG substrate structure............. 79

Figure 4-28: Photograph of the fabricated multiperiod EBG substrate fabricated. The

dimensions of this substrate are 101 x 15 mm. .................................................... 79

Figure 4-29: Simulated and measured S21 parameter for a multiperiod EBG substrate

circuit with periods, a1 = 7.5mm, a2 = 10mm. ..................................................... 80

Figure 5-1: The schematic diagram of the folded loop antenna with a 50Ω coaxial

feed line on a ground plane. ................................................................................. 85

Figure 5-2: Simulated return loss curves of the folded loop antenna operating at

5.2GHz with different w2 and constant w1=1mm............................................... 86

Figure 5-3: Photograph of the prototype folded loop antenna operating at 5.2GHz.... 87

Figure 5-4: Simulated and measured return loss curves. ............................................. 87

Figure 5-5: Simulated (+) and measured (–) radiation patterns on the X-Z plane for

(a) co-polar and (b) cross-polar............................................................................ 89

Figure 5-6: Simulated (+) and measured (–) radiation patterns on the Y-Z plane for

(a) co-polar and (b) cross-polar............................................................................ 90

Figure 5-7: Simulation model of the folded loop antenna for the (a) balanced feeding

technique and (b) unbalanced feeding technique. ................................................ 91

xii

Figure 5-8: Distribution of the surface currents on the ground plane for (a) balanced

feeding structure and (b) unbalanced feeding structure. ..................................... 91

Figure 5-9:Schematic diagram showing (a)the dielectric slab inserted into a folded

loop antenna and (b) the dielectric loaded folded loop antenna with unbalanced

feeding line on a ground plane. ............................................................................ 93

Figure 5-10: Relation between the resonant frequency and relative permittivity. ....... 93

Figure 5-11: Relation between the bandwidth and relative permittivity...................... 94

Figure 5-12: Simulated return loss curves for the folded loop antenna with and

without the dielectric slab of relative permittivity, εr=4. ..................................... 94

Figure 5-13: Simulated return loss curves for the folded loop antenna and the

proposed dielectric loaded folded loop antenna, both antennas operating at

5.2GHz. ................................................................................................................ 95

Figure 5-14: Schematic diagram of four-element diversity antenna array on a PDA in

(a) X-Z plane and (b) 3-D view. ......................................................................... 98

Figure 5-15: Photo of the prototype diversity antenna array on a PDA terminal.(a)

Front view of the PDA and (b) the feeding structures behind the PDA. ............. 98

Figure 5-16: Return loss curves from the (a) simulated and (b) measurement results.100

Figure 5-17: Isolation between each pair of antennas on the diversity antenna array

from the (a) simulated and (b) measured models............................................... 101

Figure 5-18: Simulated (+) and measured () E-plane co-polar radiation patterns of

each antenna with respect to their individual E-field polarisation plane. .......... 103

Figure 5-19: Simulated (+) and measured () E-plane cross-polar radiation patterns

of each antenna with respect to their individual E-field polarisation plane....... 104

Figure 5-20: Simulated (+) and measured () H-plane co-polar radiation patterns of

each antenna with respect to their individual H-field polarisation plane........... 105

Figure 5-21: Simulated (+) and measured () H-plane cross-polar radiation patterns

of each antenna with respect to their individual H-field polarisation plane. ..... 106

Figure 5-22: Variation of MEG with XPR in an outdoor environment evaluated

using the Gaussian statistical model. ................................................................. 110

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Figure 5-23: Variation of MEG with XPR in an indoor environment evaluated using

the Gaussian statistical model. ........................................................................... 111

Figure 5-24: Variation of MEG with XPR in an outdoor environment evaluated

using the Laplacian statistical model. ................................................................ 111

Figure 5-25: Variation of MEG with XPR in an indoor environment evaluated using

the Laplacian statistical model. .......................................................................... 111

Figure 5-26: Floor plan of the second floor of the Department of Electronic

Engineering at QMUL. The receivers are located randomly at 1000 positions in

Rooms A and B. ................................................................................................. 112

Figure 5-27: 3-D floor plan of the second floor of the Department of Electronic

Engineering at QMUL. The red dots show the random positioning of the

receivers. ............................................................................................................ 113

Figure 5-28: Full propagation paths shown in a 3-D floor plan of the Department of

Electronic Engineering at QMUL. ..................................................................... 113

Figure 5-29: Rays arriving at receivers in (a) Room A and (b) Room B are reflected

off and/or transmitted through obstacles............................................................ 114

Figure 5-30: Channel capacity performance of the proposed diversity antenna array

in Room A and Room B compared to 4 ideal dipoles in Room A and within a

SISO system. ...................................................................................................... 114

Figure 5-31: A dielectric loaded folded loop antenna is reduced in size to form a

dielectric loaded folded half-loop antenna......................................................... 116

Figure 5-32: Schematic diagram of a dielectric loaded folded half-loop antenna and

the antenna configuration on a ground plane with the feed location. ................ 117

Figure 5-33: Structure of a folded half-loop antenna constructed from a flat copper

plate. ................................................................................................................... 117

Figure 5-34: Photograph of the fabricated dielectric loaded folded half-loop antenna

with a ground plane size of 40mm x 100mm..................................................... 118

Figure 5-35: Measured and simulated return loss of the dielectric loaded folded half-

loop antenna. ...................................................................................................... 118

xiv

Figure 5-36: Simulated (+) and measured (-) co-polar radiation patterns at 5.2GHz

for (a) E-plane and (b) H-plane.......................................................................... 119

Figure 5-37: Simulated (+) and measured (-) cross-polar radiation patterns at 5.2GHz

for (a) E-plane and (b) H-plane.......................................................................... 120

Figure 5-38: Center frequency () and the -10dB bandwidth for the different ground

plane lengths, GL. .............................................................................................. 121

Figure 5-39: Center frequency () and the -10dB bandwidth for the different ground

plane widths, GW............................................................................................... 121

Figure 5-40: Location of the antenna on the ground plane in x-y direction. The

antenna’s feed location is at x=0 and y=20mm. ................................................ 122

Figure 5-41: Center frequency () and the -10dB bandwidth of the proposed antenna

at different locations in x-direction when y=0. At y=0, the antenna’s feed point

is placed 3.5mm away from the top edge of the ground plane. ......................... 123

Figure 5-42: Center frequency () and the -10dB bandwidth of the proposed antenna

at different locations in y-direction when x=20. At y=0, the antenna’s feed point

is placed 3.5mm away from the top edge of the ground plane. ......................... 123

Figure 5-43: Schematic diagram of four elements of the dielectric loaded folded half-

loop antennas on a PDA in (a) x-z plane and (b) 3D view. ............................... 124

Figure 5-44: Simulated return loss curves of each dielectric loaded folded half-loop

antenna on a PDA. ............................................................................................ 125

Figure 5-45: Computed surface currents on the ground plane of (a) dielectric loaded

folded loop antenna and (b) dielectric loaded folded half-loop antenna at

5.2GHz. .............................................................................................................. 126

Figure A-1: Illustration of the FIT discretization....................................................... 141

Figure A-2: A cell V of the grid G with the allocation of the electric grid voltage e on

the edges of An and the magnetic facet flux bn through this surface. ............... 142

Figure A-3: A cell V of the grid G with the allocation of the six magnetic facet

fluxes which have to be considered in the evaluation of the closed surface

integral for the non-existance of magnetic charges within the cell volume....... 143

xv

Figure A-4: A cell V of the grid G with the allocation of the magnetic grid voltage

h on the edges of nA and the electric facet flux dn through this surface............ 144

Figure A-5: Grid approximation of rounded boundaries: (a) standard (stair case), (b)

sub-gridding, (c) triangular and (d) Perfect Boundary Approximation (PBA).. 146

Figure A-6: TST technique. ....................................................................................... 147

Figure B-1: Computed 3D radiation patterns of the proposed four-element diversity

antenna array. ..................................................................................................... 148

Figure C-1: 3D schematic diagram of the indoor environment (i.e. second floor of

the Department of Electronic Engineering building at QMUL) modelled using

Wireless InSite as used in this thesis. The transmitter is placed at the corridor

and the receivers (RED dots) are scattered randomly in Rooms A and B. The

ceiling has been removed for visual purpose.. ................................................... 149

xvi

List of Tables

Table 2-1: Propagation models used in this thesis [55]-[56]. mV and mH are the

mean elevation angles of vertical and horizontal polarised wave distribution

respectively, σV and σH are the standard deviations of the vertical and horizontal

polarised wave distribution respectively............................................................33

Table 5-1: The dimensions of the antenna and its ground plane................................85

Table 5-2: The optimised dimensions of the dielectric loaded folded loop antenna

and its ground plane at 5.2GHz. The dielectric slab has a relative permittivity of

4..........................................................................................................................95

Table 5-3: The E- plane of each antenna on the PDA in the Cartesian co-ordinate

system...............................................................................................................102

Table 5-4: The H- plane of each antenna on the PDA in the Cartesian co-ordinate

system...............................................................................................................102

Table 5-5: Summary of the envelope correlation from the diversity antenna array on

the PDA terminal. ρenm representing the envelope correlation between antennas

n and m.............................................................................................................108

Table 5-6: Summary of the MEG for each antenna from the diversity antenna array

on the PDA terminal. .......................................................................................108

Table 5-7: Comparison of selection combiner diversity gain performance at 99%

reliability in different environments and using different statistical models.....109

Table 5-8: The optimised dimensions of the dielectric loaded folded half-loop

antenna and its ground plane at 5.2GHz. The dielectric slab has a relative

permittivity of 4. ..............................................................................................117

Table C-1: Material parameters used in the model at 5.2GHz.................................150

xvii

Abbreviations 1-D One-dimensional 1G First Generation 2-D Two-dimensional 2G Second Generation 3-D Three-dimensional 3G Third Generation 4G Fourth Generation 3GPP 3rd Generation Partnership Project AMPS Advanced Mobile Phone Service AOA Angle Of Arrival AWGN Additive White Gaussian Noise BLAST Bell labs Layered Space –Time BT British Telecom cdf Cumulative distribution function CST Computer Simulation Technology DCS Digital Cellular System DECT Digital Enhanced Cordless Telecommunications DF Degradation Factor DRA Dielectric Resonator Antenna EBG Electromagnetic Band Gap EGC Equal Gain Combining EM Electromagnetic EGC Equal Gain Combining EWC Enhanced Wireless Consortium FDTD Finite Difference Time Domain FE Finite Element FIT Finite Integral Technique IFA Inverted-F Antenna GSM Global System for Mobile Communications IFA Inverted-F Antenna ILA Inverted-L Antenna I-METRA Intelligent Multi-Element Transmit and Receive Antennas i.i.d. independent and identical distributed IS-136 Interim Standard - 136 LAN Local Area Network MAN Metropolitan Area Networks MEG Mean Effective Gain MIMO Multi-Input Multi-Output

xviii

MoM Method of Moment MRC Maximum Ratio Combining NLOS Non-Line Of Sight NMHA Normal Mode Helix Antenna Ofcom Office of Communication OFDM Orthogonal Frequency Division Multiplexing PBA® Perfect Boundary Approximation PBG Photonic Band Gap PCMIA Personal Computer Memory Card International Association PCS Personal Communication Service PDA Personal Digital Assistant PDC Personal Digital Cellular PIFA Planar Inverted-F Antenna QMUL Queen Mary University of London RT Ray Tracing SAR Specific Absorption Rate SISO Single-Input Single-Output SIMO Single-Input Multiple-Output SM Spatial Multiplexing SNR Signal to Noise Ratio SVD Singular Value Decomposition TDMA Time Division Multiple Access

TST Thin Sheet TechniqueTM

UC-PBG Uniplanar Compact PBG

ULA Uniform Linear Array

WAN Wide Area Network

WiFi Wireless Fidelity

WiMAX Worldwide Interoperability for Microwave Access

WLAN Wireless Local Area Network

XPD Cross-Polar Discrimination

XPR Cross-polarisation Power Ratio

Chapter 1: Introduction

1

Chapter 1 Introduction

1.1 Introduction

Over the last two decades, the use of mobile communication technology has

experienced a significant growth from first-generation (1G) analogue voice-only

communication to second-generation (2G) digital voice communication. These 2G

technologies became popular worldwide including GSM (Global System for Mobile

Communications) in Europe, IS-136 (also known as US-TDMA and Digital AMPS) in

the U.S., and PDC (Personal Digital Communications) in Japan. Currently, the third

generation (3G) mobile communication technology not only provides digital voice

services, also provides video telephony, internet access and video/music download

services. Further, the forthcoming fourth-generation (4G) mobile telephone

technology aims to provide on-demand high quality video and audio services.

Apart from mobile communication technology, wireless local area network

(WLAN) technology has also made a giant stride by introducing WiFi (Wireless

Fidelity). WiFi is a set of product compatibility standards for WLAN technology

based on the IEEE 802.11 specifications. It enables a person with a wireless-enabled

computer, laptop or personal digital assistant (PDA) to connect to the internet within

proximity of an access point (Hot-Spot) at a maximum data rate of 54Mbps. Recently,

WiFi is no longer used solely for internet connectivity, it is further used to broadcast

quality multimedia content throughout the entire home. WiMAX (Worldwide

Interoperability for Microwave Access) is a forthcoming wireless technology that is

designed for Metropolitan Area Networks (MAN) based on the IEEE 802.16


2

specifications. It aims to provide high speed wireless internet connections over long

distances.

Both the forthcoming 4G and WiMAX technologies require high data rates and

longer range to provide quality services to end users. In order to achieve this, wireless

communication technology has to be pushed to the physical limits of the radio

channels. Traditional communication engineering states that the channel capacity or

data rate is limited by the bandwidth and transmission power. A well known upper

bound on the maximum achievable data rate for the ideal band-limited additive white

Gaussian noise (AWGN) channel is the Shannon-Nyquist criterion [1]-[2]. Having an

available channel bandwidth, W and signal-to-noise ratio (SNR) over this bandwidth,

the maximum transmit data rate is:

From equation (1.1) shown above, data rates can only be increased by increasing the

bandwidth occupation or transmission power. However, it is very expensive to

increase the spectrum usage. Also, the signal power can not be readily increased as the

communication system is interference limited.

Until a few years ago, these limits have been expanded by introducing the

spatial domain to mobile communication antennas. By introducing an array of antenna

elements at both receiver and transmitter, the channel capacity of that system can grow

linearly with the number of antennas under ideal conditions. This system with multiple

antennas at both link-ends is termed the MIMO (i.e. multiple-input multiple-output)

system. The capacity of multi-antenna fading channels applying antenna arrays at both

link-ends was first published by Winters in 1987 [3]. However, the potential of these

systems was appreciated more than ten years later when they were re-invented by

Foschini and Gans [4]-[5], and Telatar [6]. Since then, tremendous efforts have been

put into the research and development of MIMO systems.

2log (1 ) /C W SNR bits s= + (1.1)


3

1.2 Review of the State-of-Art

Research conducted on MIMO systems has advanced greatly, with Lucent

Technologies (Bell Labs Innovations) being the leading player. Lucent Technologies

has conducted measurements on 2x2, 4x4 and 16x16 MIMO systems in an urban

environment – Manhattan, New York [7]. In the measurement campaigns, vertically

and horizontally polarised slot antenna elements were used for both the transmitter and

receiver. At the receiver (a laptop was used as the receiver terminal), the antenna

elements were spaced half-wavelength apart from each other to achieve low

correlation and high capacity. System capacities of 5.5bps/Hz, 10bps/Hz and 35bps/Hz

were reported, respectively, in 2x2, 4x4 and 16x16 MIMO systems at the 10dB system

SNR. Measurement on MIMO channels in the rural environment has also been carried

out by Lucent Technologies [8]. It was reported that the capacity in a 8x10 MIMO

system was approximately eight times the corresponding capacity in a 1x1 SISO

(single-input single-output) system, and 3.2 times the corresponding capacity in a

1x10 SIMO (single-input multiple-output) system. The measurement also found that

using antenna arrays containing antennas of both horizontal and vertical polarisations

could increase the capacity by approximately fifty percent. Measurements on 4x4

MIMO systems over a 3G wireless network have also been conducted in an indoor

environment and it was reported the overall capacity of 7.75 bps/Hz was obtained [9].

The transmitters and the receivers are shown in Figure 1.1 [10]. Furthermore, Lucent

Technologies has designed two prototype chips for mobile devices that implement the

MIMO wireless network technology [11].

In the UK, Ofcom (Office of communication – the independent regulator and

competition authority for the UK communications industries) has supported and

funded a MIMO technology research project –‘Antenna Array Technology and MIMO

Systems’ [12]-[13] , which involved Queen Mary University of London (QMUL),

University of Bristol, University of York, BT Exact Technologies, Toshiba Research

Europe Limited and Antenova Ltd. The Ofcom project has shown that MIMO

systems can provide significant capacity gains compared to SISO systems, but the

channel capacity is strongly dependent on the environments as well as the antenna

configurations. In the project, QMUL and Antenova Ltd developed two different four-

element antenna arrays on PDAs; i.e. dielectric loaded folded loop antenna arrays (the

dielectric loaded folded loop antenna presenting in this thesis were employed) and


4

dielectric resonator antenna (DRA) arrays [13]. However, in both designs the antennas

were located on each corner of the PDA terminal which was impractical as some

antennas would be covered by the user’s hand for most of the time.

(a) (b)

Figure 1-1: Experimental MIMO systems at 2.1GHz from Lucent Technologies, (a) multi-antenna base station with four omnidirectional vertically polarised ¼ dipole antennas at 4λ apart along a line at a height of about 3m, (b) receiver terminal antennas are bow-tie printed dipoles with alternating 45º polarisation fitting on the back of a laptop.(Reproduced from [10])

The European Commission has supported the I-METRA (Intelligent Multi-

Element Transmit and Receive Antennas) project [14]. The I-METRA project

consortium comprised of Universitat Politecnica de Catalunya (Spain), Aalborg

Universtiy (Denmark), Nokia (Finland) and Vodafone Ltd. The I-METRA project has

demonstrated that, by doubling the number of antenna elements at the receiver from

two to four, the system’s capacity and coverage will be significantly improved.

Today, there are some MIMO products readily available in the market for

WLAN applications (IEEE 802.11a/b/g standards). With the introduction of MIMO

technology and the OFDM (orthogonal frequency division multiplexing) modulation

scheme, WLAN can fully take advantage of high speed broadband internet

connections, accommodate bandwidth intensive applications such as video streaming

and provide reliable coverage throughout a business or residence. The latest Airgo

MIMO chipsets [15] are used in wireless routers by Netgear (Model: RangeMax 240)

[16], Linksys (Model: Wireless-G with SRX 400) [17] and Belkin (Model: Wireless


5

Pre-N Router) [18] as shown in Figure 1-2. Also, Ruckus Wireless Inc. has developed

the first wireless multimedia home distribution system using MIMO technology [19].

Airgo has reported that MIMO systems can deliver a peak capacity of 108Mbps; this

is compared to a peak capacity of 54Mbps as delivered by SISO systems. However, to

date, MIMO technology is still not being implemented on small mobile terminals (e.g.

PDAs and handsets).

(a) (b) (c)

Figure 1-2: MIMO wireless routers by (a) Netgear (Model: RangeMax 240) [16], (b) Linksys (Model: Wireless-G with SRX 400) [17] and (c) Belkin (Model: Wireless Pre-N Router) [18]. Three monopoles are used on each router and these designs are not suitable for mobile terminals.

The next generation WLAN will be extended from IEEE 802.11a/b/g standards

to the new IEEE 802.11n standard. In October 2005, twenty-seven WiFi industry

leaders have formed the Enhanced Wireless Consortium (EWC) to help accelerate the

IEEE 802.11n development process and promote a technology specification for

interoperability of next generation WLAN products [20] . This new standard is aimed

to deliver a peak capacity of 600Mbps using MIMO technology and including other

advanced technologies, e.g. beamforming.

1.3 Motivation

For a while, most of the studies on MIMO technology have focused on signal

processing algorithms and channel characteristics. Recently, the antenna’s effect on

MIMO system has been investigated by assuming the antennas are ideal half-

wavelength dipoles which radiate omnidirectionally in the azimuth plane [21]-[25].

However, when two or more dipoles/monopoles are placed closely to each other on a


6

mobile terminal, the radiation pattern of each dipole/monopole is no longer

omnidirectional due to the coupling between them. Further, it is very impractical and

unrealistic to implement a number of dipoles/monopoles on a small mobile terminal.

Therefore, in order to predict the true spectral efficiencies of MIMO systems in a real

environment whereby realistic mobile terminals are used, there is a need to design an

appropriate and realistic antenna array on mobile terminals for MIMO systems.

The diversity antenna has conventionally been implemented at the base station

for current mobile communications to mitigate the fading effects of a multipath

environment. It has also been implemented at handsets for the Personal Digital

Communications (PDC) network system in Japan [26]. Therefore, the design of

diversity antenna array at mobile terminals is carried out in this thesis to improve the

reliability and increase the capacity of MIMO systems. In order to achieve a good

diversity performance, the antennas have to meet two criteria, i.e. low correlation and

similar mean power levels between the antennas. It is a very challenging task to satisfy

both criterions given the small dimensions of mobile terminals. In this thesis, a

diversity antenna array consisting of four antenna elements on a PDA terminal having

low correlation and mean power levels between the antenna elements is proposed and

investigated.

For ideal diversity antenna array (i.e. zero correlation), the antenna elements

should have zero mutual coupling between them [27]. It has been shown that a

compact two feed ports diversity antenna can achieve a correlation of much less than

0.5 when the isolation between the antenna ports is more than 20dB [28]. The strong

surface current on the ground plane of the antennas has been one of the main reasons

for the high mutual coupling between the antennas and results in high correlation.

Conventionally, low correlation can be achieved at the base stations by spacing the

antennas an appropriate distance apart – spatial diversity. However, at mobile

terminals, the space is very limited when there are more than two antenna elements.

When the antennas are spaced closely, this will result in high correlation and no

diversity gain could be achieved. Therefore, there is a need to explore diversity

antenna arrays with different forms of diversities to achieve low correlation and

similar mean power levels between the antennas for MIMO terminals. As such, spatial,

polarisation and pattern diversities are considered in this thesis for the four-element

diversity antenna array design.


7

Electromagnetic band gap (EBG) structures have attracted a lot of attention

recently due to its ability to suppress the surface wave in a specified band of frequency

(this region of frequencies is called the ‘stopband’) [30]-[31]. EBG structure has the

potential to reduce the mutual coupling between the antennas. However, usually the

EBG structure is large in size and has a narrow stopband bandwidth. It is a very

challenging task to reduce the size of the EBG structure whilst increasing the stopband

bandwidth simultaneously. In this thesis, an easy method to reduce the size of the

EBG structures and increase the stopband bandwidth simultaneously is proposed.

1.4 Organisation of the thesis

The thesis is organised in six chapters as follows: Chapter 2: This chapter covers the introduction to MIMO systems and diversity

techniques. Two different types of coding schemes that can be used to exploit the

MIMO channels are addressed. In order to evaluate the correlation, mean effective

gain (MEG) and diversity gain of the diversity antenna in different environments, two

different propagation models that are used in this thesis are discussed.

Chapter 3: A review on small antennas for mobile terminals is detailed in this chapter.

The problems of built-in antennas implemented on current mobile terminals are

addressed. Folded loop antenna with small current leakage to the ground plane is

briefly discussed and further addressed in Chapter 5.

Chapter 4: In this chapter, three different types of EBG structures are introduced.

Amongst the three types of EBG structures, the simple planar EBG structure

comprising of a square lattice of holes etched on the ground plane of a microstrip line

is further investigated. Following this, a new multiperiod EBG structure consisting of

two different periods of square lattice of holes etched on the ground plane is proposed.

The proposed multiperiod EBG structure is studied numerically and experimentally.

Further, the capability of the multiperiod EBG structure to suppress the surface wave

on the ground plane of a microstrip patch antenna is demonstrated.


8

Chapter 5: A dielectric loaded folded loop antenna is studied in this chapter. A four-

element diversity antenna array on a PDA terminal is developed based on the proposed

dielectric loaded folded loop antenna. The diversity performance of the four-element

diversity antenna array is evaluated for two different propagation models (i.e. outdoor

and indoor environments). The effects of XPR (cross-polarisation power ratio) towards

the MEG of the four-element diversity antenna array are demonstrated. Furthermore,

the spectral efficiency of a 4 x 4 MIMO system is investigated in the indoor

environment. The size reduction for the proposed dielectric loaded folded loop

antenna is further studied in this chapter.

Chapter 6: This chapter concludes the research that has been done in this thesis.

Suggestions for future work are also given in this chapter.

References: [1] J.G. Proakis, Digital Communications, New York: McGraw-Hill, 1989.

[2] C.E. Shannon, “A Mathematical Theory of Communication,” Bell Syst. Tech. J., Vol. 27, pp. 379-423, 623-656, July & Oct. 1948.

[3] J. Winters, “On the capacity of radio communication system with diversity in a Rayleigh fading environment,” IEEE Journal on Selected Areas in Communications, vol. 5, no. 5, pp. 871-878, June 1987.

[4] G.J. Foschini, “Layered space-time architecture for wireless communication in a fading environment when using multi-element antennas,” Bell Labs Tech J., vol. 1, no. 2, 41-59, 1996.

[5] G.J. Foschini and M.J. Gans. “On Limits of wireless Communications in a Fading Environment when Using Multiple Antennas”. Wireless Personal Communications, 6:311-335, March 1998.

[6] I.E. Telatar, “Capacity of multi-antenna Gaussian channels,” European Transaction on Telecommunications, vol. 10, no. 6, pp. 585-595, Nov/Dec 1999.

[7] D. Chizhik, J. Ling, P.W. Wolniansky, R.A. Valensuela, N. Costa and K. Huber, “Multiple-input multiple-output measurements and modeling in Manhattan,” IEEE Journal on Selected Areas on Communications, vol. 21, no. 3, pp. 321-331, April 2003.

[8] D. Chizhik, J. Ling, D. Samardzija and R.A. Valenzuela, “Spatial and polarization characterization of MIMO channels in rural environment,” IEEE 61st Vehicular Technology Conference 2005, vol. 1, pp. 161-164, 2005.

[9] A. Adjoudani, E.C. Beck, A.P. Burg, G.m. Djuknic, T.G. Gvoth, D. Haessig, S. Manji, M.A. Milbrodt, M. Rupp, D. Samardzija, A.B. Siegel, T. Sizer, C. Tran,


9

S. Walker, S.A. Wilkus and W. Wolniansky, “Prototype experience for MIMO BLAST over third-generation wireless system,” IEEE Journal on Selected Areas in Communications, vol. 21, no. 3, pp. 440-451, April 2003.

[10] D. Samardzija, A. Lozano and C. Papadias, “Experimental validation of MIMO multiuser detection for UMTS high-speed downlink packet access,” IEEE Global Telecommunications Conference, vol. 6, pp. 3840-3844, 2004.

[11] Lucent press release, URL: http://www.lucent.com/press/ 1002/021016. bla.html.

[12] Ofcom research report, “Antenna arrays technology and MIMO systems” contract number: AY4476B, 2004.

[13] Ofcom research report, “Antenna designs for MIMO systems,” Contract number: AY4476A, 2004.

[14] IST-IMETRA, URL: http://www.ist-imetra.org/

[15] Airgo, URL: http://www.airgonetworks.com/

[16] Netgear, URL: http://www.netgear.com/

[17] Linksys, URL: http://www.linksys.com/

[18] Belkin, URL: http://www.belkin.com/

[19] Ruckus Wireless, URL: http://www.ruckuswireless.com/

[20] EWC URL: http://www.enhancedwirelessconsortium.org/

[21] J. P. Kermoal, L. Schumacher, F. Frederiksen, and P. E. Mogensen, “Polarization diversity in MIMO radio channels: Experimental validation of a stochastic model and performance assessment,” Vehicular Technology Conference, 2001. VTC 2001 Fall, IEEE VTS 54th, vol. 1, pp. 22-26, 2001.

[22] V. Pohl, V. Jungnickel, T. Haustein, and C. von Helmolt, “Antenna spacing in MIMO indoor channels,” in Proc. IEEE Vehicular Technology Conf., Birmingham, 2002. L. Dong, H. Ling, and R.W. Heath, “Multiple-input multiple output wireless communication systems using antenna pattern diversity,” Global Telecommunications Conference, 2002. GLOBECOM '02. IEEE, vol. 1, pp. 997-1001, 17-21 Nov. 2002.

[23] V. Jungnukel, V. Pohl, and C. von Helmolt, “Capacity of MIMO Systems with closely spaced antennas,” IEEE Communications Letters, vol. 7, no. 8, August 2003

[24] C. Waldschmidt, C. Kuhnert, S. schulteis, and W. Wiesbeck, “Compact MIMO-arrays based on polarisation-diversity,” Antennas and Propagation Society International Symposium, 2003. IEEE, vol. 2, pp. 499-502, June 22-27, 2003.

[25] M. J. Fakhereddin, K. R. Dandekar, “Combined effect of polarization diversity and mutual coupling on MIMO capacity,” Antennas and Propagation Society International Symposium, 2003. IEEE, vol. 2, pp. 495-498, June 22-27, 2003.

[26] K. Fujimoto and J.R. James, Mobile antenna systems handbook, 2nd edition, Artech House Inc., 2001.


10

[27] R.G. Vaughan and J.B. Andersen, “Antenna diversity in mobile communications,” IEEE Trans. Vehicular Technology, vol. 36, pp. 147-172, Nov 1987.

[28] S.C.K. Ko and R.D. Murch, “Compact integrated diversity antenna for wireless communications,” IEEE Trans. Antennas and Propagation, vol. 49, no. 6, pp. 954-960, June 2001.

[29] C.G. Buxton, W.L. Stutzman, R.R. Nealy and A.M. Orndorff, “The folded dipole: A self-balancing antenna”, Microwave and optical technology letters, vol. 29, no. 3, pp. 155-160, 2001.

[30] J. Shumpert, T. Ellis, G. Rebeiz, and L. Katehi, “Microwave and millimeter-wave propagation in photonic band-gap structure,” AP-S/URSI, pp. 678, 1997.

[31] Y. Qian, V. Radisic, and T. Itoh, “Simulation and experiment of photonic band-gap structures for microstrip circuits,” Asia-Pacific Microwave Conf. (APMC’97) Dig., pp 585-588, Hong Kong, 1997.

Chapter 2: MIMO and Diversity

11

Chapter 2 MIMO and Diversity

2.1 Introduction

In this chapter, the concept of the MIMO systems and its channel capacity are

introduced. Two popular signalling schemes which are used to exploit the MIMO

systems are described. The diversity theory and its technique are explained in detail

as it is the main study in this thesis. Besides the antenna’s characteristics, it is noted

that the propagation environment also plays an important role in diversity analysis.

Therefore, different propagation environments are described.

2.2 MIMO Systems

In a conventional mobile wireless communication system, there is only one

antenna at both transmitter and receiver. This system which is called the Single-Input

Single-Output (SISO) antenna system suffers a bottleneck in terms of capacity due to

the Shannon-Nyquist criterion [1]-[2]. As mentioned in Chapter 1, future wireless

mobile services demand much higher data bit-rate transmission. In order to increase

the capacity of the SISO systems to meet such demand, the bandwidth and

transmission power have to be increased significantly. Fortunately, recent

developments have shown that using MIMO (Multiple-Input Multiple-Output)

systems could increase the capacity in wireless communication substantially without

increasing the transmission power and bandwidth [3]-[4]. In the MIMO systems,

multiple antenna elements are required at both transmitter and receiver, as shown in


12

Figure 2-1. Usually the Rayleigh fading caused by the multipath signal propagation

is considered to be a severe problem in a wireless communication channel. However,

the MIMO system exploits the multipath signals to increase the system capacity.

Figure 2-1: A MIMO system created by two antenna arrays, comprising Tn transmit elements and Rn receive elements.

2.2.1 Channel Capacity

Figure 2-1 shows a MIMO system with nT transmit elements and nR receive

elements. For a narrowband channel, the complex transmission coefficient between

element k ∈ [1,….., nT] at the transmitter and element j ∈ [1,….,nR] at the receiver

at time t is represented by hjk(t). A matrix containing all channel coefficients

(channel coefficient matrix, H(t)) can be shown as:

Hence, a system transmitting the signal vector x(t) = [x1(t), x2(t),….,xnT(t)]T, where

xk(t) is the signal transmitted from the kth element would result in the signal vector

y(t) = [y1(t), y2(t),…..,ynR(t)]T being received, where yj(t) is the signal received by the

jth element, and

11 12 1

21 22 2

1 2

( ) ( ) ( )( ) ( ) ( )

( )

( )

T

T

R R R T

n

n

n n n n

h t h t h th t h t h t

H t

h h h t

⎛ ⎞⎜ ⎟⎜ ⎟=⎜ ⎟⎜ ⎟⎜ ⎟⎝ ⎠

(2.1)


13

where n(t) is the noise vector. In the rest of the thesis, the explicit time dependency (t)

of H(t) is dropped for simplicity.

The motivation for using MIMO systems is the possibility to achieve

orthogonal subchannels between the transmitters and receivers through a rich

scattering environment and consequently to increase the offered capacity.

Mathematically, the number of independent subchannels can be estimated by using

the singular value decomposition (SVD) of the channel coefficient matrix H as

where U is a unitary matrix of dimension nR x nR , V is a unitary matrix of dimension

nT x nT and Σ is a nR x nT diagonal matrices, and the superscript + denotes transpose

conjugate.

Conceptually, the MIMO system enables multiple data streams to be

transmitted simultaneously on the same frequency, hence increasing the bandwidth

efficiency by the number of data streams employed [4]. The capacity, C, for this

system is shown to be:

where I is the identity matrix, ρ is signal to noise ratio, H is the channel coefficient

matrix, the superscript + denotes conjugate transpose, and det (.) is the determinant.

Third Generation Partnership Project (3GPP) is a collaboration agreement

that brings together a number of telecommunications standard bodies to produce

globally applicable technical specifications and technical reports for a Third

Generation (3G) mobile system [5]. The 3GPP has shown that assuming flat

Rayleigh fading and zero correlation between all transmission coefficients in the

channel, the capacity of the MIMO systems is increased significantly as illustrated in

Figure 2-2.

( ) ( ) ( ) ( )y t H t x t n t= + (2.2)

H U V += Σ (2.3)

2log detT

C I HHnρ +⎛ ⎞⎛ ⎞

= +⎜ ⎟⎜ ⎟⎜ ⎟⎝ ⎠⎝ ⎠

(2.4)


14

0 5 10 15 20 25 30 350

10

20

30

40

50

60

70

80

SNR, dB

Cap

acity

, bits

/s/H

z

SISO2 by 24 by 48 by 8

Figure 2-2: Comparison of channel capacity for SISO and MIMO systems, assuming flat Rayleigh fading and zero correlation between all transmission coefficients in the channel. (Plotted in MATLAB according to the model from I-METRA [6])

The channel matrix H in the MIMO systems capacity of equation (2.4) is the

mathematical representation of the physical transmission path, which includes not

only the multipath channel characteristics of the physical environment but also the

antenna configurations. Therefore the multipath channel characteristics as well as the

antenna configurations play a key role in determining the communication

performance in a MIMO system. The channel matrix H can be evaluated by

measurement or modelling on a propagation environment [7]. In this thesis, the H is

evaluated by modelling a propagation environment using a commercial software

package (Wireless Insite) [8], which is based on a ray tracing method to solve the

multipath channels in a physical environment. Ray tracing technique [9]-[11] has

emerged as the most popular technique for analysing the physical environment due

to its ability to analyse very large structures with reasonable computational resources.

In the modelling, the antenna configurations and the multipath physical environment

including the number of multipath channels, their distribution in power, angle of

departure/arrival and delay are taken into consideration.


15

2.2.2 Spatial multiplexing

Spatial multiplexing (SM) is a signalling scheme where independent data

streams are transmitted simultaneously in parallel channels from each element in an

array of antennas. The basic principle of SM is illustrated by examining a system

with two elements at the transmitter and two elements at the receiver below.

Firstly, the bit stream of data to be transmitted is demultiplexed into two sub-

streams, then modulated and transmitted simultaneously from each transmit antenna

as shown in Figure 2-3. If the propagation channels are uncorrelated, these signals

arrived at the receive antenna are well separated. Assuming that the receiver has

knowledge of the channel, it can differentiate between the co-channel signals and

extract both signals. After demodulating the received signals, the original sub-

streams can be combined to yield the original bit stream of data. Therefore, spatial

multiplexing increases the channel capacity with the number of transmit-receiver

antenna pairs [3]. This concept can be extended to more general MIMO channels.

If the number of elements at the transmitter, nT and the receiver nR are not

equal, the maximum parallel channels that can be achieved in an ideal MIMO system

is min (nT, nR). A spatial multiplexing scheme called Bell labs Layered Space –Time

(BLAST) proposed by Bells Labs was first widely publicised in 1996 [3]. It is well

known that Shannon’s classical capacity formula indicates that in the high SNR

regime, a 3dB increase in SNR will approximately increase capacity by 1 bit/s/Hz.

However, in the MIMO systems, the capacity in the high SNR regime will increase

by min(nT, nR) bits/s/Hz with every increment of 3dB in SNR.

Considerable research activities have been carried out to show that the spatial

multiplexing concept has the potential to significantly increase spectral efficiency

[12]-[13]. Further research has been carried out on creating and evaluating

enhancements to the spatial multiplexing concepts, such as combining with other

modulation schemes like OFDM (Orthogonal Frequency Division Multiplexing)

[14]-[15]. In general, this technique assumes channel knowledge at the receiver and

the performance can be further improved when the knowledge of the channel

response is available at the transmitter. However, SM does not work well in low

SNR environments as it is more difficult for the receiver to identify the uncorrelated

signal paths [16]-[17].


16

Figure 2-3: A 2x2 MIMO system with a spatial multiplexing scheme. The original message is demultiplexed into two sub-streams (Red and Blue) and transmitted simultaneously from each transmitting antennas.

2.2.3 Space-time coding

An alternative method of exploiting MIMO channels, known as space-time

coding, has also recently generated large amounts of research [18]-[19]. This method

aims to improve the system’s performance by exploiting the multiple element

antennas for diversity gain rather than for the spatial-multiplexing gain of parallel

data streams. It increases network throughput by selecting quality signal paths such

that higher data rates can be achieved and avoiding signal paths that are likely to

produce packet errors and retransmissions.

A space-time coded transmitter differs from that of a spatial multiplexing

system in that a single data stream is encoded across both time and space to produce

the symbol streams for each transmit element as shown in Figure 2-4. Appropriate

decoding at the receiver allows a diversity gain to be achieved. This method is

particularly attractive as it does not require channel knowledge in the transmitter.

The resulting diversity gain improves the reliability of fading wireless links and

hence improves the quality of the transmission.

It is noted that the space-time coding scheme does not increase the capacity

linearly with the number of transmit/receive elements used. However, it maximises

the wireless range and coverage by improving the quality of the transmission. In

order to improve both range and capacity, a MIMO implementation requires to

support both the SM and space-time coding schemes. Therefore, the combination of

SM and diversity/space-time coding has been studied recently [20]-[21].


17

Figure 2-4: A 2x2 MIMO system with a space-time coding scheme. The original message is transmitted simultaneously from all the antennas at the transmitter without sub-streaming them.

2.3 Antenna Diversity

Multipath propagation caused by scattering had historically been regarded as

an impairment because it causes signal fading. In order to mitigate this problem,

diversity techniques were developed. The basic principle of diversity is that the

receiver should have more than one version of the transmitted signal available,

where each version is received through a different channel. Figure 2-5 illustrates the

two uncorrelated Rayleigh signals and the combined signal. If the two signals are

uncorrelated, it is rare that the two multipath fading signals will be in a deep null at

the same time. It shows that the combined signal creates a higher mean SNR at the

output compared to a single branch resulting in a diversity gain.

Apart from mitigating the signal fading problem, as previously mentioned,

diversity techniques are also used to exploit the MIMO channels using a space-time

coding signalling scheme.

There are five categories of diversities, i.e. frequency diversity, time diversity,

spatial diversity, pattern diversity and polarisation diversity. Amongst the five

diversities, only the spatial, pattern and polarisation diversity techniques are

categorised as antenna diversity and they are discussed further in detail in section

2.3.5.


18

Sign

al le

vel,

dB

Time, s

Signal 1 Signal 2 Combined Signal

Figure 2-5: Diagram showing two versions (Signals 1 and2) of transmitted signals available from two different channels in a multipath environment. Diversity technique is used to combine both the different signals. The combined signal always has the highest signal level compared to the individual signals.

Antenna diversity techniques utilise more than one antenna to receive

different (or uncorrelated) multipath fading signals in a mobile communication

channel.

Figure 2-6 shows that a dual-element diversity antenna at a receiver can

receive two different versions of transmitted signals and combine them with a

combiner circuit. For antenna diversity techniques to be successful in a mobile

fading environment, the following criteria have to be met: low correlation between

the antennas and equal power at the antennas. This is because if the correlation is too

high, then deep fades will occur simultaneously at different antennas. Also, if the

antennas have low correlation but have very different mean powers, then the signal

in a weaker antenna may not be useful although it is less faded than the other

antennas.

Figure 2-6: Diagram showing two signals are combined in a basic diversity receiver.


19

2.3.1 Diversity Combining Techniques

Figure 2-7 illustrates four different types of diversity combining techniques

can be employed in the ‘combiner’ shown in Figure 2-6 [22].

Figure 2-7: Diagram showing four types of diversity combining techniques can be employed at the receive diversity.

2.3.1.1 Switched Combining

The switched combining technique requires only one receiver radio between

the N branches as shown in Figure 2-8. The receiver is switched to other branches

only when the SNR on the current branch is lower than a predefined threshold.

Whereby, other combining techniques require N receivers to monitor the received

instantaneous signals level of every branch when there are N element antennas. Due

to size restrictions, battery life and complexity, the switched combining technique is

presently implemented in mobile terminals with diversity antennas [23]. The

optimum performance that a switched combiner can achieve is similar to that of a

selection combiner.

Figure 2-8: Block diagram of switched combining for N branches/antenna elements with only one receiver.


20

2.3.1.2 Selection Combining

The selection combining technique is similar to the switched combining

technique except that N receivers are required to monitor instantaneous SNR at all

branches. The branch with the highest SNR is selected as the output signal.

Figure 2-9: Block diagram of selection combining for N branches/antenna elements.

2.3.1.3 Equal Gain Combining

Both switched and selection combining techniques only use the signal from

one of the branches as the output signal. In order to improve SNR at the output, the

signals from all branches are combined to form the output signal. However, the

signal from each branch is not in-phase. Therefore, each branch must be multiplied

by a complex phasor having a phase -θi, where θi is the phase of the channel

corresponding to branch i (i.e. co-phased) as shown in Figure 2-10. When this is

achieved, all signals will have zero phase and are combined coherently.

Figure 2-10: Block diagram of equal gain combining for N branches/antenna elements.


21

2.3.1.4 Maximum Ratio Combining

In the equal gain combining technique, all the branches may not have a

similar SNR. Sometimes one of the branches has a much lower SNR than the other

branches and this will reduce the overall SNR to a lower value at the output. In order

to maximise the SNR at the output, each branch is applied with a weight, wi before

all the signals are combined coherently as shown in Figure 2-11. In order to

maximise the SNR at the output, a branch with a higher SNR will be given a higher

weighting.

Figure 2-11: Block diagram of maximum ratio combining for N branches/antenna elements.

2.3.2 Diversity Gain

Diversity gain is defined as the improvement in the SNR of the combined

signals relative to the SNR from a single antenna element. In this thesis, the

Rayleigh channels are assumed in the multipath environment. The cumulative

distribution function (cdf) of a Rayleigh channel is given as [12], [22] :

where Γ is the mean SNR, γ is the instantaneous SNR, P(γ<γs) is the probability

that the SNR will fall below the given threshold, γs. For a selection combiner with N

independent branches, assuming that the N branches have independent signals and

equal mean SNRs, the probability of all branches having a SNR below γs is

equivalent to the probability for a single branch raised to the power N as:

( ) 1s

sP eγ

γ γ−Γ

⎛ ⎞< = −⎜ ⎟

⎝ ⎠ (2.5)


22

where N is the number of antennas/branches.

Equations (2.5) and (2.6) are plotted in Figure 2-12 to show the reduction of

the probability of fading below a given threshold when increasing the number of

antenna, N. In this figure, diversity gain is also illustrated in terms of the increase in

SNR of a combined output compared to a single antenna. Here, the diversity gain is

marked off where P(γ<γs) of 1% (i.e. 99% reliability). The figure shows that there

is a 10dB and 13dB of diversity gain for the two branches and three branches

selection combiner respectively. For low instantaneous SNR i.e. γ <<Γ , equation

(2.6) can be approximated by [24]:

Therefore, by re-arranging the equation (2.7) the diversity gain for a 100%

efficient two branches selection combiner is 10dB with P(γ<γs) at 1%. Mobile

terminals currently available in the Japan market use two antennas for diversity. A

diversity antenna array with four antennas has been developed in this thesis for the

MIMO systems.

-40 -30 -20 -10 0 1010-4

10-3

10-2

10-1

100

Pr (S

igna

l < a

bsci

ssa)

10 log(γ/Γ), dB

1 Branch 2 Branches 3 Branches 4 Branches 5 Branches 6 Branches

Diversity Gain

Figure 2-12: Cumulative distribution function of Rayleigh fading signals for a different number of diversity branches (Plot based on equation 2.6).

( ) 1s

N

s NP eγ

γ γ−Γ

⎛ ⎞< = −⎜ ⎟

⎝ ⎠ (2.6)

( )N

ss NP γγ γ ⎛ ⎞< = ⎜ ⎟Γ⎝ ⎠

(2.7)


23

2.3.3 Correlation

In order to obtain a good diversity gain, one of the conditions is low

correlation. The correlation can be described by two metrics: complex and envelope

correlations [25].

The complex correlation, ρ12, is described as “the complex correlation

between two signal envelopes” [26]. The magnitude and phase are used to calculate

correlation. In the time domain, the complex correlation is defined in closed form as

follows:

where t is the instantaneous time point and T is the time period over which the fading

signals correlate. The two fading signals v1(t) and v2(t) are voltage functions of time

and have mean values 1v and 2v in volts, V.

However, in order to evaluate the correlation between two antennas in an

angular domain, the complex correlation is computed as follows [27]:

where the variance, 2

nσ is the variance of branch n in V2:

where XPR is the ratio of time averaged vertical power to time average horizontal

power [28] in the fading environment in linear form:

*1 21 2

012

2 21 21 2

0 0

( ( ) )( ( ) )

( ) ( )

T

T T

v t v v t v dt

v t v dt v t v dt

ρ− −

=

− −

∫

∫ ∫ (2.8)

2

* *

1 2 1 2

0 012

2 2

1 2

( , ) ( , ) ( , ) ( , ) ( , ) ( , ) sinXPR E E P E E P d dπ π

θ θ θ φ φ φθ φ θ φ θ φ θ φ θ φ θ φ θ φ θ

ρσ σ

⋅ +

=∫ ∫ (2.9)

2

2 * *

0 0

( , ) ( , ) ( , ) ( , ) ( , ) ( , ) sinn n n n nXPR E E P E E P d dπ π

θ θ θ φ φ φσ θ φ θ φ θ φ θ φ θ φ θ φ θ φ θ= ⋅ +∫ ∫ (2.10)

V

H

PXPRP

=

(2.11)


24

where Pv is the average vertical power and PH is the average horizontal power.

XPR is also referred to as the cross-polarisation power ratio or cross-polar

discrimination (XPD). Both antennas in equation (2.9) have E-fields, nEθ and nEφ

[Vm-1]. ( , )Pθ θ φ and ( , )Pφ θ φ are the angular density function of the vertical and

horizontal plane respectively. For reference purposes, θ is the angle relative to the

vertical axis z and φ is the angle in the horizontal plane as shown in Figure 2-13.

Figure 2-13: Diagram showing the relation of angular coordinates to Cartesian

coordinates.

The second type of correlation metric is the envelope correlation, eρ which is

the correlation between two signal envelopes without considering the phase [26]:

where 21 ( )R t and 2

2 ( )R t are the square envelope [V2] of 1( )v t and 2 ( )v t respectively

(i.e. 21( )v t and 2

2 ( )v t ). Envelope correlation ( eρ ) is always real as the phase is not

defined. It is assumed that with independent Gaussian sources the envelope

correlation is related to complex correlation in a Rayleigh fading environment as

follows [29]:

The previous discussion on diversity gain has assumed independent signals

received on the diversity branches, i.e. there is no correlation between the signals

( ) ( )

( ) ( )

2 2 2 21 1 2 2

02 2

2 2 2 21 1 2 2

0 0

T

e T T

R t R R t R dt

R t R dt R t R dt

ρ

⎛ ⎞⎛ ⎞− −⎜ ⎟⎜ ⎟⎝ ⎠⎝ ⎠

=⎛ ⎞ ⎛ ⎞− −⎜ ⎟ ⎜ ⎟⎝ ⎠ ⎝ ⎠

∫

∫ ∫(2.12)

212eρ ρ≈ (2.13)


25

received where 0eρ = . However, it is clear that in the majority of cases this cannot

be achieved because of insufficient antenna spacing. If the correlation coefficient is

more than zero (i.e. 0eρ > ), then it will degrade the diversity gain discussed earlier.

Hence, the correlation coefficient must be kept low enough so that the diversity is

still effective. Saunders [22] has shown that eρ needs to be less than 0.7 such that

there is not more than 3dB loss in diversity gain. The effects of envelope correlation

on diversity gain can be found in [24]. The analysis shows that where the correlation

is not too close to unity or 0.7eρ ≤ , the degradation on the diversity gain due to the

envelope correlation is by the factor of the following equation:

2.3.4 Branch Power Ratio and Mean Effective Gain

The other essential condition for good diversity is that the power levels of all

the antennas in the diversity system must not be too different. One way of illustrating

this is by using the ratio of two branch power levels, k, as follows in linear form:

where minP [W] is the power from the antenna with the lower power and maxP [W] is

the power from the antenna with the higher power in each pair of antennas. The ratio

of the two antennas’ power levels, k, is multiplied by the diversity gain to obtain the

new diversity gain for a selection combiner [24]. Hence when N = 2, equation (2.7)

becomes:

Equation (2.7) is for the ideal case when k is equal to unity. An alternative

method to obtain the branch power ratio is derived from the mean effective gain

(MEG) of the antennas as follows (assuming only two branches):

Degradation factor, 1 eDF ρ= − (2.14)

min

max

PkP

=

(2.15)

2

21( ) s

sPk

γγ γ ⎛ ⎞< ≈ ⎜ ⎟Γ⎝ ⎠ (2.16)


26

The MEG is the average gain of an antenna in a mobile environment and is

defined in [30] as the ratio between the mean received power of the antenna ( recP )

and the total mean incident power ( VP + HP ). The MEG is a figure of merit for the

average performance of an antenna on a mobile terminal taking into account the

incident radio waves in the multipath environment and the gain patterns of the

antenna. This parameter determines how effective the mobile terminal antenna will

be in a multipath environment. It is important to evaluate the MEG of the antennas to

determine their diversity performance. Taga has derived the following equation to

evaluate the MEG [30]:

where ( , )Gθ θ φ and ( , )Gφ θ φ are the θ and φ components of the antenna power

gain patterns respectively. ( , )Pθ θ φ and ( , )Pφ θ φ are the θ and φ components of the

angular density functions respectively as used in equation (2.9).

Assuming the correlation is low enough to obtain good diversity, k should be

greater than -3dB to avoid a significant loss in diversity gain. In order for diversity

gain to be achievable k should also be above -10dB, otherwise the diversity system is

not effective [31].

2.3.5 Antenna Diversity Techniques

As mentioned in section 2.3, there are only three types of diversity technique

which utilise multiple antennas to achieve diversity - spatial, pattern and polarisation

diversities. At the mobile terminal, a combination of these different types of antenna

diversity techniques is often used [31].

2 1

1 2

min ,MEG MEGkMEG MEG

⎛ ⎞= ⎜ ⎟

⎝ ⎠ (2.17)

2

0 0

1( , ) ( , ) ( , ) ( , ) sin1 1

XPRMEG G P G P d dXPR XPR

π π

θ θ φ φθ φ θ φ θ φ θ φ θ φ θ⎛ ⎞= +⎜ ⎟+ +⎝ ⎠∫ ∫ (2.18)


27

2.3.5.1 Spatial Diversity

Spatial diversity utilises more than one antenna which are sufficiently

separated from each other so that the relative phases of the multipath contributions

are significantly different at the two antennas. This is the most fundamental

technique to achieve diversity. The phase differences between the total signals

received at each of the antennas are proportional to the differences in the path

lengths from the scatterers to each antenna. When large phase differences are present,

they give rise to a low correlation between the signals at the antennas. Therefore it is

expected that the correlation decreases with an increase in the distance between the

scatterers or an increase in the distance between the antennas.

By assuming angular density function to be uniform in azimuth of the mobile

environment and no angular density function in elevation (i.e. Two-dimensional

scenario), the correlation coefficient for a distance separation d can be obtained from

the zero order Bessel function, ( )oJ x [22]:

where β is the phase constant.

The first null of 0 ( )J dβ is at d=0.4λ, as shown in Figure 2-14. As shown

graphically in Figure 2-14, the correlation coefficient starts to increase after d=0.4λ.

However, in suburban areas the measurements show that the first null appears at

about d=0.8λ [32]. This may be due to a lack of uniform angular distribution of wave

arrival. It shows that the angular distribution of wave arrival does affect the

correlation coefficient for a given spacing d, whereby if the angular density function

is restricted to a limited range then 12ρ will increase [22].

Generally, spacing, d of 0.5λ is practically used to obtain two uncorrelated

signals at mobile terminals.

Apart from the effect of the angular density functions ( , )Pθ θ φ and ( , )Pφ θ φ , it

should be noted that since the two antennas are horizontally spaced with d=0.5λ,

mutual coupling also affects the performance of diversity as well. However, equation

(2.19) does not consider the mutual coupling between the antennas. It has been

shown in theory and experimentally that mutual coupling reduces the correlation

coefficient [22]. Recently it has been reported that the MIMO capacity is still

12 0 ( )J dρ β= (2.19)


28

relatively large when the four antennas are closely spaced down to d=0.2λ in the

indoor environment [33]. In the outdoor environment, the MIMO capacity for

antennas with spacing of d=0.2λ is even larger than when the antennas spacing is

d=2.5λ [34].

0.0 0.5 1.0 1.5 2.0 2.50.00.10.20.30.40.50.60.70.80.91.0

Cor

rela

tion

coef

ficie

nt, ρ

12

Antenna spacing, d (wavelength) Figure 2-14: Diagram showing the effect of antenna spacing to correlation

coefficient.

2.3.5.2 Pattern Diversity

Pattern diversity occurs in many instances at the mobile terminals because

the antennas will pick up signals coming from different angles. Since the fading

signals coming from different directions are independent then pattern diversity can

be implemented. This has been considered at the base station in some cases and

compared with spatial diversity [35]-[36].

At the mobile terminal, two omni-directional antennas interacting with each

other whilst closely spaced can also obtain pattern diversity. Basically, the antennas

act as parasitic elements to each other and their patterns change to allow signals to be

picked up at different angles. Antennas with beam steering at the mobile terminals

(by changing the feed point impedance of parasitic elements) have been developed

[37]. Recent studies conducted on pattern diversity in the MIMO systems have

shown that with appropriate dissimilarity in the antenna pattern, the system can

achieve large channel capacity [38].


29

2.3.5.3 Polarisation Diversity

Polarisation diversity can be achieved when two or more differently polarised

antennas are used as the branches of a diversity receiver or transmitter. It has been

reported by Lee and Yeh that the horizontal and vertical polarization paths between a

mobile and a base station are uncorrelated [39]. Numerous studies on polarisation

diversity have been carried out at the base stations and have been widely applied in

practice to new base stations [40]-[42]. Previously, spatial diversity has been widely

used in base stations but the size of the antenna structures is too large. With the use

of polarisation diversity the size of the antenna structures can be reduced

significantly. Early theoretical analysis has been undertaken to show that at the base

station the largest diversity gain can be obtained when the two antennas are polarised

at ± 45° where the vertical is used as a single reference[43]-[44]. Further, it has also

been shown that polarisation diversity can be integrated with spatial diversity [45].

Polarisation diversity is an attractive option to apply at the mobile terminal

due to the reduced size of the antenna structures. Hence, recent studies on the MIMO

systems have sought to exploit the MIMO channels by using polarisation diversity

[46]-[48].

2.4 Environmental factor

In mobile wireless communication systems, most of the time the transmitted

signals are affected by buildings and other obstacles causing reflections, diffraction

and scattering. In the multipath environment, the incident radio waves arriving at the

mobile terminal antennas have various angles of arrival (AOA) and XPR. The AOA

distributions (also defined as angular density functions) at both θ and φ polarisations

and the XPR have an affect on the diversity performance of the antennas. As evident

by the correlation equation (2.8) mentioned previously, the correlation coefficient is

dependent on the multipath environment via the angular density functions

( , )Pθ θ φ and ( , )Pφ θ φ . For simplicity, the angular density function are modelled in

elevation and azimuth separately and combined according to

( , ) ( ) ( )( , ) ( ) ( )

P P PP P Pθ θ θ

φ φ φ

θ φ θ φθ φ θ φ

=

= (2.20)


30

where ( )Pθ φ , ( )Pφ φ are the angular density functions in azimuth and ( )Pθ θ , ( )Pφ θ

are the angular density functions in elevation for the θ and φ polarisations

respectively.

In order to evaluate the correlation coefficient, it is necessary to apply a

suitable statistical model that is similar to the real environment. Limited

measurements have been carried out on angular density distribution at the mobile

terminal in urban environments [49]-[51] and it is only recently that indoor

environments have been considered [52].

2.4.1 Angular density functions in outdoor urban environments

A number of models have indicated that the angular density function is

uniform in azimuth where there is a non-line of sight (NLOS) case. This is because

the base station is invisible from the mobile terminals and there is an equal chance of

a local/nearby scattering object being at any angle around the mobile. However,

cases of non-uniformity have been reported in the angular density functions

measurement recently [49], especially from non-local scatterers [50] and it has been

suggested that standard urban models have a Laplacian azimuth distribution [51].

These results warrant further investigation into appropriate standard models that can

be used for well defined types of urban environments such as street canyons, open

squares and heavily cluttered areas. In this thesis, the angular density functions are

assumed to be uniform in azimuth.

The literature reviews indicate that for the elevation, it is more complicated

due to the varying height of scatterers and the base stations. Therefore this does not

generate consistent angular distribution density statistics. Further, the measurements

at different frequencies also affect angular density functions. Despite the

inconsistency, most of the literature reviews reported that the elevation angles were

spread over the angular range from 0º to 30º. However, elevation AOA of 22.5 and

67.5 are assumed in the MIMO models from the 3GPP [53]. The spread of the

elevation angles are due to the secondary wave sources, i.e. reflection points,

diffraction points and scattering points, distributed on obstacles whose height are

spread without general rule. When the antenna moves randomly in an environment,


31

numerous sources will be observed therefore it is reasonable to assume angular

density function to be a Gaussian distribution in elevation [30].

2.4.2 Angular density functions in indoor environments

As it is only recently that research has been carried out on propagation in

indoor environments, therefore limited results are available and the results do not

give a clear indication of the type of model which can be applied. Generally, these

appear to be a uniform azimuth distribution in most cases. It has been shown that a

Laplacian distribution gives a better fit than the Gaussian model in the indoor

environment [52].

2.4.3 Proposed statistical model

It is noted that different type of distribution models has been suggested by

different researchers. This is due to different locations and environments having been

studied. When a user of a mobile terminal moves along a random route, a uniform

distribution is a reasonable assumption for the angular density functions in azimuth

direction as was assumed in [30]. However, the angular density functions in the

elevation direction are not uniformly distributed and the two most common different

distributions i.e. Gaussian and Laplacian distributions are used in this thesis:

1. Gaussian distribution:

[ ] 2

2

( / 2)( ) exp

2V

V

mP Aθ θ

θ πθ

σ

⎡ ⎤− −⎢ ⎥= −⎢ ⎥⎣ ⎦

0≤ θ ≤π

[ ] 2

2

( / 2)( ) exp

2H

H

mP Aφ φ

θ πθ

σ

⎡ ⎤− −⎢ ⎥= −⎢ ⎥⎣ ⎦

0≤ θ ≤π

(2.21)


32

2. Laplacian distribution:

where mV and mH are the mean elevation angles of vertical and horizontal polarised

wave distribution respectively, Vσ and Hσ are the standard deviations of the vertical

and horizontal polarised wave distribution respectively. Aθ and Aφ are constants

determined by the following condition:

2.4.4 Cross Polarisation Power Ratio (XPR)

The last parameter that has a significant effect on pattern and polarisation

diversity is the cross polar ratio of the fading environment, XPR. Generally, the value

of XPR is reported between 4dB and 9dB at frequencies around 900MHz in urban

macrocell environment [38], [43], [44] from the mobile terminals to the base stations.

A few different environments have been studied at 2.15GHz and the XPR varied

between 6.6 and 11.4 dB, being lowest for indoor environments and highest for

urban microcell environments [54]. All these reported results have shown that the

XPR is not constant due to varying frequencies and environments.

2.4.5 Proposed propagation model

In this thesis, both outdoor and indoor environments are used to evaluate the

diversity performance of the four-element diversity antenna array proposed in

Chapter 5. Uniform distribution is assumed in the azimuth direction whilst Gaussian

and Laplacian distributions are assumed in the elevation direction. The values of

( )2 / 2( ) exp V

V

mP Aθ θ

θ πθ

σ

⎡ ⎤− −= ⎢− ⎥

⎢ ⎥⎣ ⎦ 0≤ θ ≤π

( )2 / 2( ) exp H

H

mP Aφ φ

θ πθ

σ

⎡ ⎤− −= ⎢− ⎥

⎢ ⎥⎣ ⎦ 0≤ θ ≤π

(2.22)

2 2

0 0 0 0( , ) sin ( , ) sin 1P d d P d d

π π π π

θ φθ φ θ θ φ θ φ θ θ φ= =∫ ∫ ∫ ∫ (2.23)


33

AOA and XPR for each environment are applied on both Gaussian and Laplacian

statistical models as summarised in Table 2-1 [55]-[56].

Statistic model (Elevation/Azimuth) Scenario

Gaussian/Uniform Laplacian/Uniform

Indoor

mV = 10º mH = 10º σV = 15º σH = 15º

XPR = 5dB

mV = 10º mH = 10º σV = 15º σH = 15º

XPR = 5dB

Outdoor

mV = 20º mH = 20º σV = 30º σH = 30º

XPR = 1dB

mV = 20º mH = 20º σV = 30º σH = 30º

XPR = 1dB

Table 2-1: Propagation models used in this thesis [55]-[56]. mV and mH are the mean elevation angles of vertical and horizontal polarised wave distribution respectively, σV and σH are the standard deviations of the vertical and horizontal polarised wave distribution respectively

2.5 Summary

This chapter has shown that the MIMO system can increase the channel

capacity significantly without increasing the bandwidth and transmission power

when compared to the SISO system. The two methods of exploiting the MIMO

channels: spatial multiplexing and space-time coding have also been addressed. The

principle of correlation, diversity gain and diversity techniques have also been

discussed. Since this thesis focuses on the antenna design in mobile terminals (e.g.

laptop, PDA or handset) for the MIMO system, the space available for implementing

multiple antennas has become challenging. When the antennas are placed too close

to each other, the mutual coupling reduces the isolation between the antennas. This

leads to an increase in the correlation and reduction in diversity gain. Therefore, the

diversity antenna array on a small mobile terminal needs to have a combination of

space, polarisation and angular diversity.

Mobile terminals are used in different environment and are in motion most of

the time. Therefore, the propagation environment has to be considered when


34

evaluating diversity performance. Literature reviews have shown different incident

power distributions for different environments. Hence, both indoor and outdoor

environments with different values of mean angles and standard deviations of AOA

and XPR are used in Chapter 5 to evaluate the diversity performance.

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[19] S.M. Alamouti, “A simple transmit diversity technique for wireless communications,” IEEE J. Select Area Commun., vol. 16, pp. 1451-1458, Oct. 1998.

[20] L. Zheng and D. Tse, “Diversity and multiplexing: A fundamental tradeoff in multiple-antenna channels,” IEEE Trans. on Information Theory, vol. 49, no. 3, pp. 1073-1096, May 2003.

[21] D. Tse, P. Viswanath and L. Zheng, “Diversity-multiplexing tradeoff in multiple-access channels,” IEEE Trans. on Information Theory, vol. 50, no. 9, pp. 1859-1874, September 2004.

[22] S.R. Saunders, “Antennas and propagation for wireless communication systems”, Wiley, 1999.

[23] M. Tarkiainen and T. Westman, “Predictive switched diversity for slow speed mobile terminals,” IEEE Vehicular Technology Conference, vol. 3, pp. 2042-2044, May 1997.

[24] M. Schwartz, W. R. Bennet and S. Stein, “Communication systems and techniques”, McGraw Hill, 1996.

[25] J.S. Colburn, Y. Rahmat-Samii, M.A. Jensen, G.J. Pottie, “Evaluation of personal communications dual-antenna handset diversity performance”, IEEE Trans. On Vehicular Technology, vol. 47, no. 3, pp 737-746, August 1998.

[26] F. Adachi, M. T. Feeney, A. G. Willianson, J.D. Parsons, “Cross correlation between the envelopes of 900MHz signals received at a mobile radio base station site”, IEE Proceedings Radar and Signal Processing, vol. 133, no.6 Part F, pp 506-512, Oct 1986.

[27] G.F. Pedersen, J. Bach Andersen, “Handset antennas for mobile communications, integration, diversity and performance”, URSI Review of Radio Science, pp 119-139, 1996.


36

[28] M.A. Jensen, Y, Rahmat-Samii, “Performance analysis of antennas for hand-held transceivers using FDTD”, IEEE Trans. On Antennas and Propagation, vol. 42, no. 8, pp 1106-1113, August 1994.

[29] R.H. Clarke, “A Statistical Theory of Mobile Radio Reception”, Bell System Technical Journal, pp 957-1000, 1966.

[30] T. Taga, “Analysis of mean effective gain of mobile antennas in land mobile radio environments”, IEEE Trans. On Vech. Techn., vol. 39, no.2, pp 117-131, May 1990.

[31] T.W.C. Brown “Antenna Diversity for Mobile Terminals,” PhD thesis, University of Surrey, Sept. 2002.

[32] William C.Y. Lee, “Mobile Communications Engineering: Theory and Applications”, 2nd edition, McGraw-Hill, 1988.

[33] V. Pohl, V. Jungnickel, T. Haustein, and C. von Helmolt, “Antenna spacing in MIMO indoor channels,” in Proc. IEEE Vehicular Technology Conf., Birmingham, 2002.

[34] V. Jungnukel, V. Pohl, and C. von Helmolt, “Capacity of MIMO Systems with closely spaced antennas,” IEEE Communications Letters, vol. 7, no. 8, August 2003.

[35] R.G. Vaughan, “Pattern Translation and Rotation in Uncorrelated Source distributions for Multiple Beam Antenna Design”, IEEE Trans. On Antennas and propagation, vol. 46, no. 7, pp 982-990, July 1998.

[36] P. L. Penry, C. L. Holloway, “Angle and Space Diversity Comparison in Different Mobile Radio Environments”, IEEE Trans. On Antennas and Propagation, vol. 46, no.6, June 1998.

[37] R. G. Vaughan, “Switched Parasitic Elements for Antenna Diversity”, IEEE Trans. On Antennas and Propagation, vol. 47, no. 2, pp 399-405, Feb 1999.

[38] L. Dong, H. Ling, and R.W. Heath, “Multiple-input multiple output wireless communication systems using antenna pattern diversity,” Global Telecommunications Conference, 2002. GLOBECOM '02. IEEE, vol. 1, pp. 997-1001, 17-21 Nov. 2002.

[39] W. C. Y. Lee and Y. S. Yeh, “Polarisation diversity system for mobile radio”, IEEE Trans. Commun., vol. COM-20, no. 5, pp 912-923, 1972.

[40] A. M. D. Turkmani, A. A. Arowojolu, P. A. Jefford, C. J. Kellett, “An experimental evaluation of the performance of two-branch space and polarization diversity schemes at 1800MHz”, IEEE Trans. On Vehicular Technology, vol. 44, no. 2, pp 318-326, May 1995.

[41] K. Cho, T. Hori, K. Kagoshima, “Effectiveness of four-branch height and polarization diversity configuration for street microcell”, IEEE Trans. On Antenna and Propagation, vol. 46, no. 6, pp 776-781, June 1998.

[42] C. Beckman, U. Wahlberg, “Antenna system for polarisation diversity”, Microwave Journal, pp 330-334, May 1997.

[43] R. G. Vaughan, “Polarisation diversity in mobile communications”, IEEE Trans Vech. Tech., vol. 30, no. 3, pp 177-186, August 1990.


37

[44] S. Kozono, T. Tsuruhara, M. Sakamoto, “Base station polarization diversity reception for mobile radio”, IEEE Trans Vech Tech, vol. 33, no. 4, pp 301-306, Nov 1984.

[45] P. C. F. Eggers, J. Toftgard, A. M. Opera, “Antenna systems for base station diversity in urban small and micro cells”, IEEE Journal on Selected Area in Communications, vol. 11, no. 7, pp 1046-1057, Sept 1994.

[46] J. P. Kermoal, L. Schumacher, F. Frederiksen, and P. E. Mogensen, “Polarization diversity in MIMO radio channels: Experimental validation of a stochastic model and performance assessment,” Vehicular Technology Conference, 2001. VTC 2001 Fall. IEEE VTS 54th , vol. 1 , pp. 22-26, 2001.

[47] C. Waldschmidt, C. Kuhnert, S. schulteis, and W. Wiesbeck, “Compact MIMO-arrays based on polarisation-diversity,” Antennas and Propagation Society International Symposium, 2003. IEEE , vol. 2 , pp. 499-502, June 22-27, 2003.

[48] M. J. Fakhereddin, K. R. Dandekar, “Combined effect of polarization diversity and mutual coupling on MIMO capacity,” Antennas and Propagation Society International Symposium, 2003. IEEE , vol. 2 , pp. 495-498, June 22-27, 2003.

[49] K. Kalliola, H. Laitinen, K. Sulonen, L. Vuokko, P. Vainikainen, “Directional radio channel measurements at mobile station in different radio environments at 2.15GHz”, Proceeding of the 4th European Personal Mobile Communications Conference, Vienna, Austria, no. 113, Feb 2001.

[50] K. Kuchar, J-P. Rossi, E. Bonek, “Directional macro-cell channel characterization from urban measurements”, IEEE Trans. On Ant. and Propagation, vol. 48, no. 2, pp 137-146, Feb 2000.

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[55] V. Plicanic, “Antenna Diversity Studies and Evaluation”, Master Thesis, LUND University in cooperation with Ericsson Mobile Communications AB, May 2004.

[56] Z. Ying, “Characterization of multi-channel antenna performance for mobile terminal by using near field and far field parameters,” COST 273 TD (04) (095) Goteborg, Sweden, June, 2004. (Source: Sony-Ericsson Mobile Communications AB, Lund, Sweden)

Chapter 3: Small Antennas on Mobile Terminals

38

Chapter 3 Small Antennas on Mobile Terminals

3.1 Introduction

In Chapter 2, the MIMO system and different diversity techniques have been

introduced. It was noted that the mobile environment as well as the antenna’s

characteristics play a very important role in determining the effectiveness of the

diversity system. In this chapter, the design of antennas for small mobile terminals is

reviewed. Following the review, an advanced design of the antenna is selected for

study in MIMO systems.

3.2 Small Antennas on Mobile Terminals

Recently, the mobile terminal market has been growing rapidly globally. One

of the trends in mobile terminal technology in the past few years has been to

dramatically reduce the size and weight of the terminal. This remarkable reduction in

the terminal’s size has sparked a rapid evolution of the antennas used for mobile

terminals. Hence, the design of antennas for small mobile terminals is becoming

more challenging. The antennas are required to be small and yet their performances

have to be maintained. However, usually a degradation of the gain and bandwidth

are observed when the antenna’s size is reduced.

In MIMO systems, more than one antenna will be implemented in a mobile

terminal as mentioned earlier. As a result, the design of two or more antennas on a


39

small mobile terminal for the MIMO systems is more challenging compared to the

design of a single conventional antenna in the mobile terminal. Currently, diversity

antennas are already implemented in handsets used in Japan by utilising two

antennas on a receiver [1].

3.2.1 Design Parameters for Antennas on Small Mobile Terminals

In designing antennas for small mobile terminals, the following

characteristics are usually required to be taken into consideration: small size,

compact structure, light weight, low profile and robustness.

Generally, the antennas used for small mobile terminals are the monopole,

dipole, normal-mode helix (NMHA), planar inverted-F (PIFA), microstrip patch and

meander line antennas [1]-[5]. Most of the handsets in the PDC (Personal Digital

Cellular) system in Japan utilise a monopole as the main element, along with the

normal-mode helix and PIFA as shown in Figure 3-1. The PIFA is used as element

pairs with the monopole to form a diversity antenna. In the past few years, the GSM

(Global System for Mobile Communications) mobile terminals industry prefers to

use built-in (or internal) antennas, instead of using a monopole that sticks out.

Therefore, PIFA has been widely used as a built-in (internal) antenna. Recently, it

has been variously modified from the basic principle to obtain dual or triple

frequency bands and wider bandwidths [6]. The use of a chip antenna, which is a

very small, ceramic encapsulated antenna, has also recently become very popular

[7]-[8].

3.3 Review of Small Antennas on Mobile Terminals

In the past few years, several reviews of antennas for mobile terminals have

been released [3], [9]. The important issues that have arisen from these reviews are

summarised in this section.

In the earliest mobile communication systems (dating back to 1984), the

typical portable cellular phone was nearly 600cc in volume and approximately 850g

in weight. The antenna used for the first cellular phone terminal was a half-

wavelength monopole antenna. After many years of evolution, the volume for the


40

cellular handset had been reduced to less than 60cc and a weight of less than 60g by

1999. Therefore, built-in antennas are more preferable than the half-wavelength

monopole antenna.

Figure 3-1: Photograph of a handset’s diversity antenna used in Japan for the PDC system. (Handset model: P501i by NTT DoCoMo)

3.3.1 Monopole

The quarter-wavelength monopole antenna is the fundamental mobile antenna

and has the simplest structure as shown in Figure 3-2. However, Fujimoto and

Hirasawa showed that a quarter-wavelength monopole antenna caused large leakage

currents to the terminal case compared to the half-wavelength monopole antenna [2].

For a half-wavelength monopole, the maximum current amplitude occurs around the

center of the monopole therefore current amplitude around the feed point (between

the monopole and the terminal case) is small. However, for a quarter-wavelength

monopole the maximum current amplitude occurs around the feed point and large

current flows into the terminal case. Due to the leakage currents, the length of the

terminal case significantly changes the radiation characteristics of an antenna. In

practice, the input impedance of the quarter-wavelength monopole becomes very

high and it becomes difficult to match the input impedance with that of a feeding

cable. Instead, the 3/8 or 5/8 wavelengths monopole antennas have been employed


41

for mobile terminals as they have the appropriate input impedance for matching to

the feeding line and yet the current flow on the terminal case is very small. This

antenna is also named as the “whip” antenna.

Figure 3-2: 1/4-wavelength monopole antenna: (a) practical structure, and (b) antenna image and current distribution.

3.3.2 Normal Mode Helical Antenna (NMHA)

The monopole could be shortened by a distributed inductive loading which is

made of a spiral enclosed in plastic or rubber having a total length of 4% to 15% of a

wavelength. On communication radios, this antenna is known as a “rubber duck” [1]

but conventionally it is known as an NMHA. The commercial NMHA is shown in

Figure 3-3. Electrically this antenna is quarter-wavelength (λ/4) long, which is tuned

to shorter length by the distributed inductive loading (helically wound wire).

Usually an NMHA is connected to the center conductor of a coaxial cable at

the feed point with the outer conductor of the coaxial cable attached to the ground

plane (i.e. mobile terminal). Figure 3-4 shows the configuration of an NMHA

consists of N turns, diameter D and spacing S between each turn. The total length of

an NMHA is L = NS while the total length of the wire, Le is


42

where Lo is the length of the wire between each turn and C = πD is the

circumference of the NMHA. The pitch angle, α (i.e. the angle formed by a line

tangent to the NMHA wired and a plane perpendicular to the NMHA axis) is defined

by [10]:

Figure 3-3: NMHA used in a GSM mobile handset. (Reproduced from [1])

Figure 3-4: Geometrical configuration of a helix.

2 2e oL NL N S C= = + (3.1)

1 1tan tanS SD C

απ

− −⎛ ⎞ ⎛ ⎞= =⎜ ⎟ ⎜ ⎟⎝ ⎠ ⎝ ⎠

(3.2)


43

3.3.3 Meander line antenna

An alternative way of shortening a monopole is by using a printed meander

pattern instead of a helical wire in the NMHA. Meander line antenna is also a

physically small but electrically large antenna. For example, the meander pattern can

be printed on a small piece of flexible board which is rolled on a core like an NMHA

as shown in Figure 3-5. Recently, a planar and compact meander line antenna has

been reported [11]-[12]. Multi-band characteristics can be accomplished by

connecting two or more λ/4 meanders in parallel with each being tuned to its own

frequency as shown in Figure 3-6. Apart from the meander pattern, fractal patterns

can also achieve multi-band characteristics.

Figure 3-5: Meander printer antenna on a core. (Reproduced from [1])

Figure 3-6: Dual-band meander line antenna.


44

3.3.4 Inverted-L Antenna (ILA) and Inverted-F Antenna (IFA)

Most of the new mobile terminals (e.g. GSM phones) have built-in antennas

which are not extruded from the terminal’s exterior. Generally, the very top of the

phone is not the best position to place the antenna because the antenna element

should be kept away from the user to avoid unnecessary losses. Also, the antenna

should not be placed too low on the back of the phone either as such a position will

increase the antenna’s losses due to the users’ hand. Therefore, the surface on the

upper back of the phone is a preferable position to place the built-in antenna. The

well known built-in antennas i.e. Inverted F-antenna (IFA) and PIFA have been

widely used in most of current mobile terminals.

The IFA is originally transformed from an inverted-L antenna (ILA) which

consists of a short monopole as a vertical element and a wire horizontal element

attached at the end of the monopole as shown in Figure 3-7. The ILA is a low

profile antenna as the height of the vertical element is usually much less than a

wavelength. The horizontal element normally has a length of about a quarter

wavelengths.

The ILA has low input impedance as its input impedance is equal to that of

the short monopole plus the reactance of the horizontal element closely placed to the

ground plane. In order to increase the input impedance, another inverted-L shaped

element are attached at the end of the vertical element; therefore the ILA is modified

to the IFA as shown in Figure 3-8. Impedance matching of this antenna can be

obtained by allocating the position of the feeding point without any additional circuit

requirements.

Figure 3-7: Inverted-L antenna (ILA).


45

Figure 3-8: Wire inverted-F antenna (IFA).

However, one downside of an ILA or IFA consisting of thin wires is the

narrow impedance bandwidth. Therefore, by replacing the wire-type IFA with a

planar element will widen the bandwidth. This ILA with a planar element shown in

Figure 3-9 is termed the PIFA. The fundamental characteristics of the PIFA have

been analysed and published in [13]. The shorting pin is positioned at the corner of

the planar element to yield a maximum reduction in the antenna’s size. The narrower

the shorting plate width, W, the lower the resonant frequency of the PIFA. The

resonant frequency (f) of the PIFA can be determined from the equation below [13]:

where L1 is the width of PIFA, L2 is the length of PIFA and c is the speed of light in

free space.

Figure 3-9: Planar inverted-F antenna (PIFA).

Further research on the PIFA has been performed to determine if it achieves

wideband characteristics. In fact, an added element near to its feed point and a

parasitic element at the open end of the antenna element have been proposed [14].

1 2/ 4( )f c L L= + (3.3)


46

Parasitic elements are widely applied in the design of antennas to obtain wideband

characteristics [15]. The PIFA has also been studied to operate at two or three

frequency bands for the GSM/DCS/DECT/PCS/WLAN systems [16]-[18]. Figure

3-10 shows that by cutting slot on the planar element of a PIFA, the antenna can

work as a multi-band PIFA. The basic principle is that the longer arm resonates at

low band whilst shorter arm resonates at high band. Multi-band PIFAs have been

widely used as built-in antennas by most of the mobile handset manufacturers. For

example, the popular handset Sony Ericsson T68i shown in Figure 3-11 has used a

multi-band PIFA to operate at GSM 900/1800/1900 [19]-[20].

Figure 3-10: Top view of a dual-band PIFA with an L-shape slot on the planar element. (Reproduced from [16])

Figure 3-11: Photograph of multi-band PIFA in a Sony Ericsson T68i handset. (Reproduced from [19])

When the mobile terminals become smaller, the size of the PIFA can be

reduced either by a capacitive loading [21] or by adding inductance (a meandered

pattern as mentioned earlier) [22]. An alternative way to reduce the size of the

antenna is to design the matching circuit to match the impedance of a small sized


47

antenna. However, the matching circuit is not an attractive method to reduce the

antenna size in a mobile terminal due to its complication.

The effect of the ground plane on the performance of the antennas has always

been an issue when designing a small antenna. A PIFA on a ground plane can

achieve bandwidth of up to 17% whereas a PIFA not on a ground plane (i.e. in free

space condition) can only achieve a bandwidth of between 1% to 2% [3]. Due to the

currents on the ground plane, multi-modes are introduced into an antenna system.

Hence, variation in the antenna’s performance (such as gain and bandwidth) can be

expected. However, the antenna’s performance will be degraded by the currents on

the ground plane due to the effect of adjacent materials. More than 6dB gain

degradation has been observed when the handset is in the talk position, mainly due to

the effect of the user’s hand and head (which absorb radiation power). By increasing

the current distribution on the ground plane inside the handsets, the radiation towards

the user’s head may increase hence it is possible to increase the SAR (Specific

Absorption Rate) values in the user’s head [23]-[24]. In summary, the major reason

for the degradation of the antenna’s performance in the talk position is the current

flowing on the ground plane which couples to the user’s hand and head.

3.4 New Concept for Mobile Terminal Antenna Design

So far, a few important mobile terminal antennas have been reviewed and it is

found that the built-in PIFA is the most popular at the present time. It is noted that

the current on the mobile terminal’s ground plane is produced by the excitation of

the built-in PIFA. This is because it is fed with unbalanced feed lines such as coaxial

transmission lines. Therefore, it is realised that the degradation of the antenna’s

performance due to the variation of the surface current coupled to the user can be

avoided by decreasing the current flow on the ground plane.

In the MIMO systems, two or more antennas are required to be implemented

on a mobile terminal. Due to the current on the ground plane, not only does the

variation of the surface currents coupled to the user degrade the antenna’s

performance, different antennas on the same ground plane may also degrade the

antenna’s performance when they are placed close to one another. Therefore, the


48

antenna suitable for MIMO systems needs to be an alternative type of antenna that

does not generate high current on the ground plane.

It has been reported in [25] that the half-wavelength folded dipole is a self-

balanced antenna. A folded dipole consists of two parallel dipoles connected at the

ends forming a loop as shown in Figure 3-12, with d being much shorter than L or

much shorter than a wavelength. When a half-wavelength dipole is fed with an

unbalanced feed line as shown in Figure 3-13(a), the currents on the arm connected

to the outer conductor are not equal to the currents on the arm connected to the inner

conductor. The unequal currents give rise to undesirable radiation from the feed lines

and distort the desired antenna pattern. However, for a half-wavelength folded dipole

with an unbalanced feed line as shown in Figure 3-13(b), the currents on the top wire

and the bottom wire are unbalanced but the sum of the currents on the arms is well

balanced [25]. Hence, no currents are introduced on the coaxial feed lines for a half-

wavelength folded dipole. Without the current leakage flowing on the coaxial cable,

there would be no surface current on the ground plane when the antenna is placed on

top of a ground plane. As a result, the half-wavelength folded dipole should be a

suitable antenna where there are multiple elements on a mobile terminal.

Figure 3-12: The folded dipole antenna.


49

Figure 3-13: (a) A half-wavelength dipole with a coaxial feed line; (b) a half-wavelength folded dipole with a coaxial feed line.

However, a half-wavelength folded dipole is longer than a PIFA, which is

only about a quarter-wavelength long. Further, the size of the mobile terminals is

becoming smaller as technology advances, hence the half-wavelength folded dipole

will be too long for multiple antennas in a mobile terminal.

Recently, a one-wavelength folded loop antenna which is based on the

principle of the half-wavelength folded dipole has been proposed for handset

applications [26]. The antenna is constructed by folding the arms of the half-

wavelength folded dipole as shown in Figure 3-14. As the half-wavelength folded

dipole can be considered as a one-wavelength loop antenna, the antenna is referred to

as a folded loop. It has been shown in [27] that when a folded loop antenna with

unbalanced feed lines is mounted on a ground plane as shown in Figure 3-15, there is

no unbalanced current flow on the feed line or the ground plane which is similar to

that of the folded dipole. Figure 3-16 shows the comparison of the surface current

distribution on a ground plane for a folded loop antenna and a conventional PIFA

operating at 1.8GHz. It is shown that for the folded loop antenna case, the surface

current is localised underneath the antenna and it is not widely spread across the

ground plane like the PIFA. Therefore, the ground plane of the folded loop antenna

is acting as a reflector rather than a radiator.


50

It is clear that a folded loop antenna is shorter than a folded dipole which

makes it more practical to use in a small mobile terminal and it has also inherited the

self-balanced properties of the folded dipole. Therefore, the folded loop antenna is

studied in this thesis and is developed into a diversity antenna array for the MIMO

systems in Chapter 5.

Figure 3-14: Structure of a folded loop antenna constructed from a folded dipole where L ≈ ½ λ.

Figure 3-15: Folded loop antenna mounted on a ground plane.

PIFA

Ground plane

(a) (b) (c)

Figure 3-16: Surface current distribution on the ground plane of (a) a folded loop antenna and (b) PIFA at 1.8GHz. The simulated results are obtained from CST Microwave Studio®. The ground plane sizes are 40mm x 100mm. (c) The geometry of the PIFA is shown for clarity.


51

3.5 Summary

The design parameters for small antennas have been addressed in this chapter

including the typical antennas used in the mobile terminals since the emergence of

wireless mobile communications. These antennas include: monopole, dipole, normal

mode helix antenna (NMHA), printed meander antenna and planar inverted-F

antenna (PIFA). The built-in PIFA, being the most popular antenna is still currently

used in most handsets. Diversity antennas with two elements are already widely used

in the PDC system in Japan.

It is noted that the excitation on the PIFA has generated unbalanced current

flow on the feed line and the ground plane. Therefore, the effect of the ground plane

on the performance of the antennas has been considered. There are both advantages

and disadvantages in utilising the currents on the ground plane. An advantage in

utilising the currents on the ground plane is that the antenna’s performance is

enhanced. However, a disadvantage in utilising the currents on the ground plane is

the possible degradation of the antenna’s performance due to the effect of the human

body.

A new type of antenna (i.e. folded loop antenna) has been studied and

considered in this thesis. It has been reported that this antenna can eliminate the

unbalanced current flow on the feed line and ground plane with the conventional

coaxial cable feeding technique (unbalanced system). Due to this characteristic, the

folded loop antenna is very promising for use in designing multiple antennas on a

small mobile terminal as low correlation between the antennas could still be

achieved.

References [1] K. Fujimoto and J.R. James, Mobile antenna systems handbook, 2nd edition,

Artech House Inc., 2001.

[2] K. Fujimoto, A. Henderson, K. Hirasawa, and J. R. James, Small Antennas, Research Studies Press, 1987.

[3] H. Morishita, Y. Kim, and K. Fujimoto, “Design concept of antennas for small mobile terminals and the future perspective,” IEEE Antennas and Propagation Magazine, vol. 44, no. 5, Oct 2002.


52

[4] T. Taga and K. Tsunekawa, “Performance analysis of a built-in planar inverted F antenna for 800 MHz band portable radio units,” IEEE Journal on selected areas in communications, vol. 5, pp. 921-929, March 1987.

[5] J. R. James and P. S. Hall, Handbook of microstrip antennas, London, Peter Pereginus Ltd, 1989.

[6] M. Zheng and H. J. Shapter, “Low-profile WCDMA internal antenna,” 31st European Microwave Conference, London, 2001.

[7] D.S. Yim and S.O. Park, “Small internal ceramic chip antenna for IMT-2000 handsets,” Electronic Letters, vol. 39, no. 19, pp. 1364-1365, Sept. 2003.

[8] K.L. Wong and C.H. Chang, “WLAN chip antenna mountable above the system ground plane of a mobile device,” IEEE Trans. on Antennas and Propagation , vol. 53, no. 11, pp. 3496-3499, Nov 2005.

[9] S. Sekine, H. Shoki and H. Morishita, “Antennas for wireless terminals,” IEICE Trans. Commun., vol. E86-B, no. 3, March 2003.

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[11] K.L. Wong, G.Y. Lee and T.W. Chiou, “A low profile planar monopole antenna for multiband operation of mobile handsets,” IEEE Trans. on Antennas and Propagation, vol. 51, pp. 121-125, Jan 2003.

[12] D.U. Sim and S.O. Park, “A triple-band internal antenna: Design and performance in presence of the handset case, battery and human head,” IEEE Trans. on Electromagnetic Compatibility, vol. 47, no. 3, August 2005.

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[15] A. F. Muscat and C. G. Parini, “Novel compact handset antenna,” Antennas and Propagation, 2001. Eleventh International Conference on (IEE Conf. Publ. No. 480), vol. 1, pp. 336-339, April 2001.

[16] Z.D. Liu, P.S. Hall, D. Wake, “Dual-frequency planar inverted-F antenna,” IEEE Trans. On Antennas and Propagation, vol. 45, no. 10, pp. 1451-1458, Oct 1997.

[17] P. Song, P. S. Hall, H. Ghafouri-Shiraz, and D. Wake, “Triple-band planar inverted F antenna,” Antennas and Propagation Society, IEEE International Symposium 1999 , vol. 2 , pp. 908-911, 1999.

[18] W. S. Chen, T. W. Chiou and K. L. Wong, “Compact PIFA for GSM/DCS/PCS triple-band mobile phones,” Antennas and Propagation Society, IEEE International Symposium, 2002. , vol. 4, pp. 528-531, June 2002.

[19] C.D. Nallo, A. Faraone, M. Maddaleno and T. Galia, “Principles and applications of the folded inverted conformal antenna (FICA) technology,” Converge 2005, Milan, Italy, Nov 2005.


53

[20] Z. Yong and J. Andersson, “Multi band, multi antenna system for modern mobile terminal,” 6th International Symposium on Antennas, Propagation and EM Theory Proceedings (ISAPE’03), pp. 287-290, Beijing, China, Nov 2003.

[21] P. Ciais, R. Staraj, G. Kossiavas and C. Luxey, “Compact internal multiband antenna for mobile phone and WLAN standards,” Electronic Letters, vol. 40, no. 15, pp. 920-921, July 2004.

[22] S. Shulteis, C. Waldschmidt, C. Kuhnert and W. Wiesbeck, “Design of a miniaturized dual band planar inverted F antenna,” IEEE Antenna and Propagation Society International Symposium 2004, vol. 3, pp. 3123-3126, June 2004.

[23] J. Tofgard, S. N. Hornsleth and J. B. Andersen, “Effects on portable antenna of the antenna of the presence of a person,” IEEE Trans. Antennas and Propag., vol. 46, no. 6, pp. 739-746, 1993.

[24] M. A. Jensen and Y. Rahmat-Samii, “EM interaction of handset antennas and a human in personal communications,” Proceedings of the IEEE, vol. 83, no. 1, pp. 7-17, 1995.

[25] C.G. Buxton, W.L. Stutzman, R.R. Nealy and A.M. Orndorff, “The folded dipole: A self-balancing antenna”, Microwave and optical technology letters, vol. 29, no. 3, pp. 155-160, 2001.

[26] H. Morishita, Y. Kim, Y. Koyanagi and K. Fujimoto, “A folded loop antenna system for handsets,” IEEE AP-S Proc., vol. 3, pp. 440-443, Jul. 2001.

[27] Y. Kim, H. Morishita, Y. Koyanagi and K. Fujimoto, “A folded loop antenna system for handset developed and based on the advanced design concept,” IEICE Trans. Commun., vol. E84-B, no. 9, Sept. 2001.

Chapter 4: Electromagnetic Bandgap (EBG) Structures

54

Chapter 4 Electromagnetic Bandgap (EBG) Structures

4.1 Introduction

At the commencement of this thesis, the application of EBG material to the

antenna’s design was considered. As previously mentioned in Chapter 2, the

antennas in a diversity system cannot be placed too closely to one another. This is to

avoid high correlation between the antennas so diversity gain can be achieved.

Hence, the design of diversity antenna arrays for the MIMO systems has become

more complicated when more than two antennas are to be implemented on a mobile

terminal. One of the main reasons that the closely spaced antennas are highly

correlated is due to the surface currents on the ground plane as the antennas are

sharing the same ground plane.

Upon review of initial literature studies, EBG structure was considered for this

thesis due to its ability to suppress the surface currents on the ground plane. A brief

outline of EBG is detailed below. Further, the EBG structure that has been developed

in this thesis is discussed in this chapter.

4.2 EBG Background

Electromagnetic band-gap (EBG) formerly known as Photonic Band-gap (PBG)

structures are artificially made structures with periodicity either in two or three

dimensions. Electromagnetic wave propagation in periodic media was first studied

by Lord Rayleigh in 1887 and has long been investigated by the microwave


55

community since then. Recently new concepts and ideas on PBG have been

developed in the optical regime [1], [2] whereby they show the ranges of frequencies

in which light cannot propagate through the structure. The frequency region where

the incident waves cannot propagate through the structure is termed the ‘band-gap’

or stopband. As shown in Figure 4-1 when the wavelength is in the stopband region,

there is no transmission through the material. However, if the wavelength is in the

passband (i.e. outside stopband) region, the energy will propagate through the

material.

Figure 4-1: For frequency in the stopband, the incident wave does not travel through the media.

This idea has attracted microwave engineers to use it in microwave and

millimeter-wave regions to prevent the propagation of the electromagnetic waves in

a specified frequencies band. Extensive research has been conducted to translate and

apply these new concepts in the microwave and millimeter-wave regions [3], [4].

Hence, the formerly named PBG structure has been renamed as EBG within the

microwave engineering society. In this thesis, only the application of the EBG is be

considered and analysed. The term EBG is used for the remaining of this thesis

rather than PBG.

Manufacturing of EBG structures at microwave frequencies was initially

realised by scaling the structures used at optical frequencies. This implies

micromachining holes into a dielectric slab to create a two-dimensional and three-

dimensional periodic variation of the material refractive dielectric. Lately, some

popular EBG structure designs emerged which employ simpler and smaller

geometries.


56

4.2.1 Metallo-dielectric structures The metallo-dielectric structure is considered as a new type of metallic

electromagnetic structure and is characterised by having a high surface impedance

[5]. As shown in Figure 4-2, the high-impedance surface consists of an array of

metal protrusions on a flat metal sheet. They are arranged in a two-dimensional

lattice and can be visualised as mushrooms or thumbtacks protruding from the

surface. The protrusions are formed as metal patches on the top surface of the board,

connected to the solid lower conducting surface by metal plated vias.

Figure 4-2: (a) Cross section of a high impedance surface. (b) Top view of the high-impedance surface, showing a triangular lattice of hexagonal metal plates.

The electromagnetic properties of this structure can be described by using

lumped-circuit elements – capacitors, C, and inductors, L, when they are small

compared to the operating wavelength. They simply behave as a network of parallel

resonant LC circuits, which act as a two-dimensional electric filter to block the flow

of currents along the sheet. When the structure interacts with electromagnetic waves,

currents are induced in the top metal plates as shown in Figure 4-3 below. A voltage

applied parallel to the top surface causes charges to build up on the ends of the plates,

which can be described as a capacitance. As the charges slosh back and forth, they

flow around a long path through the vias and bottom plate. A magnetic field

associated with these currents results and hence can be described as an inductance.

Figure 4-3: Capacitance and inductance in the high-impedance surface.


57

Figure 4-4: Effective circuit used to model the surface impedance

The surface impedance is high close to the resonant frequency οω which can

be determined from the L and C as follow [5]:

The sheet impedance equals to the impedance of a parallel circuit, consisting of

the sheet capacitance and the sheet inductance [5]:

Recently this high impedance surface has been used in designing antennas for

mobile terminals [6], [7]. However, this has not been of interest to the mobile phone

manufacturers because the structures are complicated and are too costly to produce.

Hence, other simpler methods is discussed in the following section 4.2.2.

4.2.2 Uniplanar Compact PBG (UC-PBG) Structure The metallo-dielectric structure is very effective but requires a non-planar

fabrication process. Hence, research efforts have focused on the development of a

planar EBG structure that does not require metal vias. Also, planar structure can be

easily integrated in microwave and millimeter-wave circuits. One of the planar EBG

structures that has been studied is called the Uniplanar Compact PBG (UC-PBG) [8].

This structure comprises of a square lattice of square metallic pads, each one

connected to the four adjacent ones through a narrow strip to form a distributed LC

network. Figure 4-5 shows the EBG structure and its single unit cell.

The narrow strips at the connections from every single cell to its neighbour

introduce lumped inductive elements, while the gaps between neighbouring pads

introduce lumped capacitors. The effect of this two-dimensional periodic LC-

1o LC

ω = (4.1)

21j LZ

LCωω

=−

(4.2)


58

network is that of reducing the wavelength of an electromagnetic wave propagating

along the structure. The inductance and capacitance values can be calculated from

the known formulas for microstrip lines [9]. Compared to the previous structure (i.e.

metallo-dielectric structure) discussed in section 4.2.1, this structure does not require

metal vias or multilayer substrates. Given this advantage, extensive studies have

been conducted on the use of this structure in microwave circuits [8], [10], and in the

application of antennas [11]-[13].

(a)

(b)

Figure 4-5: Schematic of (a) UC-PBG structure etched on the ground plane of a microstrip line, and (b) unit cell of UC-PBG structure.

4.2.3 Square lattice of circles etched on the ground plane A simpler planar EBG structure comprising of a square lattice of holes etched

on the ground plane of a grounded dielectric slab has also been developed [14]. The

transmission characteristics of the structure is assessed by constructing a 50Ω

microstrip line with EBG structure being etched on the ground plane, as shown in

Figure 4-6.


59

Figure 4-6: Schematic diagram of a microstrip on an EBG ground plane. The EBG structure is a square lattice of etched circles.

The function of the holes is to create a spatially periodic variation of the

effective dielectric constant of the line, which produces the desired stopband. The

central frequency of the stopband can be estimated by the period of the latttice, a

using the formula below [15]:

where c is the speed of light in free space, εeff is the effective permittivity and λg is

the guided wavelength in the substrate. In order to achieve a stopband, λg must

satisfy the Bragg condition where λg = 2a. The effective permittivity (εeff) can be

estimated as that of the microstrip line with an unperturbed ground plane [15]:

where h and W are the thickness of the substrate and the width of the line

respectively and rε is the relative permittivity of the dielectric substrate as shown in

Figure 4-6. Recently, considerable research effort has been conducted in the use of

this structure in the areas of filter [14]-[17] and antenna applications [18]-[21].

1.geff

cfλε

= (4.3)

1 1 12 2 1 12 /

r reff h W

ε εε + −= + ⋅+

(4.4)


60

4.3 Multiperiod EBG structure on the ground plane

Of the three popular EBG structures discussed above, the structure of square

lattice of circles etched on the ground plane has been selected in this thesis for

further study given that it is the simplest and most effective way to etch the periodic

pattern on the ground plane of the microstrip line.

Most of the research on EBG structure with a square lattice of etched circles on

the ground plane has only a single period design [14]-[17]. Recently the study of

multiperiod EBG structures has been of interest [22]. In this thesis, a simple and

novel multiperiod EBG structure is proposed to achieve a wide stopband as well as a

reduction in the size of the structure. The proposed multiperiod EBG structure differs

from the multiperiod structures investigated by other researchers. In the proposed

multiperiod EBG structure, two different periods a1 and a2 are cascaded in series,

without changing the dimension of the unit lattice as shown in Figure 4-7.

Figure 4-7: Schematic representation of the proposed multiperiod EBG structure by cascading two single period EBG structures.

Throughout this thesis, the EBG structures are modelled and simulated using

the CST Microwave Studio® package before they are machined in the laboratory. It

has been shown that the CST Microwave Studio® package can analyse the EBG

structure efficiently and accurately with excellent agreement to the data published in

[23], [24].

The substrate used in this study is RT/Duroid 6010 with relative permittivity of

10.2 and a thickness of 0.635mm (25 mils). The periodic pattern is etched on the

ground plane of the structure by using the LPK ProtoMat 91s numerical milling


61

machine whilst the S-parameters of the circuits are measured by using a HP 8720ES

network analyser in the Queen Mary University of London (QMUL) Antenna

Laboratory.

4.3.1 1-D multiperiod EBG structure Before commencing the design of the multiperiod EBG structure, two different

single period EBG structures with periods of a1=12mm and a2=7.5mm respectively

are studied. The unit lattice used in the EBG structure is a circle with a radius, r of

3mm as shown in Figure 4-8. The 50Ω conventional microstrip line on the EBG

structure consists of a printed conductor line with a width of 0.6 mm printed on the

RT/Duroid 6010 substrate.

(a)

(b)

Figure 4-8: Photographs of the ground plane of the EBG circuits with periods of (a) a1 =7.5mm, structure’s dimensions = 104mm x 15mm and (b) a2 = 12mm, structure’s dimensions = 62.5mm x 15mm. The radius of each lattice circle, r=3mm.

From equation (4.3), the predicted central frequencies of the first stopband are

5GHz and 8GHz for the EBG structures with periods of a1=12mm and a2=7.5mm

respectively. The EBG structures were also fabricated and measured. Figure 4-9

shows that there is good agreement between the simulated and measured S-

parameters.

For the EBG structure with the period a1=12mm and the radius of circle r

being 3mm, the filling factor (r/a1) is 0.25. Figure 4-9(a) shows that there are two

narrow stopbands with the central frequencies being approximately 5GHz and

10.5GHz, respectively. The central frequency of the first stopband agrees very well

with that obtained using the equation (4.3). It is noticed that there is a passband at

8GHz between the two narrow stopbands. The 10dB bandwidth of the first stopband


62

Δfa is about 2.8GHz for both the simulated and measured S21 curves, with a slight

shifting (less than 0.1GHz); the 10dB bandwidth of the second stopband Δfb is about

2.75GHz from the measurement, and 2.65GHz from the simulation with a more

pronounced shifting (around 0.2 GHz).

0 2 4 6 8 10 12 14-70

-60

-50

-40

-30

-20

-10

0

S21,

dB

Frequency, GHz

Simulation Measurement

(a)

0 2 4 6 8 10 12 14-70

-60

-50

-40

-30

-20

-10

0

S21,

dB

Frequency, GHz


(b)

Figure 4-9: Simulated and measured S21 parameters for EBG circuits with period of (a) a1 =12mm, and (b) a2 = 7.5mm.

In the second single period EBG structure, the period was reduced to

a2=7.5mm whilst maintaining the radius of the unit lattice circle (filling factor, r/a2

=0.4). The central frequency of the stopband for the second structure is found to be


63

around 8.5GHz with 10dB bandwidth being much wider, Δf =5.3GHz for both

measurement and simulation, as shown in Figure 4-9(b). The stopband bandwidth is

determined by the filling factor (r/a) given that the larger the filling factor (r/a), the

wider the bandwidth of the stopband can be achieved [14]. The slight shifting

between the simulated and measured S21 curves in Figure 4-9 is due to the

imperfection of the unit lattice in fabrication.

It is found that the stopband of the second EBG structure is just within the

passband region of the first EBG structure. Therefore, by cascading these two

periods in series one can actually combine the stopbands of both EBG structures to

achieve a wider stopband.

Figure 4-10: Schematic representation of 2 models of the 1-D multiperiod EBG structure with 8 lattice cells. Unit lattice (circle) radius=3mm, a1=7.5mm, a2=12mm. (a) Model A and (b) Model B.

Two different possible arrangements in a multiperiod EBG structure, as shown

in Figure 4-10, have been simulated. Figure 4-11 shows that the simulated S21 of the

two different arrangements and only Model B in Figure 4-10 can achieve a wide

stopband. It is noted that a wide stopband cannot be achieved in Model A as there

are only two consecutive circles for each period of a1 and a2. This demonstrates that

in a multiperiod EBG structure, the arrangement of the lattice circles is also a factor

in achieving a wide stopband. Therefore, the EBG structure of Model B was

fabricated (as shown in Figure 4-12) and measured in the QMUL Antenna

Laboratory.


64

0 2 4 6 8 10 12 14-70

-60

-50

-40

-30

-20

-10

0

S21,

dB

Frequency, GHz

Simulation

(a)

0 2 4 6 8 10 12 14-70

-60

-50

-40

-30

-20

-10

0

S21,

dB

Frequency, GHz

Simulation

(b)

Figure 4-11: Simulated S21 curves for (a) Model A and (b) Model B in Figure 4-10.

Figure 4-12: Photograph of the ground plane of the proposed 1-D multiperiod EBG structure. The dimensions of this structure are 86mm x 15mm.


65

0 2 4 6 8 10 12 14-70

-60

-50

-40

-30

-20

-10

0

S21,

dB

Frequency, GHz


Figure 4-13: Simulated and measured S21 curves for the proposed 1-D multiperiod EBG.

Figure 4-13 shows that there is a good agreement between the simulated and

measured S21 results for the proposed 1-D multiperiod EBG structure. The S21 curves

in Figure 4-13 demonstrate that the proposed multiperiod EBG structure has

achieved a very wide stopband with a bandwidth of Δf = 8GHz for S21 ≤ -10dB and a

bandwidth of Δf = 6GHz (5.5GHz – 11.5GHz) for S21 ≤ -20dB. The stopband

bandwidth of the multiperiod EBG structure increased by approximately 34%

compared to the single period EBG structure with period a1 = 7.5mm, with both

structures having approximately the same central frequency of 8GHz. Such a wide

stopband was achieved due to the combination of the stopbands of the two single

period EBG structures in Figure 4-8.

However, it is noted that the stopband attenuation of the single period EBG

structures is less than -30dB in most of the stopband region as shown in Figure 4-9,

whilst the attenuation of the multiperiod EBG structure is shallower (~ -20dB) in

some parts of the stopband region. This is because only eight lattice circles were split

between two periods in the structure. The number of lattice circles for each period in

the multiperiod EBG structure is less than that required in a single period EBG

structure (eight circles) to generate a deep attenuation in the stopband.

4.3.2 Folded 1-D multiperiod EBG structure The 1-D multiperiod EBG structure is further studied by increasing the number

of lattice circles to obtain a deeper attenuation in the stopband. However, if the


66

number of lattice circles is doubled in a 1-D direction, the size of the structure will

be doubled. Given that size is an important factor, this design type will not be very

attractive in any microwave application. Hence, two rows of multiperiod lattice

circles from Figure 4-10(b) are placed in parallel with a separation of a1, and the

transmission line is folded back on itself as shown in Figure 4-14. Folding the

microstrip line rather than keeping it straight permits the employment of a large

number of periods in a compact size. The high confinement of the fields around the

conductor strip in the microstrip line allows the folding of the microstrip without

causing relevant variation in the filter frequency response [24]. Also, it is shown in

[24] that there is no direct coupling between meandered strip lines.

a2 a1

a1

Figure 4-14: Schematic representation of the folded 1-D multiperiod EBG structure.

Figure 4-15: Photograph of the ground plane of the folded 1-D multiperiod EBG structure measured. The dimensions of this structure are 86mm x 27mm.

The folded 1-D multiperiod EBG structure shown in Figure 4-15 was

fabricated and measured. The simulated and measured S21 results are shown in

Figure 4-16. When compared to the results in Figure 4-13, a deeper attenuation was

obtained from the folded 1-D multiperiod EBG structure shown in Figure 4-15

which has a larger number of lattice circles. The slight discrepancies between the

simulated and measured results are mostly due to the imperfection in the fabrication.


67

0 2 4 6 8 10 12 14-70

-60

-50

-40

-30

-20

-10

0

S21,

dB

Frequency, GHz


Figure 4-16: Simulated and measured S21 for the folded 1-D multiperiod EBG structure.

4.3.3 Microstrip patch antenna on the multiperiod EBG structure In section 4.3.2, it was shown that the multiperiod EBG structure could easily

be extended to a folded 1-D structure by cascading the structure in parallel. This

work is further studied for antenna applications.

Microstrip patch antennas have been widely used in mobile and radio wireless

communications as they are low profile, conformal, low-cost and robust [25].

However, the antennas have a low efficiency, a narrow bandwidth and high surface

wave losses. Hence, the EBG structure has been used as the ground plane of the

microstrip patch antennas to improve their performance [18]-[21]. In this study, the

proposed multiperiod EBG structure is used as the ground plane of the microstrip

patch antennas to show its effect on the antennas.

The microstrip patch antenna has been designed based on the transmission line

model [25]. A microstrip patch antenna operating at around 7.4GHz is designed,

where the resonant frequency falls within the stopband of the multiperiod EBG

structure in the previous sections. The substrate is the same as that used for the

multiperiod EBG structure - RT/Duroid 6010. A 50-Ω microstrip line with a width of

0.6mm is used to feed the microstrip antenna up to the 50-Ω feed point as shown in

Figure 4-17.


68

Figure 4-17: Schematic diagram of the microstrip patch antenna designed at a resonant frequency of 7.4GHz. (L= 6mm, W= 8.3mm, Wo= 0.6mm, W1= 0.7mm, yo=2.517mm)

The simplest way of applying the EBG structure to antenna applications is to

substitute the conventional PEC ground plane with the EBG ground plane. It has

been shown that the periodic circles on the EBG ground plane will create some

current leakage resulting in the increase of the back lobe of the antenna’s radiation

pattern even though the EBG ground plane can suppress the surface wave on the

substrate [18], [19]. Hence, a new method to integrate the EBG ground plane with

the antenna is proposed in this study. In order to maintain the low back lobe of the

radiation pattern exhibited by a conventional patch antenna, the EBG structure is

sandwiched between the patch and the conventional ground plane, as illustrated in

Figure 4-18. With this new design, it is expected that the back lobe of the radiation

pattern will not increase, as reported in [18], [19].

For the purpose of comparison, 3 types of patch antennas were modelled and

measured: 1) conventional patch antenna, 2) patch antenna on a sandwiched single

period EBG ground plane, and 3) patch antenna on a sandwiched multiperiod EBG

ground plane.


69

Figure 4-18: Cross section of the (a) conventional patch antenna, and (b) patch antenna on the sandwiched EBG structure.

The dimensions of the antenna are fixed at 8.3mm (W) x 6mm (L) for all the

cases. There are 8 x 6 circles with a radius of 3mm etched on the EBG ground plane.

The period of a = 12mm is used for the sandwiched single period EBG ground plane

model. For the sandwiched multiperiod EBG ground plane, the period around the

centre of the substrate remains constant at 12mm whilst the period around the edge

of the substrate is reduced to 7.5mm as shown in Figure 4-19. The antenna is placed

at the centre so that it does not overlap with any unit circle as the simulation results

show that the antenna’s performance will improve when place in this position. The

size of the ground plane for the conventional patch antenna model is kept at the same

size as the sandwiched multiperiod EBG ground plane. The input return loss (S11)

curves for these three models obtained from the simulations and measurements are

shown in Figure 4-22.

(a) Ground plane dimensions = 96mm x 72mm


70

(b) Ground plane dimensions = 78mm x 63mm

Figure 4-19: Schematics (top view) of the patch antenna on the sandwiched EBG structure with (a) single period a = 12mm, and (b) multiperiod a1 = 7.5mm, a2 = 12mm.

(a)

(b)

Figure 4-20: Photographs of the (a) single period EBG layer, and (b) multiperiod EBG layer before it is sandwiched to the ground plane.


71

Figure 4-21: Photograph of the antenna under test in the anechoic chamber.

7.1 7.2 7.3 7.4 7.5 7.6 7.7-25

-20

-15

-10

-5

0

S11,

dB

Frequency, GHz


(a)

7.1 7.2 7.3 7.4 7.5 7.6 7.7-25

-20

-15

-10

-5

0

S11,

dB

Frequency, GHz


(b)


72

7.1 7.2 7.3 7.4 7.5 7.6 7.7-25

-20

-15

-10

-5

0

S11,

dB

Frequency, GHz


(c)

Figure 4-22: Simulated (Red) and measured (Blue) input return (S11) for (a) conventional patch antenna, (b) patch antenna on a sandwiched single period EBG ground plane, and (c) patch antenna on a sandwiched multiperiod EBG ground plane.

Figure 4-22 shows that for both the simulated and measured results, the

resonant frequency of the antenna is higher when the antenna is on the sandwiched

EBG ground plane compared to when it is on the conventional ground plane. This is

due to the effective permittivity of the substrate being changed when the sandwiched

EBG layer is added between the antenna and the ground plane. The measured results

of S11 outside the resonant frequency are as low as -5dB is due to the cable loss

during the measurements.

In Figure 4-22, the measured results show that the S11 < -10dB bandwidth is

wider for the antennas on a sandwiched EBG ground plane compared to the

conventional patch antenna. The S11 < -10dB bandwidth increased from 0.47% for a

single period EBG antenna to 0.93% for a multiperiod EBG antenna. The simulated

results show that the impedance bandwidth also increased from 0.635% for a single

period EBG antenna to 0.87% for a multiperiod EBG antenna. It is noted that the

bandwidth for the antennas are very small due to the high permittivity substrate

being used in this study. A wider bandwidth can be obtained by using a lower

permittivity substrate. However, the size of the ground plane will increase as a larger

period of the lattice in the EBG design is required when a low permittivity substrate

is used so as to suppress the surface waves at the same frequency region.

An advantage of a multiperiod EBG design is that a wider impedance

bandwidth is obtained. In addition, a ground plane having smaller dimensions is just


73

required for a multiperiod EBG design compared to a single period EBG design.

Hence a figure of merit, F for these antennas is proposed in this thesis as follows:

where B is the impedance bandwidth and A is the area of the ground plane.

From the measurements, the figure of merit, F for the antenna on the

sandwiched single period EBG ground plane and multiperiod EBG ground plane is

5.06 and 14.25 respectively.

The radiation patterns of the antennas were measured in the anechoic chamber

in QMUL Laboratory. The simulated and measured results are in a good agreement,

as it can be seen from Figure 4-23 and Figure 4-24, respectively.

It is noted that due to the radiation from the microstrip feed line, the E-plane

for the antennas are asymmetrical and broader compared to the H-plane. For the

sandwiched multiperiod EBG antenna, the H-plane radiation pattern has a narrower

beam, which means it is more directional compared to the sandwiched single period

EBG antenna. The computed gain in the forward direction for the conventional

antenna, sandwiched single period EBG and multiperiod EBG antennas are 2.15dBi,

3.55dBi and 4.21dBi respectively. This shows that the multiperiod EBG antenna can

achieve a higher gain compared to the conventional and single period EBG antennas.

Due to the difficulty of measuring the surface waves in the lab, only the

simulated surface currents results generated from the CST Microwave Studio® is

shown in Figure 4-25. It can be clearly seen from Figure 4-25 that the electric field

in the conventional antenna is widely spread compared to the other two antennas

with sandwiched EBG structure. It is also shown that the multiperiod EBG structure

is more effective in suppressing the surface waves compared to the single period

EBG structure. The electric field is mostly concentrated close to the patch antenna

for the multiperiod EBG design. The effect of the surface wave suppression shown in

Figure 4-25 by utilising the 2-D multiperiod EBG structure appears to be a

promising method to suppress mutual coupling in the antenna array applications.

However, the size of the antenna with the EBG structure is not compact enough to

apply to mobile terminals.

F = B/A MHz/mm2(4.5)


74

-40 -30 -20 -10 0

30

210

60

240

90270

120

300

150

330

180

0

(a)

-40 -30 -20 -10 0

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60

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90270

120

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180

0

(b)

-40 -30 -20 -10 0

30

210

60

240

90270

120

300

150

330

180

0

(c)

Figure 4-23: Simulated (+) and measured () radiation patterns of E-plane co-polarisation for (a) conventional antenna ,(b) antenna on a sandwiched single period EBG ground plane, and (c) antenna on a multiperiod EBG ground plane.


75

-40 -30 -20 -10 0

30

210

60

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90270

120

300

150

330

180

0

(a)

-40 -30 -20 -10 0

30

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90270

120

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180

0

(b)

-40 -30 -20 -10 0

30

210

60

240

90270

120

300

150

330

180

0

(c)

Figure 4-24: Simulated (+) and measured () radiation patterns of H-plane co-polarisation for the (a) conventional antenna ,(b) antenna on a sandwiched single period EBG ground plane, and (c) antenna on a multiperiod EBG ground plane.


76

(a)

(b)

(c)

Figure 4-25: Simulated surface distribution of the electric field at 7.4GHz for the (a) conventional antenna (b) sandwiched single period EBG antenna, and (c) sandwiched multiperiod EBG antenna.


77

4.4 Multiperiod EBG structure in the substrate

In Section 4.3, the proposed multiperiod EBG structure was etched on the

ground plane. In this section, the multiperiod EBG design is studied by machining

the multiperiod pattern in the dielectric substrate rather than etching the periodic

structures on the ground plane of the microstrip circuit. Although the machining of

the EBG structure on the dielectric substrate has previously been studied [4], [26]-

[28], the investigation of utilising a multiperiod EBG structure on the dielectric

substrate is a new study.

The same principle in Section 4.3 has been applied in this case. Two different

periods, a1 and a2 which have different central frequencies respectively are cascaded

in series to form a 1-D multiperiod EBG structure. Firstly, periods of a1 = 12mm and

a2 = 7.5mm are considered. However, it is noted that the central frequencies of the

stopband of these two periods are dissimilar to that in the previous study conducted

in section 4.3. This is because the variation of the effective permittivity for the

microstrip when EBG structures are in the substrate is different to that when it is in

the ground plane. In order to obtain a wider stopband bandwidth, the stopband of

two different periods should have an overlapping region in the stopband. In this case,

periods of a1 = 10mm and a2 = 7.5mm are selected to form a 1-D multiperiod EBG

in the substrate. The lattice circles with a radius of 3mm are used. The simulated S21

curves are shown in Figure 4-25.

0 2 4 6 8 10 12 14 16 18 20-70

-60

-50

-40

-30

-20

-10

0

S21,

dB

Frequency, GHz

Simulation

(a)


78

0 2 4 6 8 10 12 14 16 18 20-70

-60

-50

-40

-30

-20

-10

0

S21,

dBFrequency, GHz

Simulation

(b)

Figure 4-26: Simulated S21 parameter for the single period EBG substrate circuit of period (a) a1 = 10mm, and (b) a2 = 7.5mm. The radius of the unit lattice circle, r is 3mm.

For the single period EBG substrate structure with period a1 = 10mm and

filling factor (r/a) is 0.3, the 10dB bandwidth of the first stopband Δfa is

approximately 4.89GHz having a central frequency of approximately 8.8GHz. As

shown in Figure 4-26(a), the second stopband has a frequency range starting from

15.5GHz and extending beyond 20GHz. In the other single period EBG substrate

structure with period a2 = 7.5mm (r/a2 = 0.3), the central frequency of the stopband

is found to be approximately 14.5GHz with the 10dB bandwidth being much wider

Δf = 8.6GHz. By cascading these two periods (i.e. a1 =7.5mm and a2 =10mm) into a

1-D multiperiod EBG structure, a much wider stopband bandwidth is expected. A 1-

D multiperiod EBG structure having these two periods was simulated and measured.

The fabrication process of creating a periodic pattern in a substrate is more

complicated than etching a periodic pattern on a ground plane. The process of

creating a periodic pattern on a substrate is shown in Figure 4-27. As the substrate

RT/Duroid 6010 was a dual sided copper laminate, a wet etching technique was first

applied to remove the copper on the substrate before creating the periodic circles.

The periodic circles were then created via machining the lattice circles through the

substrate by using a numerical milling machine. A copper plate with the dimensions

of the substrate was glued under the EBG substrate as the ground plane of the

structure. A strip line having a width of 0.6mm was cut from a 0.2mm thick copper

plate and was then glued on top of the multiperiod EBG substrate. With the presence

of the ground plane, the strip line corresponds to a 50Ω microstrip line. The


79

fabricated structure is shown in Figure 4-28 and its S21 was measured using a HP

8720ES network analyser.

Figure 4-27: Fabrication process of the multiperiod EBG substrate structure.

Figure 4-28: Photograph of the fabricated multiperiod EBG substrate fabricated. The dimensions of this substrate are 101 x 15 mm.

Figure 4-29 shows that an ultra wide stopband has been achieved with this

proposed multiperiod EBG substrate structure. In the simulation, the 10dB

bandwidth has a frequency range starting from 7GHz and extending beyond 20GHz.

However, the measured 10dB bandwidth has a frequency range starting from

7.8GHz and extending beyond 20GHz. The slight discrepancies are mainly due to

imperfections during the fabrication process. Since the EBG structure is made by

three individual parts, the microstrip line on top of the EBG substrate may have been

misaligned from the centre of the periodic circles. Both simulations and

measurements have shown that there is at least an increase of 42% in the multiperiod

structure’s stopband bandwidth compared to the single period EBG structure with

period, a1 = 7.5mm. The number of cells has increased from 8 to 11 cells in the

multiperiod EBG structure in order to achieve deeper attenuation in the whole range

of the stopband.


80

0 2 4 6 8 10 12 14 16 18 20-70

-60

-50

-40

-30

-20

-10

0

S21,

dB

Frequency, GHz


Figure 4-29: Simulated and measured S21 parameter for a multiperiod EBG substrate circuit with periods, a1 = 7.5mm, a2 = 10mm.

4.5 Summary

The main reason for studying EBG structures in this thesis is their ability to

prevent the propagation of electromagnetic waves within a specified band of

frequencies. EBG structures can be used as a filter in a microstrip circuit whilst in

antenna applications it can suppress the surface waves on the ground plate.

The three main types of EBG structures: metallo-dielectric structure, uniplanar

compact PBG (UC-PBG) structure and square lattice of circles etched on the ground

plane are introduced in this chapter. The third type of EBG structure (i.e. square

lattice of circles etched in ground plane) has been selected for this study as it is the

simplest and most effective way.

A multiperiod EBG structure has been proposed for use in a 1-D, 2-D

microstrip circuit and antenna applications. This proposed structure shows that an

ultra wide stopband bandwidth can be achieved. A very good agreement between

simulated and measured results has been shown. The performance of a microstrip

patch antenna on a conventional ground plane, sandwiched single period EBG

ground plane and sandwiched multiperiod EBG ground plane have been studied. The

microstrip patch antenna on a sandwiched multiperiod EBG ground plane has

achieved the smallest amount of surface current spreading on the ground plane

compared to the to other two cases. A figure of merit, F=B/A is proposed in this


81

thesis to quantify the microstrip patch antenna’s performance in terms of bandwidth

and size.

The multiperiod pattern was drilled into the substrate instead of simply having

been etched on the ground plane. This method allows a wider stopband bandwidth to

be achieved compared to the first proposed design. However, the fabrication process

of machining periodic patterns in the substrate is complicated. This leads to a slight

increase in the discrepancies between the simulated and measured results.

It has been shown that the size of the proposed multiperiod EBG structures is

smaller than the conventional single period EBG structures. However, it is not

sufficiently compact to be used on a small mobile terminal. Hence, the proposed

multiperiod EBG structures are more suitable for use in base stations.

References [1] E. Yablonovitch, “Photonic band-gap structures,” J. Opt Soc. Am. B., vol. 10,

no. 2, pp. 283-295, Feb. 1993.

[2] E. Yablonovitch, “Photonic Crystal: Semiconductor of light,” Scientific American, pp. 47-55, 2001.

[3] J. Shumpert, T. Ellis, G. Rebeiz, and L. Katehi, “Microwave and millimeter-wave propagation in photonic band-gap structure,” AP-S/URSI, pp. 678, 1997.

[4] Y. Qian, V. Radisic, and T. Itoh, “Simulation and experiment of photonic band-gap structures for microstrip circuits,” Asia-Pacific Microwave Conf. (APMC’97) Dig., pp 585-588, Hong Kong, 1997.

[5] D. Sievenpiper, “High-Impedance electromagnetic surfaces,” Ph.D. dissertation, Dept. Elect. Eng., Univ. California at Los Angeles, Los Angeles, CA, 1999.

[6] R. F. J. Braos, D. F. Sievenpiper, and E. Yablonovitch, “A high-impedance ground plane applied to a cellphone handset geometry,” Microwave Theory and Techniques, IEEE Transactions on ,vol. 49 , no. 7 , pp. 1262-1265, July 2001.

[7] S. Rogers, J. Marsh, W. McKinzie, and J. Scott, “An AMC Based 802.11a/b antenna for laptop computers,” Antennas and Propagation Society International Symposium, 2003. IEEE, vol. 2, pp. 10-13, June 22-27, 2003.

[8] F. R. Yang, K. P. Ma, Y. Qian, and T. Itoh, “A uniplanar compact photonic-bandgap (UC-PBG) structure and its applications for microwave circuits,” IEEE Trans. On Microwave Theory and Techniques, vol. 47, no. 8, August 1999.


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[9] I. J. Bahl, and P. Bhartia, Microwave Solid State Circuit Design, John Wiley, New York, 1988.

[10] F. R. Yang, Y. Qian, T. Itoh, “A novel uniplanar compact PBG structure for filter and mixer applications,” Microwave Symposium Digest, 1999 IEEE MTT-S International , vol. 3, pp. 919- 922, 1999.

[11] R. Coccioli, F. R. Yang, K. P. Ma ,and T Itoh, “Aperture coupled patch antenna on UC-PBG substrate,” IEEE Trans. Microwave Theory & Tech, vol. 47, no. 11, pp. 2123-2130, Nov. 1999.

[12] F. R. Yang, R. Coccioli, Y. Qian, and T. Itoh, “PBG assisted gain enhancement of patch antennas on high-dielectric constant substrate,” IEEE AP-S International Symp., pp. 1920-1923, June 1999.

[13] C. R. Simovski, and B. Sauviac, “Uniplanar PBG screens for forming antenna patterns,” Electronic Letters, vol. 39, no. 9, pp. 707-708, May 2003.

[14] V. Radisic, Y. Qian, R. Coccioli, and T. Itoh, “Novel 2-D photonic band-gap structure for microstrip lines,” IEEE Microwave guided Wave Lett., vol. 8, no. 2, pp. 69-71, Feb 1998.

[15] F.R. Yang, Y. Qian, R. Coccioli, and T. Itoh, “Analysis and application of photonic band-gap (PBG) structures for microwave circuits,” Electromagnetics, vol. 19, pp. 241-254, 1999.

[16] T. Lopetegi, F. Falcone, B. Martinez, R. Gonzalo, and M. Sorolla, “Improved 2-D photonic bandgap structures in microstrip technology,” Microw. Opt. Technol. Lett., vol. 22, pp. 207-211, 1999.

[17] T. Kim, and C. Seo, “A novel photonic bandgap structure for low-pass filter of wider stopband,” IEEE Microw. Guid. Wave Lett., vol 10, no.1, pp. 13-15, 2000.

[18] Y. Horii, and M. Tsutsumi, “Suppression of the harmonic radiation from the PBG microstrip antenna,” Microwave Conference, 1999 Asia Pacific, vol. 3, pp. 724 – 727, 1999.

[19] Y. Horii, and M. Tsutsumi, “Harmonic control by photonic bandgap on microstrip patch antenna,” IEEE Microw. Guid. Wave Lett., vol. 9, no. 1, pp 13-15, 1999.

[20] F. Zhu, S. He, Q. Lin, H. He, and Z. Ying, “Microstrip patch antenna with PBG structure,” International Conference on Telecommunications, ICT 2002, China, 2002.

[21] R. Coccioli, W. R. Deal, and T. Itoh, “Radiation characteristics of a patch antenna on a thin PBG structure,” IEEE Antennas & Propag. International Symposium, pp. 656-659, Atlanta, 1998.

[22] M. A. G. Laso, T. Lopetegi, M. J. Erro, D. Benito, M. J. Garde, and M. Sorolla, “Novel wideband photonic bandgap microstrip structure,” Microwave. Opt. Technol. Lett., vol. 24, no. 5, pp. 357-360, Feb 2000.

[23] F. Hirtenfelder, T. Lopetegi, M. Sorolla, and L. Sassi, “Designing components containing photonic bandgap structures using time domain fields solvers,” Microwave Engineering, pp. 23-29, March 2002. URL:www.mwee.com


83

[24] F. Falcone, T. Lopetgi, M. Irrisarri, M. Laso, M. Erro, M. Sorolla, “Compact Photonic Bandgap Microstrip Structures,” Microwave and Optical Techn. Letters, vol. 23, no. 4, pp. 233-236, Nov 1999.

[25] C. A. Balanis, Antenna Theory: Analysis and Design, Wiley, USA, 1997.

[26] I. Rumsey, M. Piket-May, and P. K. Kelly, “Photonic bandgap structures used as filters in microstrip circuits,” IEEE Microw. Guid. Wave Lett., vol. 8, no. 10, pp. 336-338, 1998.

[27] H. D. Yang, N. G. Alexopoulos, E. Yablonovitch, “Photonic bang-gap materials for high-gain printed circuit antennas,” IEEE Trans. Antennas Propagat., vol. 45, pp. 185-187, 1997.

[28] A. S. Andrenko, “Study and microwave applications of artificial periodic substrate PBG planar circuits,” Mathematical Method in Electromagnetic Theory, 2002, International Conference on, vol. 1, pp. 21-26, 2002.

Chapter 5: Diversity Antenna Array for MIMO Systems

84

Chapter 5 Diversity Antenna Array for MIMO Systems

5.1 Introduction

In Chapter 3, the concept of designing small antennas on a mobile terminal

for MIMO systems has been addressed. It is noted that the folded loop antenna is a

self-balancing structure where no unbalanced current is introduced on the coaxial

feed lines. As such, this antenna is used to design the diversity antenna array for

MIMO systems in this thesis. The operating frequency selected for this project is

5.2GHz which corresponds to the IEEE802.11a WLAN (Wireless Local Area

Network) applications. In this chapter, the folded loop antennas is studied and

developed into dielectric loaded folded loop antennas. Later, a diversity antenna

array for Personal Digital Assistants (PDAs) based on the dielectric loaded folded

loop antennas is studied numerically and experimentally.

5.2 Single Folded Loop Antenna

5.2.1 Single Folded Loop Antenna at 5.2GHz

Firstly, a folded loop antenna based on [1] is designed and optimised to

operate at 5.2GHz using the CST Microwave Studio® software package. The folded

loop antenna with a 50 Ω coaxial feed line on a ground plane is illustrated in Figure

5-1. The inner conductor of the coaxial feed line is connected to one of the folded

loop antenna’s arm whilst a shorting pin is connecting the other arm of the antenna


85

with the ground plane. The outer conductor of the coaxial feed line is attached to the

ground plane. The folded loop antenna is placed 1mm above the ground plane. The

dimensions of the folded loop antenna and the ground plane are detailed in Table 5-1.

The length of the antenna before both the arms are folded up is about half-

wavelength. When the arms are folded up (the antenna thereby becoming a folded

loop antenna), the length of the antenna is reduced to about a quarter-wavelength.

Figure 5-1: The schematic diagram of the folded loop antenna with a 50Ω coaxial feed line on a ground plane.

Parameter Name Value(mm) Value (λ)

a 15.25 0.2643

d 0.50 0.0087

h 3.80 0.0659

s 3.50 0.0607

w1 1.00 0.0173

w2 4.00 0.0693

GL 60.00 1.0400

GW 15.25 0.2643 Table 5-1: The dimensions of the antenna and its ground plane.

It has been shown that when the width ratio w1:w2 of the element is varied,

the input impedance of a folded dipole changes [2]. This flexibility in controlling the

antenna input impedance is a very attractive feature of the folded loop antenna. This

means that the antenna can be matched with different feed lines by changing the


86

width ratio w1:w2. It is reported in [1] that the input impedance of the folded loop

antenna is in good match with the 50Ω coaxial feeding line when the width ratio

w1:w2 is 1:4, resulting in a maximum bandwidth.

Figure 5-2 shows the simulated return loss curves of the folded loop antenna

operating at 5.2GHz with different width ratios w1:w2 whilst w1 is kept constant at

1mm. The maximum bandwidth is achieved when the width ratio w1:w2 is 1:4.

Therefore, w2 = 4mm has been selected for this study.

4.4 4.6 4.8 5.0 5.2 5.4 5.6 5.8 6.0-20

-15

-10

-5

0

Ret

urn

loss

, dB

Frequency, GHz

w2=1

w2=2

w2=3

w2=4

w2=5 w

2=6

w2=7

Figure 5-2: Simulated return loss curves of the folded loop antenna operating at 5.2GHz with different w2 and constant w1=1mm.

A prototype of the simulated antenna model was fabricated as shown in

Figure 5-3. Initially, a loop antenna was cut from a copper sheet with a thickness of

0.2mm using a numerical milling machine. Then both arms of the loop antenna were

folded up to become a folded loop antenna. In the prototype, the antenna was fed by

a 50Ω coaxial feed line. The inner pin of the coaxial feed line was connected to one

of the antenna’s arm whilst a shorting pin was connecting another arm with the

ground plane and the outer conductor of the coaxial feed line. The SMA connector

was attached to the other end of the coaxial feed line for measurement.

The return loss of the prototype antenna was measured using a HP 8720ES

network analyser. The measured results are compared with the simulated results as

shown in Figure 5-4. From the simulation results, it can be seen that the antenna can

achieve a bandwidth (S11 < 10dB) of about 420MHz (5.05-5.47 GHz). However, the


87

measured bandwidth is only 300MHz (5.08-5.38 GHz). This discrepancy is mainly

due to the imperfection of the fabrication process. There is also a possible mismatch

between the coaxial feed line and the antenna due to imperfections resulting from the

soldering process.

Coaxial feed lineFolded loop antenna

SMA connector Ground plane

Figure 5-3: Photograph of the prototype folded loop antenna operating at 5.2GHz.

4.4 4.6 4.8 5.0 5.2 5.4 5.6 5.8 6.0-20

-15

-10

-5

0

Ret

urn

loss

, dB

Frequency, GHz

Measurement Simulation

Figure 5-4: Simulated and measured return loss curves.

The radiation patterns of the prototype antenna were measured inside an

anechoic chamber with the transmitting field provided by a quad ridge horn with

dual-polarisation capability. Figure 5-5 and Figure 5-6 show the simulated and


88

measured radiation patterns of the antenna. Generally speaking, the co-polar

measured radiation patterns are in good agreement with the simulated patterns.

The radiation patterns in Figure 5-5 is plotted along the XZ-plane which

corresponds to the E-plane of the antenna. The radiation patterns are asymmetrical in

the positive and negative X-directions which is due to the unbalanced feed line being

used. The measured co-polar radiation pattern has a null at about 130º which does

not appear in the simulation. This null is caused by the long and bended coaxial feed

line being used in the measurement as shown in Figure 5-3. It is noted that the

agreement between the simulated and measured results for the cross-polar radiation

pattern is not as good as that for the co-polar radiation pattern. As the signal level for

the cross-polar pattern is 10dB lower than that for the co-polar pattern, the ripples

shown in the measured cross-polar patterns (which has not been predicted from the

simulation) is due to the interference/noise in the anechoic chamber.

Figure 5-6 shows the radiation patterns in the YZ-plane which corresponds to

the H-plane of the antenna; and an omni-directional pattern is achieved in both

measurement and simulation. The discrepancies on the cross-polar patterns between

measurement and simulation are mainly due to the long and bended coaxial feed line

as mentioned earlier.

The surface currents on the ground plane of the folded loop antenna with

balanced and unbalanced feeding lines are accessed numerically and compared. For

the folded loop antenna with balanced feeding lines, the coaxial cable in the

unbalanced model as shown in Figure 5-1 was replaced with a voltage source in the

simulation as shown in Figure 5-7. The distribution of surface currents for both cases

with balanced and unbalanced feeding structures is shown in Figure 5-8. It is noted

that the surface currents on both cases are confined to the area underneath the folded

loop antenna; there is little current flow to the other area of the ground plane. This

has further shown that the surface current generated by a folded loop antenna with an

unbalanced feed is not widely spread on its ground plane.


89

(a)

(b) Figure 5-5: Simulated (+) and measured (–) radiation patterns on the X-Z plane for (a) co-polar and (b) cross-polar.

-40 -30 -20 -10 0

30

210

60

240

90270

120

300

150

330

180

0

x

z

-40 -30 -20 -10 0

30

210

60

240

90270

120

300

150

330

180

0

x

z


90

(a)

(b) Figure 5-6: Simulated (+) and measured (–) radiation patterns on the Y-Z plane for (a) co-polar and (b) cross-polar.

-40 -30 -20 -10 0

30

210

60

240

90270

120

300

150

330

180

0

y

z

-40 -30 -20 -10 0

30

210

60

240

90270

120

300

150

330

180

0

y

z


91

(a)

Coaxial feed line

Shorting pin

(b)

Figure 5-7: Simulation model of the folded loop antenna for the (a) balanced feeding technique and (b) unbalanced feeding technique.

Figure 5-8: Distribution of the surface currents on the ground plane for (a) balanced feeding structure and (b) unbalanced feeding structure.


92

5.2.2 Single Dielectric Loaded Folded Loop Antenna at 5.2GHz

Inside a dielectric, the effective wavelength of an electromagnetic wave is

shorter than that in free space. Therefore, the folded loop antenna can be made

smaller in size by loading a dielectric slab inside the antenna as shown in Figure 5-9.

It is found that the folded loop antenna is not a planar structure after folding and

some foam is required inside the antenna to act as a mechanical support. Hence, as

well as reducing the antenna’s size, the dielectric slab also functions as a mechanical

support to the antenna. When mass produced, this antenna could be simply mask

metallised on the dielectric slab rather than folding up the metal sheet as in the

prototype antenna. This proposed antenna is termed as a dielectric loaded folded

loop antenna throughout the thesis.

However, the dielectric material tends to reduce the bandwidth of the

antennas. Hence, the effects of the dielectric material on the performance of the

folded loop antenna are investigated.

A dielectric loaded folded loop antenna with an unbalanced feeding line on a

ground plane shown in Figure 5-9(b) has been modelled and simulated according to

the parameters in Table 5-1. The effects of a dielectric slab on the antenna with

different relative permittivity, εr of 2, 4, 6, 8, and 10 are investigated. Figure 5-10

shows the effect of the dielectric slab on the resonant frequency of the antennas.

When the relative permittivity is equal to 1, this means that there is no loading of

dielectric. It is shown that the resonant frequency of the antenna is reduced with the

increase of the relative permittivity values. This is because the effective wavelength

of the electromagnetic waves decreases with increasing values of relative

permittivity. The effect on the bandwidth of the antenna is plotted in Figure 5-11.

The bandwidth is reduced more significantly compared to the resonant frequency of

the antenna. When the relative permittivity is equal to 6, the bandwidth has been

reduced by 60% compared to when no dielectric is loaded inside the antennas. In

order to have a wider bandwidth, a dielectric slab only with a relative permittivity of

4 is selected in this thesis.

The return loss performance of the folded loop antenna with a dielectric slab

is compared to one without a dielectric slab. The results are plotted in Figure 5-12.

The resonant frequency of the dielectric loaded folded loop antenna decreased from


93

5.24GHz to 4.29GHz when the antenna is not loaded. Further, the -10dB bandwidth

has reduced from 8% (5.05GHz -5.47GHz) to 5.7% (4.18GHz – 4.424GHz).

Figure 5-9:Schematic diagram showing (a)the dielectric slab inserted into a folded loop antenna and (b) the dielectric loaded folded loop antenna with unbalanced feeding line on a ground plane.

0 1 2 3 4 5 6 7 8 9 103.0

3.5

4.0

4.5

5.0

5.5

6.0

3.253.52

3.85

4.29

4.88

Res

onan

t req

uenc

y, G

Hz

Relative permittivity, εr

5.24

Figure 5-10: Relation between the resonant frequency and relative permittivity.


94

0 1 2 3 4 5 6 7 8 9 1050

100

150

200

250

300

350

400

450

70

115

170

245

325

Ban

dwid

th, M

Hz

Relative permittivity, εr

425

Figure 5-11: Relation between the bandwidth and relative permittivity.

2.8 3.2 3.6 4.0 4.4 4.8 5.2 5.6 6.0-25

-20

-15

-10

-5

0

Ret

urn

Loss

, dB

Frequency, GHz

Folded loop antenna Folded loop antenna

with dielectric loaded, er=4

Figure 5-12: Simulated return loss curves for the folded loop antenna with and without the dielectric slab of relative permittivity, εr=4.

As the resonant frequency of the proposed dielectric loaded folded loop

antenna decreased to 4.29GHz, the antenna was redesigned and optimised to operate

at 5.2GHz. The optimised dimensions of the proposed antenna are summarised in

Table 2. It is noted that the size of the ground plane used in Section 5.2.1 would be

too small compared to the typical size of a handset. Hence, the size of the ground

plane for the proposed antenna is 100mm x 40mm which is closer to the typical size

of a handset.


95


a 13.40 0.2323

d 0.50 0.0087

h 3.00 0.0520

s 2.90 0.0503

w1 1.00 0.0173

w2 4.00 0.0693

GL 100.00 1.7333

GW 40.00 0.6933 Table 5-2: The optimised dimensions of the dielectric loaded folded loop antenna and its ground plane at 5.2GHz. The dielectric slab has a relative permittivity of 4.

4.6 4.7 4.8 4.9 5.0 5.1 5.2 5.3 5.4 5.5 5.6 5.7 5.8-25

-20

-15

-10

-5

0

Ret

urn

loss

, dB

Frequency, GHz

Folded loop antenna Dielectric loaded folded loop antenna

Figure 5-13: Simulated return loss curves for the folded loop antenna and the proposed dielectric loaded folded loop antenna, both antennas operating at 5.2GHz.

Figure 5-13 shows the return loss curve of the optimised dielectric loaded

folded loop antenna operating at 5.2GHz. The return loss performance of the

proposed antenna is compared to the performance of the original folded loop antenna

(also operating at 5.2GHz) from Section 5.2.1. It is shown in Figure 5-13 that the

bandwidth of the proposed antenna has reduced to 190MHz due to the dielectric

loading and the reduction of the antenna’s size. Despite this, the bandwidth of


96

190MHz still meets the required bandwidth of 120MHz for WLAN applications. On

the other hand, the volume of the original folded loop antenna studied in Section

5.2.1 has decreased by 30% due to the effect of dielectric loading.

5.3 Four-Element Diversity Antenna Design

In this section, a four-element diversity antenna array on a PDA has been

designed using the proposed dielectric loaded folded loop antennas. For simplicity, a

metal box with dimensions of 110.0mm x 75.0mm x 13.4mm was used to represent a

PDA. These dimensions are similar to the typical size of a PDA and have also been

used in industry development, e.g. Antenova Ltd U.K [3].

The proposed design involves four elements of dielectric loaded folded loop

antennas being located in the upper part of the PDA to avoid the antennas being

covered by the user’s hand. The performance of the antennas could be degraded if

they are covered by the user’s hand.

By placing four antennas on a PDA terminal, especially when only the upper

position of the terminal is considered, the diversity antenna array has to be carefully

designed to achieve low correlation between the antennas. As the dielectric loaded

folded loop antenna is a linearly polarised antenna, mutual coupling between the

antennas can be reduced by arranging the antennas orthogonally to each other.

Simultaneously, polarisation diversity could also be achieved.

The optimised design of the diversity antenna array on a PDA is shown in

Figure 5-14. Antenna 1 is located on top of the PDA, antennas 2 and 3 are placed at

the side of the PDA and finally antenna 4 is placed on the back of the PDA.

The distance between antennas 2 and 3 is approximately 75mm which

corresponds to 1.3λ at 5.2GHz. The spacing of 1.3λ is sufficiently far apart to

achieve high isolation between antennas 2 and 3. Further, the boresight of the

antennas are opposing each other in the x-direction; hence they tend to pick up

signals received from different directions leading to strong pattern diversity.

Therefore, low correlation can be achieved between antennas 2 and 3 as they are

spatially apart and are positioned opposite from each other.

Antennas 1 and 4 are placed between antennas 2 and 3 as shown in Figure

5-14. Antennas 1 and 4 are arranged in a way that they are orthogonally polarised to


97

any other antenna resulting in polarisation diversity. The distance between antennas

1 and 4 in the z-direction is 26.75mm which corresponds to 0.46λ. As the distance

between antennas 1 and 4 is approximately half-wavelength and the antennas are

arranged orthogonally polarised to each other, low correlation between the antennas

is expected.

The prototype of the proposed diversity antenna array shown in Figure 5-15

was made in the QMUL Antenna Laboratory. The metal box was constructed from a

copper sheet with a thickness of 0.3mm. 50Ω semi-rigid coaxial cables are used to

feed the antennas. It is noted that it is relatively difficult to make the dielectric

loaded folded loop antenna in the QMUL laboratory. Hence, every antenna made is

not perfectly identical.

(a)


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(b)

Figure 5-14: Schematic diagram of four-element diversity antenna array on a PDA in (a) X-Z plane and (b) 3-D view.

110 mm

75 mmz

xy

(a) (b)

Figure 5-15: Photographs of the prototype diversity antenna array on a PDA terminal. (a) Front view of the PDA and (b) the feeding structures behind the PDA.


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5.3.1 Return loss and isolation performances

The simulated and measured return loss performances of each antenna (from

the proposed four-element diversity antenna array on a PDA) are compared as shown

in Figure 5-16.

It is noted that the return loss curves are slightly different for each antenna.

This is due to the different size of the ground plane as each antenna is located at

different locations on the PDA. The ground plane of antennas 2 and 3 is similar in

size hence they should have the same return loss performance as predicted from the

simulations. However, the measured results show that there are discrepancies

between antennas 2 and 3 due to the fabrication imperfection of the antennas.

The measurements show that antenna 1 (which is positioned on top of the

PDA) has a -10dB bandwidth of approximately 200MHz whilst the other antennas

have a bandwidth of approximately 300MHz. This is probably due to the width of

the ground plane for antenna 1 being equal to the length (i.e. parameter a) of the

antenna element as shown in Figure 5-14(b). Generally speaking, all the antennas

have achieved enough bandwidth to meet the 120MHz bandwidth requirement of the

IEEE 802.11a wireless applications.

4.6 4.7 4.8 4.9 5.0 5.1 5.2 5.3 5.4 5.5 5.6 5.7 5.8-30

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urn

loss

, dB

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S11 S22 S33 S44

(a)


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-25

-20

-15

-10

-5

0

Ret

urn

loss

, dB

Frequency, GHz

S11 S22 S33 S44

(b)

Figure 5-16: Return loss curves from the (a) simulated and (b) measurement results.

Figure 5-17 shows the isolation between each pair of antennas obtained from

the simulations and measurements. Isolation of more than 20dB for each pair of

antennas has been achieved. In other words, mutual coupling is very low between the

antennas therefore the resonant frequencies of each antenna is not detuned

significantly as shown in Figure 5-16. Therefore, a low correlation between the

antennas could be realised and would lead to good diversity gain.

4.6 4.7 4.8 4.9 5.0 5.1 5.2 5.3 5.4 5.5 5.6 5.7 5.8-50-45-40-35-30-25-20-15-10-50

Isol

atio

n, d

B

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S12 S13 S14 S23 S24 S34

(a)


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4.6 4.7 4.8 4.9 5.0 5.1 5.2 5.3 5.4 5.5 5.6 5.7 5.8-50-45-40-35-30-25-20-15-10

-50

Isol

atio

n, d

B

Frequency, GHz

S12 S13 S14 S23 S24 S34

(b)

Figure 5-17: Isolation between each pair of antennas on the diversity antenna array from the (a) simulated and (b) measured models.

5.3.2 Radiation patterns

The radiation patterns of the prototype diversity antenna array on a PDA were

measured in an anechoic chamber in QMUL Antenna Measurement Laboratory. The

simulated and measured results are compared and detailed in this section.

The E-plane of each antenna in the Cartesian co-ordinate system is

summarised in Table 5-3. In Figure 5-18 and Figure 5-19, the radiation patterns of

each antenna are plotted with respect to their own E-plane. There is a very good

agreement between the simulated and measured radiation patterns. It is noted in

Figure 5-18 that the back-lobe of the radiation patterns for each antenna is

approximately 10dB less than the peak in the front-lobe (i.e. radiation above the

ground plane of each antenna). This is due to the ground plane of each antenna

operating as a reflector thereby reducing the back-lobe radiation (i.e. radiation

behind the ground plane of each antenna). Figure 5-19 shows the E-plane cross-polar

radiation patterns of each antenna on the PDA. It is found that the cross polarisation

is less than -15dB in most angles.

The H-plane radiation patterns of the antennas on the PDA were also

measured and compared with the simulated results, as shown in Figure 5-20 and

Figure 5-21. The H-plane of each antenna on the PDA in the Cartesian co-ordinate

system is summarised in Table 5-4. Generally speaking, the simulated and measured


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results agree very well to each other with respect to the H-plane radiation patterns. It

is noted that the cross-polar patterns are quite strong in the H-planes. However, the

peak in the front-lobe of the co-polar radiation patterns is still 10dB stronger than the

cross-polar patterns.

Antenna E-field plane 1 y-z plane 2 x-z plane 3 x-z plane 4 x-y plane

Table 5-3: The E- plane of each antenna on the PDA in the Cartesian co-ordinate system.

Antenna H-field plane 1 x-z plane 2 x-y plane 3 x-y plane 4 y-z plane

Table 5-4: The H- plane of each antenna on the PDA in the Cartesian co-ordinate system.

The radiation patterns are not symmetrical due to the imbalanced feeding

lines used in the design. It is noted that the radiation patterns of each antenna has its

boresight at different directions. Hence, the RF signals received at the PDA terminal

from different angles/directions after experiencing multipath propagation will be

received by a different antenna element. There is a very rare chance that all the

received signals will be in a deep null at the same time. As a result, the combined

signal would have a higher mean SNR at the output compared to a single antenna

receiver.


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Figure 5-18: Simulated (+) and measured () E-plane co-polar radiation patterns of each antenna with respect to their individual E-field polarisation plane.


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Antenna 1 Antenna 2

Antenna 3 Antenna 4

z

x

z

y

y

x

z

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1

2 4

x

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y

Figure 5-19: Simulated (+) and measured () E-plane cross-polar radiation patterns of each antenna with respect to their individual E-field polarisation plane.


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Antenna 3 Antenna 4

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1

2 4

x

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x

Figure 5-20: Simulated (+) and measured () H-plane co-polar radiation patterns of each antenna with respect to their individual H-field polarisation plane.


106

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Figure 5-21: Simulated (+) and measured () H-plane cross-polar radiation patterns of each antenna with respect to their individual H-field polarisation plane.


107

5.4 Diversity Performance of the Four-element Diversity Antenna Array

The diversity performance of the proposed four-element diversity antenna

array on a PDA is studied in this section. The diversity performance has been

evaluated by calculating the correlation, mean effective gain (MEG) and diversity

gain of the antennas.

5.4.1 Correlation and MEG

3-D radiation patterns are required to evaluate the correlation using equation

(2.9). As there is a lack of facility to conduct the measurement of 3-D radiation

patterns at QMUL, only simulated results have been used. The results from the

simulations conducted so far have agreed very well with the measured results;

therefore it is reliable to use the simulated 3-D radiation patterns to evaluate the

diversity performance of the antennas. The simulated 3-D radiation patterns have

been obtained with 5º step elevation cuts while the PDA is positioned vertically in

free space.

The correlations for each pair of antennas on the PDA are evaluated using

equation (2.9) and the results are summarised in Table 5-5. The impact of the

outdoor and indoor environments on the envelope correlation has been evaluated

using two different statistical models as discussed in Chapter 2 (i.e. Gaussian and

Laplacian distribution). Table 5-5 shows that the envelope correlation of less than

0.1 (as evaluated by two different statistical models) has been achieved for each pair

of antennas, in both the outdoor and indoor environments. It is noted that there is not

much difference between the correlation values when evaluated by the two statistical

models. The results from Table 5-5 show that different values of correlation are

obtained in the different environments for the same pair of antennas.

The MEG of each antenna within the different environments is evaluated

from the 3-D gain patterns using equation (2.18) and the results are tabulated in

Table 5-6. For the same antenna in the same environment, the MEG value for the

Gaussian and Laplacian statistical models vary less than 0.1dB. However, the MEG

of each antenna within the different environments can vary up to 1.5dB. In the

outdoor environment, the MEG for different antennas can vary up to 2.5dB. On the


108

other hand, the MEG for different antennas has a difference of less than 1dB in the

indoor environment.

Envelope correlation, ρe Propagation Model ρe12 ρe13 ρe14 ρe23 ρe24 ρe34

Outdoor 0.0444 0.0378 0.0092 0.0597 0.0015 0.0122

Gau

ssia

n

Indoor 0.0102 0.0063 0.0040 0.0021 0.0020 0.0077

Outdoor 0.0458 0.0400 0.0151 0.0800 0.0014 0.0128

Lap

laci

an

Indoor 0.0163 0.0112 0.0006 0.0013 0.0046 0.0097

Table 5-5: Summary of the envelope correlation from the diversity antenna array on the PDA terminal. ρenm representing the envelope correlation between antennas n and m.

MEG/dB Propagation

Model Antenna 1 Antenna 2 Antenna 3 Antenna 4

Outdoor -2.1434 -3.1915 -3.1771 -4.7733

Gau

ssia

n

Indoor -2.7694 -3.1326 -3.1420 -3.1108

Outdoor -2.1056 -3.2365 -3.2197 -4.7178

Lap

laci

an

Indoor -2.7527 -2.9759 -2.9817 -3.0172

Table 5-6: Summary of the MEG for each antenna from the diversity antenna array on the PDA terminal.

Equation (2.14) shows that zero correlation will lead to a maximum diversity

gain. Therefore, minimum correlation is desired to achieve a maximum diversity

gain. In the proposed diversity array antenna, a very low correlation (i.e. less than

0.1) for each pair of antennas was obtained in both indoor and outdoor environments

using different statistical models. These low correlation values will result in high

diversity gains. The ratio of the MEG (i.e. branch power ratios) between each pair of

antennas also contributes to the diversity gain as shown in equations (2.16) and

(2.17). A maximum diversity gain will be achieved when the branch power ratios for

each pair of antennas is one. Table 5-6 shows that in the indoor environment, all the


109

antennas have very similar MEG values, which is ideal to achieve a high diversity

gain. Both the effects of correlation and MEG on diversity gain are elaborated in the

next section.

5.4.2 Diversity Gain

After assessing the correlation and MEG results, the diversity gain of the

proposed four-element diversity antenna array on the PDA can be evaluated. The

selection combiner diversity gain results are shown in Table 5-7. The diversity gain

results have included the degradation factor (DF) from equation (2.14) due to the

correlation and the branch power ratio (k) from equation (2.17).

In an ideal 4-branch selection combiner, the diversity gain at 99% reliability

is 15dB as shown in Figure 2-12. Table 5-7 shows that the diversity gains of the

proposed four-element diversity antenna array are less than 15dB. As the correlation

is less than 0.1 in all cases as shown in Table 5-5, the degradation in the diversity

gain is mainly due to unequal mean branch power. The branch power ratio (k) is

higher in an outdoor environment compared to an indoor environment as the XPR is

higher (i.e. XPR = 5dB for outdoors and XPR = 1dB for indoors). Therefore, the

lowest diversity gain obtained is around 5dB less than the maximum possible

diversity of 15dB. The best diversity gain performance is achieved in an indoor

environment with only around 0.8dB less than the maximum value.

Propagation Model Degradation Factor, DF/dB k/dB Diversity Gain/dB

Outdoor -0.20 -4.71 10.09

Gau

ssia

n

Indoor -0.04 -1.08 13.88

Outdoor -0.22 -4.86 9.92

Lap

laci

an

Indoor -0.06 -0.72 14.22

Table 5-7: Comparison of selection combiner diversity gain performance at 99% reliability in different environments and using different statistical models.


110

5.4.3 Effect of cross polar ratio (XPR)

In the above studies, XPR = 5dB and XPR = 1dB have been assumed for the

outdoor and indoor environments respectively. The XPR could vary from -10dB to

10dB in different fading environments [4]. Figure 5-22 to Figure 5-25 show a

comparison of MEG with varying XPR in different environments.

When the XPR is higher than 0dB, the average vertical power of the RF

signals is stronger than the average horizontal power and vice versa. The comparison

results show that Antennas 1-3 are dominant to vertical polarisation due to their

arrangement on the PDA terminal. Antennas 2 and 3 have the same performance as

they are arranged in a similar way but just facing away from each other on the PDA

terminal. Antenna 4 is arranged orthogonal to Antennas 2 and 3; therefore, the MEG

results show that antenna 4 is dominant to horizontal polarised signals as predicted.

All antennas have similar MEG values when the XPR is around 1dB especially

within an indoor environment. The further the XPR values are away from 1dB, the

larger the branch power ratio which will lead to a greater degradation in diversity

gain.

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-7

-6

-5

-4

-3

-2

-1

0

MEG

,dB

XPR,dB

Antenna 1 Antenna 2 Antenna 3 Antenna 4

Figure 5-22: Variation of MEG with XPR in an outdoor environment evaluated using the Gaussian statistical model.


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-7

-6

-5

-4

-3

-2

-1

0

MEG

,dB

XPR,dB


Figure 5-23: Variation of MEG with XPR in an indoor environment evaluated using the Gaussian statistical model.

-10 -9 -8 -7 -6 -5 -4 -3 -2 -1 0 1 2 3 4 5 6 7 8 9 10-8

-7

-6

-5

-4

-3

-2

-1

0

MEG

,dB

XPR,dB


Figure 5-24: Variation of MEG with XPR in an outdoor environment evaluated using the Laplacian statistical model.

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-7

-6

-5

-4

-3

-2

-1

0

MEG

,dB

XPR,dB


Figure 5-25: Variation of MEG with XPR in an indoor environment evaluated using the Laplacian statistical model.


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5.5 Channel Capacity

After evaluating the diversity performance, the channel capacity of a MIMO

system using the proposed diversity antenna array is also assessed. Wireless InSite

[5], a ray tracing (RT) simulation tool is used in this thesis to construct realistic

realisations of the channel matrix H in an indoor wireless environment. The indoor

wireless environment model used in this thesis is the second floor in the Department

of Electronic Engineering at QMUL as shown in Figure 5-26. The indoor

environment model consists of wooded doors, glass windows and concrete walls.

The base station (transmitter) consists of four ideal dipoles (with half-wavelength

spacing) placed on the ceiling of the corridor outside Room A as shown in Figure

5-26. The transmit power level from the transmitter is 20dBm and the bandwidth is

20MHz. The receivers are clustered randomly in 1000 positions near the desktop

height in Rooms A and B. This propagation model has been verified with the IEEE

802.11 MIMO model [6] using a uniform linear array (ULA) of four ideal dipoles at

both the transmitter and receiver for a 4 X 4 set up [7].

In this thesis, the receivers are the proposed four-element diversity antenna

array on the PDA terminal. The 3-D simulated radiation patterns of the antennas

obtained from the CST Microwave Studio® were used as the receivers in the indoor

propagation model in the Wireless InSite.

16 m

eter

s

Figure 5-26: Floor plan of the second floor of the Department of Electronic Engineering at QMUL. The receivers are located randomly at 1000 positions in Rooms A and B.


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Figure 5-27: 3-D floor plan of the second floor of the Department of Electronic Engineering at QMUL. The red dots show the random positioning of the receivers.

Figure 5-28 and Figure 5-29 show the simulated rays arriving at the receivers

in Rooms A and B. For clarity, only a few ray paths are drawn in Figure 5-29. It is

shown that the signals launched from the transmitter are reflected off and/or

transmitted through obstacles before reaching the receivers. The information of each

path i.e. the time delay, the length of the path and the received power at the receiver,

are obtained from the Wireless Insite simulation. The channel response is modelled

as the vector sum of all the rays arriving at the receivers which can be computed by

the following equation [8]:

where M is the number of rays, fo is the carrier frequency, Pk is the received power, lk

is the length of the kth ray and τk is the time delay of the kth ray. The channel matrix H

is computed using equation (5.1) where Pk, lk, and τk are obtained from the Wireless

InSite simulation. After obtaining the channel matrix H of the indoor environment,

the eigenvalues from the channel matrix H is evaluated mathematically by the

singular value decomposition (SVD) method and the channel capacity is then

computed using equation (2.4).

Figure 5-28: Full propagation paths shown in a 3-D floor plan of the Department of Electronic Engineering at QMUL.

( )2 / 2

1

k o kM

j l j fij k

kh P e eπ λ π τ

== ⋅ ⋅∑ (5.1)


114

(a)

(b)

Figure 5-29: Rays arriving at receivers in (a) Room A and (b) Room B are reflected off and/or transmitted through obstacles.

0 5 10 15 20 25 300

5

10

15

20

25

30

35

SNR, dB

Capa

city

, bits

/s/H

z

4 ideal dipoles in Room ARoom ARoom BSISO

Figure 5-30: Channel capacity performance of the proposed diversity antenna array in Room A and Room B compared to 4 ideal dipoles in Room A and within a SISO system.

Figure 5-30 shows the MIMO channel capacity obtained from the realistic

propagation environments (i.e. Room A and Room B) with the proposed diversity

antenna array on the PDA as the receivers. The channel capacity results from Rooms


115

A and B are compared to that within the SISO system which consists of only a single

dipole at both the transmitter and receiver. When SNR = 20dB, the channel capacity

increased from 7bps/Hz in the SISO system to 18bps/Hz in the MIMO system in

both Room A and Room B using the proposed diversity antenna array, as shown in

Figure 5-30. It is also noted that the channel capacity performances at Rooms A and

B are similar despite the different sizes of the rooms. A 4 x 4 MIMO system with

ULA of ideal dipoles at the receiver is also compared in Figure 5-30. In the ULA

ideal dipoles model, the mutual coupling and PDA case effects had not been taken

into consideration. It is shown that the channel capacity is slightly smaller in the

proposed diversity antenna array when the mutual coupling and case effects are

taken into consideration. However, the difference is not significantly large.

5.6 Dielectric Loaded Folded Half-loop Antenna

In the Sections 5.3 to 5.5, the proposed dielectric loaded folded loop antenna

has been used to develop a four-element diversity antenna array on a PDA terminal

for MIMO systems. However, the dielectric loaded folded loop antenna may be too

large to be placed inside a mobile handset or Personal Computer Memory Card

International Association (PCMIA) card if four elements are needed. Therefore, the

dielectric loaded folded loop antenna is further studied in this section for the purpose

of size reduction.

5.6.1 Design Concept

In general, a quarter-wavelength monopole is about half the size of a half-

wavelength dipole. The size reduction of a monopole is due to the image effect from

the ground plane of the monopole as shown in Figure 3-2. In this section, the size of

the dielectric loaded folded loop antenna is further reduced using the same principle

as that used in a monopole. Figure 5-31 shows that the size of a dielectric loaded

folded loop antenna has been reduced by replacing one side of the antenna with a

copper plate. It is noted that the copper plate acts as an electric wall or image to

reduce the antenna size by half. Therefore, the new proposed antenna is termed the

‘dielectric loaded folded half-loop antenna’.


116

The copper plate on the dielectric loaded folded half-loop antenna has been

designed to have the same size as the dielectric slab’s side wall. Therefore, the width

and height of the antenna remain unchanged when the copper plate is introduced.

The antenna is placed 1mm above a ground plane with dimensions of GW = 40mm x

GL = 100mm. The antenna is fed by a 50Ω coaxial cable with the inner conductor of

the coaxial cable directly connected to the folded arm whereby the copper plate is

shorted to the ground plane through a shorting pin as shown in Figure 5-32. The

optimised dimensions of the antenna operating at 5.2GHz are detailed in Table 5-8.

The width ratio w1:w2 of the antenna is fixed at 1:4, similar to that of the dielectric

loaded folded loop antenna. It is noted that the length of the dielectric loaded folded

half-loop antenna, a is only 7.85mm (i.e. ≈1/8 of wavelength) compared to the

length of the dielectric loaded folded loop antenna which is about a quarter-

wavelength.

The experimental antenna was constructed from a copper sheet with a

thickness of 0.2mm using a numerical milling machine. The copper plate was folded

up to become a folded half-loop antenna as shown in Figure 5-33. A dielectric slab

with relative permittivity, εr of 4 is then inserted into the folded half-loop antenna to

form a dielectric loaded folded half-loop antenna. Figure 5-34 shows a prototype of

the proposed dielectric loaded folded half-loop antenna fabricated at QMUL.

Dielectric Slab

Copper Plate

Dielectric loaded folded loop antenna

Dielectric loaded folded half-loop antenna

14 1

8

(a) (b) Figure 5-31: A dielectric loaded folded loop antenna is reduced in size to form a dielectric loaded folded half-loop antenna.


117

Figure 5-32: Schematic diagram of a dielectric loaded folded half-loop antenna and the antenna configuration on a ground plane with the feed location.


a 7.85 0.1361

d 0.50 0.0087

h 3.50 0.0607

s 2.00 0.0347

w1 1.00 0.0173

w2 4.00 0.0693

GL 100.00 1.7333

GW 40.00 0.6933 Table 5-8: The optimised dimensions of the dielectric loaded folded half-loop antenna and its ground plane at 5.2GHz. The dielectric slab has a relative permittivity of 4.

Figure 5-33: Structure of a folded half-loop antenna constructed from a flat copper plate.


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Figure 5-34: Photograph of the fabricated dielectric loaded folded half-loop antenna with a ground plane size of 40mm x 100mm.

5.6.2 Return loss performance

The return loss of the proposed antenna was measured using a HP8720ES

network analyser at QMUL. The measured result is compared with the simulated

result as shown in Figure 5-35. The measured return loss shows that the proposed

antenna has a -10dB bandwidth of more than 200MHz with a center frequency of

around 5.2GHz to cover the IEEE802.11a band. However, the antenna element is

found to be very small in size, hence, imperfection during the fabrication process has

caused a slight shifting of the resonant frequency and a reduction in the bandwidth

compared to the simulated result as shown in Figure 5-35.

4.6 4.7 4.8 4.9 5.0 5.1 5.2 5.3 5.4 5.5 5.6 5.7 5.8-20

-15

-10

-5

0

Ret

urn

loss

, dB

Frequency, GHz

Measurement Simulation

Figure 5-35: Measured and simulated return loss of the dielectric loaded folded half-loop antenna.

5.6.3 Radiation patterns and gain

The radiation patterns of the dielectric loaded folded half-loop antenna were

measured inside an anechoic chamber with the transmitting field provided by a quad


119

ridge horn with dual-polarisation capability. Figure 5-36 and Figure 5-37 show the

simulated and measured co-polar and cross-polar radiation patterns of the antenna.

Generally speaking, the measured radiation patterns are in good agreement with the

simulated results. The small discrepancy at the back lobe of the radiation patterns is

due to the presence of a long coaxial cable being used to feed the antenna from

behind the ground plane.

The radiation patterns are plotted along the XZ-plane and YZ-plane as shown

in Figure 5-36 and Figure 5-37. It is noted in Figure 5-36 that the co-polar radiation

patterns are not symmetrical as the antenna is not symmetrical from the feed point.

The front to back ratio of the radiation patterns is about 10dB in the co-polar patterns.

Further, the radiation patterns show that the proposed antenna has stronger radiation

intensity in the positive Z-direction. This indicates that the ground plane is behaving

like a reflector rather than a radiator.

Figure 5-37 shows the measured and simulated cross-polar patterns of the

antenna in both the XZ-plane and the YZ-plane. It is noted that the proposed antenna

can achieve cross-polar patterns of -10dB or lower in most directions, so the antenna

remains a linear polarisation structure.

The antenna gain of the antenna was measured inside the anechoic chamber.

The measured maximum gain of the antenna at 5.2GHz is 2dBi.

(a) (b)

Figure 5-36: Simulated (+) and measured (-) co-polar radiation patterns at 5.2GHz for (a) E-plane and (b) H-plane.

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(a) (b) Figure 5-37: Simulated (+) and measured (-) cross-polar radiation patterns at 5.2GHz for (a) E-plane and (b) H-plane.

5.6.4 Effect of the ground plane

It is known that the characteristic of a conventional PIFA strongly depends on

the ground plane size as the ground plane is used as part of the radiator [9], [10].

Hence, usually the PIFA has to be redesigned when the size of the ground plane is

changed. It has been shown in Section 5.2 that the ground plane for the folded loop

antenna is behaving more like a reflector rather than a radiator. Therefore, the

surface current on the ground plane is mostly confined in the area underneath the

antenna. In this section, the study of ground plane effects on the proposed dielectric

loaded folded half-loop antenna is conducted.

5.6.4.1 Effect on the size of the ground plane

The proposed antenna is firstly fixed at the location as shown in Figure 5-32.

The ground plane width, GW is fixed at 40mm whilst the ground plane length, GL is

changed from 20 to 100mm with a 10mm interval. Figure 5-38 plots the changes of

the center frequency and the -10dB bandwidth with the changes of the GL. The

minimum and maximum of the vertical bars show the lower and upper frequencies of

the -10dB bandwidth respectively. The bandwidth is not greatly affected when the

ground plane length GL is changed. The center frequency of the proposed antenna is

also very stable from GL = 40mm to 100mm as shown in Figure 5-38. There is only

less than a 100MHz shift of center frequency and bandwidth when the GL is 20mm.

-40 -30 -20 -10 0

30

210

60

240

90270

120

300

150

330

180

0

-40 -30 -20 -10 0

30

210

60

240

90270

120

300

150

330

180

0


121

Figure 5-39 shows the changes of the centre frequency and bandwidth when

the ground plane width, GW is varied from 20mm to 40mm whilst the ground plane

length, GL is fixed at 100mm. It is noted that the ground plane width, GW can be

reduced from 40mm to 30mm without a significant shift in the center frequency and

reduction in the bandwidth performance. The bandwidth becomes about 100MHz

narrower and the center frequency shifts approximately 50MHz when GW is reduced

to 25mm. At 20mm,GW = the bandwidth reduces significantly compared to GW =

40mm. However, practically the ground plane width (GW) of 40mm is sufficiently

small for a handset. A ground plane width that is smaller than 30mm will not be

practical for a current handset. This study has shown that the proposed antenna does

not require a new design when the size of the ground plane is changed.

0 20 40 60 80 100 1205.00

5.05

5.10

5.15

5.20

5.25

5.30

5.35

5.40

Freq

uenc

y, G

Hz

Ground plane length (GL), mm Figure 5-38: Center frequency () and the -10dB bandwidth for the different ground plane lengths, GL.

15 20 25 30 35 40 455.00

5.05

5.10

5.15

5.20

5.25

5.30

5.35

5.40

Freq

uenc

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Ground plane width (GW), mm Figure 5-39: Center frequency () and the -10dB bandwidth for the different ground plane widths, GW.


122

5.6.4.2 Effect of the antenna’s location on the ground plane

After studying the effect of the ground plane size, the performance of the

antenna at different locations on the ground plane is studied in this section. The

ground plane size is fixed at GL = 100mm x GW = 40mm in this study. The location

of the antenna on the ground plane in x-y coordinate system is shown in Figure 5-42.

The origin of the coordinate system is 3.5mm away in y-direction from the top edge

of the ground plane as shown in Figure 5-42.

Figure 5-40: Location of the antenna on the ground plane in x-y direction. The antenna’s feed location is at x=0 and y=20mm.

In the first study, the antenna is moved along the ground plane width in the x-

direction whilst keeping y = 0. Figure 5-41 shows that the center frequency does not

change much when the antenna is moved from x = 2mm to 30mm along the ground

plane width. However, the optimal bandwidth is achieved and stays almost constant

when the antenna is placed around the center of the ground plane width (e.g. from x

= 14mm to x = 20mm). The antenna’s -10dB bandwidth reduces to zero when it is

moved further away from the center of the ground plane, i.e. x ≤ 8mm.

Figure 5-42 demonstrates the performance of the antenna when the antenna’s

feed location is moved along the ground plane length in the y-direction from y = 0 to

80mm whilst x = 20mm. The center frequency is quite stable when the antenna is

moved along the ground plane length and the bandwidth does not change much as

well. It is noted that the antenna’s performance is affected more when it is moving

along the ground plane width as the antenna is polarised in the direction parallel to

ground plane width.


123

0 2 4 6 8 10 12 14 16 18 20 22 24 26 28 30 325.00

5.05

5.10

5.15

5.20

5.25

5.30

5.35

5.40

Freq

uenc

y, G

Hz

Location in x-direction, mm Figure 5-41: Center frequency () and the -10dB bandwidth of the proposed antenna at different locations in x-direction when y=0. At y=0, the antenna’s feed point is placed 3.5mm away from the top edge of the ground plane.

-10 0 10 20 30 40 50 60 70 80 905.00

5.05

5.10

5.15

5.20

5.25

5.30

5.35

5.40

Freq

uenc

y, G

Hz

Location in y-direction, mm Figure 5-42: Center frequency () and the -10dB bandwidth of the proposed antenna at different locations in y-direction when x=20. At y=0, the antenna’s feed point is placed 3.5mm away from the top edge of the ground plane.

5.6.5 Four elements of Dielectric Loaded Folded Half-loop Antennas on a PDA

Four elements of the dielectric loaded folded half-loop antennas on a PDA

terminal were modelled and simulated as shown in Figure 5-43. The dimensions of

the PDA terminal are 110.0mm x 75.0mm x 13.4mm, which have the same

dimensions as the PDA terminal used in Section 5.3. The antennas are also placed at

the same locations as the proposed four-element diversity antenna array in Section


124

5.3. All four elements of the dielectric loaded folded half-loop antennas have the

same dimensions as summarised in Table 5-8.

(a)

(b)

Figure 5-43: Schematic diagram of four elements of the dielectric loaded folded half-loop antennas on a PDA in (a) x-z plane and (b) 3D view.


125

Figure 5-44 shows the simulated return loss results for each antenna on a

PDA. It is noted that Antenna 1 is detuned and no longer has a -10dB bandwidth.

The resonant frequency of Antennas 2 and 3 is detuned to approximately 5.1GHz

with a -10dB bandwidth of 130MHz. The exception is Antenna 4 which is still

performing well at a resonant frequency of 5.2GHz due to its large ground plane. In

Section 5.6.4, the resonant frequency and bandwidth of a dielectric loaded folded

half-loop antenna is shown to decrease when either the ground plane size is reduced

or the antenna is moved closer to the edge of the ground plane. Therefore, Antenna 1

(which has the narrowest ground plane width) has the worst performance amongst

the antennas on the PDA terminal.

Figure 5-45 shows that there are more surface currents on the ground plane of

the dielectric loaded folded half-loop antenna compared to the surface currents on

the ground plane of the dielectric loaded folded loop antenna as studied in Section

5.2.2 . This is because the dielectric loaded folded half-loop antenna is not a self-

balanced antenna, due to one of the dielectric loaded folded loop antenna’s arms

being replaced by a copper plate. When the size of a dielectric loaded folded half-

loop antenna’s ground plane is significantly reduced, the surface currents on the

ground plane are disturbed which will change the antenna’s resonant frequency.

Therefore, the dielectric loaded folded half-loop antenna does not perform well on

the edge of the PDA where the width of the ground plane is very narrow. Hence, the

dielectric loaded folded half-loop antennas are only suitable to be placed on the back

of a PDA terminal and not ideal to be used as MIMO array elements in the

configuration as shown in Figure 5-43.

4.6 4.7 4.8 4.9 5.0 5.1 5.2 5.3 5.4 5.5 5.6 5.7 5.8-20-18-16-14-12-10-8-6-4-20

Ret

urn

loss

, dB

Frequency, GHz

S11 S22 S33 S44

Figure 5-44: Simulated return loss curves of each dielectric loaded folded half-loop antenna on a PDA.


126

Figure 5-45: Computed surface currents on the ground plane of (a) dielectric loaded folded loop antenna and (b) dielectric loaded folded half-loop antenna at 5.2GHz.

5.7 Summary

A new type of antenna – dielectric loaded folded loop antenna has been

developed for the MIMO system. The carrier frequency chosen for this study is

5.2GHz as used by the IEEE802.11a WLAN system. It is found that the dielectric

loaded folded loop antenna is a self-balanced antenna which generates only small

amounts of surface currents on the ground plane.

Four dielectric loaded folded loop antennas arranged orthogonally to each

other have been used to design a diversity antenna array on a PDA terminal. With

such an arrangement, the isolations of more than 20dB for each pair of antennas have

been achieved from the measurements. Also, all the antennas have a bandwidth of

more than 120MHz to meet the requirement of the IEEE802.11a applications. The

radiation patterns of the antennas are different to each other and the boresight of

patterns are facing different directions. Therefore, each antenna could pick up RF

signals coming from different directions.


127

The diversity gain of the proposed four-element diversity antenna array has

been evaluated by assessing the correlation and MEG of the antennas in both outdoor

and indoor environments. Generally speaking, the envelope correlations of the four-

element diversity antenna array are less than 0.1. The MEG of the antennas has a

significant effect on the diversity gain for the proposed diversity antenna array as the

antennas are arranged orthogonally (i.e. in polarisation diversity). Hence, the

unequal branch power is large when the XPR is high. As a result, it has been shown

that the proposed four-element diversity antenna array performs best in an indoor

environment with a 14.22dB diversity gain achieved for 99% reliability. In the

outdoor environment, diversity gain of around 10dB is still achieved despite the

large unequal branch power.

The channel capacity of a 4 x 4 MIMO system is evaluated by modelling the

system in a realistic environment. The proposed four-element diversity antenna array

is used as the receiver whilst a ULA of four ideal dipoles is used as the transmitter in

the MIMO system. The study has shown that the channel capacity increased from

7bps/Hz in the SISO system (i.e. a single dipole at both the transmitter and receiver)

to 18bps/Hz in the 4 x 4 MIMO system with the proposed four-element diversity

antenna array as the receiver.

Further, the size of the dielectric loaded folded loop antenna is reduced to

approximately half its size and the new proposed antenna is termed as the ‘dielectric

loaded folded half-loop antenna’. The size reduction is achieved by replacing one of

the folded arms of the dielectric loaded folded loop antenna with a copper plate to

provide an image effect for the other folded arm. Whilst the antenna’s size is reduced,

the antenna still maintains a good performance at resonant frequency of 5.2GHz with

a -10dB bandwidth of 200MHz. It has also been shown that the dimensions of the

ground plane do not impact much on the performance of the dielectric loaded folded

half-load antenna until the ground plane’s width is narrower than 30mm. As such,

the performance of the dielectric loaded folded half-loop antenna deteriorates when

it is placed on the edge of a PDA terminal where the ground plane’s width is very

narrow. Hence, the proposed dielectric loaded folded half-loop antennas are not

suitable to be used as MIMO array elements with the configuration in Section 5.6.5.


128

References [1] S. Hayashida, H. Morishita, Y. Koyanagi and K. Fujimoto, “Wideband folded

loop antenna for handsets”, IEEE AP-S Int. Symp. Proc., pp. 440-443, 2002.

[2] R.W. Lampe, “Design formulas for an asymmetric coplanar strip folded dipole”, IEEE Transaction on Antennas and Propagation, vol. 33, no. 9, pp. 1028-1031, 1985.

[3] Ofcom research report, “Antenna designs for MIMO systems,” Contract number: AY4476A, 2004.

[4] T. Taga, “Analysis of mean effective gain of mobile antennas in land mobile radio environments”, IEEE Trans. On Vech. Techn., vol. 39, no.2, pp 117-131, May 1990.

[5] Remcom, “Wireless InSite user manual version 1.5.1,” 2003.

[6] V. Erceg, Indoor MIMO WLAN Channel Models, IEEE 802.11-03/87r0, November 2003.

[7] Y. Gao, X. Chen and C.G. Parini, “Experimental evaluation of indoor MIMO channel capacity based on ray tracing”, London Communication Symposium, University College London, pp. 189-192, Sept. 2004.

[8] C.N. Chuah, G.J. Foschini, R.A. Valenzuela, D. Chizhik, J. Ling and J.M. Kahn, “Capacity growth of multi-element arrays in indoor and outdoor wireless channels”, Wireless Communications and Networking Conference, 2000, WCNC 2000 IEEE, vol. 3, pp. 23-28, Sept 2000.

[9] M. Hunynh and W. Stutzman, “Ground plane effects on planar inverted-F antenna (PIFA) performance,” IEE Proceedings Microwaves, Antennas & Propagation, vol. 150, no. 4, pp. 209-213, Aug. 2003.

[10] M.F. Abedin and M. Ali, “Modifying the ground plane and its effect on planar inverted-F antennas (PIFAs) for mobile handsets,” IEEE Antenna Wireless Propagation Letters, vol. 2, pp. 226-229, 2003

Chapter 6: Conclusions and Future Work

129

Chapter 6 Conclusions and Future Work

6.1 Summary

The MIMO system has already been employed in WLAN and will be

implemented in future wireless mobile communications (e.g. WiMAX, 3G and 4G).

This is due to its ability to significantly increase capacity without increasing the

transmitter signal power and spectrum usage. The throughput and range of a MIMO

system can be improved by using spatial multiplexing (SM) and space-time coding,

respectively, to exploit the MIMO channels. Significant amounts of research have

been carried out on signal processing algorithms for MIMO systems but limited

research has been done on antenna designs for MIMO systems. Therefore, the

antenna designs for MIMO systems were carried out in this thesis.

The diversity technique (which utilises more than one antenna to receive or

transmit signals), being a well known method to solve the signal fading problem in a

multipath environment, is utilised in space-time coding to exploit the MIMO

channels. Therefore, the study of three diversity techniques: spatial, pattern and

polarisation diversity has been undertaken in this thesis. In respect of antenna

designs, the antennas have to be placed sufficiently far apart from one another or

arranged orthogonally to one another to achieve a low correlation. Apart from the

antenna’s parameters, environment is also another factor that affects the diversity

performance of diversity antennas on a mobile terminal. Information on the average

distribution of incident power at the mobile terminal in different propagation

environments is required in the evaluation of the diversity performance. Two


130

different environmental models (i.e. indoor and outdoor environments) were chosen

in this thesis to evaluate the diversity performance of the diversity antenna arrays on

a PDA terminal.

It is noted that the built-in PIFA used in most handsets nowadays generates

unbalanced current flow on the feed line and ground plane. Hence, the ground plane

is acting as a radiator rather than a reflector. A balanced folded loop antenna which

generates only a small amount of current on the ground plane was addressed in this

thesis. When two or more balanced folded loop antennas are placed on a ground

plane, a low correlation between the antennas could be achieved. Therefore, this

antenna was adopted in this thesis to reduce the coupling between the antennas

caused by the surface current on the ground plane.

EBG structure has also been considered in this thesis as an alternative method

to solve the mutual coupling problem when the diversity antenna arrays experience

severe correlation. A new multiperiod EBG structure which can achieve a wide

stopband bandwidth was proposed and studied in this thesis. However, it was found

that the size of the multiperiod EBG structure was not sufficiently compact for use

on a mobile terminal. Further, the increasing cost of manufacturing the mobile

terminals would be of a concern to the manufacturers. Therefore, it is noted that the

multiperiod EBG structure is more suitable for use in large terminals or base stations.

Further, the multiperiod EBG structure can be adopted for future UWB (Ultra-

Wideband) antenna arrays technology where the operating frequencies range from

3.1 – 10.6 GHz.

It was proposed that the size of the balanced folded loop antenna be further

reduced so that a four-element diversity antenna array can be placed on a PDA

terminal. This size reduction was achieved by loading a dielectric slab into the

antenna. The diversity performance was evaluated with selection combiner technique

and it was found that a diversity gain of 14.42dB at 99% reliability was achieved in

an indoor environment. A 4 x 4 MIMO system in a real indoor environment (the

realistic four-element diversity antenna array on a PDA terminal being the receiver)

has achieved twice the channel capacity obtained from a SISO system. The size of

the proposed dielectric loaded folded loop antenna was further reduced to

approximately half the original size by replacing one of the folded arms of the

antenna with a copper plate.


131

6.2 Key Contributions

The major contributions in the thesis are detailed in the three sections below:

Multiperiod EBG Structure

• A new multiperiod EBG structure with smaller dimensions and a wider stopband

bandwidth was proposed and presented. This multiperiod EBG structure is simple

in design and easy to manufacture.

• The proposed multiperiod EBG structure is capable of suppressing the surface

current on the ground plane more effectively than the conventional single period

EBG structure which has larger dimensions and a narrower stopband bandwidth.

• The microstrip patch antenna on a sandwiched multiperiod EBG ground plane has

a wider impedance bandwidth compared to a microstrip patch antenna on a

sandwiched single period EBG ground plane. Therefore, the ratio of impedance

bandwidth (B) over the dimensions area of the EBG ground plane (A), F = B/A, is

proposed in this thesis as a figure of merit to quantify the performance of the

antenna with different EBG ground planes.

Four-element diversity antenna array on a PDA

• A self-balanced type antenna (i.e. folded loop antenna) was realised. The size of

the folded loop antenna was reduced by loading a dielectric slab and this antenna

was termed the ‘Dielectric Loaded Folded Loop Antenna’. The ground plane of the

self-balanced antenna acts as a reflector rather than a radiator to the antenna.

• A four-element diversity antenna array (four elements of the dielectric folded loop

antennas were used) on a PDA terminal operating at 5.2GHz was proposed. The

antennas were arranged orthogonally polarised to one another and each antenna’s

boresight faced different directions. Therefore, an isolation of more than 20dB for

each pair of antennas was achieved.


132

• The diversity performance of the proposed diversity antenna array was evaluated in

both the outdoor and indoor environments. A maximum diversity gain of 14.22 dB

at 99% reliability was achieved in the indoor environment.

• A 4 x 4 MIMO system in an indoor environment (i.e. 2nd floor in Department of

Electronic Engineering at QMUL) was modelled with the proposed four-element

diversity antenna array on a PDA terminal as the receivers. The 4 x 4 MIMO

systems achieved twice the capacity of a SISO system.

Miniature Dielectric Loaded Folded Half-loop Antenna

• The size of the dielectric loaded folded loop antenna was further reduced to

approximately 50% by replacing one of the folded arms of the antenna with a

copper plate to provide an image effect for the other folded arm. Whilst the

antenna’s size is reduced, the antenna still maintains a good performance at a

resonant frequency of 5.2GHz with a -10dB bandwidth of 200MHz. However,

there are more surface currents on the ground plane of the dielectric loaded folded

half-loop antenna compared to the surface currents on the ground plane of the

dielectric loaded folded loop antenna. Therefore, the performance of this antenna

starts deteriorating when the width of its ground plane is narrower than 30mm.

6.3 Future work

Based on the conclusions drawn and the limitations of the work presented,

further work can be carried out in the following areas:

• The capability of the proposed multiperiod EBG structure to suppress the surface

current on a ground plane of a microstrip patch antenna has been demonstrated in

Chapter 3. Alternatively, two microstrip patch antennas on the same ground plane

could also be used to study the performance of the multiperiod EBG structure. The

isolation between the antennas should increase if the operating frequency is within

the stopband range of the multiperiod EBG structure.


133

• As the dimensions of the proposed multiperiod EBG structure are too large to be

used in a mobile terminal, it could be applicable for use on a base station given that

more space would be allowed on the base station for the EBG structure.

• In the future, more PDAs will integrate with mobile phone functions. Therefore,

the user’s effect on the diversity antenna’s performance should be studied.

Different situations have to be considered, e.g. talk position and where as a PDA is

only used for writing or reading. In the talk position, the SAR (Specific Absorption

Rate) value of the antenna arrays on the user should also be considered.

• The 4 x 4 MIMO channel capacity in an indoor environment presented in Chapter

5 was carried out by simulation only. Therefore, measurement campaigns on the

MIMO channel capacity with the proposed four-element diversity antenna array as

a receiver could be carried out in different environments.

• The future IEEE 802.11n standard is going to support the current WiFi frequency

bands at 2.4GHz and 5.2GHz. The proposed dielectric loaded folded loop antenna

could be further developed to operate at 2.4GHz band or to operate at both bands.

• Recently, the dual-antenna design has been considered for GSM networks with the

aim to reduce the impact of radio interference to bit rates. This effect would

increase average bit rates and extend radio coverage. Therefore, research on the

design of a two-element diversity antenna array on a handset at GSM bands could

be carried out.

Publications: 134

Publications

Journal Papers:

1. L. Guo, J. Liang, C.C. Chiau, X. Chen, C.G. Parini and J. Yu, “Performances of

UWB disc monopoles in time domain”, IEE Proceedings Microwaves, Antennas

& Propagation, 2005 (Submitted).

2. S. Yeap, X. Chen, J. Dupuy, C.C. Chiau and C.G. Parini, “Low profile diversity

antenna for MIMO applications”, IEE Electronic Letters, vol. 42, no. 2, pp. 69-

70, 2006

3. X. Chen, J. Liang, P. Li and C.C. Chiau, “UWB Electric and Magnetic

Monopole Antennas”, African Journal of Information & Communication

Technology, vol. 2, no. 1, 2006.

4. C.C. Chiau, X. Chen and C.G. Parini, “A Miniature Dielectric-Loaded Folded

Half-Loop Antenna and Ground Plane Effects”, IEEE Antennas and Wireless

Propagation Letters, vol. 4, pp. 459-462, 2005.

5. S. Yeap, X. Chen, C.C. Chiau, J. Dupuy and C.G. Parini, “Integrated Diversity

Antenna for Laptop and PDA terminals”, IEE Proceedings Microwaves,

Antennas & Propagation, vol. 152, no. 6, pp. 495-504, Dec 2005.

6. J. Liang, L. Guo, C.C. Chiau, X. Chen and C.G. Parini, “Study of CPW-Fed

circular disc monopole antenna”, IEE Proceedings Microwaves, Antennas &

Propagation, vol. 152, no. 6, pp. 520-526, Dec 2005.

7. J. Liang, C.C. Chiau, X. Chen and C.G. Parini, “Study of a Printed Circular Disc

Monopole Antenna for UWB Systems”, IEEE Transactions on Antennas and

Propagation, vol. 53, no. 11, pp. 3500-3504, Nov 2005.

8. C.C. Chiau, X. Chen and C.G. Parini, “A Sandwich Multi-period EBG Structure

for Microstrip Patch Antennas”, Microwave and Optical Technology Letter, vol.

46, no. 5, pp. 437-440, 5 Sept 2005.

9. Y. Gao, C.C. Chiau, X. Chen and C. G. Parini, “Modified PIFA and its Array

for MIMO Terminals”, IEE Proceedings Microwaves, Antennas & Propagation,

vol. 154, no. 4, pp. 255-259, Aug 2005.

Publications: 135

10. J. Liang, C.C. Chiau and X. Chen, “Printed circular ring monopole antennas”,

Microwave and Optical Technology Letters, vol. 45, no.5, pp. 372-375, 2005.

11. C.C. Chiau, X. Chen and C.G. Parini, “A compact four-element diversity

antenna array for PDA terminal in a MIMO system”, Microwave and Optical

Technology Letters, vol. 44, no.5, pp. 408-412, March 2005

12. J. Liang, C.C. Chiau, X. Chen and C.G. Parini, 'Printed circular disc monopole

antenna for ultra wideband applications', IEE Electronic Letters, vol. 40, no. 20,

pp.1246-1248, Sept 2004.

13. C.C. Chiau, X. Chen and C.G. Parini, “A Multi-period EBG Structure for Wide

Stopband Circuits”, IEE Proceedings on Microwaves, Antennas and Propagation,

vol.15, no.6, 2003 pp.489-492, Dec 2003.

Conference Papers:

1. C.C. Chiau, J. Dupuy, X Chen and C.G. Parini, “Design of UWB antenna for

Autonomous Interference Monitoring System (AIMS)”, IEEE AP-S International

Symposium on Antennas and Propagation, USA, 2006. (Accepted)

2. J. Liang, K. Wu, C.C. Chiau, X. Chen, C.G. Parini, “Printed UWB elliptical disc

monopole”, Loughborough Antennas and Propagation Conference 2006,

Loughborough, UK, 2006. (Accepted)

3. Z. Wang, C.C. Chiau, X. Chen, B.S. Collins, S.P. Kingsley and S.C. Puckey, “A

miniature broadband Dielectric Resonator Antenna (DRA) operating at 2.4GHz ”,

IEEE International Workshop on Antenna Technologies (iWAT 2006), New

York, USA, pp. 104-107, 2006.

4. X. Chen, J. Liang, P. Li, L. Guo, C.C. Chiau and C.G. Parini, “Planar UWB

monopole antennas”, Asia-Pacific Microwave Conference, SuZhou, China, 4-7

Dec 2005 (Invited paper).

5. Z. Wang, C.C. Chiau, X. Chen, B.S. Collins, S.P. Kingsley and S.C. Puckey,

“Broadband dielectric loaded trapezoidal planar antenna”, Asia-Pacific

Microwave Conference, SuZhou, China, 4-7 Dec 2005.

Publications: 136

6. J. Liang, C.C. Chiau and X. Chen, “Time domain characteristics of UWB disc

monopole antennas”, 35th European Microwave Conference, Paris, France,

October 4-6, 2005.

7. C.C. Chiau, X. Chen and C.G. Parini, “A Novel Dielectric Loaded Folded Half-

Loop Antenna for Mobile Terminals”, IEEE AP-S International Symposium on

Antennas and Propagation, Washington DC, USA, vol. 1A, pp. 499-502, 3-8 July,

2005.

8. Y. Gao, C. C. Chiau, X. Chen and C. G. Parini, “A Modified PIFA with a Small

Ground Plane”, IEEE AP-S International Symposium on Antennas and

Propagation, Washington DC, USA, vol. 2B, pp. 515-518, 3-8 July, 2005.

9. J. Liang, C.C. Chiau, X. Chen and C.G. Parini, “CPW-Fed Circular Ring

Monopole Antenna”, 2005 IEEE AP-S International Symposium on Antennas

and Propagation, Washington DC, USA, vol. 2A, pp. 500-503, 3-8 July, 2005.

10. Y. Gao, C. C. Chiau, X. Chen and C. G. Parini, “A Compact Dual-Element

PIFA Array for MIMO Terminals”, Loughborough Antennas & Propagation

Conference, U.K., 4-5 April 2005.

11. Z. Wang, C.C. Chiau, X. Chen, B.S. Collins, S. P. Kingsley, S. C. Puckey,

“Optimisation of A Broadband Dielectric Antenna”, Loughborough Antennas &

Propagation Conference, U.K., 4-5 April 2005.

12. J. Liang, C.C. Chiau and X. Chen, “Design analysis in a planar UWB circular

ring monopole”, Loughborough Antennas and Propagation Conference 2005,

Loughborough, UK, 4-5 April, 2005.

13. C.C. Chiau, Y. Gao, X. Chen and C.G. Parini, “Evaluation of Indoor MIMO

Channel Capacity with a realistic four-element diversity antenna array on a PDA

terminal”, IEEE International Workshop on Antenna Technologies (iWAT 2005),

Singapore, pp. 454- 457, 7-9 March 2005.

14. Z. Wang, C.C. Chiau, X. Chen, B. S. Collins, S. P. Kingsley, S. C. Puckey and J.

R. Thorpe, “Study and Optimisation of A Broadband Dielectric Antenna”, IEEE

International Workshop on Antenna Technologies (iWAT 2005), Singapore, pp.

125-128, 7-9 March 2005.

Publications: 137

15. J. Liang, L. Guo, C.C. Chiau and X. Chen, “CPW-Fed Circular Disc Monopole

Antenna for UWB Applications”, IEEE International Workshop on Antenna

Technologies (iWAT 2005), Singapore, pp. 505-508, 7-9 March 2005.

16. J. Liang, C.C. Chiau, X. Chen, J. Yu, “Effect of the ground plane on the

operation of a UWB monopole”, 2004 Progress in Electromagnetics

Research Symposium, Nanjing, China, 28-31 Aug 2004.

17. C.C. Chiau, S.B. Yeap, X. Chen, C. G. Parini, “A Four Element Diversity

Antenna Array for a MIMO PDA Terminal”, 2004 International Symposium on

Antennas And Propagation, Sendai, Japan, 17-21 Aug 2004.

18. J. Liang, C.C. Chiau, X. Chen, J. Yu, “Study of a circular disc monopole

antenna for ultra wideband applications”, 2004 International Symposium on

Antennas and Propagation, Sendai, Japan, 17-21 Aug 2004.

19. J. Liang, C.C. Chiau, X. Chen and C.G. Parini, “Analysis and Design of UWB

Disc Monopole Antennas”, IEE International Workshop on Ultra Wideband

Communication Technologies & System Design, London, UK, 8 July 2004.

20. X. Chen, Y. Alfadhl, C.C. Chiau, and Z. Wang, “Numerical Dosimetry on the

Scales of Biological Body, Tissue and Cell”, 2004 URSI International

Symposium on Electromagnetic Theory, Italy, May 2004. (Invited paper)

21. C.C. Chiau, X. Chen, C.G. Parini, “EBG Structure on Substrate and Ground

Plane for Ultra-Wide Stopband Circuits”, 27th ESA Antenna Technology

Workshop on Innovative Periodic Antennas: Electromagnetic Bandgap, Left-

handed Materials, Fractal and Frequency Selective Surfaces, Spain, 8-11 March

2004.

22. C.C. Chiau, X. Chen, C.G. Parini, ‘A Multi-period EBG Structure on Substrate

for Ultra-wide Stopband Circuits’, IEE Seminar on Metamaterials for Microwave

and (Sub) Millimetre Wave Applications: Photonic Bandgap and Double

Negative Designs, Components and Experiments, DTI, London, U.K, 24th Nov.

2003.

23. C.C. Chiau, X. Chen, C.G. Parini, ‘A Microstrip Patch Antenna on the

Embedded Multi-Period EBG Structure’, ISAPE’03, Beijing, China, pp. 96-99,

1-5 Nov, 2003.

Publications: 138

24. Y. Alfadhl, C.C. Chiau, Z. Wang, X. Chen, C. Gabriel and A. Peyman,

“Evaluation of electromagnetic fields interactions with animals as a function of

tissue dielectric properties”, BEMS’03, Hawaii, USA, 22-26 June, 2003.

25. C.C. Chiau, X. Chen and C.G. Parini, “A multi-period EBG structure for

Microstrip Antennas”, 12th International Conference on Antennas and

Propagation, Exeter, UK, 31 April - 3 May 2003.

26. Y. Alfadhl, C.C. Chiau, Z. Wang, X. Chen, C. Gabriel and A. Peyman,

“Numerical Dosimetry on 10, 30, and 70 days old rat models exposed to a range

of RF: influences of model size, tissue dielectric properties, and direction of

exposure”, IoP/IEE International Conference on RF Interaction with Humans:

Mechanism, Exposure and Medical Applications, London, UK, 27-28 February,

2003.

27. Y. Alfadhl, C.C. Chiau, Z. Wang, X. Chen, and C. Gabriel, “Assessment of

SAR Distribution inside 10 and 70 days Old Rats Exposed to 900 MHz RF”,

International Symposium on Antennas and Propagation –i02, Yokosuka

Research Park, Japan, 26-28 November 2002.

Appendix A 139

Appendix A

Electromagnetic (EM) Numerical Modelling

The technology of wireless communications is established on the principles

of electromagnetic (EM) fields and waves. It is facilitated by the underlying concept

of EM radiation, which is mediated by antennas and associated wave propagation.

The basis of EM theory is based on the relationship between the electric and

magnetic fields, charges and currents. In 1886, James C. Maxwell has assembled the

Faraday’s Law, Ampere’s Law, Gauss’ Law and magnetic field law into a set of

equations which form the basis of EM theory [1]-[3]. The Maxwell’s equations can

be written in the differential form:

BEt

∂∇× = −

∂ Faraday’s Law (A.1)

0B∇⋅ = Magnetic Field Law (A.2)

DH Jt

∂∇× = +

∂ Ampere’s Law (A.3)

D ρ∇⋅ = Gauss Law (A.4)

and the equivalent integral form are:

BE d s d At

∂⋅ = − ⋅

∂∫ ∫ Faraday’s Law (A.5)

0B d A⋅ =∫ Magnetic Field Law (A.6)

DH d s J d At

⎛ ⎞∂⋅ = + ⋅⎜ ⎟∂⎝ ⎠

∫ ∫ Ampere’s Law (A.7)

D d A dVρ⋅ =∫ ∫ Gauss Law (A.8)

Appendix A 140

In addition to the above four Maxwell’s equations, there are three material

equations

where

E is the electric field intensity (v/m)

H is the magnetic field intensity (A/m)

D is the electric flux density

B is the magnetic flux density

J is the electric current density (A/m²)

σ is the electric conductivity (S/m)

rεεε 0= Electrical permittivity (F/m)

rμμμ 0= Magnetic permeability (H/m)

Numerical techniques have been used to solve EM field problems especially

when the problems’ complexity increases. There are a number of numerical

techniques available to solve EM problems e.g. Finite Element method (FE) and

Method of Moments (MoM) solve the EM problems in frequency domain whilst

Finite-Difference Time-Domain method (FDTD) and Finite Integration Technique

(FIT) solve the EM problems in time domain instead. Each numerical technique is

well suited for the analysis of a particular type of problem. Different numerical

software packages have been compared and it has shown that FDTD/FIT is fast in

computation and the resolution is better than other available numerical software

packages [4]. Therefore, CST Microwave Studio® which is based on the FIT

numerical method is used in this thesis.

Finite Integral Technique (FIT)

The FIT was first proposed by T. Weiland in 1977 as a time-domain

numerical technique for solving Maxwell’s equations in their integral form [5]. FIT

D Eε= (A.9)

B Hμ= (A.10)

J Eσ= (A.11)

Appendix A 141

is equivalent to the FDTD, however, it discretises the integral form rather than the

differential form of Maxwell’s equations.

The first step of the FIT discretisation is to define the computation domain

which contains the space region of interest. The computation domain is enclosed by

the restriction of the electromagnetic field problem, which normally represents an

open boundary problem to a bounded space region. The next step is the

decomposition of the computation domain into a finite number of the simplicial cell

complex G, which serve as a computational grid. The primary grid G can be

visualised in the CST Microwave Studio®, whilst internally a dual grid G is set up

orthogonally to the primary grid. In the Cartesian system, the dual grid G is defined

by taking the foci of the cells of G as grid points for the mesh cells of G as shown in

Figure A-1. The electric voltages e and magnetic fluxes b are allocated on the

primary grid G whilst the dielectric fluxes d and the magnetic voltages h are

allocated on the dual grid G . A voltage is defined as the integral of a field strength

value (electric or magnetic) along a (dual) mesh edge whilst a flux is defined as the

integral of a flux density value (electric or magnetic) across a (dual) mesh cell facette.

A cell of Dual Grid G

A cell V of Grid G Computation Domain

Grid G

V

Figure A-1: Illustration of the FIT discretization.

Faraday’s Law

By considering a single cell V of the grid G as shown in Figure A-2, the

integration form of Faraday’s Law (A.5) can be rewritten for a facet An as a sum of

four grid voltages as shown in Figure A-2:

i j k l nde e e e bdt

+ − − = − (A.12)

Appendix A 142

where the scalar value e is the electric voltage along one edge of the surface An,

representing the exact value the integral over of the electric field along this edge.

The scalar nb represents the magnetic flux though the cell facet An.

ej

ek

ei

elbn

An

Figure A-2: A cell V of the grid G with the allocation of the electric grid voltage e on the edges of An and the magnetic facet flux bn through this surface.

The differential approximation can be represented as a process of storing the

factors -1,1:

Therefore, the discrete form of Faraday’s Law can be expressed in the general

form:

where C is a matrix coefficient C ∈-1,0,1 which contains the incidence relation of

the cell edges within G and on their orientation.

The Magnetic Law

The second Maxwell’s equation to be considered is the magnetic law’s

equation (A.6). For a cell V as shown Figure A-3, the non-existance of magnetic

charges in the cell volume can be evaluated in the following:

(1 1 1 1)C = − − (A.13)

dCe bdt

= − (A.14)

1 2 3 4 5 6 0b b b b b b− + − + − + = (A.15)

Appendix A 143

Figure A-3: A cell V of the grid G with the allocation of the six magnetic facet fluxes which have to be considered in the evaluation of the closed surface integral for the non-existance of magnetic charges within the cell volume.

Again the relation in (A.15) can be expanded to all the available cells and

expressed in a general form as:

where S is a matrix which contains the incident relation of the cell facet, representing

the discrete divergence-operator for grid G.

Ampere’s Law

The discretisation of the two remaining Maxwell’s equations (A.7)and (A.8)

using the FIT requires dual grid G which is dual to the primary cell complex G as

shown in Figure A-1.

On a facet A of a dual grid cellV , the summing of the magnetic grid voltages

to obtain the displacement current and the conductive current through the facet can

be rewritten as follows:

0Sb = (A.16)

1 2 3 4 ndh h h h d jdt

+ − − = + (A.17)

Appendix A 144

h2

h3

h1

h4dn

Figure A-4: A cell V of the grid G with the allocation of the magnetic grid voltage h on the edges of nA and the electric facet flux dn through this surface.

The discrete matrix equation for the relationship in (A.17) is:

where C is a matrix which contains the incident relation of the cell edges within

G and their orientation.

Gauss’s Law

The integral form of Gauss’s Law (A.8) can be discretised for the dual grid

cells and its discrete matrix form is:

where S is a matrix which contains the incident relation of the cell facet,

representing the dual discrete divergence-operator for grid G .

Maxwell’s Grid Equations (MGE’s)

It has been shown above that in the FIT discretisation, the integral form of

Maxwell’s equations (A.5) - (A.8) is transformed into a complete set of discrete

matrix equations (A.14), (A.16), (A.18) and (A.19), termed the Maxwell Grid

Equations (MGE’s). It can be shown that the curl (C, C ) and divergence (S, S )

matrices from the MGE’s have the following properties:

dCh jdt

d= + (A.18)

S qd = (A.19)

Appendix A 145

The relations in equations (A.20) and (A.21) have ensured that there is no electric or

magnetic charges arise during the computation due to the numerical algorithm.

Finally, the material equations (A.9)-(A.11) are also expressed in terms of

material matrices Mε, Mμ and Mσ as shown below:

Advanced techniques in CST Microwave Studio®

The most common disadvantage of the FIT in three dimensional is the usage

of Yee-type Cartesian grids [6]. The standard gridding scheme introduces errors to

the geometry representation of the curved structure surface due to the staircase

approximation as shown in Figure A-5(a). In order to reduce the errors, a fine mesh

is usually introduced around the curved surface. However, it leads to an overall fine

mesh in the whole structure. Therefore, sub-gridding technique has been introduced

where it is more efficient to refine the mesh density only within the desired area (e.g.

curve surface) instead of the whole structure as shown in Figure A-5(b). Figure

A-5(c) shows the triangular filling, which is another approach that has been

introduced to overcome the geometry approximation problem. However, most of

these techniques have stability problems or low efficiency.

A more accurate and efficient technique termed Perfect Boundary

Approximation (PBA®) as shown in Figure A-5(d) has been implemented in the

commercial EM modelling package, CST Microwave Studio® [7][7]. Using this

technique the computational grid does not have to conform to the curved

surface/boundaries. Instead, the sub-cellular information is taken into consideration

0SC = (A.20)

0SC = (A.21) T

C C= (A.22)

d M eε= (A.23)

b M hμ= (A.24)

j M eσ= (A.25)

Appendix A 146

resulting in an algorithm with second order accuracy for arbitrary shaped boundaries.

Unlike other techniques, PBA only requires slightly higher numerical cost during the

iteration. The algorithm of PBA has never been published by CST due to

commercial reasons.

However, PBA can only define one field value within PEC partially filled

cells. There is still fine mesh to be defined in the thin PEC region of the structure. As

such, Thin Sheet TechnologyTM (TST) has been introduced in the CST Microwave

Studio® to solve the problem. It is possible for TST to handle two different field

values within one FDTD cell as shown in Figure A-6.

Original Object inside Cartesian grids

Coventionally filled cells

`

PBA filled cells

(c) (d)

(a) (b)

Figure A-5: Grid approximation of rounded boundaries: (a) standard (stair case), (b) sub-gridding, (c) triangular and (d) Perfect Boundary Approximation (PBA).

Appendix A 147

PEC Partially filled cells – Perfect Boundary Approximation (PBA)

Figure A-6: TST technique.

References:

[1] David K. Cheng, Field and Wave Electromagnetics, 2nd edition, Addison Wesley.

[2] Matthew N.O. Sadiku, Numerical Techniques in Electromagnetics, 2nd edition, CRC Press LLC.

[3] John D. Kraus and Daniel A. Fleisch, Electromagnetics with Applications, 5th edition, McGraw-Hill.

[4] Z. Wang, Design of Low-SAR Antennas for Mobile Communications Devices, PhD Thesis, 2001.

[5] M. Clements and T. Weiland, “Discrete electromagnetism with the Finite Integration Technique”, Progress In Electromagnetics Research, pp. 65-87, 2001

[6] K.S. Yee, “Numerical solution of initial boundary value problems in isotropic media”, IEEE Trans. On Antennas and Propagation, vol. 14, pp. 302-307, 1996.

[7] B. Krietenstein, R. Schuhmann, P. Thoma and T. Weiland, “The Perfect Boundary Approximation technique facing the challenge of high precision field computation”, Proceeding of the XIX International Linear Accelator Conference (LINAC’98), pp. 860-862, Chicago, 1998.

(a) PBA example: each cell consists of single non-PEC area

(b) TST example: each cell can consist of two non-PEC areas

Appendix B 148

Appendix B

3-D Radiation patterns of the proposed diversity antenna array

The computed 3D radiation patterns of the proposed four-element diversity

antenna array used in Chapter 5 is shown in Figure B-1.

1

23

4

Figure B-1: Computed 3D radiation patterns of the proposed four-element diversity antenna array.

Appendix C 149

Appendix C

Material properties used in the indoor environment

In Chapter 5, the indoor environment (i.e. second floor of the Department of

Electronic Engineering at QMUL) was modelled using Wireless InSite. Three

different materials (i.e. concrete, wood and glass) have been used to model the

indoor environment and all materials are assumed to be homogenous. Concrete walls,

ceiling and floors were used in the model. Wooden doors and glass windows are

also modelled as shown in Figure C-1. The properties of the materials are

summarised in Table C-1[1].

Figure C-1: 3-D schematic diagram of the indoor environment (i.e. second floor of the Department of Electronic Engineering building at QMUL) modelled using Wireless InSite as used in this thesis. The transmitter is placed at the corridor and the receivers (RED dots) are scattered randomly in Rooms A and B. The ceiling has been removed for visual purpose.

Appendix C 150

Material Permittivity Conductivity

Walls, floor and ceiling (Concrete) 5.5 0.3

Door (Wood) 2.5 0.03

Window (Glass) 6.0 0.04

Table C-1: Material parameters used in the model at 5.2GHz.

References:

[1] F. Layer, R. Kattenbach and H. Fruchting, “Modelling and anaylsis of dominant propagation effects in real indoor environment at 5.2GHz,” The ninth IEEE International Symposium on Personal, Indoor and Mobile Radio Communications, vol. 2, pp. 630-634, Sept. 1998.

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