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Electronics Projects chosen for approval by P. MUZHUMADHI

B. Tech – ICE Dept

Advanced LED temperature indicator Project Option I

APPLICATION OF THIS PROJECT

This project illustrates the use of a V/F converter in monitoring temperature in degrees Fahrenheit (0F). The

block diagram of the temperature indicator is shown in Figure 1-1. The indicator is composed of a

temperature sensor, amplifier, V/F converter, three-digit binary-coded-decimal (BCD) counter, time base, and

LED display In addition to the 9400 V/F converter, other ICs needed for this project include the LM334

temperature sensor, LF353 dual op-amp, NE555 timers, 74LS00 NAND gate, MC14553 three-digit BCD

counter. MC14543 BCD-to-seven segment decoder/driver/latch and three seven-segment (common anode or

common cathode) LED displays with three PNP switching transistors.

Working of the system

Figure 1-2 shows the schematic diagram, which is designed to display temperatures from 0° to

100°F. Operation of the circuit is as follows. The output of the temperature sensor changes linearly

as a function of temperature (10 mV/ K). This output is an input to the summing amplifier, which is

used to calibrate the output of the temperature sensor for a desired temperature type (K, 0C, or 0F)

and an intended range. That is, to display the temperature in either K, °C, or 0F, potentiometer R4 is

adjusted accordingly so that a suitable voltage appears at the output of the summing amplifier. Since

the output of the temperature sensor is directly proportional to temperature changes-, R4 needs to be

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adjusted at only one temperature. The output of the summing amplifier then drives the inverting amplifier. The purpose of the inverting amplifier is twofold: (1) to invert the input so that its output voltage is positive,

which is necessary for the V/F converter, and (2) to provide a suitable gain, which depends on the voltage-to-frequency

scaling used for the V/F converter.

The output of the inverting amplifier is the input to the V/F converter; therefore, the output frequency of the converter is

directly proportional the output voltage of the inverting amplifier. For example, as the temperature goes up the output

voltage of the summing amplifier increases in the negative direction, Whereas that of the inverting amplifier increases in

the positive direction, which in turn causes the frequency of the V/F to increase in the positive direction.

The output frequency of the converter is then ANDed with the gating signal to produce the clock signal for the three-digit

BCD counter. The BCD output of the counter drives the three LED displays sequentially via the BCD-to-seven segment

decoder/latch/driver stage, and the temperature is displayed on the LEDs, depending on the relationship between the

frequency of the V/F converter and the gate signal. The gate, latch, and reset signals are generated by the time-base

circuit, which consists of a free-running multivibrator and two one-shot multivibrators.

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PARTS LISTS Resistors (all ¼-watt, ± 5% Carbon)

R1 = 1 kΩ potentiometer at 230 Ω

R2, R7, R12, R13, R21 = 10 kΩ

R3, R5, R6, R11 = 100 kΩ

R4 = 10 kΩ potentiometer

R8= 1 MΩ potentiometer

R9 = 180 kΩ

R10, R15 = 50 kΩ potentiometer

R14 = 510 kΩ

R16 = 3 kΩ potentiometer

R17 = 15 kΩ

R18 = 20 kΩ

R19 = 10 kΩ potentiometer

R20 = 1 kΩ potentiometer

R22 – R28 = 220 Ω

R29 – R31 = 1k Ω

Capacitors

C1 = 1000 pF

C2 = 100 pF

C3, C6, C9 = 1 µF

C4, C5, C7, C8, C10 = 0.01 µF

C11 = 0.001 µF

Semiconductors

IC1 = LM334 temperature sensor

IC2 = LF353 dual op-amp

IC3 = Teledyne 9400 V/F convertor

IC4 = 74LS00 NAND gate

IC5, IC6, IC7 = NE/SE 555 timers

IC8 = MC 14553 three-digit BCD counter

IC9 = MC 14543 BCD-to-seven segment decoder/driver/latch

Q1, Q2, Q3 = 2N1305 switching transistors

D1, D2 = 1N914 signal diodes

Three seven-segment common anode LEDs: MAN72A or equivalent

Circuit Description

Next let us examine the design considerations and procedures for each of the sections in the

temperature indicator of Figure l-2. The temperature sensor LM334 is a three-terminal adjustable

current source whose current can be programmed from 1µ to 10 mA with one external resistor R1.

The three terminals are labeled +V, R, and –V. The pin diagram of the LM334 is shown separately in

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Figure 1-3.

The LM334 has a wide operating voltage range of 1 to 40 V. It can also withstand reverse voltage of

up to 20 V(terminal + V is negative with respect to –V). It is designed to operate over a temperature

range of 0° to 70°C. For a wider temperature range, such as -550 to 150°C, Intersil’s AD590

temperature sensor is recommended.

For the values indicated in Figure 1-2, the output of the LM334 changes 10 mV/K. This means that

at 00F = 255.22 K the output of the sensor will be 2552.2 mV, which must be scaled down to 0 V so

that the temperature displayed will be in degrees Fahrenheit. This is accomplished by the use of the

summing amplifier. Specifically, potentiometer R4 of the summing amplifier is adjusted so that the

output is 0 V. The same procedure is used to calibrate the output of the summing amplifier at any

other value of 0F. Table 1-1 shows the relationship between K, °C, 0F, and the output of the

temperature sensor and the summing amplifier at corresponding values of temperature. Because the

output of the sensor directly proportional to the temperature, the output of the summing amplifier

needs to be calibrated at the temperature at which the circuit is initially started up (refer to Table 1-

1).

Table 1-1

RELATIONSHIP BETWEEN DIFFERENT TEMPERATURE UNITS AND OUTPUTS OF THE SENSOR

AND SUMMING AMPLIFIER

Kelvin (K) Degrees Celsius Degrees Fahrenheit Output of the

temperature sensor

(mV)

Output of the

summing amplifier

(mV) to be adjusted

to

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255.22 -17.78 0 2552.2 0

273 0 32 2730 -177.8

298 25 77 2980 -427.8

310.78 37.78 100 3107.8 -555.6

Note that the output of the summing amplifier is a negative dc voltage since the net input voltage is

always positive for temperatures >00 (see Table 1-1). However, the 9400 V/F requires a positive

input voltage. The summing amplifier must be therefore be followed by an inverting amplifier. The

gain of the inverting amplifier, however, depends on the voltage-to-frequency scaling of the

convertor. The V/F convertor of figure 1-2 is calibrated for the maximum frequency of 50 KHz,

which represents a temperature of 1000F when the input voltage is 10 V maximum. Since the output

of the summing amplifier is -555.6 mV at 1000F, the gain of the inverting amplifier must be equal to

10V/555.6mV = 17.9985

The output frequency of the V/F converter is then ANDed with the output frequency (called the gate

signal) of the 555 free-running multivibrator to produce the clock signal for the three-digit BCD

counter. Since the maximum 50-kHz output frequency of the converter represents 1000F, the three-

digit BCD counter must be clocked 100 times to display I00°F. To accomplish this, the pulse width of

the free-running multivibrator must be 100/50 K = 2 ms so that 100 pulses will be produced in 2

ms. At the end of 2 ms, the count of the counter is latched and displayed on the LED display. After

the count is displayed as a temperature on the LEDs, the BCD counter is reset and the cycle repeats.

In other words, the counter continuously cycles through three states: count, latch, and reset.

Therefore, the free-running multivibrator (gate signal) must provide for the time period required to

count, latch, and reset the BCD counter. The latch enable and master reset pulses for the BCD

counter MCl4553 are produced by using two 555 one-shot multivibrators. Where the time period of

the free-running multivibrator is approximately 12.5 ms with a pulse width of 2 ms. The pulse width

of the latch enable pulse is approximately 10 ms, and the master reset pulse width is approximately

0.5 ms. To accomplish 2-, 10-, and 0.5-ms pulse widths, adjust potentiometers R16, R18, and R20,

respectively (Figure l-2).

The three-digit BCD counter used in Figure 1-2 is the MC14553. The MCl4553 consists of three

negative-edge-triggered BCD counters with a quad latch at the output of each counter, which

enables the storage of any given count. The outputs of the latches are time multiplexed so that one

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BCD digit at a time is produced. The operation of the multiplexer output selector is controlled by the

on-chip oscillator, whose frequency depends on the external capacitor.

As shown in Figure 1-2, the master reset (MR) and latch enable (LE) pulses for the MC14553 are

produced by using two 555 one-shot multivibrators. In addition, the clock signal for the MC14553 is

produced by ANDing the output of the V/F converter with the gating signal, which is obtained by

using the 555 free-running multivibrators. The digit select outputs DS1, DS2, and DS3 sequentially

drive the 2N1305 PNP transistors T1, T2, and T3, which in turn control the three LED displays. The

BCD outputs of the MCl4553 are connected to the BCD inputs of the MCl4543, which is a BCD-to-

seven segment latch/decoder/driver. The seven-segment outputs of the MCl4543 then drive the

seven segments of the LED selected by the digit-select of the MC14553.

The MCl4543 is designed to provide three functions: a 4-bit storage latch, an 8421 BCD-to-seven

segment decoder, and a driver. The device is capable of driving LCD and LED displays.

The PH pin 6 of the MC14543 is connected to VDD (logic 1) because the LED displays are the

common anode type. To limit the current through each of the LED segments, a separate resistor is

used in series with each segment. Note, however, that only seven resistor are required for all the

segment of LEDs. This is possible because the digit-select output of the MC14553 function

sequentially. In addition, the BCD output are also multiplexed, one BCD digit at a time.

Finally, since the accuracy of the temperature displayed depends mainly on the frequency stability of

the free-running multivibrator, all resistors must be of 5% or better tolerance, and capacitor must be

either Mylar of tantalum types. Also remember that the temperature indicator must be calibrated at

a temperature at which it is initially turned on.

Digital Mains Failure/Resumption Alarm Project Option II

APPLICATION OF THIS PROJECT

AC mains fails when over load is connected and this problem is common in now days. Here is the

simple circuit using optocoupler Digital Mains failure and resumption alarm, for indicating AC mains

fails or resumes by producing alarm sound.

Circuit Description of digital mains failure alarm

The circuit digital mains failure alarm is built around optocoupler. The resistor R1, capacitor C1 & C2,

with diode D1 &D2 provide sufficient voltage to glow internal LED of optocoupler. Here the IC2

CD4011 is used as oscillator to generate low frequency of 0.662 Hz to 1.855 KHz controlling with

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preset VR1. Audio sound is generator by timer IC NE555 (IC2). The generated frequencies from IC2

vary from 472 Hz to 1.55 KHz controlling with preset VR2. For sensing mains fails position of switch

SW1 to point 1 and for sensing mains resumption change the position of switch SW1 to point 2.

PARTS LIST

Resistors (all ¼-watt, ± 5% Carbon)

R1, R4 = 1 KΩ

R2, R5 = 10 KΩ

R3 = 22 KΩ

VR1 = 50 KΩ

VR2 = 47 KΩ

Capacitors

C1 = 0.22 µF

C2 = 1 µF/16V

C3, C4 = 10 µF/16V

C5 = 0.04 µF

C6 = 0.01 µF

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C7 = 100 µF/16V

C8 = 470 µF/16V

Semiconductors

IC1 = MCT2E (optocoupler)

IC2 (N1-N3) = CD4011

IC3 = NE555 (Timer IC)

D1, D2, D3 = 1N4001

Miscellaneous

SW1 = SPDT (Single Pole Double Throw) Switch

SW2 = ON/Off Switch

LS1 = 8Ω/0.5W

9V Battery

Circuit Description of twilight lamp blinker Project Option III

APPLICATION OF THIS PROJECT

The entire circuit of twilight lamp blinker is designed and fabricated around LDR (Light Detector

Resistor) and IC CD4093 (IC1). The preset VR1 is used to control brightness. For sensor LDR1 is used

that has a high resistance during night (i.e. dark) and a low resistance at day time (i.e. light). The

NAND gates (N3 and N4) of IC1 is used as oscillator where high input from NAND gate (N1) makes

the output of NAND gate (N2) low and vice-versa. The high at NAND gate N2 result LED1 blinks by

conducting transistor T1 where transistor T1 is the LED driver transistor. For more brightness more

LEDs is connected parallel to LED1.

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PARTS LIST

Resistors (all ¼-watt, ± 5% Carbon) R1 = 10 KΩ/10W

R2 = 1 MΩ/1W

R3 = 100 KΩ

R4 = 100 Ω

VR1 = 100 KΩ

Capacitors

C1 = 0.68 µF/400V

C2 = 100 µF/40V

C3 = 10 µF/35V

Semiconductors

IC1 (N1 – N4) = CD4093

T1 = BC547

D1, D2 = 1N4007

ZD1 = 5.6V/1W

ZD2 = 15V/1W

Miscellaneous

LED1 = Blinker

LDR1 = Light Detector Resistor

Battery = 4.8 V/500 mAh battery pack

SW1 = SPST (Single Pole Single Throw)

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Power Supply for Integrated circuit (ICs) and Microprocessor Project Option IV

APPLICATION OF THIS PROJECT

Since a power supply is a vital part of all electronics systems. Most digital ICS, including

microprocessor and memory ICS, operate on a ± 5-V supply, while almost all linear ICS (op-amps and

special-purpose ICS) require ± 15-V supplies. Therefore, the power supply presented in this section

will have ±5 and ±15 V.

Working of the system

Figure 1-1 shows the block diagram of a typical power supply. The schematic diagram of the power

supply that provides output voltage of ±5-V at 1.0A and ± 15 V at 0.500 A is shown in Figure 1-2. In

this figure two separate transformers are used because they are readily available; however, it is

possible to custom design a single transformer with the same specifications to replace the two. The

supply voltages are obtained from a 26.8-V center-tapped (CT) transformer, and the supply voltages

are obtained from the 12.6-V CT transformer. The output of these secondaries is then applied to the

bridge rectifiers, which convert the sinusoidal inputs into full-wave rectified outputs. The filter

capacitors at the output of the bridge rectifiers are charged to the peak value of the rectified output

voltage whenever the diodes are forward biased. Since the diodes are not forward biased during the

entire positive and negative half-cycle of the input waveform, the voltage across the filter capacitors

is a pulsating dc that is a combination of do and a ripple voltage. From the pulsating dc voltage, a

regulated dc voltage is extracted by a regulator IC.

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Circuit Description

Consider first how the ± 15-V supply voltages are obtained in the circuit of Figure 1-2. The 7815 is a

+15-V regulator, the 7915 is a -15-V regulator, and both can deliver output current in excess of 1.0

A, They will hence perform satisfactorily in the circuit of Figure 1-2 by providing ±15 V at 0.500 A.

However, since the drop-out voltage (Vin – V0) is 2 V, the input voltage for the 7815 must be at least

+17 V and that for the 7915 must be at least — 17 V. This means that the rectified peak voltage

must be greater than +17 V and — 17 V, which in turn implies that the secondary voltage must be

larger than 34 V peak or 24 V rms. The voltage across the center-tapped secondary in Figure 1-2 is

26.8 V rms, thus satisfying the minimum voltage requirement of 24 V rms. Also, the peak voltage

between either of the secondary terminals and the center-tap (ground terminal) is 18.95 V peak,

which is less than the maximum peak voltages of +35 V and -35 V for the 7815 and 7915,

respectively.

Note that the voltages across the two halves of the center-tapped secondary are equal in amplitude

but opposite in phase. During the positive half-cycle of the input voltage, diode D1 conducts and

capacitor C1 charges toward a positive peak value =18.95 V. At the same time, diode D3 is also

conducting; hence capacitor C3 charges toward a negative peak value = -18.95 V. This means that

the voltage across nonconducting diodes D2 and D4 is 37.90 V peak, which implies that the peak-

reverse-voltage (PRV) rating of the bridge rectifiers must be larger than 37.90 V peak or 26.8 V rms.

The PRV rating of the bridge rectifier diodes, also known as a working inverse voltage (WIV), is

specified on the data sheets. The bridge rectifier, MDA200 (Mot0r0la’s rectifier) in Figure 1-2, has a

PRV rating of 50 V, which is higher than needed. This bridge rectifier is, in fact, used here because it

is readily available and more commonly used.

During the negative half-cycle of the input waveform, diodes D2 and D4 conduct and charge

capacitors C1 and C3 toward the peak voltage of 18.95 V with indicated polarities. Note, however,

that the diode pair that conducts during either the positive or negative half-cycle does not do so for

the entire half-cycle. The diodes conduct only during the time when the anodes are positive with

respect to the cathodes. In other words, when the diodes are forward biased, the capacitors are

charged by current pulses. Data sheets give the maximum average rectified current I0max that the

diode can safely handle. For the MDA200,Iomax is 2.0 A. In addition, when the power supply is first

turned on, the initial charging of the capacitor causes a large transient current called the surge

current to pass through the diodes. The surge current IFS flows only briefly and is therefore much

larger than the maximum average current I0max. The maximum surge-current IFSM is normally

included on the data sheets; it is 60 A for the MDA200.

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Finally, the size of the filter capacitor depends on the secondary current rating of the transformer. As

a rule of thumb, a 1500-µF capacitor should be used for each ampere of current. The working

voltage rating (WVDC) of the capacitor, on the other hand, depends on the peak rectified output

voltage and must be at least 20% higher than the peak value of the voltage it is expected to charge

to. Capacitors C1 and C3 satisfy these requirements (see Figure 1-2). Capacitors C2 and C4 at the

output of 7815 and 7915 regulators, respectively, help to improve the transient response and should

be in the range of 1µF.

Next consider the ±5-V supply. The circuit arrangement of the ±5-V supply is identical to that of the

±15-V supply except that here the specifications for the transformer T2 secondary are different.

Therefore, the operation and considerations for the ±5-V supply are the same as those presented for

the ±15-V supply.

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The voltage regulators in Figure 1-2 will require heat sinks. Let us examine why. The power

dissipated by the 15-V regulators is as follows:

Power dissipated = (dropout voltage) (current) = (18.95 — I5) (0.5) = 1.98 W

Similarly, the power dissipated by the 5-V regulators is (8.91 – 5)(1.0) = 3.91 W

Therefore, for the proper operation the regulators must be heat-sinked in order to keep their

temperature down. If a regulator is a metal package (TO-3 type), the appropriate heat sink is

mounted on the case of the package. However, if the regulator is an epoxy package, silicon grease

may be used on the back of the package, and then the package can be bolted to the chassis of the

power supply cabinet with insulating hardware.

Besides the ±15 and ±5-V regulated supply voltages, there is often a need for a 60-Hz square-wave

signal, which is used as a time base in scanning the digital displays and as a trigger for sequential

and timing circuits. If needed, a 1-Hz (1-s) signal for the real-time clock can be readily obtained

from the 60-Hz signal by using a divide-by-60 network. Although not commonly done, a higher-

frequency Signal can also be obtained from the 60-Hz signal by using e multiplier. For these reasons,

in Figure 1-2 a 60-Hz square-wave signal is produced by using two small-signal diodes and a 555

timer as the Schmitt trigger

PARTS LISTS

Resistors (all ¼-watt, ± 5% Carbon)

R1 = 10 kΩ

Capacitors

C1, C3 = 1500 µF

C2, C4, C6, C8 = 1 µF

C5, C7 = 3000 µF

Semiconductors

IC1 = MC7815

IC2 = MC7915

IC3 = MC7805

IC4 = MC7905

IC5 = NE555 timer

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D1 – D2 =MDA200 PVR = 50 V, I0max = 2.0 A, IFSM = 60 A

D5 – D8 = MDA970A1 PVR = 50 V, I0max = 4.0 A, IFSM = 100 A

D9, D10 = 1N914 signal diodes

Miscellaneous

Transformer T1 = Primary: 117 V, 60 Hz: Hobart P-300

Secondary: 26.8 V CT, 1.0 A

Transformer T2 = Primary: 117V, 60 Hz: Hobart P-305

Secondary: 12.6 V CT, 2.0 V

Fuse 0.750 A slow blow

Switch On-off toggle type

Silicon grease with insulating hardware or four het sink for Voltage regulator

Electronics Cricket on board Project Option V

This is very interesting project not for only who love cricket for also who love to watch this game

because thousands of us want to play this game but some time it is not possible because of this busy

life, lack of ground etc. The game electronics cricket on board can be played by anyone even in their

home by sitting on table. The interesting fact is even a single player can play this game.

Circuit description of electronics cricket on board

This electronics game circuit cricket on board is design by most popular IC LM555 and decade

counter IC CD4017 from CMOS family. IC1 555 timer IC, forms the heart of circuit used as clock

pulse generator. Generated clock pulse is fed to pin 14 of IC2 CD4017. Output is obtained from pin

number 3, 2, 4, 7, 10, 1, 5, 6, 9, 11 by connecting LED to each pin as shown in circuit diagram. To

play this game switch SW1 are placed in on position (or pushed). All 10 LEDs are in on mode. But

when we release switch SW1 last pulse only lit up one LED which is the game result. Now compare

the result of cricket on board to chart and write your score in score boards, lastly count your all run.

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PARTS LIST

Resistors (all ¼-watt, ± 5% Carbon)

R1, R2, = 10 KΩ

Capacitor

C1 = 0.1 µF/50V

Semiconductors

IC1 = NE555 Time IC

IC2 = CD4017

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GAME RESULT

Catch Out LED 1

Stamp Out LED 2

Bold Out LED 3

Leg By LED 4

Single Run LED 5

Two Run LED 6

Four Run LED 7

Sixer LED 8

Wide Ball LED 9

No Ball LED 10

Electronics Lucky Number Project Option VI

The circuit is simple and self-explanatory. IC1, a NE555 timer, is used as astable mode to generate

clock pulses at a rate of about 20 Hz. The frequency can be varied with the help of potentiometer

VR1.

The clock pulses from the timer are fed to clock input of IC2, a decade counter. The outputs of the

counter are decoded by IC3, (BCD to seven segment decoder/driver) which drives a common anode

display (DIS1) to show from figure 0 to 9.

On pressing switch SW1, IC1 starts working, with the display changing from 0 to 9. Capacitors C1 is

charged during this time.

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Once SW1 is released, the capacitor discharges through resistor R3, VR1, R1 and R2. Thus, the

frequency of IC1 decrease (and hence the rate of changing figures on the display) and ultimate

becomes zero, once C1 has totally discharged thereby stopping the displayed is enhanced by the time

for which SW1 is pressed. So, the random effect is natural outcome.

PARTS LIST

Resistors (all ¼-watt, ± 5% Carbon)

R1 = 100 KΩ

R3 = 1 KΩ

R4 – R10 = 330 Ω

VR1 = 100 KΩ

Capacitors

C1 = 47 µF/10V

C2 = 0.47 µF

C3 = 0.01 µF

Semiconductors

IC1 = NE555

IC2 = 7490

IC3 = 74247

D1 = 1N4001

DIS1 = FND507 or LTS542

Miscellaneous

SW1 = push to on switch

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Programmable electronics dice project Option VII

Dice game is very interesting indoor game mostly played in childhood. Here is verified game

project programmable electronics dice useful in many game. With the help of this project we can

display any number between 1 - 9 according to our dip switch setting

Circuit description of programmable electronics dice

The project programmable electronics dice comprises three ICs as heart and for output a common

anode display. Here, IC1 used is a dual 4-input Schmitt trigger NAND gate IC where gate N1 used as

frequency generator which generate the clock frequency of 70kHz with the help of resistor R2 and

capacitor C1 and gate N2 load data at the input of IC2, Where IC2 is a presettable binary counter with

the facilities of parallel loading. Lastly the output of IC2 is displayed on common-anode, 7-segmant

display with the help of IC3 which is BCD-to-7-segmant decoder and the resistor R8 is used as

current limiter.

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Setting for the 4-way DIP switch for display range

Dice Range Close the inner switch Open the inner switch

1 to 2 B and A D and C

1 to 3 C only A, B and D

1 to 4 A and C B and D

1 to 5 B and C A and D

1 to 6 A, B and C D only

1 to 7 D only A, B and C

1 to 8 A and D B and C

1 to 9 B and D A and C

PARTS LIST

Resistors (all ¼-watt, ± 5% Carbon)

R1 = 1 KΩ

R2 = 100 Ω

R3 – R7 = 4.7 KΩ

R8 = 220 Ω

Capacitor

C1 = 0.1 µF

Semiconductors

IC1 = 74LS13, dual 4-input Schmitt trigger NAND gate IC

IC2 = 74LS191, presettable binary counter with parallel facility

IC3 = 7474, BCD-to-7-segmant decoder

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Miscellaneous

DIS-1 = LTS542 common anode display of equivalent

SW1 = ON/OFF switch

SW2 = 4-way dip switch