IEEE COMMUNICATIONS MAGAZINE

Minimum-Shift Keyed Modem Implementations for High Data Rates RODGER E. ZIEMER AND CARL R. RYAN

An examination of methods for the implementation of modulators and demodulators suitable for high-data- rate applications.

IGITAL communications links capable of convey- ing data at hundreds of megabits per second (Mb/s) are becoming increasingly important in various

applications. An example is the use of time division multiple access (TDMA) transmission through a satellite relay wherein several data channels are routed through single uplink and downlink paths (antenna beams). Separation of the channels from each other is accomplished through interleaving and deinterleaving in time, with possible rerouting taking place in the satellite. When carrying traffic from large metropolitan areas, the required data rates can exceed 100 Mb/s. One such implementation, currently under study by NASA,' involves data transmission at rates of up to 550 Mb/s.

Such applications require the use of modulation schemes that use the available bandwidth efficiently. Furthermore, because power is at a premium on board a satellite, the utmost in power efficiency is desired of the chosen modulation scheme. This implies the use of a constant-envelope modulation scheme since amplifiers, such as traveling wave- tube amplifiers, are most efficient when operated near saturation.

Three popular constant-envelope modulation schemes for efficient transmission of digital data are quadriphase-shift keying (QPSK), offset (or staggered) quadriphase-shift keying (OQPSK), and minimum-shift keying (MSK). Two excellent recent articles [1,2] in IEEE Communications Magazine have discussed their general properties and attributes as well as those of other modulation schemes with constant (or nearly constant) envelopes. Other recent papers [3-61 have presented analytical results and computer

..

'This is NASA's Advanced Communications Satellite Program which involves development of Proof o f Concept hardware for several components of a TDMA common carrier satellite system. See Microwaue Systems News. vol. 10, pp. 19-26, Oct. 1980; or The National Telesystems Conference Record, pp. A2.4.1-A2.4.6. Nov. 1982.

simulations comparing the performances of these three modulation schemes in bandlimited and/or nonlinear chan- nels. Generally, the choice between QPSK, OQPSk, and MSK is not clearcut in environments typical of satellite or line-of-sight communications links. Each has its merits when system parameters are optimized for the environment of interest. The choice of a particular modulation technique in a given application will quite often involve considerations other than communications or bandwidth efficiency,Z such as ease of implementation. The objective of this paper is to examine methods for the implementation of modulators and demodu-

Fig. t. Parallef MSK modulator and demodukator structures. (a) Modulator (b) Demodulattir

I .

?See the inset for a discussion of communications and bandwidth efficiency.

0163-6804/83/1000-0028 $01.00 0 1983 IEEE

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(PliASE(ERd0R e, 2cos 2nt t+e)

IDEALLY ZERO) Fig. 2. lmpfementatlon of a serlal MSK modem utilking

bandpass conversion and matched filters. (a) modulator (b) demodulator. ’

lators for MSK signals that are suitable for high data-rate applications.

MSK Modulation and Demodulation The modulation and demodulation of MSK can be accom-

plished in two equivalent fashions, namely parallel and serial. We will concentrate on the serial technique in this paper, since it is advantageous for high data-rate implementa- tions. However, for completeness, parallel modulator and demodulator block diagrams are shown in Fig. 1. In parallel modulation of MSK (Fig. l(a)), the serial data stream, d(t), can be thought of as being “demultiplexed” into its even-and- odd-indexed bits to produce two bit streams, a,(t) and aa(t), which are staggered 1/2 symbol and then used to biphase modulate the signals x(t) and y(t) shown in the top figure. Each symbol of al( t ) and aa(t) is then weighted by a half- sinusoid. If the symbol weighting functions cos (7rt/2T) and sin (7rt/ZT) are replaced by rectangular shaping functions, MSK becomes staggered or offset quadriphase shift keying. Without staggering by 1/2 symbol and sinusoidal weighting, QPSK results.3

Because MSK is a quadrature-multiplexed modulation scheme, it can be optimally detected by coherently demodu-

%ee [ I ] for a more extensive discussion of the parallel approach to modem implementation for QPSK, OQPSK. and MSK.

‘0 fo .o.zs f + L E O T

f

(a) BPSK power spectrum

fo -0.7s f0+F f, i LE” f

(b) Frequency response of conversion filter-magnitude squared

f ~ ~ 0.75 Po f +0.75 0 - 7 -

f

(c) MSK power spectrum Fig. 3. Formation of an MSK spectrum by passing a BPSK

signal through a conversion filter

lating its inphase and quadrature components separately, as shown in Fig. l(b). Clearly, the quadrature channels of the modulator and demodulator must be time synchronized, amplitude balanced, and in phase quadrature to minimize overall system degradation. This becomes more difficult as the data rate increases. The serial method is an alternative approach to parallel modulation and demodulation of MSK which avoids these problems. It was first publicized by Amoroso and Kivett [7] in 1977, and later implemented at 760 Mb/s by Ryan, Hambley, and Vogt [8].

The serial modulation of MSK is somewhat more subtle to grasp than the parallel method. A serial modulator structure for serial MSK is illustrated in Fig. 2(a). I t is seen to consist of a biphase-shift keyed (BPSK) modulator with carrier frequency of f,, -1/4T Hz, and a bandpass conversion filter

I I Communications and Bandwidth Efficiency

Communications efficiency is defined as the signalsto-noise given BER, a point of reference being that a BER of ratio (SNR) required to achieve a given bit error rate (BER) requires an &/No of about 10.6 dB. In contrast, the bandwidth performance. In an additive white Gaussian noise (AWGN) efficiency of each, which is a measure of bandwidth occupancy of channel with single-sided noise spectral density No, the SNR is the modulated signal in b/s/Hz, differs considerably. Bandwidth often expressed as the ratio of the energy per bit, Eb, to No. efficiency is not as easily specified as communications efficiency, QPSK, OQPSK, and MSK all have the same communications for it depends on the bandwidth occupancy criterion used. See [Z] efficiency in AWGN in terms of the E b / N o required to achieve a for a discussion of bandwidth measures for digital data signals.

I 1

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with impulse response

g(f) = T sm 27r(f0 + 1/4T)t, O<t<T (1) r 0, otherwise

which corresponds to a (sin x)/x-shaped transfer function. W e will return shortly to a more complete discussion of why these operations produce an MSK modulated signal.

The serial demoddlator structure for MSK is essentially the reverse of the serial MSK modulator structure, and is illustrated in Fig. 2(b). It consists of a bandpass conversion filter followed by a coherent demodulator and lowpass filter which eliminates double frequency components at the mixer output. A major difference between the serial modulator and demodulator (other than the reversal of operations) is the matched filter, which has a transfer function proportional to the square root of the power spectrum of the MSK signal (with a linear phase response included). While the response of the matched filter to a single bit of duration T at the modulator input lasts for 2 T seconds, the response due to a preceding symbol is ideally zero at the optimum sampling instant for the present symbol, thereby producing zero intersymbol inter- ference [7].

Serial MSK modulation and demodulation have the advantage that 311 operations are performed serially, and therefore offer significant implementation tradeoffs at high data rates. The precise synchronization and balancing required for the quadrature signals of the parallel structures are no longer present. The critical system components now are the biphase modulator, the bandpass conversion and matched filters, and the coherent demodulator.

Although conversion and matched filters have been implemented with surface acoustic wave (SAW) devices, the maximum bandwidth of SAW devices is about 10-30% of the center frequency empioyed. For production SAW’S, center frequencies of a few hundred MHz represent the upper limit, assuming normal fabrication techniques, with 1 GHz representing the upper limit if laser trimming and other special

30

(P AND ,9 ARE ~nme ERRORS w n m ARE IDEALLI ZEROJ

(B)

Fig. 5. Implementation of a serial MSK modem utiiizin! lowpass I/Q equivalents for the conversion and niatchec filters. la) modulator (b) demodulator

techniques are used. Thus, the use of SAW filters implies an upper limit on data rate of about 100 Mb/s, assuming the use of normal fabrication techniques. We will examine alterna- tives to SAW implementations for serial MSK modems in this paper.

A frequency-domain proof of the validity of the serial modulation technique for MSK is provided by considering the product of the power spectrum of the BPSK signal (shown in Fig. 3(a)) and the magnitude-squared of the transfer function

OCTOBER 1983

. . - . ..

1.5; I1

Fig. 7. Frequency response functions tor the seriql MSU matched fitter.

of the conversion filter (Fig. 3(b)), whose center frequency is offset from the BPSK carrier by one-half data rate. The resulting MSK power spectrum is shown in Fig. 3(c). When mathematically expressed in terms of single-sided spectra, the BPSK spectrum is proportional to

SBpSK(f) = 2 sinc2 [ ( f - f , )T + 0.251 (2)

where sinc(x) = (sin T X ) / ( T X ) . The conversion filter transfer function is

C( f ) = sinc [ ( f - f , ) T - 0.251 exp (--jZ.rrft,) (3)

where t , represents a n arbitrary delay for the filter. The product of SBpSK(f) and I C(f) I can be simplified to

which is the result for the power spectrum of MSK [ 11. At the receiver, essentially the reverse process takes place,

except that the matched filter transfer function is proportional to the square root of the MSK power spectrum expressed by (4). We will discuss methods for approximating the transfer functions for the conversion and matched filters shortly, but we first discuss terminology and the concept of a trellis diagram. This will be convenient in a later discussion of carrier synchronization for serial MSK demodulators.

Terminology and Trellis Diagrams Terminology which has been used in the literature [9] for

the various frequencies appearing in (2)-(4) is the following:

0 Space frequency-the frequency fl = fo + 1/4T which corresponds to an alternating 1-0 data sequence.

That these data sequences do indeed give these frequencies will now be demonstrated with a convenient tool referred to as a trellis diagram. The trellis diagram of the excess phase of an angle-modulated signal is a useful way of visualizing the signal’s behavior for various data inputs. We define the instantaneous phase of any modulated signal as the argument, O(t), of the sine or cosine representing the signal. In general, the instantaneous phase of a modulated signal consists of two terms, that due to the carrier, 27rfOt, where f, is the carrier frequency in Hz; and that due to the modulation, 4(t), called the excess phase, The resultant modulated signal is

m(t) = A cos [w,t + 4(t)], wo = 2 d 0 . (5)

For MSK, assuming parallel modulation, the modulated signal can be written as

m(t) = A[a/(t) cos (7rt/2T) cos (27rf,t)

+ aQ(t) sin (7rt/2T) sin (27rf,t)] (6)

where a/(t) and aa(t) are the k 1 -valued data sequences for the I and Q channels. Using trigonometric identities, we can write (6) in the form (5) where

4(t) = -tan“ [aa(t) sin (7rt/2T) / al(t) cos (zrt/eT)]

--bk ( t ) (Tt/2T) + 4 b (7)

where bk( t ) = -a / ( t ) U Q ( ~ ) and @k = 0 or T corresponding

0 Apparent carrier-frequency fo at which the maximum

0 Mark frequency-the frequency fl = fo - 1/4T which of the signal spectrum occurs.

corresponds to an all 1’s or all 0’s data sequence.

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r IEEE COMMUNICATIONS MAGAZINE

FREOUEFICY

Fig. 9. Details on fabrication of quadrature hybrids.

to a / = 1 or - 1 . Therefore, if the signs of aQ( t ) and al(t) are the same (that is, d(t) is an all 1’s or all 0’s data sequence), the instantaneous phase of the carrier is’

e(t) = ~ ~ f , t - r t / 2 ~ = 27r(f0 - 1 / 4 ~ ) t (8)

which corresponds to a sinusoid of frequency f, - 1/4T Hz. On the other hand, if the signs of aQ(t) and a l ( t ) are opposite (that is, d(t) is an alternating sequence of 1’s and O’s), then

e(t) = 27rfot + 7rt/2T= 27r(f0 + 1/47) (9) which corresponds to a sinusoid of frequency f, + 1/4 T Hz. If we plot the excess phase of an MSK signal versus t, as shown in Fig. 4, it follows that it is piecewise linear and increases or decreases exactly 7r/2 radians each T s. Lines with positive slope represent alternating 1-0 sequences and lines with negative slope represent all 1’s or all 0’s sequences. Examples of each of these situations are shown as the heavy lines in Fig. 4.

Lowpass Realization of Conversion and Matched Filters It is convenient from an implementation standpoint to

realize the bandpass conversion and matched filters of a serial MSK modem as lowpass filters in the inphase (I) and. quadrature ( Q ) arms of the parallel structures shown in Fig. 5. In addition to the realization advantage, which we will explore further in the next section, the I/Qreceiver structure shown in Fig. 5(b) allows carrier synchronization by means of a Costas loop structure with the addition of another multiplier. This will also be discussed later.

That bandpass filters can be implemented by means of parallel I / Q demodulator-modulator cascades can be shown either by considering the response to an ideal impulse, as in [lo], or by the use of complex envelope representation of the

signals involved [ l l ] . Because of the additional mixing operations at the transmitter and receiver, a demodulator operation is removed from the transmitter I / Q filter imple- mentation, and a modulator operation is removed from the receiver l /Q filter implementation.“

Once one is convinced of the validity of the modulator and demodulator structures shown in Fig. 5, it is straightforward to obtain the transfer functions of the ideal I- and Q-arm filters. The mathematical derivations are given elsewhere [ 1 1, 121, and will not be repeated here. Figure 6 shows the frequency response functions of the lowpass equivalents of the ideal serial MSK conversion filter, and Fig. 7 shows the frequency response functions of the lowpass equivalents of the ideal serial MSK matched filter. The equivalent bandpass

“Stripline implementations of bandpass conversion filters have also been implemented. See D. J. Rasmussen, “Generation of Serial MSK Signals with Nan-Ideal Conversion Filters,” M.S.E.E. Thesis, Arizona State University, Tempe, AZ, May 1981.

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frequency response function is obtained by adding the /- and Q-channel frequency responses in phase quadrature and shifting the result up in frequency by f,-1/4T= f l Hz. (Note that the plot of the Q-channel frequency response includes a factor of -j.) The corresponding bandpass frequency response functions are shown in Figs. 6(a) and 7(a) for the conversion and matched filter, respectively.

Our objective is to approximate these lowpass I- and Q- arm transfer functions with filters that are easily realized and result in low overall system degradation. In addition, any realization should be compatible with the high data-rate requirement. Stripline filter realizations are ideal in light of this requirement. I t should be noted that any deviation from the desired phase-quadrature relationship between the two channels results in distortion of the bandpass frequency response functions. This results in performance degradation Hardware implementations that keep the total of all these degradations below acceptable levels have been built and will be described later in this paper.

To introduce two candidate realizations, we take what may at first appear to be an unrelated path.

Transversal Filter Realizations Examining Figs. 6 and 7 more closely, we note that the

/-arm filter frequency response fpnction is lowpass and an even function of frequency, while the Q-arm response is highpass and odd. The realization of finite impulse-response (FIR) digital filters with these two characteristic transfer functions is a well established theory [ 131. The simplest structures involve a single delay and are shown in block diagram form in Fig. 8. Using the fact that a delay of T corresponds to a transfer function of the form exp(-j2&), it is easy to show that the structure of Fig. 8(a) has the transfer function

H , ( f ) = a [ ~ + e - j ~ ~ ' ~ ] = ~acos( . r r f~)e- j~" (10)

while that of Fig. 8(b) has the transfer function

H b ( f ) = p[ 1 -e-j'rfi] = j2psin( ( 1 I )

Except that their frequency responses are periodic and extend indefinitely, these transfer functions have exactly the even and odd character desired of the /-arm and Q-arm filters for the serial MSK conversion and matched filters. By increasing the number of delays, we could approximate arbitrarily closely, in the sense of minimum mean-square error, the ideal frequency responses of Figs. 6 and 7. In order to suppress the undesired periodic lobes of the periodic frequency responses, a bandpass filter can be employed at the output of the modulator, and a lowpass filter can be employed at the output of the demodulator.

The realization of stripline compatible filters with transfer function given by (1 0) is accomplished in a direct fashion by means of an open transmission line, while the realization of a filter with transfer function given by ( 1 1 ) can be achieved with a shorted transmission line. This follows from the observation that the driving point impedance of a lossless, open transmission line is

ZLO = -j Z, cot pQ and that of a shorted line is

ZLS = j Z, tan pQ where Z, is the characteristic impedance of the transmission line, Q is its length, and p = ~ A / A where A is the wavelength of the propagating signal. If driven by a source of impedance R , = Z,, the resulting frequency responses for the open and shorted lines are exactly of the forms given by ( 10) and ( 1 1 ), respectively. Computer evaluation of modulators and de- modulators using this approach have resulted in degradations of less than 0.5 dB [ 1 1,121.

At lower data rates, baseband conversion filters based on this principle can be implemented by using a shift register and summing network, as illustrated in Fig. 8(c). This simple procedure using digital logic devices can be extended to obtain more precision by employing an N-stage shift register, operating it at the data clock multiplied by N , and properly summing the shift register outputs to shape the I and Q data spectra into the quadrature modulators.

In the next section, we examine another realization which is easily implemented as a stripline filter.

Quadrature Hybrid Realizations A quadrature hybrid is a four-port coupled microwave

structure with transfer characteristics closely approximated by the following transfer functions (see inset,page 36):

H12(f) = 0 (12) (input-to-isolated ports)

H13(f) = cos 8 exp(-jpP) (13) (input-to-dc ports)

Hl4(f) = j sin 8 exp(-jpQ) (14) (input-to-coupled ports)

where

8 = Omaxsin pQ (15)

is the coupling angle. The constant Om,,, referred to as the maximum coupling

angle, is geometry dependent; with emax = 45O, one-half of the input power is coupled to the dc port and one-half appears at the coupled port. The frequency dependence of the quadrature hybrid results from the coupling angle argument, which is

pQ = 27rQ/A = (27rQ/u)f (16)

where p = 2 ~ / h , A is the wavelength, Q is the coupling length, u is the speed of electromagnetic propagation in the medium, and f is the frequency of interest. From ( 1 3), ( 15), and ( 1 6) it is seen that H13(f) has the even response characteristic desired of the /-arm filter, and from (14) it follows that H14(f) has the odd response characteristic desired for the Q-arm filter. By appropriately choosing the coupling length to adjust the bandwidth of the quadrature hybrid; and following it with a bandpass filter in the case of the modulator or with a

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IEEE COMMUNlCATiONS MAGAZINE

COURIER mlSE ERROR ca.s,ee.i

Fig. f l . Degradation in signal-to-noise ratio versus carrier phase error for BPSU, serial'MSK, convbntionat MSK, and offset QPSK;

lowpass filter in the case of the demodulator, a close approximation to the ideal serial MSK conversion and matched filters can be obtained. Computer simulations [ 1 1,151 and hardware implementations [ 161 have resulted in overall performances within 0.5 dB of the ideal for quadrature-hybrid implemented conversion and matched filters.

Carrier Synchronization

SO far, the discussion has dealt solely with the realization of conversion and matched filters for the serial MSK modem. At least two other operations must be implemented at the receiver, namely, carrier and bit synchronization. Once a reliable carrier reference has been attained, one of several standard techniques can be implemented for bit synchroniza- tion [ 171.

Two approaches can be used in establishing carrier synchronization for the MSK signal. One involves squaring the received signal to produce spectral components at 2f l and 2f2, as mentioned by Pasupathy [ 11. Since we are interested in high data-rate systems here, this approach is not satisfactory as it involves doubling the frequency of a spectral component which must be greater than the half bandwidth of the wideband modulated signal. In the high data-rate applications of interest here, frequencies in excess of 1 GHz would be required.

Another approach, which is more satisfactory for high data-rate applications, makes use of a Costas loop structure as illustrated in Fig. 10. It is especially applicable to the I/Q demodulator structure illustrated in Fig. 5(b), since the I - and Q-channel demodulated signals are already available for the matched filter implementation. All that is required to derive

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OCTOBER 1983

* ”

?kg. 13. Probability of error uersue @gnabto-hofse ratic) for a- pr6of ot canoept (POG) hardware implementation-of .a serial MSK modem at 27.5 Ub/s.

an error signal that is an odd function of the phase error is an additional multiplier which results in the Costas loop implementation shown in Fig. 10.

To give a heuristic verification that the Costas loop implementation can be used to obtain carrier synchroniza- tion, consider the following two hypothetical cases for the data:

0 d( t ) consists of long strings of 1’s or 0’s; 0 d(t) is a sequence of alternating 1’s and 0’s.

In the former case, the received signal, assuming a noiseless environment, can be written as

m(t) = A cos 2 r ( f0 - 1/4T)t = A cos (2rf1t)

where (8), which was developed in relation to the trellis diagram of Fig. 4, was employed. For this condition, the outputs of the I / Q arms of the demodulator of Fig. 5(b), after removal of double frequency terms, are

I(t) = A cos 4 (17)

35

and

Q ( t ) = A sin d, (18)

respectively. Their product is

e 1 ( t ) = YzA2 sin 24 (19)

which can be used as the control signal for a voltage- controlled oscillator for the reference signals into the I/Q mixers of the demodulator.

On the other hand, for the case where d ( t ) is an alternating 1-0 sequence, the input signal can be written (again using the trellis diagram of Fig. 4)

m(t) = A COS 27r(f0 + 1/4T)t = A cos (2nf2t

which results in the I/Q detector outputs

I(t) = A cos ( r t / 2 T +@)

and

Q(t) = A sin (7rt/2T +d,)

respectively. The product of these two signals is

ez(t) = %A2 sin (7rt/T+ 2 4 )

which corresponds to a spectral component at one-half the bit rate. Since no dc component is present, it is not suitable for obtaining phase correction of the tracking loop and, in fact, can be considered to be a data-dependent noise term.

The actual data sequence can be considered to be a combination of the two cases just considered (that is, no data transitions for half the time and an alternating 1-0 sequence the other half of the time). The net result is a two-to-one loss in gain for the tracking loop along with the data dependent noise already mentioned.

Note that, because of A appearing squared in (19), a 1 80° phase ambiguity exists in the carrier acquisition loop. (One cannot tell whether the loop locked on d ( t ) or -d(t).) This ambiguity also manifests itself in (19) due to the factor of 2 multiplying the phase error 4. (Stable lock points of the phase-locked loop exist for 4 = 180° as well as for 4 = OO.)

Two schemes that can be used to remove this ambiguity are differential encoding of the data [ 181 and injection locking of the carrier acquisition loop [ 191. Injection locking pre- supposes the availability of an unmodulated carrier (or at least a strong spectral line at the carrier frequency). This can be achieved in a TDMA system by including an unmodulated carrier as a portion of the preamble. Once carrier lock is achieved, the loop can be switched to the normal Costas loop mode. Note that only two states of ambiguity exist, rather than the four inherent in QPSK or some implementations of parallel MSK demodulators.

The effect of phase error jitter on serial MSK modem performance can be estimated from Fig. 11, which compares degradation due to static phase error for conventional MSK, serial MSK, OQPSK, and BPSK [20]. I t is seen that when demodulated serially, MSK degradation due to static phase error is comparable to BPSK.

Example of Hardware Implementations of Serial MSK Modems

Figure 12 shows a photograph of a 550 Mb/s serial demodulator which is part of the Proof of Concept hardware for the Advanced Communications Satellite Program TDMA baseband processor mentioned at the beginning of the article. I t employs Costas loop carrier demodulation and a quadrature hybrid-implemented matched filter. The stripline implementation allows a compact realization of a complex demodulator and detector. A 550 Mb/s serial MSK modulator was implemented using similar construction techniques. A complete satellite baseband processor requires three 550 Mb/s modems and 260 medium rate (27.5 Mb/s and 110 Mb/s) modems, which shows the importance of a highly modular design. A custom medium-scale integrated circuit was developed to perform several of the signal processing functions of the medium rate demodulators in order to reduce their size, weight, and power. Back-to-back performance of the medium rate modem is shown in Fig. 13 in terms of bit error probability, PE, as a function of bit-energy to noise-density ratio. Data for tests both with and without two adjacent channels present at spacings of 1.5 times the data rate are shown. For no adjacent channel interference, the degradation at PE = 1 Op6 is about 1 dB, while with adjacent channels occupied, the degradation is less than 2 dB. It is emphasized that these degradation measurements include all sources of error, including phase and timing jitter, since the tests were made with back-to-back modulators and de- modulators.

Conclusion The advantage of serial implementations of MSK modems

for high data-rate applications is that the balancing and timing requirements of parallel implementations are avoided. The serial implementation does require close approximation of the required bandpass conversion and matched filters, however. Implementations making use of inphase and quadrature channel mixers to realize the conversion and matched filters

as lowpass equivalents are particularly advantageous be- cause of their compatibility with microwave integrated-circuit fabrication techniques.

Acknowledgment The theory and results presented in this paper were

developed under sponsorship of the “Advanced Communica- tions Satellite Baseband Processor Subsystem,” sponsored by NASA Lewis Research Center under contract NAS3- 22502. Discussions with the following persons involved in the modem development for that effort are gratefully acknowledged: James Stilwell, Don Rasmussen, and Brian Mussatti.

References [ 11 S. Pasupathy, “Minimum shift keying: a spectrally efficient modula-

tion,” I€€€ Communications Magazine, vol. 17, pp. 14-22, July 1979.

[Z] F. Amoroso, “The bandwidth of digital data signals,” l E E E Communications Magazine, vol. 18, pp. 13-24, Nov. 1980.

[3] R. J . F. Fang, “Quaternary transmission over satellite channels with cascaded nonlinear elements and adjacent channel interference,” IEEE Trans. Commun.. COM-29, pp. 567-581, May 1981.

[4] V. K. Prabhu, “MSK and offset QPSK modulation with bandlimiting filters,” IEEE Trans. Aerosp. Electron. Syst., AES-17, pp. 2-8, Jan. 1981.

[5] D. H. Morais and K. Feher, “Bandwidth efficiency and probability of error performance of MSK and offset QPSK systems,” IEEE Trans. Commun., COM-27, pp. 1794-1801. Dec. 1979.

[6] S. A. Gronemeyer and A. L. McBride, “MSK and offset QPSK modulation,” IEEE Trans. Commun., COM-24, pp. 809-819, Aug. 1976.

[7] F. Amoroso and J. A. Kivett, “Simplified MSK signaling technique,” IEEE Trans. Commun., COM-25, pp. 433-441, April 1977.

[SI C. R. Ryan, A. R. Hambley, and D. E. Vogt, “760 Mb/s serial MSK microwave modem,” I€€€ Trans. Commun., COM-28, pp. 771-777, May 1980.

[9] R. deBuda. “Coherent demodulation of frequency-shift keying with low deviation ratio,” I€€€ Trans. Commun., COM-20, pp. 429-435, June 1972.

[ lo] J. M. Wozencraft and I . M. Jacobs, Principles of Communication Engineering, New York: John Wiley, pp. 496-497, 1965.

[ 1 I ] R. E. Ziemer, C. R. Ryan, and J. H. Stilwell, “Conversion and matched filter approximations for serial minimum-shift keyed modulation,” IEEE Trans. Commun., COM-30, pp. 495-509, Mar. 1982.

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[I21 R. E. Ziemer and C. R. Ryan, “Near optimum delay-line detection filters for serial detection of MSK signals,” ICC ’81 Conf. Rec., pp. 56.2.1-56.2.5, June 1981.

[13] R. E. Ziemer, W. H. Tranter, and D. R. Fannin, Signals andSystems: Continuous and Discrete, chap. 8, New York: Macmillan, 1983.

[I41 G. Matthaei, L. Young, and E. M. T. Jones, Microwave Filters, Impedance-Matching Networks, and Coupling Structures, chap. 13, Dedham, MA: Artech House Books, 1980.

[I51 J. M. Liebetreu and C. R. Ryan, “Performance slmulation of receiver filters for serial detection of MSK signals,” Allerton Conf. Proc., pp. 35 1-358, Oct. 1980.

[ 161 J. H. Stilwell, “Serial MSK modem for the Advanced Communications Satellite Program,” National Telesystems Conf. Rec., pp. A2.5.5, Nov. 1982.

[17] J. J. Spilker, Satellite Communications, chap. 14, Englewood Cliffs: Prentice Hall, 1977.

[I81 R. E. Ziemer and W. H. Tranter, Principles of Communications: Systems, Modulation, and Noise, Boston: Houghton-Mifflin, p. 324, 1976.

[I91 F. M. Gardner, Phase-Locked Loops, New York: John Wiley, 1979. [20] D. J. Rasmussen, “The Effects of Phase Error on Conventional and

Serial MSK Demodulator Implementations,” Technical Memoran- dum, Motorola GEG, Communications Research Facility, Gilbert, AZ, Dec., 1981.

Rodger E. Zierner was born in Sargeant, M N , on August 22, 1937. He received the B.S., M.S.E.E., and Ph.D. degrees from the University of Minnesota, Minneapolis, M N in 1960, 1962, and 1965, respectively.

He has been on the faculty of the University of Missouri-Rolla (UMR) since 1968, where he is currently a professor of Electrical Engineering. In addition to teaching courses and performing research in communications and

signal processing, he has been a consultant with several industries and government agencies on problems involving signal processing in communi- cations and radar systems. During the academic year 1980-1981 he was on leave from UMR while doing research and development on high data-rate communications systems at the Communications Research Facility of Motorola Government Electronics Division at Scottsdale, AZ. ‘

Dr. Ziemer has published numerous papers in his areas of research interest and has coauthored two textbooks: Principles of Communications: Systems, Modulation, and Noise (Boston, MA: Houghton-Mifflin); and Signals and Systems: Continuous and Discrete (New York: Macmillan). He is a member of Tau Beta Pi, Eta Kappa Nu, Sigma Xi, and the American Society for Engineering Education. Dr. Ziemer is a Registered Professional Engineer, and a Fellow of the IEEE.

Carl R. Ryan was horn in Gateway, AR, on March 3,1938. He received the B.S., M S . , and Ph.D. degrees in Electrical Engineering from the University of Arkansas, Fayetteville, AR; Iowa State University, Ames, IA; and the University of Missouri-Rolla, in 1962, 1963, and 1969, respectively.

He has been with the Motorola Government Electronics Division at Scottsdale, AZ, since 1963, involved in various aspects of high data-rate signal processing and communications systems. From 1977 to 1979, he was on the faculty at Michigan Technological University, Houghton, MI, as a professor of Electrical Engineering. He is currently manager of the Communications Research Facility of Motorola.

Dr. Ryan has published numerous articles and papers, and holds eight patents relating to communication signal processing and high data-rate circuit and system design. He is a member of Eta Kappa Nu, Sigma Xi, and Phi Kappa Phi, and a Senior Member of the IEEE.

CALL FOR PAPERS Special Issue on Telecommunications Standards

IEEE Communications Magazine

The lEEE Communications Magazine will be publishing a Special Issue on Telecommunications Standards in January

Original papers, surveys, and tutorial articles concerning telecommunications standards are solicited for this special 1985. The guest editor will be J. S. Ryan of Bell Laboratories.

issue. Possible topics include, but are not limited to:

0 Why standards? International standards Regional standards

0 National standards 0 Standards organizations 0 Relationships between standards organizations

The players Anatomy of a standard Military standards

The papers should be in keeping with the general spirit of the Magazine.

followed by five copies of the proposed article by June 15, 1984. A 100-200 word summary/abstract should be sent to the address given below, by March 1, 1984. This should be

J. S. Ryan-2C620 Bell Laboratories Crawfords Corner Road Holmdel, NJ 07733 USA

Tel. (201) 949-5813

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