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PAM4 Analysis and Measurement Webinar Slidedeck

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WEBINAR P AM4 Analysis and Measurement Considerations 1 6/14/2017 Mike Hertz [email protected]
Transcript
Page 1: PAM4 Analysis and Measurement Webinar Slidedeck

WEBINARPAM4 Analysis and Measurement Considerations

16/14/2017

Mike [email protected]

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About Us: Teledyne LeCroyo LeCroy founded in 1964 by Walter LeCroy o Origins are high speed digitizers for particle physics

researcho Teledyne LeCroy corporate headquarters is located in

Chestnut Ridge, NYo Teledyne LeCroy has the most advanced technology

and widest line of Real-Time digital oscilloscopes (from 40 MHz to 100 GHz)

o Long History of Innovation in Digital Oscilloscopeso Teledyne LeCroy became the world leader in protocol

analysis with the purchase of CATC and Catalyst, and creating a protocol analyzer division based in Santa Clara, CA.

o In August 2012, LeCroy was acquired by Teledyne Technologies and was renamed Teledyne LeCroy

o In April 2016 we acquired Frontline Test Equipment and Quantum Data to add wireless and video to our protocol analyzer portfolio

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About the Presenter

1. Field Applications Engineer with Teledyne LeCroy in Michigan for over 16 years

2. BSEE from Iowa State University and an MSEE from the University of Arizona

3. Awarded six U.S. patents for oscilloscope measurement design

Mike HertzSenior Field Applications Engineer Teledyne [email protected]

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PAM4 Analysis and Measurement ConsiderationsUsing Oscilloscopes

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Agenda

• Quick review: eye patterns, bit rate, baud rate, PAM4 nomenclatures

• PAM4 test configurations• PAM4 compliance measurements• PAM4 debug techniques• Real time and sampling scopes for PAM4• Questions

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What Is An Eye Pattern?

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Real Time Eye Pattern

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PAM4: Pulse Amplitude Modulation 4-Level

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Bit rate and baud rate

Bit rate ( 𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠

) = Baud rate (𝑏𝑏𝑠𝑠𝑠𝑠𝑏𝑏𝑠𝑠𝑠𝑠𝑏𝑏𝑏𝑏𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠

) ∗ ( 𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑠𝑠𝑠𝑠𝑏𝑏𝑠𝑠𝑠𝑠

)

Bit rate is #𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑏𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠

Baud rate is #𝑏𝑏𝑠𝑠𝑠𝑠𝑏𝑏𝑠𝑠𝑠𝑠𝑏𝑏𝑏𝑏𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠𝑠

PAM4 encodes 2 bits into each symbol, therefore a 56 Gb/s throughput requires a PAM4 signal at 28 Gbaud. 56 Gb/s is often referred to as “50G” because coding overhead results in a reduced data payload capacity. PAM4-based Ethernet standards for 100G, 200G and 400G use 2, 4, or 8 lanes of 28 Gbaud PAM4.

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PAM4 Eye Patterns

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Ethernet interfaces and nomenclatureElectrical Optical

Source: Ethernet Alliance

Note that NRZ electrical signaling is used at 10 and 25 Gb/s, and PAM4 is exclusively used for rates 50 Gb/s and higher

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Ethernet technologies nomenclature (followed by most of the high-speed Ethernet standards)

400GBASE-CDAUI-8

100GBASE-KR2

Aggregate bit rate of entire link after coding overhead is removed Interface type

Number of Lanes

Attachment Unit Interface

Backplane interface

Roman numerals for aggregate bitrate in Gb/s are appended when followed by “AUI”

400 Gb/s across 8 lanes, or 50 Gb/s on each lane

100 Gb/s across 2 lanes, or 50 Gb/s on each lane

Emerging standards such as 50GBASE-KR and 100GBASE-CR2 achieve 50 Gb/s per lane by using PAM4 signaling. Ethernet standards which have a speed of 50 Gb/s per lane are using 28 Gbaud PAM4 signaling (includes overhead coding), with a bitrate of 56 Gb/s (data throughput of 50 Gb/s)

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OIF and IEEE OIF-CEI56G-*-PAM4, where “*” is a variant:

OIF-CEI56G-XSR-PAM4, for 0-50mm traces OIF-CEI56G-VSR-PAM4, for 125mm host + 25mm

module traces OIF-CEI56G-MR-PAM4, for 500mm trace + 1 connector OIF-CEI56G-LR-PAM4, for 1000mm trace + 2 connectors

Two IEEE 802.3 (Ethernet) amendments: 802.3bs, for 400 GbE (8 lanes of 25G PAM4) 802.3cd, for 50/100/200 GbE (1/2/4 lanes of 25G PAM4)

IEEE 802.3 bs

IEEE 802.3 cd The VSR variant involves most of the test requirements for PAM4 because:

VSR has normative (compliance) tests which are somewhat unique to it and to XSR But it also has informative tests which are measurements that also appear in –MR, -LR

and the IEEE specs

Most measurements in this presentation refer to –VSR unless specified otherwise Most of the measurement concepts can be easily generalized

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PAM4 Test Configurations

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Test point TP1a“The output of the Host Compliance Board (HCB) provides access to the host-to-module electrical signal (host electrical output) defined at TP1a.”

Signals from the host Tx are measured after passing through the host’s PCB trace and a defined Host Compliance Board (HCB). Signals at TP1a are expected to be measured using a standard receiver CTLE for CEI-56G.

The HCB can either be a physical compliance board, or emulated using S-parameters. A .s4p file is used for the HCB, and CDR and CTLE are emulated in the scope. Device

under test

HCB

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Test point TP4“The output of the Module Compliance Board (MCB) provides access to the module to host electrical signal (module electrical output) defined at TP4.”

The signal from a module Tx is measured after passing through the host’s PCB trace and a defined Module Compliance Board (MCB).

Signals at TP4 are expected to be measured using a standard receiver CTLE for CEI-56G, and the MCB can either be a physical compliance board, or emulated using S-parameters. A .s4p S-parameter file is used for the MCB, and, CDR and CTLE are implemented within the scope software.

Device under test

MCB

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Test setup for TP0a

“TP0a is defined to be separated from TP0, the ball of the package performing the host-to-module transmit function, by 1 dB of PCB attenuation at 14 GHz.”

TP0a does not have a well-defined test setup because it is not at or near a point of connection in an actual interface. The goal for TP0a is to connect as close to the output pin of the Tx chip as possible. This point is typically accessible for silicon/IP developers who are working on a development board.

The exact connection setup and embedding or de-embedding requirements will vary depending on what development or evaluation board connections are available. Signals at TP0a are expected to be measured without any receiver equalizer emulation, and VSR tests at TP0a are informative only.

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PAM4 Compliance Measurements

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PAM4 Differential Voltage, Common Mode Voltage, Common Mode Noise

Differential voltage pk-pk = max(Dp - Dn) – min(Dp - Dn) Max Common mode voltage = max(Dp + Dn) Min Common mode voltage = min(Dp + Dn)

RMS Common mode noise = sdev(Dp + Dn)

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Transition Time MeasurementThe Transition Time measurement is described in the standard as:

“Transition times (rise and fall times) are defined as the time between the 20% and 80% times, or 80% and 20% times, respectively, of isolated -1 to +1 or +1 to -1 PAM4 edges. Using the QPRBS13-CEI test pattern the transitions within sequences of three -1s followed by three +1s, and three +1s followed by three -1s, respectively, are measured. These are PAM4 symbols 1820 to 1825 and 2086 to 2091, respectively, where symbols 1 to 7 are the run of seven +1’s. In this case, the 0% level and 100% level may be estimated as the average signal within windows from -1.5 UI to -1 UI and from 1.5 UI to 2 UI relative to the edge.”

Histogram of rise times

Histogram of fall times Histogram of slew rates

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CEI-56G-VSR-PAM4 and 106

Signaling speed Baud rate of 18 – 29 Gbaud Most devices operate at 28 Gbaud This corresponds to a raw bit rate

of 56 Gb/s

Forward Error Correction (FEC) CEI-56G-VSR-PAM4 only requires

a raw BER of only 10-6 (very different from NRZ signals)

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Eye Diagram measurements and 106 UI

Most of the eye diagram measurements are made with specific reference to 10-6

contours because the threshold for FEC in use by these standards is a BER of 10-6 at the physical layer. Therefore, unlike NRZ analysis, extrapolation is not needed for PAM4 analysis, because 106 symbols can be easily captured in a single acquisition on a real-time oscilloscope.

Acquire a sufficient number of UI’s, at least 106, and ideally 107.

10-6 contours are displayed here in green.

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BER Contour Description

Click here

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Eye Diagram measurements: TmidTmid is the midpoint of the maximum horizontal eye opening of the 10-3

(red) inner eye contour of the middle eye.

Tmid is the expected time position for the hardware receiver to sample the signal.

The Tmid calculation is the starting point for many other eye-diagram based measurements.

Tmid

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Eye Diagram Measurements: EH6 (Eye Height @ 10-6)

EH6 represents the height of the respective eye at a BER of 10-6, (green contour) determined from voltage CDFs in a +/-0.025 UI time window centered on Tmid.

Tmid

EH6 upp

EH6 mid

EH6 lowNote that EH6 is not necessarily measured at the point of maximal eye height, since EH6 must be measured at Tmid, the midpoint of the 10-3 (red contour). EH6 is the distance between intersection points of Tmid and the 10-6

contour ring for each eye.

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Eye Diagram measurements: EW6

EW6 is the width of the respective eyes determined from CDFs of eye edges halfway between the 10-6 points of the voltage CDFs of the middle eye (EH6/2).

Note that EW6 is not necessarily measured at the point of maximal eye width (especially observable for the upper and lower eyes), since the EW6 measurement location is (EH6/2) for each eye.

(EH6 mid)/2

EW6 upp

EW6 low

EW6 mid

(EH6 upp)/2

(EH6 low)/2

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Eye Diagram Measurements: Eye LinearityEye Linearity is an alternative to the RLMmeasurement (discussed later). Eye Linearity is defined as:

Where AVupp, AVmid and AVlow are the average of the eye amplitudes (not heights), defined as the difference of the mean levels of the upper and lower level voltage histograms in a +/- 0.025 UI time window centered at Tmid.

The measurement of Eye Linearity determines symmetry of the three eyes. An ideal signal has an eye linearity of 1.000.

AVupp

AVmid

AVlow

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Vertical Symmetry and Eye Linearity

Examples of good eye linearity Examples of problematic eye linearity

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Eye Diagram measurements: Mask test

The VSR mask is purely horizontal and is defined as:“…an Eye Width mask centered on Tmid having a width from the relevant table which extends above and below the waveform for the upper and lower PAM4 eyes. The EW6 low, middle and upper eye edges shall be outside this Eye Width mask.”

Because the mask is centered on Tmid, it’s possible for the upper and lower eyes to pass the EW6 test (wide enough at 10-6) but fail the mask test (not sufficiently centered on Tmid)

Mask

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Horizontal Symmetry and Mask Testing

Horizontally-symmetrical eyes passing mask test: Asymmetrical eyes failing mask test:

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Skew Between Generated Bits Affecting Horizontal Symmetry

Asymmetrical eyes failing mask test:

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Background on Jitter Methodology for PAM4

Many NRZ signal standards require extrapolation of Total Jitter (Tj) to BERs of 10-12. This required fitting values to a model (dual-dirac), and the terms involved (Tj, Rj, Dj) are associated with those models and extrapolation methods.

PAM4 technologies require only a BER of 10-6 at the physical layer. Since oscilloscopes can easily acquire 106 bits in a single acquisition, traditional methods of Rj/Dj extrapolation are not needed, and new methods and terminology is used for PAM4 signaling.

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Jitter Methodology for PAM4 UUGJ (Unbounded, Uncorrelated Gaussian Jitter) – Conceptually similar to random jitter

UBHPJ (Uncorrelated, Bounded High-Probability Jitter) – Conceptually similar to deterministic jitter

EOJ (Even-Odd Jitter) – Systematic jitter occurring between even- and odd-numbered transitions. This was usually called “F/2 jitter” in an NRZ context, and was often mistaken for DCD.

UJ4 – Measured peak-peak uncorrelated jitter at the 10-4 probability level UJ6 – Measured peak-peak uncorrelated jitter at the 10-6 probability level

J4 – Measured peak-peak jitter at the 10-4 probability level J6 – Measured peak-peak jitter at the 10-6 probability level

Note: J4 and J6 are deprecated terms that were used in jitter calculations before UJ4 and UJ6 were adopted in more recent revisions

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Uncorrelated jitter (UJ4 and UJ6)

A repeating pattern must be used

For each transition in the pattern, form a histogram of its edge arrival times

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Uncorrelated jitter (UJ4 and UJ6) Remove the

mean from each edge histogram and sum it with all other edges from the same eye

Now we have an uncorrelated jitter (UJ) histogram for each of the three eyes

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Calculating UUGJ and UBHPJ

A jitter CDF is derived from each histogram

J4 and J6 are calculated as the width of the CDF at 10-4 and 10-6

respectively

UUGJ and UBHPJ are calculated from:

*Equation from CEI-56G-MR-PAM4 spec

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Even-odd jitter

“Even-odd jitter is measured using two repetitions of a QPRBS13-CEI test pattern with FIR off. The deviation of the time of each transition from an ideal clock at the signaling rate is measured.Even-odd jitter is defined as the magnitude of the difference between the average deviation of all even-numbered transitions and the average deviation of all odd-numbered transitions, where determining if a transition is even or odd is based on possible transitions but only actual transitions are measured and averaged.”(from CEI-56G-MR-PAM4 spec)

Note QPRBS13-CEI is a pattern with an odd number of symbols – so in any repetition, each symbol will land on the “opposite” even/odd clock edge than it did in the previous repetition.

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Linear Fit Method

1. Acquire waveform on scope.2. Resample the waveform using an integer number of samples.3. Generate an ideal waveform with the same pattern.4. Deconvolve to obtain a pulse response of a full-swing 0-to-3-to-0 transition to

determine the impulse response of the system.

The mathematical definition is described in OIF-CEI-03.1, section 11.3.1.6.4.

The linear fit error is calculated as the difference between the pulse response and the actual signal for each resampled point.

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Linear fit pulse response - examples

This signal is very clean

Note how controlled the pulse response is

The impulse response is derived from the waveform.

Linear fit pulse response

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Linear fit pulse response - examples

This signal has twice as much noise.

Note the pulse response hasn’t changed.

Linear fit pulse response

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Linear fit pulse response - examples

This signal has low noise but is severely bandwidth-limited.

Note the pulse response reflects the signal shape.

Linear fit pulse response

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Linear fit pulse response - examples

Since LFPR is sampled an integer number of times per UI, we can decimate it to get one amplitude value per UI.

This can be used to optimize transmitter emphasis coefficient values.

Removes all noise, all pattern dependent artifacts to produce ideal normalized coefficient values.

Linear fit pulse responsedecimated to one point per UI

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SNDR

The linear fit error is the difference between the actual transmitter output signal and the ideal signal, producing an error vector e(k). SNDR is calculated using the maximum value of the pulse peak, pmax, and the linear fit error, e(k). Note the RMS deviations of the voltage levels are not used in the calculation.

*from CEI-56G-MR-PAM4 spec

Signal-to-noise-and-distortion ratio (SNDR) is calculated using the linear fit pulse response and the linear fit error:

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Transmitter Linearity (RLM)

This measurement is not required in normative or informative –VSR tests, but is required by most other variants

It is conceptually similar to Eye Linearity but is measured in a different way (and the resulting values are not directly comparable) A perfect signal has an RLM of 1 but it

does not scale the same as eye linearity

The definitions of the signal levels V-1, V-1/3, V+1/3, V+1 have changed as the standard evolved

RLM is also referred to as “Level Separation Mismatch Ratio”

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RLM – the “old” way In older specifcation revisions, RLM

required a special pattern to be used.

Note in this pattern, each symbol is repeated for 16 UIs to ensure settling.

The level values were determined as the voltage at the middle of the run of 16 symbols.

This pattern was difficult for many device vendors to generate. *from IEEE p802.3bj

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RLM – the “new” way

Now as described in the –VSR spec (and referenced by the others), RLMderives the voltage levels directly from a QPRBS13 signal (which is much easier to generate)

The voltage level at the center of each symbol in the pattern is measured, and these are used to calculate a mean value for each level.

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RLM – what is measured currently

The RLM calculation was developed when the “old” way was in use, but was generalized to estimate RLM for arbitrary patterns: Find the longest run of each level

(must be >6 symbols) Use the center point of this run to

calculate the voltage value for that level

The result is the correct calculation on the (now-deprecated) special pattern, and good on other patterns

Considerations:

If the pattern does not have a run of >6 UI of all 4 levels, no values are produced

The measurement can change substantially if the pattern changes

Long patterns with long consecutive symbols yield the best results

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RLM variation as a function of pattern length

PRBS7 PRBS13

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PAM4 Debugging Techniques

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QuickStart for PAM4

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PAM4 multiple eye pattern view

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Fixture de-embedding with S-parameters, virtual probing, and receiver equalization

Physically probed

Virtuallyprobed

Virtuallyprobed Equalized

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Virtual Probing with PAM4

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Linear Equalization on PAM4

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Intersymbol Interference

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Intersymbol Interference

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Intersymbol Interference Measurement on PAM4

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Data-Dependent Jitter Plot on PAM4

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BER Contour on PAM4

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BER Contour on PAM4

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Jitter Sim Operator with PAM4

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Oscilloscope Bandwidth Selection:Power Spectral Density Example of 28 Gb/s SERDES

The Power Spectral Density of a 28 Gb/s serial data waveform is plotted above, with Power (dB) on the Y-axis and Frequency (GHz) on the X-axis. For a 28 Gb/s signal: the fundamental frequency is centered at 14 GHz, there is a null at 28 GHz, the third harmonic is centered at 42 GHz, and the next null is at 56 GHz. Therefore, an oscilloscope with at least 56 GHz bandwidth is needed in order to capture all of the power spectral density of the third harmonic of a 28 Gb/s signal, and all of the PSD associated with the third harmonic will be captured by a LabMaster 10-60Zi 60 GHz oscilloscope. The darker blue area is the extra power spectral density provided by a 60 GHz oscilloscope compared to a 32 GHz oscilloscope.

Illustration of harmonic content forming a bit pattern

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Impact of Bandwidth Reduction on a PAM4 Signal

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Recommended Oscilloscope Hardware and Software for PAM4 Testing

Serial Data Analyzer oscilloscope (probably 65 GHz bandwidth)

PAM4 compliance test softwarefor conformance (and debug)

PAM4 signal analysis softwarefor PAM4 debug

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Real Time and Sampling ScopesRealtime ScopeA realtime scope (bandwidths up to 100 GHz) typically triggers on a waveform event, then collects many sample points (millions, billions) from the single trigger event. The sample resolution of a realtime acquisition can be as low as 4.16 ps/pt, which is the inverse of the sample rate (up to 240 GS/s)

Sampling ScopeA sampling scope (bandwidths up to 80 GHz) typically triggers on a reference clock, then collects one sample point per trigger. The typical sampling scope maximum sample rate is 200 kS/s (slower for long patterns).

• Teledyne LeCroy has the world's highest bandwidth realtime scope (100 GHz) with world’s highest sample rate (240 GS/s)

Sampling oscilloscopeBandwidths up to 80 GHzMax sample rate: 200 kS/s (very undersampled)Not able to capture one-time eventsEvents must be repetitiveTrigger source is mandatoryNo software clock recovery for jitterLimited debug and analysis

Real time oscilloscopeBandwidths up to 100 GHz real time Max sample rate: 240 GS/s real timeAble to capture one-time events: transients, runts, glitches, etc.Trigger source not requiredSoftware clock recovery for jitterAdvanced debug and analysis

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Real Time Scopes and Sampling Scopes for PAM4 Testing

A real time scope is able to sample at rates up to 240 Gigasamples per second real time, while a sampling scope is limited to undersampling the signal at a maximum rate of approximately 200 Kilosamples per second (one million times slower than a real time scope). Unlike a real time scope, a sampling scope cannot capture a contiguous block of data, so the risk is that low-frequency anomalies cannot and will not be detected or measured.

A real time scope does not require an external clock source, which completely eliminates trigger jitter from the measurement. A sampling scope always uses an external clock, which adds jitter that is not real, into the measurement.

A real time scope is able to detect periodic jitter on contiguous waveform edges and demodulate it, then display the modulation profile. Unlike a real time scope, a sampling scope cannot capture contiguous data and therefore cannot identify the source of jitter.

A real time scope is able to display the periodic jitter spectrum in the frequency domain. A sampling scope cannot do this. By displaying the periodic jitter spectrum, the frequency content of jitter sources is revealed.

A real time scope can detect one-time glitches in a PAM4 signal. Undetected glitches can cause bit errors at the receiver. A sampling scope cannot detect one-time glitches.

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Real Time Scopes and Sampling Scopes for PAM4 Testing

A real time scope can detect one-time runts in a PAM4 signal. Undetected runts can cause bit errors at the receiver. A sampling scope cannot detect one-time runts.

A real time scope can detect one-time non-monotonic edges in a PAM4 signal. Undetected non-monotonic edges can cause bit errors at the receiver. A sampling scope cannot detect one-time non-monotonic edges.

A real time scope can detect a signal dropout in a PAM4 waveform. Signal dropouts can result in system malfunction. A sampling scope cannot detect a signal dropout.

A real time scope is able to isolate intersymbol interference (ISI) and display the effects of ISI on the PAM4 pattern. A sampling scope does not have an ISI plot, and therefore cannot determine the effect of bit order on jitter independently of the serial data pattern.

A real time scope is able to isolate data dependent jitter (DDj) and display the effects of DDj on the PAM4 pattern. A sampling scope does not have a DDj plot, and therefore cannot determine the effect of bit order on jitter relative to the serial data pattern.

A real time scope can generate a jitter simulation signal in software, used to verify PAM4 setups without the need for a physical signal. A sampling scope cannot do this.

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Real Time Scopes and Sampling Scopes for PAM4 Testing

Not all devices under test have access to the clock signal. In this case, a sampling scope cannot be used since it requires access to the clock signal. A real time scope does not require a clock signal.

A real time scope implements a software clock recovery, completely eliminating the effects of CDR jitter. A sampling scope does not use a software CDR; it uses a hardware CDR, always introducing hardware CDR jitter.

A sampling scope cannot be used for debug in the event of a PAM4 compliance test failure. A real time scope provides detailed analysis capabilities including advanced math, measurements, and custom algorithms to identify root cause of failure.

A real time scope allows for simplified deskewing of a differential signal pair using an automated process. A sampling scope often requires expensive phase adjusters and a tedious manual deskewing process.

A real time scope can perform timing measurements between data and clock while viewing both waveforms, and can analyze the system clock. Since a sampling scope requires using the same system clock as the reference, it cannot detect problems with the system clock. This is important to identify and troubleshoot clock-related problems.

A real time scope is able to generate an eye pattern up to 1,000 times faster than a sampling scope allowing for faster accumulation of meaningful, statistical data. Testing PAM4 using a real time scope results in faster time to insight.

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Questions?


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