Performance Improvement of Shunt Active_TPE_2007

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ASIMINOAEI et al.: PERFORMANCE IMPROVEMENT OF SHUNT ACTIVE 249

TABLE ICALCULATION OF THE LINE CURRENT

I

FOR EACH CASE IN FIG. 3

For the feedback APF

(5)

which gives

(6)

Thus, one may describe the resultant line current as a func-

tion of the initial harmonic current from the ASD and the ex-

isting source voltages ( and , i.e., fundamental respective

harmonic voltages from the grid side). The analytical expression

of the line current determined for all six topologies in Fig. 3

is given in Table I.

As it can be noticed the line current in case-(a) does not

depend on the existing voltage harmonic distortion, and the re-

sultant transfer function is the direct effect of each individualtransfer function of the active power filters. Case-(c) is useful

for damping the resonances between the grid and possible ac-ca-

pacitor at the load side, because both and give the

effect of a damping in series with the grid impedance. However,

the harmonic current mitigation directly depends on the filter’sinductances, which may create design restrictions depending on

the grid impedance value . For case-(b) and case-(d), the com-

mand law depends on the imposed reference and

for each filter, which is useful for sharing the loading between

filters, during different operating conditions. Case-(f) achieves

better harmonic mitigation compared to case-(e) regarding the

influence of the harmonic grid voltages .

Furthermore, for case-(b), case-(d) and case-(f), the schemesuse just one pair of harmonic detection sensors, which reduce

the complexity of the hardware design, the number of analog

digital (AD) converters and possible the speed of the calcula-

tions.

However, case-(b) and case-(d) require a careful design re-

garding the imposed harmonic current references. In the sim-

plest way the reference can be given as half of the measuredharmonic current for each inverter, but then both inverters must

be rated at the same nominal power. For other references values,

additional processing is necessary in order to correctly balance

the reference signals and as it is shown in Fig. 3.

Finally, the actual study selects only the topology shown in

case-(f).

B. Description of the Selected Topology (Case-f)

The proposed topology is shown in Fig. 2. As it can be no-

ticed, the APF is composed of two separate inverters, each of

them using independent current and voltage controllers. To min-

imize the cost this structure may be furthermore reduced by

connecting the inverters together on the dc-link [4]. Thus, onlyone dc-link capacitor would be needed and implicitly a single

voltage controller. However, it makes the hardware design and

the control more complicated, as a zero sequence current cir-

culates between inverters [7]. In order to break the circulation

path of the zero sequence currents between inverters, a galvanic

isolation is inserted by means of either isolating transformer or

separate dc-capacitors. Thus, the inverters can be independently

controlled.

The load current sensor for harmonic detection is placed in

between the connection points of the inverters, thus achieving

a feedback path for the first inverter (PWM inverter 1), and a

feedforward path for the second inverter (PWM inverter 2). This

new topology may allow each inverter to perform different tasks,depending of the imposed current references.

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250 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 22, NO. 1, JANUARY 2007

Thegenerationofharmonicreferencecurrentisdoneinmaster-

slave configuration, as thefirstinverter dictates theharmonic ref-

erence of the second. Two separate master-slave configurations

can be implemented,one is the harmonic-sharing and the other is

the load-sharing control. In the harmonic-sharing mode the first

invertermitigates only severalharmonic currents, and the second

inverter takes care of the rest of the harmonic spectrum.

It is also possible to operate the filters in load-sharing mode,

when both filters compensate the same harmonic spectrum but

with different imposed reference amplitudes. If the harmonic

reference is equally divided between inverters then each takes

50% of the total required power. Other ratio of harmonic refer-

ences in each inverter is also possible and useful for inverters

with different rated powers or when it is desired to control the

distribution of the power losses, but the reference generation al-

gorithm is more complicated.

A third possibility is to configure the proposed topology in a

redundant-mode,whichisabenefitprovidedbytheproposedcas-

caded configuration. The redundant-mode is a particular case of

the load-sharing mode, when the harmonic reference is not di-vided between inverters but setat maximum foronly oneinverter.

Thus,APF1hasthetaskofthemainharmoniccompensator.Since

APF2 follows APF1, when APF1 ceases its harmonic mitigation,

APF2automaticallycantakeover,ifitisconfiguredtodoso.This

offers a wireless independent control feature, because there are

no connectivity wires between the inverters to change informa-

tion aboutthe internalvariableor statesof each control loop.Each

APF’s control is only based on information available locally at

the inverter terminals.However,the redundant-modeimplies that

the rated power of APF2 is suf ficiently large to cover the overall

harmonic mitigation, which may not be cost-effective.

In order to optimizethe cost, onemay usethe feedback inverter

(PWMinverter1)tocompensateonlyapartoftheharmonicspec-trum, for instance, the low order harmonics as most of the power

densityiscontainedhere.Asthecompensationiscarriedoutinthe

feedback loop, the steady state response is expected to be much

more tolerant to nonlinearities and parameter variation [6].

Consequently, since the inverter is of high rated power a lower

switching frequency is desirable. Regarding the selection of the

switching frequency one choice is to use different switching

frequencies (i.e., APF1 low, APF2 high) in order to have re-duced power losses but still a good overall harmonic mitigation

performance. Another choice is to use the same switching fre-

quency for both, which allows implementation of interleaving

techniques to reduce the switching frequency ripple, and conse-

quently EMI and the line filter.In this paper, the second choice was used due to some hard-

ware limitation (APF1 and APF2 have the same switching fre-

quency), but a discussion presented in Section VI analyzes the

potential operational ef ficiency improvement if APF1 switches

at a lower frequency. In the case when both power inverters

have the same switching frequency, by interleaving their carriers

by 180 it may significantly reduce the line switching current

ripple. Thus, the highest ripple amplitude is seen at a doubled

switching frequency, which is much easier to filter out. As the

frequency of the switching current is higher it gives lower EMI

and is also damped much faster in the power lines due to their

inductive effect.

The feedforward inverter (PWM inverter 2) receives the in-formation of the harmonic content only after the first inverter

Fig. 4. Principle algorithm of harmonic extraction and reference current con-trol in synchronous fundamental frame.

has already mitigated part of the harmonic spectrum. The cur-

rent reference for the feedforward APF contains only the har-

monics left in the line currents. The second inverter is designed

for mitigation of high order harmonics. As a result, the second

inverter is configured for a faster dynamic response, thus com-

pensating the slower dynamic of the first inverter (which may

react slowly and imprecisely in transient conditions especially

if its switching frequency is lower).

The entire topology has the advantage of a good harmonicmitigation in stationary conditions (due to the feedback in-

verter), but also very good response in transient conditions (due

to the faster feedforward inverter) [8]. The power losses of the

second inverter are kept at a lower value since the high order

harmonics are not so large. Another advantage of this topology,

but not tested here, is that the feedback APF can provide an

active damping of existing resonances between the grid and a

possible installed ac-capacitor at the load side, e.g., EMI filter

or custom passive harmonic filter, although this requires an

increased bandwidth of the APF current controller.

III. CONTROL ALGORITHM

The control algorithm is developed in the synchronous funda-mental reference dq-frame. The input signals, measured in abc-

coordinates (i.e., stationary reference frame), are transformed

into the fundamental -rotating reference frame by means of

the Park transformation

(7)

where , and , , are the currents in the -frame, re-

spective in -frame; and is the angular position of the ref-

erence frame.

The frame rotates at fundamental angular frequency that

makes the fundamental current to appear as dc-component and

the harmonics as ac-signals. Thus, harmonic detection becomes

a matter of removing the dc-signal by means of a high-order

high pass filter [(HPF) in Fig. 4 with a cutoff frequency between

25 Hz and 120 Hz] [9]. The HPF outputs the harmonic current

to be compensated by the active filter. This is a fourth order

filter implemented in fundamental frame, which removes the

fundamental current, i.e., the dc-signal

(8)

where the cutoff frequency is 300 rad/s, and 0.8.





Fig. 5. Control block diagram of the proposed topology in dq-frame. APF1 hasa single inner harmonic current controller tuned for mitigation of the 6th har-

monic order in dq-frame (i.e., fifth and seventh in stationary frame). APF2 hasa multituned inner current controller realized by superposing several harmoniccontrollers tuned at orders of 6

k

ind q

-frame.

For line synchronization purposes, the line voltages aremeasured, and the line frequency and phase are extracted

by means of a phase locked loop (PLL).

The block diagram of the proposed current control topology

is shown in Fig. 5.

The control contains one dc-voltage control loop for each

APF. The dc-voltage controllers are classical pro-

portional–integral (PI) units, which receive as inputs the dif-

ference between the reference voltage and the measured

dc-voltage . The output represents the fundamental active

current references, on the -axis of the synchronous frame

(9)

where and are the proportional and the integral

gains of each dc-voltage PI controllers.

The current controllers are, on the other hand, divided for

each APF into two distinct paths: the fundamental current con-

trol and the harmonic current control [11].

The APF reference voltage is the sum of all current controller

outputs and it is realized by a space vector modulation strategy

(SVM in Fig. 4).

The fundamental current controller is of a feedback type,

which provides pole-zero cancellation for this plant, the line

inductor. Vector model of the line inductor, in synchronous fun-

damental frame, is

(10)

where and are the resistance and inductance of the line

inductor, is the filter voltage, is the instantaneous filter

current, and is the line voltage vector.

A complex-coef ficient PI controller ( and ),

with cross-coupling decoupling and line voltage feedforward

compensation can be used

(11)

where and are the proportional and the integralgains of each complex current controllers.

The reference for the fundamental current controller is re-

ceived from the dc-voltage controllers. The fundamental current

controllers regulate the -axes active current, while

the reactive current on the -axes is set to zero, as in this appli-

cation no reactive power needs to be controlled.

The harmonic current control is realized by separate control

of each harmonic, each controller for one pair 6 1 of

positive and negative sequence harmonics. The total harmonic

reference is the superposition of commands produced by all cur-

rent controllers. Harmonic current controllers are equiv-

alent harmonic integrators, one controller being tuned here for

each harmonic pair 6 1 [10].

In order to control at the same time, a pair of both positive

and negative characteristic harmonics from a typical ASDs with

six-pulse diode based front-end rectifier, with a single controller,

the controller transfer function is selected as in [11]

(12)

where and are the proportional and the integral gains

of the selected harmonic integrator.

Notably, represents the transfer function of an equiv-

alent harmonic integrator (resonant controller) tuned for fre-

quency (for both negative and positive sequences), which

provides zero gain for the dc-component and infinite gain at the

selected frequency . Such type of controller is implemented

in fundamental reference frame for each harmonic order , up

to the 31st harmonic [12].

Finally, the total harmonic current controller is realized as

the superposition of individual controllers given by (12). In a

general form the resultant harmonic controller is

(13)

Controller gains are selected so as , and

of small value in order to create a narrow selectivity, thus

avoiding the overlap between the neighbor harmonic controllers.

The number of harmonic pairs , which can be compensated

in this way, can be selected based on the available computa-

tional resources and on the sampling and switching frequency

[13]–[16].

The feedback APF1 harmonic current controller is imple-

mented as only for equal to 6. For APF1 there

is only a single harmonic controller, i.e., 6, sixth harmonic

order in -frame meaning compensation of fifth and seventhharmonic currents in stationary frame.

The feedforward APF2 has with

6,12,18,24,30 . The superposition of the individual har-

monic controllers is done by summing the output voltage

reference of each

(14)

(15)

if APF2 is set for redundant-mode 1, else 0.





Fig. 6. Bode plots of (a) open loop transfer functionsH ( s )

for APF 1 (b)openloop transfer function

H ( s )

for APF 2, and (c) overall closed loop system

transfer function as in Table I.

By replacing (14) and (15) into the initial open loop transfer

functions of each active filter in (1) and (2), one can deter-

mine the Bode plots of each system as it is given in Fig. 6. The

behavior of the open loop transfer functions , of

each active filter APF1, respectively, APF2 is dominated by the

transfer functions of the harmonic integrators re-

spectively . Therefore, the close loop Bode plot of

the overall system (APF1 together with APF2) looks as a mul-

tiple notch filter tuned for the selected frequency in -frame

[see Fig. 6(c)].Thus, APF1 mitigates only the lower harmonic currents (fifth

and seventh) the controller having a very high selectivity (i.e.,

low value for gain) in order to determine a precise mitiga-

tion of these harmonics. However, it is known that lower values

of give also a slow dynamic response.

The filter APF2 mitigates the rest of the characteristic har-

monic currents up to 31st order. The selectivity of the equivalent

harmonic integrators is set wider (i.e., higher values for the

gains) in order to cope with the variation of the line frequency.

As the line frequency varies, the harmonics suffer also a small

frequency shift, which it is more severe as the harmonic goes

higher. Thus, in order to provide a better harmonic tracking, it

is advisable to have a wider selectivity of the higher order res-onant controllers. As the selectivity of the controller is wider,

the dynamic response of the APF2 becomes faster compared to

APF1.

The redundant-mode feature is implemented in the APF2

control by setting 1 [see Fig. 5 and (15)] which enables the

sixth harmonic inner current controller in . This can be

seen in Fig. 6(b) where the open loop transfer function

is plot in two cases, with and without a harmonic integrator

tuned for 300 Hz. If is not tuned for 300 Hz it means

that the filter APF2 cannot mitigate the same harmonics as the

first filter APF1. If is tuned for 300 Hz, it allows APF2

to mitigate also the fifth and seventh harmonics. But since its

mitigation follows after APF1, the effect is similar as sharingthe harmonic load when the first filter APF1 does not fulfill its

Fig. 7. Picture of laboratory setup where the proposed topology is tested.

TABLE II

PARAMETERS USED IN CONTROL FOR BOTH INVERTERS APF1 AND APF2

task. This may happen in transient conditions due to the slow

response of the APF1 as previously presented.

Another particular case is when the APF1 is programmed to

reduce (but not completely) the amount of mitigated harmonics

due to different operating conditions, which automatically en-

ables the second filter APF2 to take over the harmonic miti-

gation without any special communication between both active

power filters. However, as initially mentioned, in this case the

power rating of the second filter should be sized for that pur-

pose.

IV. IMPLEMENTATION

The proposed topology and control method are tested on a

laboratory setup (Fig. 7) composed of two Danfoss inverters

VLT 5006 rated as 400 V/7.6 kVA each. The switching fre-

quency was set to 12.6 kHz for both inverters. The control algo-

rithm is implemented with a dSpace system DS1103, using the

Matlab/Simulink Real Time Workshop toolbox [17], having the

control parameters set as in Table II.

In practice a proper the design of the inductors depends on

the mitigated harmonic current spectrum and its power level.

A higher power inverter requires a lower inductance in order

to reach lower iron losses. A faster inverter (i.e., compensationof higher harmonic orders) requires also a lower inductance to





Fig. 8. General diagram of the laboratory setup.

TABLE IIIMEASURED HARMONIC CURRENT OF ASD LOAD AT 5.9 kVA POWER

achieve a higher output current gradient . APF1 compen-

sates here only fifth and seventh, but due to the higher power

level of the first inverter its boost inductance is of a lower value.

The second inverter APF2 compensates here the high order har-

monic currents, and even if the inverter is of a smaller power

it still requires a lower inductor to achieve a high bandwidth

and high . The selected values of the boost inductors are

5.6 mH for the feedback APF1 and 3.0 mH for the feedforward

APF2 (Fig. 8).

The nonlinear load that creates the harmonic currents is a

three-phase diode rectifier, which replicates the behavior of atypical adjustable speed drive (ASD). The rectifier may be set

as dc-smoothed capacitor or inductor by switching “sw1” and

“sw2” such that different cases may be created.

The load is a variable resistor with values between 50–200 .

This way one may create different levels of harmonic currents.

Harmonic distortion created by the ASD is measured as indi-

cated in Table III and Fig. 9, for a typical industrial ASD of a

rated power of 5.9 kVA.

A. Feedback Compensation

This section describes the mitigation performance of only the

APF1-feedback compensation loop (the APF2-feedforward is

disconnected). Here the reference is set for mitigation of onlythe fifth and seventh harmonics. As it can be seen in Fig. 10,

Fig. 9. Measured line current and supply voltage when noneof theactive filtersis running. The line current consists of the ASD current.

Fig. 10. Measured steady-state response of the APF1—feedback compensa-tion. The harmonic compensation reference is set for mitigation of fifth andseventh harmonics.

and more detailed in Table V column 3, the reduction of the

fifth and seventh harmonics currents is from the initial values

of 25% respective 8.7% down to 0.2% respective 0.6%. As the

higher harmonics are not compensated, the resultant waveform

of the line current is still distorted (see Fig. 10).

B. Feedforward Compensation

This experiment was carried out with only the APF2-feed-

forward loop (the APF1-feedback is disconnected). The refer-

ence is set for mitigation of the high order harmonics, from 11th

up to 31st. As discussed in Section III the feedforward inverter

is able to take over all harmonic currents, including the fifth

and seventh by enabling the redundant mode [ 1 in (15)].

Two tests are performed with and without this functionality en-

abled, in order to show its harmonic mitigation performance. It

is clear that, if the filter is able to compensate for high order har-

monics, it is also able to compensate for low orders as the fifth

and seventh. However, this feature would require a considerableincrease of the inverter power rating, not evaluated here.





TABLE IVCHARACTERISTICS OF THE PROPOSED TOPOLOGY DEPENDING ON THE DESIRED COMPENSATED

SPECTRUM AND SELECTED SWITCHING FREQUENCY OF EACH INVERTER

Fig. 11. Measured steady-state response of the APF2—feedforward compen-sation. The harmonic compensation reference is set for mitigation of the highorder (11th to 31st) harmonic currents.

A typical stationary response for the feedforward APF2 is

given in Fig. 11, where the total line current still contains a sig-

nificant quantity of harmonic distortion, but with smoother cur-

rent gradients. As the current spectrum indicates (see Table V

column 5) the APF2 filter compensates only for the harmonic

currents higher than 11th order.

C. Combined Compensation

Different tests can be performed with the proposed topologydepending on the imposed compensated spectrum and the se-

lected switching frequency of each inverter, as it can be seen in

Table IV. In this paper only Case-1 and Case-3 are tested.

First, both feedback and the feedforward loops are connected

together and set as in Case-1, Table IV. The result, seen in the

line current waveform, represents the cumulative effect of both

loops. Thus, both low order and high order harmonics are com-

pensated, as shown in Fig. 12. Table V compares the harmonic

current spectrum for each of the measured cases. The proposed

topology brings the THD from the initial value of 27%, without

any filtering, down to less than 2% with both APF filters active.

The control method uses harmonic integrators, which ideally

present infinite amplification at the selected harmonic. There-fore, the mitigation performance of each branch should theo-

Fig. 12. Measured steady-state response of the proposed topology. Both loworder and high order harmonic currents are mitigated down to 2% THD , by

enabling APF1 and APF2 together.

retically be identical (see columns 2 and 4). Furthermore, it is

proven in [6], that although some of the gains are different,

the transfer functions of the feedforward and feedback loops

are equivalent. This should give the same harmonic mitigation

performance of the feedback compared to the feedforward re-

spective the proposed topology. However, it is experimentally

noticed that the proposed topology provides a better harmonic

current mitigation, presumably due to the cascade configuration

(see columns 6 respective 7 of Table V). In order to measure thedata presented in columns 2 and 4 of Table V, the control of each

branch APF1 or APF2 is configured as in (13) by enabling all

five orders of the harmonic integrators, 6,12,18,24,30 .

The second test is with the proposed topology set as in Case-3

Table IV, but in a particular case as redundant-mode, which is

done by setting the parameter 1 in (15) and Fig. 5. The

steady state harmonic mitigation performance can be seen in

Table V column 7, which is the best compared to the other pre-

sented cases. Fig. 13 shows the dynamic response of the entire

APF for a load step of 5 kW due to the ASD. Active filter APF1

responds slower because of the lower gain of its equiva-

lent harmonic integrator. Active filter APF2 responds faster and

takes over the compensation of the fifth and seventh harmonicsduring the transient. This is possible due the enabled redun-





TABLE VCOMPARISON OF THE APF’S HARMONIC MITIGATION PERFORMANCE FOR EACH MEASURED

CASE: F EEDBACK, FEEDFORWARD AND THE PROPOSED PARALLEL TOPOLOGY

Fig. 13. Dynamic response of the proposed topology in redundant-mode, for aload step of 5 kW created by the ASD. APF1 responds slower than APF2 dueto different parameters set for the current controller.

dant-mode as it is also shown in Fig. 6. As APF1 reaches the

steady-state conditions, APF2 ceases to mitigate the fifth and

seventh harmonic currents.

Another measurement with the redundant-mode is presented

in Fig. 14, with APF1 starting up (reproducing a possible

maintenance intervention when APF1 had to be stopped and

restarted). Fig. 14 shows the dynamic response of the proposed

topology when APF1 starts. The active filter APF2 is running,replacing the harmonic mitigation tasks of APF1. As soon as

Fig. 14. Dynamic response of the proposed topology in redundant-mode, when

the APF1 is powered on while APF2 was compensating the overall harmoniccurrent spectrum. Once APF1 becomes active APF2 keeps its main task of com-pensating only the higher harmonic orders.

APF1 becomes active, both active filters return to normal oper-

ation (i.e., APF1 takes out lower harmonics and APF2 higher

harmonics), due to the load-sharing feature. As APF1 has to

start from zero-conditions, its dc-voltage has to be increased

to the reference value, which explains the fundamental current

drained at the beginning. As it can be seen the output current of

both APF inverters are similar, which means that both must besized to withstand the maximum harmonic current.







Fig. 15. Estimated ef ficiency limit based on the switching frequency and theload current THD .

is output inverter current, which can be expressed as a function

of the load current and filter THD , is a constant

including all other constant parameters, and is the abso-

lute value function.

Assuming that in the harmonic-sharing mode the inverter

switches at a lower frequency, the switching losses of APF1

inverter are reduced by a factor of

THD

THD (21)

Switching losses in harmonic-sharing mode are

THD THD

(22)

Switching losses in load-sharing mode are equally distributed

between inverter

THD (23)

The ef ficiency improvement of the harmonic-sharing compared

to load-sharing mode is estimated as

THD THD

THD (24)

Fig. 15 gives a graphical interpretation of (24), as a function of

the switching frequency factor .

Thus, assuming a reduction by half of the switching fre-

quency, i.e., 0.5, the ef ficiency improvement of the

harmonic-sharing is

for theoretical

ASD model for measured data (25)

One may determine the boundary limit of the switching fre-

quency where the ef ficiency of the harmonic-sharing is lower

compared to the load-sharing. Considering also the data inTable VI, it seems that in load-sharing the inverters have an

Fig. 16. Steady-state response of theAPF filters’ currents and theirduty-cycles.

overall smaller power, even though the ef ficiency may not be

very high. The ef ficiency of the harmonic-sharing mode is

attractive only if the switching frequency of the APF1 inverter

(mitigating the lower order harmonics) is at maximum two

times smaller than the switching frequency of the APF2 inverter

(mitigating the higher order harmonics).

As it was previously mentioned, another advantage of the

topology is that, having two separately controlled inverters, they

can be interleaved by 180 , to reduce the line switching current

ripple.

Fig. 16 shows the filter currents and along with their

duty-cycles. It can be noticed that, even the output currents aredifferent, the duty-cycles look similar. This is because the in-

verter voltage must be comparable to the line voltage in order

to make the current flowing through the line inductors. Thus,

the duty-cycles contain a fundamental component that makes

their waveforms to resemble. This property is used to reduce

the switching ripple of the line current if one uses two carriers

shifted by 180 . A simplified illustration of the interleaving is

provided in Fig. 17 for one phase only. The amplitude of the

current ripple in each inverter depends on the used inductors.

Furthermore, even though the carriers are interleaved by

180 , the duty-cycles are ac-signals, therefore determining

different pulse width lengths in each inverter. Thus a complete

cancellation of the current ripple at the switching frequencyis not possible. However, the interleaving helps in minimizing

the total line current ripple. The switching frequency is only

partially canceled, as it can be seen in Fig. 18, which reduces

the requirement of a stronger line filter against EMIs. A smaller

EMI filter is assumed to be inexpensive, as it requires smaller

line connected capacitors. Furthermore, smaller capacitors

mean an increase of the system stability, because possible

power system resonances between the capacitors and source

impedances are shifted to higher frequencies. The resonances

may be easily damped and may not interfere with the control.

If the rated power of APF2 is of a small value, as it is cal-

culated in (17), then the load-sharing feature cannot be used.

Therefore, one may need to assign a higher rated power for thesecond inverter (APF2), in order to operate in load-sharing. This





Fig. 17. Simplified description of the current ripple based on the on-off con-

duction instants with the carriers interleaved by 180 .

Fig. 18. Harmonic spectrum of thefilters’ currents and total line current. Spec-trum shows the reduction of the switching frequency

f

in the source current

I

due to 180 interleaved carriers.

means a higher cost for the second inverter, but the advantage

is the achievement of redundancy and ride-through functions

when one inverters fails. The ratio of the inverter’s size becomes

a matter of practical design. For instance if APF2 is only used

for mitigation of transient conditions due to slower APF1 re-

sponse, then its rated power may be selected in the range of

25%–50% of APF1 power. If APF2 is going to be permanently

used for load-sharing, then its rated power would be comparableto APF1.

VI. CONCLUSION

This paper describes a new topology for dual operation of ac-

tive power filters. The novelty consists in a parallel connection

of two active filters, one working in feedforward current loop

and the other in feedback current loop, both sharing the same

load current sensor. The proposed topology is theoretically an-

alyzed and then practically tested on a laboratory platform.The most significant features of the proposed solution are as

follows.

— It improves the stationary performance compared to an

APF based on a single inverter.

— It provides an improved dynamic response in frequency-

sharing operation due to the fact that one filter is pro-

grammed for fast compensation, while the other is effec-

tive in stationary operation.

— Load distribution with or without redundancy is possible

in load-sharing operation.

—Wireless operation of the twofilters is an intrinsic property

of this topology in all operating situations.

—Current ripple reduction is possible and was confirmed ininterleaving operation.

— The overall achieved THD performance is below 2% from

uncompensated THD value of 27%.

The proposed solution is targeted for medium power Ad-

justable Speed Drives applications where the harmonic pollu-

tion is a significant issue.

REFERENCES

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Lucian Asiminoaei (S’03–M’06) was born inGalati, Romania, on April 3, 1973. He received theB.Sc. and M.Sc. degrees in electrical engineeringfrom “Dunarea de Jos” University of Galati, in 1996and 1997, respectively, and is currently pursuing thePh.D. degree at the Institute of Energy Technology,Department of Power Electronics and Drives, Aal-borg University, Aalborg, Denmark.

From1996 to1999, hewas withIron& Steelworks

Sidex S.A., Galati, as a Maintenance Engineer and in1999 moved to the IT Department, IspatSidex LNM

Group, Galati, as a Hardware Engineer. In February 2003, he joined the Instituteof Energy Technology, Department of Power Electronics and Drives, AalborgUniversity, and is involved in projects sponsored by Danfoss Drives A/S, Den-mark and Power Lynx A/S, Denmark. His areas of interests include harmonicmitigation, harmonic measurement, and design of active and hybrid filters.

Cristian Lascu received the M.S. and Ph.D. de-grees in electrical engineering from the UniversityPolitehnica Timisoara, Timisoara, Romania, in 1995and 2002, respectively.

In 1995, he joined the Department of ElectricalEngineering, University PolitehnicaTimisoara wherehis research was focused on power electronics andhigh performance electrical drives. He was a VisitingResearcher at the Institute of Energy Technology,

Aalborg University, Aalborg East, Denmark, in 1997and 2005, and with the Department of Electrical

Engineering, University of Nevada, Reno, from 1999 to 2000. From 2002 to

2004, he was with SIEI S.p.A., Italy, working on advanced power electronicsand drives for electrical vehicles under a European Marie Curie Fellowship.

Dr. Lascu received the IEEE IAS Prize Paper Award in 1998.

Frede Blaabjerg (S’86–M’88–SM’97–F’03) wasborn in Erslev, Denmark, on May 6, 1963. Hereceived the M.Sc.EE. and Ph.D. degrees fromAalborg University, Aalborg, Denmark, in 1987 and1995, respectively.

He was with ABB-Scandia, Randers, Denmark,from 1987 to 1988. He became an Assistant Pro-fessor in 1992 at Aalborg University, in 1996 an

Associate Professor, and in 1998 a Full Professorin power electronics and drives. In 2000, he was aVisiting Professor with the University of Padova,

Padova, Italy, as well as a part-time Programme Research Leader in windturbines at the Research Center Risoe. In 2002, he was a Visiting Professorat Curtin University of Technology, Perth, Australia. He is involved in morethan ten research projects within the industry. Among them is the DanfossProfessor Programme in Power Electronics and Drives. He is the author orcoauthor of more than 350 publications in his research fields including Controlin Power Electronics (New York: Academic, 2002). He is an Associate Editorfor the Journal of Power Electronics and Elteknik . He has been very involved inDanish Research policy in the last ten years. His research interests are in powerelectronics, static power converters, ac drives, switched reluctance drives,modeling, characterization of power semiconductor devices and simulation,wind turbines, and green power inverters.

Dr. Blaabjerg received the 1995 Angelos Award for his contribution inmodulation technique and control of electric drives, the Annual Teacher Prize

from Aalborg University, in 1995, the Outstanding Young Power ElectronicsEngineer Award from the IEEE Power Electronics Society in 1998, five IEEEPrize paper awards during the last five years, the C. Y. O’Connor fellow-ship from Perth, Australia in 2002, the Statoil-Prize for his contributions inpower electronics in 2003, and the Grundfos-prize for his contributions inpower electronics and drives in 2004. He is an Associate Editor of the IEEETRANSACTIONS ON INDUSTRY APPLICATIONS and the IEEE TRANSACTIONS ON

POWER ELECTRONICS. He is a member of the Danish Academy of TechnicalScience, the European Power Electronics and Drives Association, and theIEEE Industry Applications Society Industrial Drives Committee. He is also amember of the Industry Power Converter Committee and the Power Electronics

Devices and Components Committee, IEEE Industry Application Society.

Ion Boldea (M’77–SM’81–F’96) received the B.S.

and Ph.D. degrees in electrical engineering from theUniversity Politehnica, Timisoara, Romania, in 1967and 1973, respectively.

He is currently a Full Professor at the UniversityPolitehnica Timisoara. He has worked and publishedextensively on linear and rotary electric machinesand their power electronics control, with and withoutmotion sensors. He co-authored Induction MachineHandbook (Orlando, FL:CRC, 2001), Linear Motion

Electromagnetic Devices (London, U.K.: Taylor &Francis, 2001), Electric Drives (Boca Raton, FL: CRC, 2006), and The ElectricGenerators Handbook (Boca Raton, FL: CRC, 2006). He spent about five yearsas a Visiting Scholar in the U.S. and U.K., and presented keynote addresses,intensive courses, and does consultant work in the U.S., Europe, and Asia. He is

Associate Editor of the Electric Power Components and Systems Journal, andDirector and Founder of the Internet-only, Journal of Electrical Engineering.

Dr.Boldeais an active memberof the Industrial Drivesand Electric Machines

Committee, IEEE Industry Applications Society, and Co-chairman of the IEEEIAS Technically Sponsored OPTIM 96, 98, 00, 02, 04, and 06 InternationalConferences.

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