UNIVERSITA DEGLI STUDI DI FIRENZE
DIPARTIMENTO DI FISICA
DOTTORATO DI RICERCA IN FISICA
Performances and Tests on the forwardsensors of the CMS Silicon Tracker
Tesi di Dottorato di Ricerca in Fisica di
Simone Busoni
Relatore Dott. Carlo Civinini
Relatore esterno Dott. Andrea Vacchi
XIII Ciclo di Dottorato
Coordinatrice Prof. Anna Cartacci
Firenze, 30 Dicembre 2000 Anno Accademico 1999/2000
Contents
Introduction 1
1 CMS experiment at LHC 51.1 The Large Hadron Collider LHC . . . . . . . . . . . . . . . . . . . . . . . . . 51.2 Physics at LHC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
1.2.1 The SM Higgs sector . . . . . . . . . . . . . . . . . . . . . . . . . . . 91.2.2 Standard processes . . . . . . . . . . . . . . . . . . . . . . . . . . . . 101.2.3 bbb quark physics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111.2.4 SUSY sector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
1.3 CMS detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 121.3.1 Magnet . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 141.3.2 Muon spectrometer . . . . . . . . . . . . . . . . . . . . . . . . . . . . 151.3.3 Calorimetry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
1.4 Data Acquisition and trigger . . . . . . . . . . . . . . . . . . . . . . . . . . . 171.5 Tracker . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
1.5.1 The Pixel Subdetector . . . . . . . . . . . . . . . . . . . . . . . . . . 221.5.2 The Silicon Microstrip Tracker . . . . . . . . . . . . . . . . . . . . . . 24
2 Silicon microstrip detectors 272.1 Silicon properties . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 272.2 The p-n junction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 312.3 Principle of operation of silicon detectors . . . . . . . . . . . . . . . . . . . . 35
2.3.1 Energy loss of high energy charged particles in silicon . . . . . . . . . 362.4 Silicon microstrip detectors . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
2.4.1 Single sided device . . . . . . . . . . . . . . . . . . . . . . . . . . . . 382.5 The Florence detector prototypes . . . . . . . . . . . . . . . . . . . . . . . . . 41
2.5.1 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . 442.6 Signal and Noise evaluation . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
2.6.1 Charge collection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 512.6.2 Noise evaluation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52
3 Irradiated silicon microstrip detectors 593.1 Radiation damage in silicon detectors . . . . . . . . . . . . . . . . . . . . . . 60
3.1.1 Surface damage effects . . . . . . . . . . . . . . . . . . . . . . . . . . 603.1.2 Bulk damage effects . . . . . . . . . . . . . . . . . . . . . . . . . . . 603.1.3 The absorbed dose expressed as 1 MeV neutron equivalent fluence . . . 65
3.2 Irradiation of silicon detectors and dosimetry . . . . . . . . . . . . . . . . . . 66
i
3.3 Characterization of irradiated detectors . . . . . . . . . . . . . . . . . . . . . . 693.3.1 Leakage current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 693.3.2 Bulk capacitance and full depletion voltage . . . . . . . . . . . . . . . 703.3.3 Bias resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 703.3.4 Coupling capacitance . . . . . . . . . . . . . . . . . . . . . . . . . . . 713.3.5 Interstrip capacitance . . . . . . . . . . . . . . . . . . . . . . . . . . . 723.3.6 Total capacitance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74
4 The APV6 front-end chip 774.1 The APV6 chip . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 77
4.1.1 Analogue stages . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 804.1.2 Control interface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 814.1.3 Operation modes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82
4.2 APV6 chip response . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 844.2.1 APV6 characterization . . . . . . . . . . . . . . . . . . . . . . . . . . 86
4.3 The APV25 read-out chip . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 90
5 The laboratory setup 915.1 The Florence laboratory setup . . . . . . . . . . . . . . . . . . . . . . . . . . 915.2 The Tracker Interface Card . . . . . . . . . . . . . . . . . . . . . . . . . . . . 935.3 The Sequencer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 94
5.3.1 The timing circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 965.4 The timing sequence . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 99
5.4.1 Internal Calibration Mode . . . . . . . . . . . . . . . . . . . . . . . . 1005.4.2 DAQ mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 101
5.5 The Data Storage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1035.5.1 The FED ADC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103
6 The laser test station 1056.1 The laser source . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1066.2 The laser driver . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1086.3 The optical and the positioning systems . . . . . . . . . . . . . . . . . . . . . 1116.4 System performances . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 113
7 Performances of the detector prototypes 1177.1 Off-line analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 117
7.1.1 Cluster and total charge reconstruction . . . . . . . . . . . . . . . . . . 1227.2 The 300 µµµm detectors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 123
7.2.1 The βββ source measurements . . . . . . . . . . . . . . . . . . . . . . . 1237.2.2 The Beam Test measurements . . . . . . . . . . . . . . . . . . . . . . 1267.2.3 Results summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 129
7.3 The 500 µµµm detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1327.3.1 The modules . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1327.3.2 Florence laboratory results . . . . . . . . . . . . . . . . . . . . . . . . 133
Conclusions 137
ii
Appendix 141
A APV6 response parametrization 141
B The effect of deconvolution on noise 145
C The Sequencer schematic 149
Acknowledgements 157
iii
IntroductionThe new generation of high energy experiments at colliders will use silicon tracking detectors
in a heavier way than in the past. High particle production rate requires a high granularity and
fast detector response. The use of gas detectors in this environment is often discouraged due to
the high density of particle tracks and response speed.
In particular the CMS [1] experiment at LHC [2] will build an “all silicon” tracking detec-
tor [3] [4]. This choice is dictated by the necessity of a robust tracking and a detailed vertex
reconstruction in a very dense particle environment due to the high luminosity necessary to ac-
cess the full physics range of proton-proton collisions at the LHC energy. The main challenge
involved in such a project is the construction of a large area silicon detector, with a surface of
230 m2 to be compared with about 0.7 m2 of LEP vertex silicon devices, equipped with fast
electronics and that has to face with as low as possible material budget. In addition, the CMS
silicon tracker will have to operate in a heavy radiation environment, with a high bunch cross-
ing rate and a strong magnetic field. All these constraints have to be fulfilled by a tracker that
should be built, due to the very high number of detectors needed, using an industrial approach.
The instrument that should match the requirements of this scenario will consist of an inner
part, very close to the interaction point, built with silicon pixel technology and of an outer part
with silicon microstrip devices. After a long and careful R&D program, carried out both on
detectors and electronics, the first production phase is going to start.
The work and results described in this thesis are located in the larger contest of the activity
performed by the Florence1 CMS group, responsible for the construction of part of the mi-
crostrip detector, for the Silicon Microstrip Tracker collaboration. The main target of the thesis
is the study of the performances of several full size detector modules, with different geometry
(pitch and thickness) and electrical characteristics, in order to obtain the best sensor definition
for the tracker. Some of the detector crystals were heavily irradiated to simulate and study the
effects of 10 years of operation in the hostile LHC radiation environment. In addition, a deep
study of the operation principle and characteristics of the CMS microstrip silicon tracker front-
1INFN and University of Florence
1
end chip prototype (APV6) [5] has been performed, together with the setting up of a reliable
data acquisition system (DAQ in the following). The DAQ must be flexible enough to allow
to perform electronics and sensors tests in laboratory, maintaining, at the same time, the basic
characteristics of the CERN Beam Test readout chain. The pre-production phase start will re-
quire a fast way to test detector; in this perspective a laser test station that can check out the
response of a complete module in a very short time has been built in Florence in the context of
the activities related to this thesis.
In the first chapter the main physics goals of LHC are summarized, together with their
implications on the characteristics and performances of the Silicon Tracker. Then the CMS
experiment and its components are described, with particular attention to the Silicon Microstrip
subdetector and its new layout after the 1999 winter revision.
In the second chapter the principle of operation of silicon microstrip detectors is reported,
together with main electrical and geometrical parameters that drive their performances in terms
of signal to noise (S/N) response to minimum ionising particles (MIPs). Laboratory measure-
ments on sensors, designed by the CMS Florence group, are presented. Finally, a review of all
the fullsize modules prototypes built from these sensors and tested during this thesis work is
given and their characteristics are critically discussed.
Most of the work done in this thesis is a comparison between irradiated and non irradiated
detectors performances and the study of the survival of silicon devices in a LHC like radiation
environment. In chapter 3 the neutron irradiation procedure performed on a set of detectors is
described and their characterization is briefly summarized.
The studies performed on the front-end read out chip APV6 are presented in chapter 4.
Special care has been given to the definition of the steps necessary to test the full functionality
of the chips sitting on the front-end read out hybrid.
The project and the construction of a custom sequencer board used to drive the readout chips
are described in chapter 5, together with the front-end interface card used to connect the detector
to the digitisation system, whose main block is the official CMS Tracker ADC, the Front End
Driver (FED) [6]. The system allows to test the hybrid alone as well as the fullsize detector; in
the latter case a β source is used to study particle detection and S/N performances directly in
our laboratory.
2
Based on the same DAQ system, a laser test station, whose main component is a 1064 nm
pulsed laser diode, has been implemented. An original laser driver and an optical system have
been built to obtain the desired performances in terms of time response and laser spot size. The
detector can be placed on a pair of orthogonal axis that can be positioned by remote control
under the laser spot. This system, described in details in chapter 6, allows a fast check of the
response of all the strips and electronic channels of the device.
The results in terms of charge collection and noise are given in chapter 7 for fullsize modules
exposed to MIPs, emphasizing the dependence with respect to crystal orientation, substrate
resistivity, thickness as well as to irradiation effects. The results are compared with the expected
values according to the APV6 performances and characterization measurements performed on
the sensors. Part of these results have been published or submitted for publication [7] [8]. Beam
tests performed during this work using a particle beam at CERN SPS have confirmed the results
obtained in laboratory, although a more complex experimental setup closer to the final one is
used. This gives us confidence from one side that Florence setup can manage in performing
most of the preliminary testing measurements on complete detectors and on the other side that
the tested fullsize detectors prototypes well behave with respect to the constraints of CMS
experiment.
Notes.
A system of units in which = c = 1 is used in the following. Mass, energy and momentum
are expressed in GeV, unless otherwise stated.
The reference system adopted to describe the detector layout has a cylindrical geometry, with
the rotation axis referred to as z.
3
Chapter 1
CMS experiment at LHC
1.1 The Large Hadron Collider LHC
High energy physics has based many of its research fields on experiments at accelerator. In
the last decade a great contribution has come from LEP collider located at CERN, which has
been taking its last data in November 2000 at the project energy limits (∼208 GeV centre of
mass energy for electron positron collisions). A big effort is being made to build a new hadron
collider that will be housed in the 27 Km long LEP tunnel. This project, known as Large Hadron
Collider (LHC) [2], will provide proton-proton collisions with an unprecedented centre of mass
energy of 14 TeV, as well as heavy ions collisions (lead-lead) up to 1312 TeV, covering in this
way several physics fields. The two collider beams will counterrotate in separated pipes, bent
by superconducting magnets and accelerated by superconducting RF cavities. They will interact
in four points corresponding to the experiments sites approved by LHC committee: CMS [1],
ATLAS [9], LHCb [10], Alice [11] (see Fig. 1.1 ).
More in detail, 1238 magnetic dipoles generating a magnetic field up to 8.4 Tesla, 386
quadropoles, 360 sextopoles and 360 ottopoles will be used to steer the particles on their tra-
jectory. Protons will be injected in the LHC rings after three pre-acceleration stages that will
sequentially use the ”Proton Linac”, able to accelerate protons up to 50 MeV, the PS complex,
up to 26 GeV, and the SPS up to 450 GeV.
One of the challenges related to experiments with proton storage rings is the need to increase
the luminosity proportionally to the square of the centre of mass energy. This is necessary in
order to keep abreast of the cross section for the processes that are of interest to high energy
physicists, which falls as the square inverse of the mass of the particle one wishes to discover
5
Figure 1.1: LHC experiment sites and accelerator complex at CERN.
[12]. To this end LHC luminosity L, defined for a collider as:
L =N1N2nbf
4πσxσy(1.1)
will reach the value of 1034 cm−2s−1 after a first start-up period, defined as low luminosity
phase, lasting a few years at 1033 cm−2s−1. In equation 1.1 N1,2 is the number of particle
per bunch, nb the number of bunches, f the bunch orbit frequency, σx,y the bunch transverse
dimensions. For heavy ions collisions L will reach the value of 1027 cm−2s−1.
To achieve such an unprecedented value the two beams will contain up to 2835 bunches,
each filled with 1.1 · 1011 protons. The bunch crossing rate will be 40 MHz, corresponding to
25 ns collision time.
The high bunch crossing rate and the high energy of the particles involved in the collisions
are expected to create a very dense background due principally to low energy secondaries from
proton-proton collisions and, especially in the outer region of the tracker, neutron albedo from
the calorimeters. All these characteristics impose severe constraints on electronics, which must
be fast enough to keep up with crossing rate in order to avoid pile-up from more than one
6
bunch crossing, and on implementation, where necessary, of radiation hard devices both for
subdetectors and electronics.
1.2 Physics at LHC
The Large Hadron Collider, operating at a centre-of-mass energy of 14 TeV with a design lu-
minosity of 1034cm−2s−1, will be the first machine to probe parton-parton collisions directly at
energies ≈ 1 TeV [2]. The Standard Model (SM) [13] of particle physics, the theory of elec-
troweak and strong forces, provides a remarkably successful theoretical picture. This theory
has been tested rigorously at LEP, the Tevatron and the linear collider at SLAC. Nevertheless
there is a key question that is waiting for an answer: the discovery of the scalar Higgs boson,
predicted by the SM in the mass generation section and last element still experimentally miss-
ing. Other fields of interest in particle physics that can be investigated at the LHC are the test
of Supersymmetry theory (SUSY) or any extension of the SM, the search for new particles and
the study of CP violation in B system.
The total proton-proton cross section at hadron colliders is very large, about 100 mb at the
LHC as can be extrapolated from the results obtained at lower energies from previous experi-
ments (UA1,CDF, etc.). The corresponding inelastic cross section is ≈ 70 mb. An average value
of 25 minimum bias events1 piled up at every bunch crossing are expected for high luminosity
runs at the LHC if we consider empty bunches (∼ 20%). The expected energy dependence
of the total cross section and of some interesting physics processes produced by proton-proton
collisions is shown in Fig. 1.2, together with the event rates foreseen at LHC during high lumi-
nosity phase.
The minimum bias events are an unavoidable background for all the processes involved in
LHC physics and affects most of the detector design choices.
From the same plot it appears that the Higgs boson production, in the mass range of about
500 GeV and at LHC energy, is of the order of ∼ 105 events per high luminosity year2. The
experimentally easiest discovery signature, H → ZZ → 4l±, has a branching ratio ≈ 3 · 10−4,
and thus requires a large integrated luminosity.
1A minimum bias event is a single proton-proton interaction, regardless of being or not selected by the experi-ment trigger.
2Each high luminosity year corresponds to 107 s with a 1034 cm−2s−1 luminosity.
7
Figure 1.2: Energy dependence of some characteristic cross-section at hadron colliders.
LHC allows a deep study of b-quark physics as is shown by the rise of the bb cross section,
up to a value of 106 events per second. This property will make of LHC an unique instrument
to perform a broad B-physics program, starting from tagging and reconstruction of b-jets and B
hadrons within these jets, mainly using the information coming from the tracker.
A similar favourable situation is tt production, with 106 quark pair per year already from low
luminosity running. Top physics will play an important role at LHC since from first year the top
quark properties will be measured with excellent precision, starting from its mass, production
cross section, branching ratios couplings and exotic decay channels searches.
Main physics field can be divided into standard processes, Higgs boson search, B meson
and SUSY sector studies.
8
1.2.1 The SM Higgs sector
The main goal of LHC is the search of the Higgs boson within the Standard Model (SM) in
order to investigate the process of the electroweak symmetry breaking. The Higgs mass is not
predictable within the SM but theoretical constraints related to perturbative consistency lead to
an upper bound of about 1 TeV. Arguments of vacuum stability suggest a lower Higgs mass
limit [14], depending also strongly on the top mass. Taking the measured value of the top mass
(mt = 174.3± 5.1 GeV) [15] and assuming that no new physics exists below the Planck scale,
the Higgs mass should be around 160±20GeV. At present the four LEP experiments have ruled
out the existence of a Higgs with a mass of less then 107.9 GeV at 95% confidence level [16].
The LHC allows to cover the SM Higgs mass range from the expected LEP200 limit all the
way up to about 1 TeV. At hadron colliders the basic Higgs production mechanisms, sketched
in Fig. 1.3, are gluon-gluon fusion, WW(ZZ) fusion, tt fusion and W(Z) bremsstrahlung pro-
duction [17].
H0
t
tt
g
g
gg fusion
H0
WW ZZ fusion
W,Z
W,Z
q
q
q
q
W,Z
t
t
H0
tt fusion
g
g
t
t
H0
W,Z bremsstrahlung
W,Z
q
q
Figure 1.3: Main Higgs production mechanisms at LHC.
At the LHC the gluon-gluon fusion provides the dominant contribution over most of the
accessible mass range, but at the highest masses (mH ≥ 0.7 TeV) the WW(ZZ) fusion becomes
comparable and provides an additional event signature thanks to the two energetic and forward
9
tagging jets. At the lower end of the Higgs mass range, the WH (and ZH) bremsstrahlung mech-
anism provides an additional signature owing to the accompanying W (or Z). A Higgs boson
production about 106 ÷ 104 events per year is foreseen at a luminosity of L= 1034 cm−2s−1.
CMS detector is designed to access the entire SM Higgs mass range taking into account the
different decay modes and experimental signatures. Depending on the Higgs mass value, there
are different best experimental signature for Higgs discovery.
In the mass range between 80 and 130 GeV the most promising signature is the decay
H → γγ, with a branching ratio of only ≈ 2·10−3. The natural width of the Higgs in this case is
very narrow (<10 MeV) and thus the observed signal is entirely dominated by the experimental
γγ mass resolution. Taking into account that the signal is superimposed to a large irreducible
QCD diphoton background a mass resolution better than 1% is required for the electromagnetic
calorimeters. The signal significance is greatly enhanced by an efficient reconstruction of all
the hadronic tracks down to pt of 2 GeV thus allowing the rejection of π0 − π0 (jet - jet) and
the π0 − γ (jet) background. Such isolation criteria rely strongly on performances of tracking
detector too.
In the mass range 130 GeV < MH < 800 GeV, where the Higgs total width reaches the
value of a about 200 GeV, the decay H → ZZ → 4l+l− and H → ZZ → 4l+l− provide
the experimentally easiest discovery signature as the events should contain four isolated high
pT leptons. These decay channels require good integration of data from both tracker, muon and
electromagnetic calorimeter detectors.
For Higgs masses above ≈ 500 GeV (in this case the width ΓH varies as ΓH ≈ 0.5 TeV ·(MH/1 TeV)
3), additional signature involving hadronic W and Z decays as well as invisible
Z decays like H → ZZ → l+l+νν should be also used. These high mass Higgs signatures
involve missing transverse energy and jet-jet masses and require thus hermetic detectors with
good jet-energy reconstruction.
1.2.2 Standard processes
The analysis of standard processes includes all those phenomena that need a deeper investiga-
tion or further confirmation at LHC energies. In particular a great interest will be devoted to the
total cross section for p−p collisions that at present is extrapolated from previous experiments.
10
The same extrapolation leads to expect a ratio between elastic to total cross section of the order
of 0.26. This information has a direct consequence on LHC detectors since it is related to the
minimum bias events present at LHC. The understanding of minimum bias events in a hard
scattering process is important as they can limit or spoil the calorimeter resolution and increase
the tracking detector occupancy and thus degrading the lepton and photon isolation criteria,
crucial in order to discover new particles. At LHC high pt jets, produced in anelastic collisions,
are accessible up to the TeV range. An important goal of jet physics is to look for possible de-
viations at high transverse energy from expected QCD point-like behaviour, which could reveal
a possible composite structure of the quarks. Another field of interest is the direct photon pro-
duction, foreseen at the rate of one photon per day with EγT = 1 TeV. At the low end of the Eγ
T
spectrum, as gq → qγ is the dominant production mechanism, direct photon production allows
investigation of the poorly known low-x behaviour of the gluon structure function. The high
production rate of W and Z bosons is a source of a substantial background in the Higgs boson
search but, on the other hand, allows investigation of coupling characteristic of the electroweak
SM.
1.2.3 bbb quark physics
LHC is a powerful tool for observing particles containing heavy quarks, when such particles
have a measurable decay length. At a centre of mass energy Ecm=14 TeV the bb cross section
allows the production of ∼ 1013 b-quark pairs per high luminosity year. The main issue in B
physics at the LHC is the observation of CP violation in the B system, and the ultimate goal is
to measure the three interior angles of the Cabibbo-Kobayashi-Maskawa (CKM) matrix unitary
triangle. CP violation, initially discovered in the K 0 meson decay, can be studied in the B0B0
system also. Resolution obtained with silicon detectors allows the characterization of displayed
secondary vertices and separation of tracks coming from multiple vertices. In Fig. 1.4 the decay
of B0 meson in two muons and two pions, through the creation of a J/ψ and a K 0s meson, is
shown. The identification of B0 displayed vertex allow b tagging, as well as the identification
of a semileptonic decay like b → clν.
Furthermore b jets can result from decays of new particles or in associated production via
gluon-gluon fusion mediated by b-quark exchange, thus giving access to the study of new
11
π—
π+
p p
µ—
B0
b
µ— µ+
jet
b
Figure 1.4: Representation of an event involving a bb decay.
physics. All these experimental signatures rely deeply on excellent tracking performances, two
tracks separation capability and secondary vertex resolution, combined with a muon detector.
1.2.4 SUSY sector
In the supersymmetric extension of the SM a set of new particles should exist with a mass scale
around 1 TeV. The minimal version of the supersymmetric SM (MSSM) contains three neutral
and two charged Higgs bosons, and one of the neutral ones is expected to have a mass around
100 GeV. For the lightest MSSM Higgs boson h the most appropriate decay channel to inves-
tigate is the same as the SM Higgs boson, i.e. h → γγ, and the experimental requirements
and expected backgrounds are similar to the SM H → γγ decay channel. Also the four lepton
channel is crucial for the discovery of a Higgs boson in the MSSM. These two decay chan-
nels,together with others involving the τ lepton, are able to cover most of the theory parameter
space. The energies available at LHC can explore this sector as well as the “sparticle” sector.
1.3 CMS detector
CMS is a general purpose proton-proton detector designed to run at the highest luminosity at
LHC, but it is also well adapted for studies at the initially lower luminosities. Like almost every
high energy detector at colliders it has a cylindrical geometry, covering with its subdetectors as
much a solid angle as possible. The overall dimensions are a length of about 22 m, a diameter
of 14.6 m and a total weight of 14500 tons. A CMS detector layout is shown in Fig 1.5.
12
Return yoke
Tracker
Superconducting coil
Crystal ECAL
Forward calorimeter
Muon chambers
HCAL
Figure 1.5: Layout of CMS detector.
Each subdetector is composed of a cylindrical part, referred to as barrel, coaxial to the beam
pipe, and a disk shaped one, called endcap, installed perpendicular to the beam axis at both
ends of the cylinder. This layout covers the full detectable volume and ensures a high detection
hermeticity. Large particle fluxes will make track reconstruction difficult and consequently a
high granularity and good time resolution, especially for inner detectors are needed.
Main design goals, in order to cleanly detect the different signatures of new physics at LHC,
are robust tracking, calorimetry and vertex reconstruction within a strong magnetic field to
identify and precisely measuring muons, photons, electrons and jets over a wide energy range.
In addition, a good impact-parameter resolution and secondary vertex reconstruction will play
an important role for b-tagging. The Tracker Detector, together with the muons chambers and
the e.m. and hadronic calorimeters, will provide this information.
The 4 Tesla magnetic field necessary to measure the particle momenta is provided by a large
13
superconducting solenoid. Muon chambers are located outside the coil while hadronic and
electromagnetic calorimeters are inside the magnet. Closest to the interaction point we found
the Silicon Tracker.
The overall layout aims to a compact, but with excellent performances, design for the muon
spectrometer, hence the name CMS (Compact Muon Solenoid). The experiment goal is to mea-
sure photons, muons, electrons energy with a resolution of about 1% over a wide momentum
range [18]. A further challenge is the implementation of a trigger system able to select, out of
the 4 · 107 Hz bunch crossing rate present at LHC, the most interesting physics events, with an
expected rate of about 100 Hz. At the first level, this task is accomplished by means of pipelined
front-end electronics.
1.3.1 Magnet
An important aspect of the overall detector design is the magnetic field configuration. Large
bending power is required to precisely measure high-momentum muons and other charged par-
ticles. The choice of the magnet structure strongly influences the remaining detector design.
The CMS magnetic field is provided by the largest and most powerful superconducting
solenoid ever designed with its 2.5 GJ stored energy [1]. The solenoid, working at liquid helium
temperature, will provide a very uniform magnetic field up to 4 Tesla over a cylindrical volume
of 13 m length and 5.9 m radius. The magnetic flux is returned through a 1.8 m thick saturated
iron yoke (with a 1.8 T return field). The return yoke is interleaved with four layers of muon
chambers. The overall design allows housing calorimeters and tracking detectors inside the coil.
Main result obtained by this configuration, thanks to the high magnetic field and favourable
length/radius ratio, is that bending power for charged particle tracking and muon detection up to
pseudo-rapidities 3 of 2.5 is provided without the need of forward toroids, simplifying the de-
tector design. An important aspect of a solenoidal magnet compared to a toroid is that the first
provides bending in the transverse plane and facilitates the task of triggering on muons, which
are pointing to the event vertex, so that one can take advantage of the small transverse dimen-
3Pseudo-rapidity η is a kinematics quantity defined as:
η = −ln(tgθ
2)
where θ = arccos(pz/p), p is the particle momentum and pz its projection along beam direction.
14
sions of the beam (∼ 20µm) [19]. The drawback of the degradation of momentum resolution in
the forward direction is overcome by the large length of CMS magnet design.
Since the magnet is the main element of CMS in terms of size, weight and structural rigidity,
it is used as the principal structural element to support all other barrel detector components.
1.3.2 Muon spectrometer
The muon detector should fulfil three basic tasks: muon identification, trigger and momentum
measurement. The high field solenoidal magnet and its instrumented iron flux return, which
also serves as the absorber for muon identification, ensure the performance of these tasks. The
muon detector has a geometric coverage up to pseudorapidity |η| = 2.4, since at the LHC
efficient detection of muons from Higgs bosons, W, Z and tt decays requires a large rapidity
acceptance. A track is identified as a muon candidate if it has penetrated through at least 16
interaction length (λ) of material.
Both barrel and endcap regions are equipped with four muon stations interleaved with the
iron return yoke plates. The overall geometry provides redundancy in track reconstruction
and reliability of the system. Several technologies have been adopted to provide the required
position determination (∼ 100µm). In the barrel, where the expected occupancies and rates are
low (< 10 Hz/cm2) and there is no appreciable radial magnetic field in the vicinity of most of
the muon stations, a system of drift tubes will be used. Each drift chamber module consists of
twelve planar layers of aluminium drift cells: eight layers of tubes parallel and four layers of
tubes perpendicular to the beam to provide respectively precise measurements along the rφ and
z coordinate.
In the endcap cathode strip chambers have been chosen because of their capability of func-
tioning in a highly non-uniform magnetic field. Furthermore such detectors can withstand high
rate and the signals from anode wires provide good time resolution for tagging the beam cross-
ing. Each chamber contains six layers with cathode strips oriented radially to measure the
azimuthal coordinate.
In addition, both barrel and endcap regions are equipped with resistive plate chambers lay-
ers to have dedicated trigger detectors with excellent timing capability (1 ns) and reasonable
position resolution. The precise muon chambers and fast dedicated detectors provide a trigger
15
with transverse momentum selection up to 100 GeV.
1.3.3 Calorimetry
Photons, electrons and hadrons energy measurement is accomplished by an inner high resolu-
tion electromagnetic calorimeter (ECAL) and an outer sampling hadron calorimeter (HCAL),
both housed inside the superconducting coil and subdivided in barrel and endcap regions. In the
endcap regions the electromagnetic calorimeter extends up to rapidity |η| = 2.6 and the hadron
calorimeter up to |η| = 3.0. This central calorimetry system is supported, in missing transverse
energy measurements and forward jets identification, by a very forward calorimeter that covers
the pseudorapidity range 3.0 < |η| < 5.0 and is located ±11 m from the interaction point.
The physics process that imposes the strictest performance requirements on the electro-
magnetic calorimeter is the Higgs boson decay in two photons (H → γγ) in the mass region
100 ≤ mH ≤ 140 GeV, where the Higgs width is only a few MeV and therefore the measured
mass resolution is entirely dominated by the experimental resolution. The CMS collabora-
tion has chosen a homogeneous electromagnetic calorimeter, made of Lead Tungstate (PbWO4)
crystals, to optimize energy resolution within the overall detector design. This choice is dic-
tated by the PbWO4 short radiation length (X0=9 mm) and small Moliere radius (2.2 cm) thus
leading to a compact ECAL. Other reasons are a short scintillation decay time constant (∼ 10
ns), which matches the LHC bunch crossing time of 25 ns, and a good radiation hardness. The
drawback of low light yield is effectively overcome by the use of new generation large area sil-
icon avalanche photodiodes. Crystals have a length of 23 cm ( 25.8 X0) in the barrel and 22 cm
in the endcap. The front face of each crystal is 20.5× 20.5 mm2 in the barrel and from 27×29
to 18× 20 mm2 in the endcaps. A preshower device, 3 X0 thick, is placed in front of crystals to
enhance neutral pion rejection in the end-cap region.
If we parametrize energy resolution as:
σE
E=
[a√E
⊕ σn
E⊕ b
]
where a is the stochastic term, b a constant and σn is the energy equivalent of noise, and E
is given in GeV, we expect an energy resolution of σE/E ∼ 0.6% for electrons and photons
of E=120 GeV. In Table 1.1 the contributions to energy resolution from ECAL and HCAL are
summarised.
16
Parameter ECAL HCALa ≤ 0.03 ∼ 0.8b ∼ 0.005 ≤ 0.03σn 0.15 ∼ 1.0
Table 1.1: Contributions to energy resolution in CMS calorimeter system for the barrel regionat small η.
The hadron calorimeter surrounds the ECAL and acts, in conjunction with it, to measure the
energies and direction of jets, also providing hermetic coverage for measuring missing trans-
verse energy. In the central region around η = 0 a hadron shower tail catcher is installed
outside the solenoid coil to ensure adequate sampling depth and reduce the hadron quenching
in the muon chamber region. The active elements of the barrel and endcap HCAL consist of
plastic scintillator tiles with wavelength-shifting fibre readout and copper absorbers. The tiles
are arranged in projective towers with fine granularity (lateral segmentation of ∆η × ∆φ ≈0.09 × 0.09) to provide good di-jet separation and mass resolution. HCAL performances play
an essential role in detection of the Higgs in the mass range mH 500 GeV, in both squark and
gluino searches, in QCD jet studies, in t-quark physics and in channels involving τ leptons in
the final state.
1.4 Data Acquisition and trigger
One of the main challenges at the LHC will be the reduction of 40 MHz interaction rate to about
100 Hz output rate of data recording for further off-line analysis while keeping high efficiency
on all interesting physics channels . The on-line data reduction will proceed via different trigger
levels. At the first level, local pattern recognition and energy evaluation on prompt macro-
granular information will provide “object” identification such as high-pt electrons, muons, jets
and missing transverse energy from muon and calorimetry system. Level-1 will select events at
105 Hz. In order to eliminate the Level-1 trigger dead time it is necessary to have a pipeline in
the front-end electronics so to be able to store events at bunch crossing rate for a time up to 3.2
µs. This feature is accomplished by the tracker APV6 chip that has a programmable register,
called latency , that allows to select the signals corresponding to the triggered event in the
chip analogue pipeline . For level-2 trigger, finer granularity and more precise measurements
17
will be used together with event kinematics and topology. By matching different subdetectors,
clean particle signatures will be selected resulting in a level-2 rate of 103 Hz. Finally, event
reconstruction and on-line analysis will result in physics process identification, leading to an
output rate of about 100 Hz. Except for the level-1 trigger, the remaining are software triggers.
1.5 Tracker
The detection and study of the different signature for new physics at the LHC will rely on the
clean identification and precise measurements of leptons, photons and jets. Robust tracking and
detailed vertex reconstruction within a strong magnetic field are essential tools to reach these
objectives. The CMS silicon inner tracking system provides precise momentum and impact
parameter and secondary vertex measurements for charged particles. It is also essential for e
and τ identification, and for the calibration of the electromagnetic calorimeter with electrons,
using the p/E matching.
The LHC environment imposes stringent requirements on the tracking detector with respect
to granularity, timing and radiation hardness. Another strong constraint is that pattern recogni-
tion and momentum resolution is affected by photon conversion and bremssttrahlung, so a low
material budget is desirable also in order to fully exploit the ECAL performance. This limits the
number of active layers and selects both the amount and type of material and the cable routing
layout.
A Monte Carlo study of Higgs to γγ decays shows that in 46 % of such decays both photons
leave the Tracker volume without converting and the loss of efficiency for this Higgs search
channel does not exceed other irreducible inefficiencies [20]. In addition, the unprecedented ef-
fort to build the biggest silicon sensitive surface ever realized makes indispensable an industrial
approach both to the tracker construction and to the financial resources managing.
The main challenge for a tracker at LHC is pattern recognition within a highly congested
environment. In the volume covered by the tracker, a background of about 500 soft charged
tracks, coming from ∼ 25 minimum bias events, is foreseen every bunch crossing at a luminos-
ity of 1034 cm−2s−1. To isolate interesting events and overcome pattern recognition problem,
low cell occupancy and large hit redundancy are required. Low occupancy can be obtained by
working with small detection cell size (high granularity) and fast primary charge collection. Re-
18
dundancy relies on the largest number of measured points per track, in line with an acceptable
material budget as mentioned before.
The very high magnetic field of CMS affects event topologies, by confining low pt charged
particles to small radius helical trajectories. Coupled with the steeply falling pt spectrum char-
acteristic of minimum bias events, this results in a track density which rapidly decreases with
increasing radius. This is illustrated in Fig. 1.6, where typical primary charged particle densities
are shown for different radii with 0 and 4 T solenoidal field, at η = 0. In the absence of a mag-
Figure 1.6: Primary charged particle density per cm2 at η = 0, for 20 minimum bias eventssuperimposed.
netic field, the charged track density simply falls off as 1/r2. Under the effects of the 4 T field,
the decrease in charged track density with radius is initially more gradual and then significantly
more pronounced than 1/r2. This has important implication for the architecture of CMS tracker.
In particular, granularity is such that typical single channel occupancy at high luminosity, for
detectors with at least one hit on them, is kept between 1% and 3% everywhere in the Tracker.
Two detector technologies, each best matched to the task of satisfying the stringent resolu-
tion and granularity requirements in the higher and lower particle density regions, have been
chosen. The inner part of the tracker is equipped with Pixel Detector, from a radius of 4 cm
19
from the interaction point to a radius of 19 cm, while in the outer region a Silicon Microstrip
Detector will be used up to a radius of 120 cm (see Fig. 1.7).
r
z
η
η
Figure 1.7: Longitudinal view of one quarter of the all silicon tracker.
The detector types chosen are both fast on the scale of 25 ns, allowing event pile-up to be
confined within to a single bunch crossing.
The Tracker baseline design has changed with respect to the TDR in December 1999, after a
deep R&D program and a suffered decision to abandon MSGC technology. The motivation for
changing the baseline design in favour of an all silicon tracker have to be found in the following
items [4]:
• The first key element is manufacturing sensors using new 6” instead of 4” industrial pro-
duction lines, of at least equal quality and high volume capacity. This will allow the use
of large area modules in the outer part of the tracker, with comparable dimensions as the
MSGC ones. Therefore in the outer region there will be a similar number of both modules
and read-out channels as initially foreseen.
20
• Second is the recently tested automation of module assembly, as well as the possibil-
ity of exploiting the recent generation of high throughput wire bonding machines, with
consequent time saving.
• Third a successful implementation of the front-end read-out chip in deep sub-micron
technology (APV25), cheaper and with improved S/N performances.
• Moving from two technologies to a single one is a unique opportunity to concentrate all
the efforts on a reduced set of problems.
• The reduction in surface, driven by budgetary constraints, allows to build an all silicon
tracker to very good approximation cost neutral with respect to the previous baseline
design. Due to the faster response and better charge localization of silicon compared to
MSGC’s, the tracker performance remains practically unchanged.
The main change in the overall mechanical design is the removal of the central support tube
between the central tracker and MSGC’s. This requirement was dictated by the necessity of
providing well separated thermal volumes for the silicon (to be operated at -10 C) and MSGC
tracker (to be operated at room temperature). In the actual all-silicon tracker this constraint no
longer exists.
One of the key element in tracker realization is its survival to heavy irradiation during the full
experiment lifetime. The region closer to the interaction point is strongly affected by radiation,
as it is shown in Fig. 1.8.
The radiation field within the tracker volume is characterised by two distinct sources. At
the inner layers the dominating contribution comes from secondaries from the pp interactions,
the products of their interaction in the structures and some decay products. Almost all of the
charged hadron fluence originates from the vertex. On the other hand, most of the neutron
radiation is due to albedo from the surrounding electromagnetic calorimeter. To survive this
high radiation environment, the whole tracker needs to be kept cold (see section 3.1). For this
reason the entire tracker volume will be permanently maintained at -10C and only for limited
periods of time it will be allowed to reach temperatures above 0C for maintenance purposes.
Next paragraphs describe more in detail the pixel detector and the new silicon microstrip
detector. Both apparatus are arranged in a barrel geometry in the central rapidity region, while
21
10 4
10 5
10 6
0 100 200
z (cm)
Dose (Gy)
7 cm
21 cm
49 cm
75 cm
111 cm 10 13
10 14
0 100 200
z (cm)
Neutrons (cm-2)
7 cm
21 cm
49 cm
75 cm
111 cm
10 13
10 14
10 15
0 100 200
z (cm)
Ch. Hadrons (cm-2)
7 cm
21 cm
49 cm
75 cm
111 cm
Figure 1.8: Expected value for absorbed dose and neutron and charged hadrons (and neu-tral kaons) fluences at selected radii. All values correspond to an integrated luminosity of5 · 105pb−1, that is expected over ten years of LHC operation. The neutron fluences includeonly the spectrum above 10 keV.
at higher values of rapidity they are deployed as end-cap disks.
1.5.1 The Pixel Subdetector
The CMS pixel system [3] consists of two barrel layers and two end layers (end disks) on each
side of the barrel. The barrel layer extends in radius from 4 cm up to 7 cm at low luminosity
and from 7 cm up to 11 cm at high luminosity and is 60 cm long, the end-caps cover radii from
6 cm to 15 cm and longitudinally the region ± 50 cm around the interaction point (see Fig. 1.9).
The layers are made of modular detector units, each one containing a thin (∼ 200−250µm)
segmented sensor plate with highly integrated read-out chips connected to them using bump
bonding technique. Since the main task of the pixel detector is to find and localize secondary de-
cay vertices, both rφ and z (r) hit coordinates are important in the barrel (end disks). Therefore
a square pixel shape has been chosen so to provide 3D point information with high resolution
22
Figure 1.9: Perspective view of the CMS pixel system in the high luminosity configuration.
for both coordinates simultaneously. The pixel size is around (150 µm)2, mostly dictated by the
minimal circuit area needed to accommodate each readout channel. The sensors are n+ pixels
on initial n type silicon substrate. Charge collection is strongly affected by the large Lorentz
drift angle of electrons (32 in a 4 T magnetic field). The barrel detectors are arranged such that
the drift angle induces significant charge sharing across neighbouring cells in the rφ plane, so
to improve resolution and cluster size conditions. Charge sharing in the barrel is also present
along the z direction for inclined tracks. Electric and magnetic field are parallel in the end-cap
disks, and most tracks are close to normal incidence. To benefit of charge sharing in this region
the detectors are rotated by 20 around their central radial axis. The induced Lorentz effect
improves charge sharing among adjacent pixels both in r and rφ directions. In this respect n+
implants are preferred because electrons drift angle is three times larger than the holes one.
The readout is of analogue type to benefit from position interpolation, where effects of
charge sharing among pixels are present, so to improve final resolution. Resulting hit resolution
is approximately 10 µm and 15 µm in the φ and z coordinates respectively, to be compared with
the 150 µm pixel dimensions. Similar resolution, between 15 µm and 20 µm are obtained in the
end-caps. In the high luminosity configuration the Pixel detector has an active surface of 0.92
m2, instrumented with about 40 · 106 channels.
23
The Pixel detector will instrument the most hostile region, from a radiation point of view,
of the whole CMS detector volume. Therefore the pixel system, although designed with radi-
ation hard criteria, must be replaced at least once during the experiment lifetime to maintain
acceptable performances.
1.5.2 The Silicon Microstrip Tracker
Microstrip silicon detectors are the natural choice to instrument the intermediate and outer re-
gions of CMS tracker system due to its high spatial and time resolution characteristic, radiation
hardness, high detection efficiency if compared with gaseous detectors with the same dimen-
sions. The excellent spatial resolution required in the CMS central tracking volume is ensured
by the fine strip pitch that can be carried out in microstrip devices and the fast charge collection
time in silicon allows single bunch crossing identification.
The CMS Silicon Strip Tracker (SST), based on microstrip silicon devices, will instrument
the intermediate and outer region of the Central Tracker. It will cover a cylindrical volume of
1.2 m radius and 6 meter length, corresponding to a pseudo-rapidity up to |η| = 2.5. Beyond
|η| ∼ 2.5 the radiation level and the track density becomes too high to operate silicon detectors
reliably. From a conceptual point of view it can be subdivided in a barrel region and a forward
region. With respect to the distance from beam pipe the detector can subdivided in an inner
tracker and an outer tracker.
The fundamental units of the tracker are silicon sensors, organized in modules of different
shapes and dimensions in order to properly match the different regions of the detector. To
provide the second coordinate a certain number of detectors are equipped with double modules.
The modules are then included in detectors, composed by one or two layers of sensors and the
front-end electronics mounted on a ceramic hybrid circuit.
The outer tracker sensors are processed on 500 µm thick silicon substrates while the inner
ones are 320 µm thick. In fact in the outer part of the tracker, to reduce the number of electronic
channels, larger detectors are needed. A wafer thickness of 500 µm is currently an industry
standard for 6” production lines and can compensate the reduction in signal to noise ratio, due
to the outer sensors increased strip length (and consequent increase in noise), with a better
charge collection. Part of the work performed for this thesis regards the study of signal to noise
24
ratio (S/N) and charge collection in 500 µm thick detectors compared to the 300 µm ones.
The inner barrel consists of four layers equipped with rectangular thin detectors tilted by an
angle of 9 along the symmetry axis parallel to the beam to compensate for the Lorentz angle.
The strips are parallel to the beam axis and provide r−φ information; this choice is dictated by
the fact that the coordinate perpendicular to the magnetic field directly measures the transverse
momentum resolution and its precision determines the resolution. In addition, the first two
layers are double sided equipped with stereo detectors whose strips are tilted by 100 mrad angle
for reconstruction of the z coordinate. The inner barrel is built with a shell type mechanics, and
is cut in two at z = 0. Each half contains 6 detectors in z and a number varying from 28 to 56
in φ. The outer barrel is composed by 6 layers, with number 1,2 and 4 double sided. It is built
with a rod type mechanics and each layer is made of several rods containing 6 detectors in z.
The forward region consists of 9 big disks, each one made of 7 rings. The 3 outermost
are equipped with thick detectors, the 4 innermost with thin detectors. To match the circular
geometry of the forward part, sensors have a wedge shape, with strips arranged radially, pointing
to the nominal beam position, to optimize φ coordinate measurement. Stereo angle detectors,
located in rings 1,2 and 5, improve track finding and vertex measurements by providing the r
coordinate read-out. In this case too, detectors which provide the stereo coordinate are tilted
by an angle of 100 mrad with respect to the detectors that give the φ coordinate. The inner
end-cap is instrumented with 3 small disks, with sensors identical to those of the outer forward,
that close the inner barrel (see Fig. 1.7). The forward support structure is based on a sector
mechanics principle, whose basic element has a petal shape.
A single hit resolution of better than 20 µm in the inner part (40 µm in the outer tracker) and
a two track resolution better than 200 µm are required from the SST to allow an efficient overall
pattern recognition. These requirements reflects in a pitch ranging from 80 µm to 180 µm in the
barrel and from 80 µm to 200 µm in the end-caps. One related design goal is to maintain single
cell occupancies at the level of a per cent that, together with the noise requirements, determines
the maximum acceptable strip length (12 and 18 cm for inner and outer region respectively).
Another strong constraint in the detector design is the long term survival after heavy irradia-
tion. Radiation hardening is strongly affected by geometry design and by silicon bulk properties
and is necessary to keep signal to noise ratio above 10:1. With this value it is possible to obtain
25
a single hit efficiency close to 100% and has been set as minimum value for silicon detector
operations after 10 years of LHC running.
Sensors are processed from 6” silicon wafers and are manufactured in nine different geome-
tries to equip the different tracker layers and rings. A small detector overlap, both in r − φ
and z coordinate, is foreseen to avoid dead regions and optimize alignment. The Silicon Strip
Tracker has a total number of roughly 10 millions electronic channels, read-out by 78256 front-
end chips and consists of approximately 170 m2 of instrumented silicon microstrip detectors
and 225 m2 silicon surface taking into account the contribution of double sided detectors [21].
Actual microstrip vertex detectors at LEP cover a surface two orders of magnitude smaller. This
difference and the fact that CMS SST must operate in hard radiation environment and with a
completely new generation of front-end electronics, points out the hard work necessary in terms
of research and development in order to be confident that required performances can be obtained
and maintained during the full experiment lifetime. The items described in this work have con-
tributed to study several aspects of SST project and to provide solid bases for the production
phase.
26
Chapter 2
Silicon microstrip detectors
Silicon electrical and physical properties make this material one of the best candidates not only
for microelectronics applications but also for particle detection purposes. The planar technology
development [22] in the last two decades of XX century has allowed to build segmented p-n
junctions and to design detectors able to spatially localize the particle position with an accuracy
down to a few microns. In this field great importance has the possibility to use artificially grown
silicon crystals, with the requested purity and dimensions.
In this chapter the silicon properties are briefly summarized and the principle of operation of
a silicon microstrip detector is described. Main electrical and geometrical parameters involved
in the detector performances (see chapter 7) are analyzed and a prediction of the signal to noise
ratio is deduced for the detectors built by the Florence group. Furthermore a review of all the
module tested is given.
2.1 Silicon properties
Silicon is a semiconductor element belonging to the fourth group of the periodic table. The
crystal structure of silicon, that consists of a regular repetition in three dimensions of a unit cell
having the form of a tetrahedron with an atom at each vertex, strongly affects its electrical and
physical properties. The Si atom is tetravalent and, in its crystalline structure, shares each of the
four valence electrons with its neighbours forming covalent bonds. At a temperature different
from zero few of these bonds are broken and some electrons are free to contribute to conduction,
thus the material is classified as semiconductor.
This property is evident in terms of energy levels. For a perfect crystal (free of impurities
27
and geometrical defects) the energy levels of outermost electrons are distributed in bands of
closely spaced energy states, separated by forbidden energy regions [23]. For a temperature
T=0 all the valence band levels are filled by electrons and the conduction band is empty. In
this situation the material is a perfect insulator. If T =0 some electrons can be excited in the
upper band, referred to as conduction band, and are free to migrate through the crystal lattice
while the lower band, called ”valence band”, is occupied by electrons that are bound to specific
sites in the crystal. The energy amplitude of the band forbidden to electrons that are included
between the energy of the highest valence band and the energy of the lowest conduction one
is known as bandgap Eg. Only electrons with energy greater than Eg are excited from the
valence band up to the conduction band leaving an empty site in the lower levels. This vacancy
is called a hole and, from a conduction point of view, it is a carrier of electricity comparable in
effectiveness with the free electron. In terms of atom bonds this mechanism can be depicted as
broken covalent bonds; the two representation are shown in Fig. 2.1.
Ec
EF
Ev
Forbidden region
Hole
Conduction Band
Intrinsic semiconductor
Valence Band
E
E=0E
e-
g
(a)
Si+4
Si+4
Si
+4
Si+4
Si+4
Si+4
Si+4
Si+4
Si+4
Valence electrons
Covalent bond
Hole
Silicon ions
Broken covalentbond
Free electron
(b)
Figure 2.1: (a) Band structure for outer shell electron energies in silicon. Electron-hole pairsproduction is shown together with the Fermi energy level EF . (b) Silicon crystal with a brokencovalent bond.
In crystalline silicon Eg = 1.12 eV at room temperature, to be compared with the 5 eV
or greater typical of insulators. The low value of the band gap makes silicon one of the best
candidates for particle detection purposes when small material volumes are necessary. In fact
28
one gets, on average, an electron-hole pair for every 3.6 eV released by a particle crossing the
medium while 30 eV are required to ionize a gas molecule in a gaseous detector [24].
Furthermore the high density of the medium reduces the range of energetic secondary elec-
trons, produced by the incoming particle, allowing good spatial resolution.
In a crystal of pure silicon, referred to as intrinsic semiconductor, the thermally produced
hole and electron densities are equal. Most of semiconductor devices base their operation on
the introduction of a small (1 part in 106-108), carefully controlled, amount of impurities into
the intrinsic material. The addition of impurities forms an extrinsic or doped semiconductor
with the results that allowed energy states in the forbidden gap are generated and the number of
thermally produced carriers is increased. Usually silicon is doped with elements from the III or
the V groups thus resulting in p and n-type materials respectively. Typical impurity densities in
n-type materials range between 1012 and 1015 cm−3.
In n-type semiconductors some of the lattice sites are occupied by a pentavalent impurity,
usually phosphorus or arsenic. Four of the five valence electrons interact with the crystal silicon
atoms through a covalent bond while the fifth electron is weakly bound and is available as
a charge carrier. In fact the energy required to detach the fifth electron from an atom is of
the order of 0.05 eV in silicon and the thermal excitation at room temperature is sufficient to
break the bond and leave a positively charged ion in the crystal lattice. The net result is that
n-type silicon has an excess of electron carriers while the number of holes decreases because
the large number of electrons present causes the rate of recombination of electron-hole pair to
increase (according to the mass-action law). Consequently pentavalent impurities are referred
to as donors and electrons are the majority carriers .
In term of energy levels the n-type doping process generates a new energy level ED in the
forbidden gap just below the lower conduction band level EC , as shown in Fig. 2.2.
The Fermi level EF is positioned between ED and EC . For phosphorus doped silicon the
difference EC −ED is about 45 meV.
In p-type semiconductors elements of the third group (usually boron, gallium or indium)
are added to the silicon and they can fill only three covalent bonds. The unsaturated bond can
easily attach an electron from silicon atoms thus generating a hole. Therefore elements from
the third group are called acceptors and the majority carriers are holes. In terms of energy
29
v
Forbidden region
E
E
F
D
Holes
-e
E
E=0
Valence band
E g
Conduction band
+ + + + + + + + + +
n-type semiconductor
cE
E
(a)
Si+4
Si+4
Si
+4
Si+4
Si+4
Si+4
Si+4
Si+4
Pentavalent
Valence electrons
Free electron
Covalent bond
+5
Silicon ions
impurity ion
(b)
Figure 2.2: (a) Band structure for outer shell electron energies in n-type silicon. Electron-hole pairs production is shown. (b) Crystal lattice with a silicon atom displaced by pentavalentimpurity atom.
levels the impurities create new states in the forbidden gap just above the higher valence band
level. At room temperature the unsaturated bonds are filled by thermally excited electrons thus
generating fixed negative charges in the silicon lattice.
In both the p and n-type materials there are carriers (called minority carriers) coming from
thermal excitation of silicon atoms. They play an important role in the behaviour of semicon-
ductor devices based on p-n junction (section 2.2).
A silicon electrical parameter that plays an important role in the design and characterization
of silicon microstrip detectors is the material resistivity ρ. The resistivity is related to the carrier
density and mobility µ by the following equation:
ρ =1
q(µnn+ µhp)(2.1)
where q is the electron charge and µn and µh are mobilities of electrons and holes respectively,
whose density is n and p. Mobility is defined as:
µ =v
E(2.2)
with E being the electric field and v the drift velocity, and has, at room temperature, the value of
1350 cm2/Vs for electrons and 480 cm2/Vs for holes. For intrinsic silicon at room temperature
30
ρ 235 KΩ·cm; for doped materials this value is much lower because of a higher carrier
density.
2.2 The p-n junction
The p-n junction is the basic building block on which the operation of all semiconductor devices,
and in particular the silicon microstrip detectors, is based. A p-n junction is formed when a
single crystal of semiconductor is doped with acceptors on one side and donors on the other.
In order to easily depict the processes involved in such a device we will consider a planar step
junction built by ideally connecting two semiconductor crystals of p and n type as shown in
Fig. 2.3.
- -----
- + + + + + +
+ + + + + +
- -----
- -----
+ + + + + +
+ + + +
Donor ion
+ +
- +
+ + + + +
++++++
+-----
- - - - - -
------
p-type p n-type n
depleted region
- -+ + + +
+ + + +
- -
- -- -
p-n junction
+ Hole
Acceptor ion
- Electron
(a)
F
E v
E c
φe
E
n-type regionzona p zona n
p-type region n-type region
(b)
Figure 2.3: Representation of the planar p-n step junction (a) and energy level scheme (b).
Initially a concentration gradient exists across the junction and holes diffuse towards the n-
type region while electrons start to migrate in the opposite direction. Consequently the positive
holes which neutralized the acceptor ions near the junction in the p-type silicon have disap-
peared as a result of the combination with electrons which have diffused across the junction.
Similarly, electrons in the n-type silicon combine with holes which have crossed the junction
from the p material. The migration of free carriers creates at the same time a potential barrier φ
that contrasts the diffusion process until a stationary state is reached. In steady state conditions
31
the region in the neighbourhood of the junction is depleted of mobile charges and therefore it
is called depletion region or space-charge region. During this process the Fermi energy levels
(EF ) in the two regions overlap and the overall result is a modification of the conduction and
valence band (see Fig. 2.3).
The voltage corresponding to the potential barrier, known as built-in voltageVbi, is of the
order of a few hundred millivolts at room temperature and for typical doping densities ND ∼1015 cm−3 (donor density) and NA ∼ 1017 cm−3 (acceptor density).
The electric field distribution E(x) along a direction orthogonal to the junction can be de-
rived by solving the Poisson equation with the condition E(xp) = E(xn) = 0, where xp and xn
are the limits of the depleted region in the p and n material respectively. One further constraint
is imposed by the neutrality of the depleted region, where the condition NAxp = NDxn must
be satisfied. The electric field behaviour turns out to be:
E(x) =
−eNA
ε(xp + x) −xp ≤ x ≤ 0
−eND
ε(xn − x) 0 ≤ x ≤ xn
(2.3)
where e is the electron charge and ε is the silicon dielectric constant. The potential in the
depleted region, with the condition V (xp) = 0, is:
V (x) =
eNA
2ε(xp + x)2 −xp ≤ x ≤ 0
−eND
2ε(xn − x)2 + φ 0 < x ≤ xn
φ = e2ε(NAx
2p +NDx2
n)
(2.4)
The width W of the space charge region is given by:
W =
√2εVbi(NA +ND)
eNAND(2.5)
For particle detector devices based on p-n junction, an external voltage Vbias of the same sign
as the built-in one is usually applied to the junction in order to increase the depletion region.
Under this circumstance the junction is said to be reverse biased and in the formula 2.5 Vbi must
be replaced by:
Vtot = Vbi + Vbias (2.6)
32
In case the density of one type of doping element is much higher than the other, the depletion
zone extends mainly on the side with the lower doping concentration. This is an usual situation
in the design of silicon detectors which base their operation principle on a p-n junction (see
section 2.3). In this condition the expression for the depleted zone width can be approximated
by:
W =
√2εVtot
eNx(2.7)
where Nx is the lower doping density. The same quantity can be expressed in terms of the
material resistivity ρ (see eq. 2.1) and of the majority carriers mobility µ as:
W =√2ερµVtot (2.8)
It is evident from eq. 2.7 and eq. 2.8 that the depleted zone increases with the applied bias
voltage until the free carriers are removed from the whole silicon volume. The voltage value
that makes the detector completely depleted is known as full depletion voltage Vd:
Vdepl =eNxW
2max
2ε− Vbi (2.9)
From eq. 2.8 we can see that, for equal thickness, a lower full depletion voltage is obtained
in a high resistivity material. This is an important parameter in the CMS detector design since
silicon devices have to be fully depleted during the entire tracker lifetime, especially when
radiation effects deeply modify the silicon effective resistivity.
For example, for a 300 µm thick p-n junction, with a ND doping density giving a resistivity
of 4 KΩ·cm, the full depletion voltage is about 80 V.
The same equation allows to compute the full depletion voltage in terms of the material
resistivity, whose value is usually provided by the silicon device manufacturer. Vice versa,
from the Vd measurement the doping density Nx and the material resistivity (see eq. 2.1) can be
deduced.
It is worth noting that when a junction side contains both donors and acceptors with con-
centrations of the same order of magnitude, Nx represents the effective doping density Neff
defined as the difference between the numbers of donors and acceptors:
Neff = |ND −NA| (2.10)
33
This quantity is one of the main parameters to be considered when dealing with irradiated de-
tectors as will be explained in section 3.1.
A fundamental quantity in a reverse biased junction, that is worth examining since it strongly
influences the silicon detector performances, is the leakage current . The depletion region is free
of majority carriers but under equilibrium conditions electron-hole pairs are thermally generated
everywhere within the crystal volume. The electric field generated by the bias voltage makes the
electrons and holes drift towards their electrodes giving rise to the leakage, or reverse, current.
Neglecting the diffusion current, that comes from charge generated in the neutral silicon
and diffusing to the space charge region, the generation current is the only contribution to the
leakage current. In this case the leakage current has a density given by:
Jgen =1
2qni
τ0W (2.11)
where ni is the carrier density in intrinsic silicon, τ0 is the effective lifetime of minority carriers
within the depletion zone and W the depletion zone width. The lifetime τ0 is inversely pro-
portional to the density of impurities or traps that are involved in generation and recombination
processes. Therefore special care must be taken in order to keep the silicon crystal “clean” in all
applications where the leakage current effect must be reduced, as in silicon detector production.
Moreover the lifetime is strongly affected by the presence of deep impurities, whose energy
levels lay in the central region of the forbidden gap. This effect explains the increase in leakage
current measured in the irradiated detectors tested in the framework of this work (section 3.1).
This current component is proportional to the depth of the depletion zone W and conse-
quently to√Vbias. It is worth noting that the generation current will stop to increase after the
bulk is fully depleted. Furthermore the dependence from ni makes necessary to keep the tem-
perature constant for a stable operation of the detector based on a reverse biased p-n junction.
Since there is a voltage dependent charge associated with the depletion zone the p-n junc-
tion shows a capacitor behaviour. The junction capacitance per unit area (also known as bulk
capacitance) is defined as [23]:
C =dQ
dVB=
dQ
dW
dW
dVB(2.12)
34
where dW is the widening of the depletion region caused by an increase of the barrier voltage
dVB and dQ is the corresponding charge variation on both side of the junction. Modeling the
junction as a parallel plate capacitor and in the hypothesis of a junction side much more doped
than the other (Nx Ny), from Eq. 2.7 the junction capacitance per unit area is obtained:
C =
√qεNx
2VV ≤ Vdepl
εddepl
V > Vdepl
(2.13)
where Vdepl is the full depletion voltage and ddepl is the maximum allowed depth for the
depletion layer.
The junction capacitance decreases with increasing bias voltage, reaching a constant value
when the depletion layer reaches the back of the crystal. In this case a further increase of the
bias voltage would not change the charge on the junction side.
The depletion voltage can be measured from the voltage dependence of the junction capaci-
tance and, as we will see in next sections, is an useful estimation of the voltage to be applied to
the detector in operating conditions.
2.3 Principle of operation of silicon detectors
In pure silicon the intrinsic carrier density is about 1.45 · 1010 cm−3 at room temperature. The
total number of free carriers in a 300 µm thick silicon material, with a volume comparable to
the usual detectors, is of the order of 108, so four orders of magnitude higher than the expected
signal (see section 2.3.1).
The way to collect all the charge released by a particle crossing the silicon is to deplete
the detector volume from free carriers through a reverse biased p-n junction. In this situation
the depleted region draws only a little reverse current under the applied voltage, but any charge
deposited within its volume drift towards the junction and can be collected. The charge released
in the non-depleted zone quickly recombines with the free carriers and is lost. This implies that
silicon detectors should operate with an applied voltage sufficient to fully deplete the entire
crystal volume.
In Fig. 2.4 the principle of operation of a silicon detector is summarized. In this case the
p-n junction is carried out with a n-type silicon bulk, acting as detector volume, and a heavily
35
doped, shallow, p+ implantation.
traj
ecto
ry
bias
bias
part
icle
p+ implant
metal
-
-
--
-+
++
++
- h
Bulk n
+-e
V
R
+
Figure 2.4: Principle of operation of a silicon detector.
The detector is depleted by applying a reverse bias voltage Vbias, by means of a bias resistor,
to a pair of electric contacts on the two sides of the junction. A charged particle that crosses the
detector generates along its path electron-hole pairs that drift towards the electrodes following
the electric field present in the depleted region. The charge is collected by the read-out electron-
ics and amplified. The resulting signal is proportional to the number of generated electron-hole
pairs and therefore to the energy loss of the particle.
2.3.1 Energy loss of high energy charged particles in silicon
High energy charged particles traversing crystalline silicon lose energy mainly by ionization.
The energy loss distribution for highly relativistic charged particles in thin absorbers is de-
scribed by the Landau theory [25], in the hypothesis of free electrons. In Fig. 2.5 the energy loss
distribution measured with MIPs (Minimum Ionizing Particles) crossing a thin silicon detector
(300 µm thick bulk) is shown. The shape of the distribution follows the Landau curve behaviour
but it is wider than expected. The discrepancy is overcome taking into account the effects of
the bindings of atomic electrons. A phenomenological calculation leads to derive the simplified
distribution, commonly used to fit experimental data, known as Moyal distribution [26]:
f(λ) =P1
2πe−
12(λ+e−λ) (2.14)
36
P1 7987.P2 234.5P3 32.01
Signal (ADC counts)
Num
ber
of e
vent
s/bi
n
0
250
500
750
1000
1250
1500
1750
2000
2250
0 100 200 300 400 500 600 700 800
Figure 2.5: MIP experimental energy loss distribution in a 300 µm silicon detector. The resultof the fit with the curve described in eq. 2.14 has been superimposed.
with:
λ =E − P2
P3(2.15)
where P1 is a normalization factor, P3 is related to the distribution width, E is the released energy
and P2 is the most probable value of the energy distribution. For thin samples, as the wafers
commonly used for detectors, the average energy loss is significantly higher (about 50 %) than
the most probable one. This asymmetry is due to processes in which high energy electrons (δ
rays) are emitted in the material. These electrons themselves are able to generate up to several
times the mean energy loss. It is worth noting that ejected electrons with high energy have
a not negligible range inside silicon thus spoiling the spatial resolution performances of the
detector [27].
The average energy loss is about 390 eV/µm [15] for a MIP in silicon and is independent on
the thickness of the crossed material. Taking into account that the mean energy needed to create
an electron-hole pair is 3.6 eV, this value gives about 108 carrier pair for micron of traversed
silicon. The most probable value for the energy loss per unit length is about 280 eV/µm but
scales within 10 % for detector thickness ranging from 20 µm to 300 µm.
37
2.4 Silicon microstrip detectors
In silicon detectors there is no multiplication of primary charge, as happens in gaseous devices,
and the collected signal is, in principle, a linear function of the detector thickness. In high
energy physics experiments silicon detectors are usually installed very close to the interaction
point where a large amount of material would spoil the track parameters measurement and the
electromagnetic calorimeter energy resolution. Thus the detector thickness should be as low
as possible, down to a practical limit set by the requirements on signal-to-noise ratio and, also
important, the related services (electronics, mechanics, cooling) should be carefully optimized.
The best compromise adopted for the CMS Silicon Microstrip detectors placed in the inner
region is a thickness of 320 µm for which one obtains on average 3.4 · 104 electron-hole pairs, a
signal easy detectable with low noise electronics. In the outer part of the tracker, to reduce the
number of electronic channels, larger detectors are needed. In this case the noise contribution
coming from the capacitive load to the front-end electronics is higher, consequently thicker
detectors (500 µm) are employed to keep the S/N in a reasonable range.
For tracking purposes, in order to obtain a position sensitive device, one side of the junction
must be divided into smaller elements. In the silicon microstrip detectors described in this thesis
the geometry adopted is the segmentation of the p+ junction side in an array of narrow strips.
Each strip has its own bias circuit and, together with the n-type bulk, behaves as a reverse biased
diode. Under the influence of the electric field the holes released by a ionizing particle crossing
the detector drift towards the closer strips. Coupling a read-out channel to each strip, a charge
measurements provides information about the coordinate of the particle position while crossing
the detector.
2.4.1 Single sided device
Track reconstruction requires detectors able to provide three dimensional information, at least
for some fraction of the Tracker layers. Referring to the cylindrical reference system of the CMS
Tracker (see section 1.5.2) the radial coordinate (axial in the end-caps) is directly obtained by
the mechanical position of the silicon layers, while the two remaining must be measured with
an appropriate detector design choice. The CMS collaboration has decided to use pairs of
single-sided detectors, coupled back to back, with the strips slightly tilted in order to obtain two
38
coordinates information. This solution has been preferred to the double sided technology be-
cause, after an accurate R&D program [28], has showed to be the best choice from an industrial
production point of view, while maintaining the material budget under the requested limits.
The fundamental unit of a detector designed with a single segmented side is referred to as
single sided crystal. It is built starting from a n-type substrate (bulk). An array of narrow p+
strips, which provides one dimensional information, is implanted on one side (junction side).
The p+ on n junction based detectors offer a great advantage in terms of cost and industrial
production capacity since they are the simplest device that can be manufactured using the usual
semiconductor electronics production lines. However this choice has a drawback because, after
type inversion of the bulk induced by radiation, the detector must be over-depleted in order to
maintain satisfactory performance (see section 3.1).
The distance between two adjacent strips is called strip pitch and their width is referred to as
implant width. On the other side (ohmic side or backplane) the silicon bulk has an n+ implant
layer to ensure a good ohmic contact with the metal electrode and prevent minorities carriers
injection in the bulk [3]. In Fig. 2.6 the principle of operation of a silicon microstrip detector is
sketched, stressing the design geometry.
One of the most important design choice is the way the bias voltage is provided to each
strip. The solution adopted by CMS collaboration is mainly motivated by the requirements im-
posed by the hostile radiation environment the silicon tracker will have to operate in. Acquired
experience has shown that the best choice is to use sensor integrated resistors connecting each
strip with a common bias ring surrounding the detector active area. The bias voltage is applied
from the supply lines through microbondings to the bias ring and to the backplane and is dis-
tributed to the strips. The bias resistor value is a compromise between two contrasting needs;
it must be sufficiently high in order to keep low the thermal noise on the front-end electronics
(section 2.6.2) but, in the meantime, it has to maintain low the voltage drop across the resistance
due to the leakage current, which increases when dealing with irradiated detectors (section 3.1).
The implant strips have a voltage close to the bias ring (ground) since for typical values
(512 strips detectors, 1µA mean leakage current, 2 MΩ bias resistors), the voltage drop across
the resistor is only a few mV, negligible with respect to the applied bias during operation.
After irradiation this value increases, up to three orders of magnitude, still remaining into an
39
Implant,p -type+
SiO +2
Si N3 4
t
S
EBulk,n-type
Pre-amplifiers/Shapers
Strip pitch, P
( 30
0um
)
electrons
holes
Particle
Backplane, n - type silicon++ Bias Voltage
Metallization
Implant width, W
Principle of operation
Figure 2.6: Principle of operation of a silicon microstrip detector. The bias section is not shown.
acceptable range.
A further p+ ring (guard ring) surrounds the bias ring in order to separate the space charge
region from the heavily damaged region along the cutting edge. In this way the active area of
the detector is isolated from potentially dangerous injection of charges from the cut region. The
CMS collaboration has allowed the detector manufacturer to use a multi-guard design provided
that the breakdown requirements are maintained. This is the case of the 500 µm thick detector
tested in the framework of this thesis (see section 7.3).
For what concerns the coupling capacitors, needed to insulate the read-out electronics from
the leakage current, the CMS collaboration has decided to integrate them directly on the detec-
tor. The easiest solution is to separate each implant strip from the read-out metal electrode by
a thin insulating layer, as shown in Fig. 2.6. This integrated capacitor is usually built by means
of a double layer of silicon oxides (SiO2 and Si3N4) to reduce the risk of pinholes.
The metal strips are then connected to the read-out electronics by means of ultrasonic mi-
crobondings.
40
2.5 The Florence detector prototypes
The detectors described in this thesis have been completely designed, characterized and tested
by the Florence CMS group. They are part of a more extensive work performed from sev-
eral groups working for the Tracker in the framework of the so called “Milestone 1999” and
concluded with beam test measurements during summer 2000.
The sensors have been manufactured by CSEM, Switzerland, starting from a n-type sub-
strate 4” wafers. Since the final version of these prototypes will be installed in the end cap
region of the tracker, they have been designed adopting a wedge geometry layout. The detector
thickness is 300 µm and corresponds to a MIP most probable signal of 24000 electron-hole
pairs (see Section 2.3.1).
In order to compare the performances of different detectors, with particular emphasis placed
on the radiation tolerance studies, the same design has been carried out on two different sub-
strate types, a low resistivity one with <100> crystal lattice orientation, and a high resistivity
one with <111> orientation. The main differences between the two type of substrates, with
respect to the silicon detectors requirement, is that the <111> detectors have a greater density
of silicon-oxide interface charges and defects and this affects the quality of the oxide too. A
deep characterization is therefore needed in order to understand which kind of orientation is
best suited for the CMS experiment.
A parallel analysis has been carried out, in the framework of this thesis, with respect to the
substrate resistivity, that strongly affects the depletion voltage before and after irradiation. In
the following I will refer to high resistivity (HR) and low resistivity (LR) for substrates with
initial measured resistivity of about 6 KΩ·cm and about 1 KΩ·cm respectively.
The fundamental unit of the Silicon Microstrip Tracker is a module made by one or two
sensors; in this second case the strips are daisy chained together to obtain a larger detecting
surface while keeping the total number of electronic channels into an affordable range. The
wedge modules assembled in the framework of this thesis are composed of two sensors with an
overall strip length of about 12.7 cm.
Since the two crystals have different dimensions, due to the wedge geometry, in the follow-
ing they will be referred to as F4 for the crystal closer to the read-out electronic and F5 for
the other one. Each crystal has 512 p+ strips implanted with a constant angular pitch of 0.23
41
mrad. The strips point to the nominal beam position. The main geometrical parameters are
summarized in Table 2.1, where N is the number of strips, L their average length, P the strip
pitch, W the constant strip width (∼ 25µm), Wm the metal implant width on the junction side,
A the detector active area.
Detector N L P W/P Wm A(cm) (µm) (µm) (cm2)
F4 512 6.59 108.5 - 123.7 0.2 - 0.23 33 39.26
F5 512 5.64 124.3 - 137.4 0.18 - 0.20 33 37.83
Table 2.1: Main detector geometrical parameters.
The layout of the two detector crystals daisy chained together is sketched in Fig. 2.7; the
resulting single sided module is called an rφ module and, if coupled to a sensor with strips
slightly tilted with respect to the radial direction, provides a double coordinate information.
pitch 137.4 um
F5
F4
512 strips
junction side
junction side
73.12 mm
58.87 mm
68.45 mm
57.97 mm
pitch 108.5 um
pitch 124.3 um
Figure 2.7: Module layout with two crystals daisy chained together.
42
A module, connected to the front-end electronics and glued on a carbon fiber support is
shown in Fig. 2.8. It is clearly visible the hybrid that houses four APV6 chips and the pitch
adapter connecting the shorter side of a F4 crystal to the read-out chips. The bias voltage
connections are placed on the right bottom corner of the module. A kapton cable (on the
bottom) provides all the electrical lines needed to operate the detector and to acquire the signals.
Figure 2.8: The complete detector module.
The implant strips are coupled to the readout chip through integrated capacitors as described
in section 2.4.1. In order to reduce the risk of metal-implantation short circuits (pinholes) the
detector has been designed with a multi-layer structure. The dielectric is deposited as a double
layer made of SiO2 and Si3N4 with the goal of decoupling single layer intrinsic defects.
Each metal strip has two aluminium pads on both side of the sensor in order to allow the
micro-bonding connection. Due to the role they play they are called AC pads. The spare pads
are reserved in case of failure during the first micro-bonding procedure.
43
The elements connecting the bias ring with the p+ implant strips are polysilicon resistors
with a winding structure in order to obtain larger resistance values. The polysilicon material
has been chosen by the CMS collaboration for its good radiation hardness.
All the structures described above are depicted in Fig. 2.9, which shows the original design
of the detector junction side corner.
p+ strips
Guard Ring
Bias Ring
AC Pads
DC Pads
Bias Resitors
Figure 2.9: Design of the detector junction side corner.
It is worth noting that bias resistors are distributed every second strip on both ends of the
sensor in order to satisfy the stringent space requirements. Furthermore the metal pads contact-
ing the node between the bias resistor and the implant strip are visible. These structure, called
DC pads, can be contacted at the surface of the detector and are used for testing purpose and
characterization described in next section.
2.5.1 Electrical characteristics
Since the detector performances depend on the electrical impedances of the sensors it is worth
defining the main capacitive and resistive components that characterize such devices. In Fig. 2.10
44
the cross section of a microstrip device with the main capacitances present between the detector
elements is shown.
Bulk n
insulator
CCCAC AC AC
Cb bC bC
Cimp C Cimp
Cmet Cmet Cmet
imp
CAC
Cb
metal (Al)
p+ implant
n+ implant
Figure 2.10: Cross section of a microstrip device and main capacitances involved in its charac-terization.
The definitions of the components are the following:
• The coupling capacitance CAC is the capacitance between each implant strip and its re-
lated aluminium readout strip. Its value depends on the strip width and on the dielectric
material and thickness. The coupling capacitance is responsible for the induction of the
electronic signal on the front-end chip input when charge is collected on the implant strip.
• The back plane capacitance Cb is the capacitance between the implant strip and the back
plane. It behaves as the typical capacitor associated to a reverse biased p-n junction, as
mentioned in section 2.2. Cbulk is the total bulk capacitance measured between the bias
ring and the back plane.
• The interstrip capacitances are the ones between two consecutive implant or metals strips,
Cimp and Cmet respectively. They depend mainly on the strip pitch and width. Further-
more each strip shows capacitive couplings, although of lower intensity, with the next
order neighbour strips.
45
All these impedances have been measured directly on detectors with a probe station in a clean
room. Furthermore a complete characterization of bias resistor, metal strip resistance and leak-
age current has been carried out in order to understand the device behaviour and quantify the
contribution of each component to the signal to noise ratio (Section 2.6). Thus we are able to
know the best operational conditions for the final module and to predict its performance.
In the following the clean room characterization results are summarized, with particular em-
phasis on the measurements interesting the quantities that affect the final module performances.
Leakage current
The leakage current value as a function of the applied voltage depends mainly on the silicon
bulk impurity and defect concentration . Such measurement is a good indicator of the fabri-
cation process quality and allows to select detectors with lower leakage current thus reducing
shot noise and the heating of the device that can eventually result in thermal runaway. The
measurement setup is shown in Fig. 2.11 together with a typical total leakage current vs. bias
voltage curve. The current shows a behaviour proportional to√Vbias, as expected from Eq. 2.11
p
Aluminium metal strip
Silicon Silicon
p
Bias resistor
Bias ring
Guard ring
Implant p+Implant p
n
Bias ring
Bias resistor
Oxide
0.3
mm
Aluminium back-plane~ 60 mm
Keithley 237
H L
+n
+Implant
Voltage generator
AC padDC Pad
(a)
Vbias(V)
I(A
)
T = 21oC
10-8
10-7
10-6
10-5
0 200 400 600 800 1000
(b)
Figure 2.11: (a) Scheme of the experimental setup used for the leakage current measurement.(b) Leakage current as a function of the applied voltage for a F4 detector.
and 2.8, before reaching the full depletion voltage. This confirms that the generation current is
the main component of the leakage current. After a knee in the current-voltage curve, corre-
46
sponding to the full depletion voltage, the current increases slowly due to the overall effect of
the edge current arising from lateral bounds of the depleted zone, the interface current due to
electron hole-pair generation at the silicon-oxide interface and the injection of majority carriers
from the metal backplane. An abrupt increase of the current takes place in correspondence of
a voltage known as breakdown point which ultimately limits the detector bias voltage. The I-V
measurement allows to select the detectors with a breakdown voltage far beyond the expected
operational bias voltage. This parameter is 500 V for the devices to be used in CMS, even after
the heavy irradiation accumulated in 10 years of LHC operation. Typical values of the strip
leakage current is of the order of a few nA for non irradiated devices.
In some detectors the leakage current shows one or more discontinuities after the full deple-
tion voltage plateau , as shown in Fig. 2.12(a). This current increase is localized to a few strips,
as emerged by scanning the strip currents one at a time with a probe card, and is mainly related
to localized defects in the n+ layer between the metal backplane and the bulk. Such a mea-
Vbias(V)
I(A
)
T = 21oC
10-8
10-7
10-6
10-5
10-4
0 200 400 600 800 1000
(a)
strip
I stri
p(A
)
T = 23oC
Vbias = 350 V
10-12
10-11
10-10
10-9
10-8
10-7
10-6
200 400
(b)
Figure 2.12: (a) Leakage current for a detector with a faulty strip. (b) Single strip leakagecurrents for a F5 detector. The faulty strip is clearly visible.
surement allows to disconnect the faulty strips from the front-end electronics and eventually to
reject a detector if their number exceeds a certain threshold. This last operation is necessary to
47
fulfil the stringent requirements of the CMS experiment that imposes a maximum of 1% broken
strips on a single crystal.
Bulk capacitance and full depletion voltage
The bulk capacitance measurement is fundamental for the detector characterization because
from its dependence on the bias voltage it is possible to measure the full depletion voltage and,
furthermore, it is strictly related to the single strip backplane capacitance Cb. This last quantity
is obtained, in the model of parallel plate capacitor, simply dividing the Cbulk value by the
total number of strips. All the values reported have been measured with a LCR meter at low
frequency, where the impedance shows a purely capacitive behaviour.
In Fig. 2.13 the bulk capacitance versus bias voltage is shown and the full depletion voltage
is extracted, following Eq. 2.13 (b), from the plot 1/C2bulk versus bias voltage as the intersection
of the plateau value and a linear fit at lower voltages.
Vbias(V)
Cbu
lk(p
F)
f = 1 KHzCbulk
0
2000
4000
6000
8000
10000
12000
14000
0 20 40 60 80 100
(a)
Vbias(V)
1/C
2 bulk
(pF
-2)
f = 1 KHz
Vdepl = 53 V
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
x 10-6
0 20 40 60 80 100
(b)
Figure 2.13: (a) Bulk capacitance as a function of the bias voltage for a F5 high resistivitydetector. (b) Extrapolation of the full depletion voltage for the same detector. All measurementare taken at 1 KHz.
The mean value obtained for the full depletion voltage of the high and low resistivity detec-
tors are (54± 10)V and (240± 30)V respectively.
48
From the depletion voltage measurements and writing Eq. 2.8 in the form:
ρ =W 2
s
2εVsµ(2.16)
the resistivity turns out to be ρ = (5.8±1.1)KΩ·cm for HR detectors and ρ = (1.1±0.2)KΩ·cm
for LR ones, where Ws is the crystal thickness. Furthermore from Eq. 2.7 the effective donor
density can be measured and results (7.9±1.5)·1011 cm−3 for the HR substrate and (35±4)·1011
cm−3 for the LR one.
Strip and bias resistances
All the metal electrodes and polysilicon resistance have been systematically measured on the
detectors using a probe card. The mean values are presented in Table 2.2.
Coupling and implant capacitances
The coupling capacitance value affects the charge collection and noise of the detector, as will
be explained in section 2.6. The measurement of this quantity, fundamental to properly charac-
terize the detector, is reported in Table 2.2. We can observe that the CAC value is higher for the
<100> detectors since for such crystal orientation the dielectric layer is 20% thinner.
Furthermore broken capacitors have been revealed by contacting all the metal strips with a
probe card and by applying a direct polarization (1 V) between the metal strips and the back-
plane, as shown in Fig. 2.14 (a).
When the metal strip is in direct contact with the p+ strip due to a pinhole in the double layer
oxide, the measured current is several orders of magnitude higher than in case of good coupling
capacitor and the defect is easily detectable (Fig. 2.14 (b)). The strips with pinholes are less
than 1% for all the detectors tested. These strips are not bonded to the front-end electronics
when a module is assembled.
The measurement of the implant interstrip capacitance (Cimp) is necessary since it is the
main contribution to the total capacitance that the pre-amplifier senses at its input and that must
be minimized in order to reduce the noise (see section 2.6). From Table 2.2 we observe that
both the metal and implant capacitances depend on the strips pitch and width, as expected from
geometrical consideration, but not on the substrate resistivity and crystal orientation.
49
L
PinholeI
Switch array
Voltage generator
Probe Card
Keithley 7002
HP 4142B-MPSMU
H
(a)
strip
I(A
)
Direct bias1 V
10-13
10-12
10-11
10-10
10-9
10-8
10-7
10-6
10-5
10-4
100 200 300 400 500
(b)
Figure 2.14: (a) Experimental setup to detect pinholes in the double oxide layer. (b) Evidenceof oxide defects in a F5 HR damaged detector.
Summary of the main electrical parameters
In Table 2.2 the main electrical quantities involved in the detector characterization and noise
evaluation are summarized. Their values, when possible, are presented as value per strip and
per unit length.
Detector Type Rbias Rmet Cb CAC Cimp Cmet
(MΩ) (Ω/cm) (pF/cm) (pF/cm) (pF/cm) (pF/cm)
F4 <111> 1.76 6.7 0.38 33.3 0.227 0.080
F5 <111> 1.76 6.7 0.41 32.8 0.206 0.074
F4 <100> 2.17 6.5 0.38 38.6 0.223 0.080
F5 <100> 2.17 6.5 0.43 39.0 0.207 0.074
Table 2.2: Main electrical parameters for non irradiated devices measured with the detectorsoverdepleted.
It is worth noting that the Cb value is greater for the F5 geometry, according to the increase
in the strip pitch with respect to the F4 detectors, while it is independent on the substrate type.
50
2.6 Signal and Noise evaluation
Performances in terms of signal to noise ratio are strongly affected by detector electrical param-
eters, as well as by the readout electronics. For tracking purpose the S/N is the main responsible
for an optimal detection of the passage of a particle through the sensor. An evaluation of the
charge collected from the detector and of the noise affecting the measurement is fundamental
during the design phase and as cross check with experimental data to understand if the device
is working properly and where possible improvements can be made.
The electrical parameters of the detector are fundamental with respect to the noise consid-
eration too. In fact the read-out electronics noise is the main component to the total noise and it
heavily depends on the impedances seen by the pre-amplifier.
2.6.1 Charge collection
The charge released by a particle crossing the detector is not completely collected by the read-
out electronics. In fact, in order to evaluate the signal read-out by the front-end chip, it is
necessary to take into account the complex network of sensor capacitances that describes the
silicon detector.
In Fig. 2.15 a circuit sketch of a silicon sensor with its pre-amplifier stages is shown; the
black dots are the metal read-out strips, kept at a virtual ground by the front-end electronics,
while the open points are the implant strips.
The charge released by a particle crossing the detector near the strip called ”B” has been
schematized as a current source between the node ”B” (hit strip) and ground. Since the points
”A” (neighbour metal electrodes) are connected to the pre-amplifier inputs and can be consid-
ered fixed to ground, the charge is shared between the coupling capacitance CAC and the sum
of all the other capacitances (Cstrip)that the implant strip ”B” sees to ground. The fraction of
charge collected by the coupling capacitor is therefore given by the ratio:
CAC
CAC + Cstrip(2.17)
where Cstrip, the strip capacitance, once solved the circuit in Fig. 2.15 turns out to be:
Cstrip = Cb + 2Cimp(CAC + Cb)
CAC + Cimp + Cb(2.18)
51
ground ground
ground
backplane
B
AAmetC
AC
virtual
virtualvirtual
C imp
C
C
b C
C
b
C
C
C
b
met
impC
AC AC
Figure 2.15: Schematic representation of the impedances involved in charge collection in asilicon detector. Black dots are the metal read-out strips, open circles the implant strips (Cross-section view).
It is evident that in order to reach the best charge collection efficiency the following condition
must be satisfied:
CAC Cstrip (2.19)
From the measured values, summarized in Table 2.2, it appears that during the operational
conditions (detector fully depleted) the coupling capacitance is large with respect to the implant
one and the following approximation can be made with an error less than 1%:
Cstrip Cb + 2Cimp (2.20)
The second neighbour metal strip contribution C2imp is of the order of 10%Cimp. In order to
take into account the effects of the second neighbour metal strips on the Cstrip it is sufficient to
substitute Cimp with 1.1 · Cimp [29]. From Eq. 2.20 it is evident that the coupling capacitance
must be larger than the back plane and the implant capacitance in order to optimize the charge
measurement by the front-end amplifier.
2.6.2 Noise evaluation
The noise that affects the performance of a silicon detector is usually expressed in terms of
equivalent noise charge (ENC), that is the charge to be injected in the input of an ideal noiseless
52
amplifier so to have an output equal to the r.m.s. fluctuation value of the real amplifier. In terms
of signal to noise ratio is the charge that corresponds to S/N= 1.
The total noise present at the front-end chip output is the resulting effect of several contribu-
tion, originating from the electrical components of the detector and from the read-out electronics
noise. In order to quantify the noise contributions, a single p+ strip can be schematized as the
series of a diode, with a capacitance Cstrip in parallel with it, and the series resistance Rs seen
by the pre-amplifier. (see Fig. 2.16).
R biasIf
APV6
C R sBVbias
Cstrip
AC
(a)
i n
enC tot
(b)
Figure 2.16: (a) Electrical scheme representing a single strip connected to a front-end chipchannel used to evaluate the noise contributions. (b) Detector noise sources.
The series resistance Rs is the sum of two impedances: the pitch adapter resistance (Rpa ∼20Ω) between the detector metal strip and the electronic channel, and the metal strip resistance,
being negligible the bonding wire contribution. Since the metal strip resistance Rmet is dis-
tributed along the strip, a transmission line effect occurs and the effective value for the series
resistance turns out to be [30]:
Rs =Rmet
3+Rpa (2.21)
The diode is reverse biased through a bias voltage Vbias and a bias resistor Rbias and is
coupled to the front-end chip pre-amplifier by means of the coupling capacitance CAC . The
leakage current is taken into account by a current generator in parallel with the diode. According
to this model the physical noise sources can be identified in the strip leakage current (shot noise)
and in the bias resistor and series resistor (thermal noise), to be added to the contribution of the
read-out electronics.
53
Summarizing, the noise sources mentioned above can be modelled as a current generator in,
in parallel with the diode representing the strip, related to the shot noise and to the bias resistor
thermal noise, and a voltage generator en, related to the series resistance, in series between the
amplifier and the detector, as shown in Fig. 2.16(b). Ctot is the total capacitance as seen from
each front-end chip channel input.
The noise expression depends on the effects of the pre-amplifier and shaping sections of the
read-out chip. In the case of the CMS front-end chip, the APV6 (see chapter 4), we can consider
a CR-RC semigaussian formation with a shaping time τ of 50 ns. If we express the contribution
to the series and parallel noise in terms of ENC (electrons number), we have [30]:
(ENC)2series =
e2
8q2C2tot
1τe2n
e2n = 4KTRs
(2.22)
and (ENC)2parallel =
e2
8q2 τi2n
i2n = 2qIf + 4KT/Rbias
(2.23)
where q is the electron charge, e is the Euler constant and τ is the shaping time. The expression
for the total strip capacitance seen at the readout input with respect to ground is obtained from
Fig. 2.15 and is:
Ctot = 2Cmet +CACCstrip
CAC + Cstrip(2.24)
The capacitance values in Eq. 2.24 are referred to the full module strip, which is composed
by the contribution of two crystals. Since the two consecutive strips are connected through the
AC pads by a microbonding, and in the same way are connected the bias rings and the ohmic
side metal implants, each kind of capacitance is in parallel with the corresponding one on the
adjacent crystal and their effect on Ctot must be summed.
The behaviour of the total capacitance, calculated from the laboratory measurement, is
shown in Fig. 2.17 as a function of the bias voltage for two crystal orientations. We can observe,
as expected, that Ctot is approximately constant for a bias voltage greater than the depletion volt-
age. In fact the total capacitance depends on the Cstrip capacitance which, in turn, is dominated
54
Vbias(V)
Cto
t(pF
)
Module <111>
non irradiated
5
10
15
20
25
30
35
40
45
25 50 75 100 125 150 175 200
(a)
Vbias(V)
Cto
t(pF
)
Module <100>
non irradiated
5
10
15
20
25
30
35
40
45
100 200 300 400 500 600 700 800
(b)
Figure 2.17: Total capacitance behaviour vs. bias voltage for a) <111> e b) <100> crystalorientation.
by Cimp and Cb. From the measurement performed in laboratory, summarized in Table 2.2, it
has emerged that, once the bulk is overdepleted, the implant and bulk capacitance are mainly
functions of the design geometry, that is the same for the different orientation prototypes.
This fact explains why the Ctot is similar for the two detectors, the only difference depending
on the CAC value. This last contribution is more strongly dependent on the <100> or <111>
orientation and the oxide thickness.
The Ctot value is of fundamental importance in the noise evaluation since is the only external
parameter that appears in the amplifier noise expression and that can be partially under the
designer control. In our case the APV6 chip noise in peak mode can be parametrized as [3]:
ENCAPV 6 = a + b · Ctot(pF ) (2.25)
with a = 510 e− and b = 36 e−/pF.
The APV6 chip can be operated in a second distinct mode, as will be explained in chapter 4,
that is called deconvolution mode and that effectively reduces the time shaping constant [31]
at the expense of a larger amplifier noise. In this case we have, referring to Eq. 2.25, that the
coefficients are a = 1000 e− and b = 46 e−/pF. The shorter time constant obtained by the decon-
55
volution algorithm has an effect on the series and parallel noise too. It can be demonstrated [31]
that for a CR-RC pulse shape with a nominal peaking time of 50 ns and a sampling time of 25
ns the effect of the deconvolution method on the noise is taken into account by multiplying the
expressions (2.22) and (2.23) with appropriate weights:(ENC)deconvseries = (ENC)peakseries · 1.45
(ENC)deconvparallel = (ENC)peakparallel · 0.45(2.26)
We observe an increase of the series noise and a lowering in the parallel noise contribution.
So the deconvolution method is well suited for application where large leakage current are
present, as happens for irradiated detectors (see Appendix B).
The total noise is the quadratic sum of all the contribution mentioned above (parallel, series,
APV6) and, for the detectors described in this work, is principally affected by the amplifier
noise. The expressions (2.22) and (2.23) can be formulated in a simplified form parametrized
as a function of the measured quantities. For a temperature of -10C, corresponding to the
operating conditions at CMS, they are:
ENCIf 107 ·
√If (µA)τ(ns) e− (2.27)
ENCRs 23 · Ctot(pF ) ·√
Rs(Ω)/τ(ns) e− (2.28)
ENCRbias
√2 · 23 ·
√τ(ns)/Rbias(MΩ) e− (2.29)
In the last equation the term√2 arises from the quadratic sum of the bias resistor contribu-
tion for each crystal making up the module.
From the above relations it turns out that the parameter involved in charge and noise mea-
surement are the capacitance components, the bias and metal strip resistances and the leakage
current. All these quantities have been measured as described in section 2.5.1 (the leakage cur-
rent has been measured during operation too), so that the signal to noise ratio expected can be
compared to the experimental value. This will be the subject of section 7.2.
The module expected total noise, expressed in terms of ENC (electrons), as a function of the
bias voltage is shown in Fig. 2.18 for the<111> HR non irradiated detector and the <100> LR
irradiated detector. Both the peak and deconvolution noises are calculated taking into account
56
the measured electrical components (see section 2.5.1 and section 3.3 for the irradiated modules)
and the operational leakage current. The front-end chip, series and parallel noise contributions
to the total noise are also shown.
57
Vbias(V)
EN
C(e
- )
ENCIf
ENCRpoli
ENCRs
ENCAPV6
ENCtot
<111> HR non irr. detector
peak mode
0
250
500
750
1000
1250
1500
1750
2000
20 40 60 80 100 120 140 160 180 200
(a)
Vbias(V)
EN
C(e
- )
ENCIf
ENCRpoli
ENCRs
ENCAPV6
ENCtot
<111> HR non irr. detector
deconvolution mode
0
500
1000
1500
2000
2500
3000
20 40 60 80 100 120 140 160 180 200
(b)
Vbias(V)
EN
C(e
- )
ENCIf
ENCRpoli
ENCRs
ENCAPV6
ENCtot
<100> LR irradiated detector
Peak mode
0
500
1000
1500
2000
2500
3000
100 200 300 400 500 600 700 800
(c)
Vbias(V)
EN
C(e
- )
ENCIf ENCRpoli
ENCRs
ENCAPV6
ENCtot
<100> LR irradiated detector
Deconvolution mode
0
500
1000
1500
2000
2500
3000
100 200 300 400 500 600 700 800
(d)
Figure 2.18: Expected total noise (ENCtot) and APV6 chip, series and parallel noise contri-butions to the total noise as a function of the bias voltage for the <111> HR non irradiatedmodule ((a) peak mode (b) deconvolution mode) and for the <100> LR irradiated module((c) peak mode (d) deconvolution mode). The noise is expressed in electrons. The total noiseincrease due to the deconvolution algorithm is clearly visible (b) (d).
58
Chapter 3
Irradiated silicon microstrip detectors
The CMS microstrip silicon detectors will operate in an unprecedented radiation environment,
characterized both by particles produced in the primary proton-proton interaction and by albedo
neutrons emitted by backscattering from the electromagnetic calorimeter surrounding the tracker.
An average fluence of 1.6 · 1014 1 MeV equivalent n/cm2 is envisaged on the devices closer to
the interaction point, after 10 years of LHC operations. It is then evident that one of the most
critical issue of the silicon tracker is the long-term survival after heavy irradiation and that a
detailed study of radiation effects on detector performances is required in order to guarantee an
optimal behaviour during the full experiment lifetime. Furthermore, from the characterization
of the sensors after irradiation, the expected signal to noise ratio and the optimum operating
conditions can be foreseen. For these reasons a set of detectors has been irradiated up to a flu-
ence of 1.1 · 1014 1 MeV equivalent n/cm2, corresponding to the foreseen radiation value after
10 years of LHC operations in the region where they will be installed, and their performances
have been compared with similar non-irradiated modules. Finally, results obtained with the sev-
eral specimens built in Florence allow to compare the irradiation effects with respect to crystal
orientation and bulk resistivity.
We will show that the main macroscopic effects after heavy irradiation reflect in a full de-
pletion voltage change and in an increase of the leakage current. In order to have the detector
properly operated the temperature must be kept low and the bias voltage must be adjusted so to
overdeplete the bulk in its new conditions.
59
3.1 Radiation damage in silicon detectors
At the microscopic level the radiation damage suffered by the detectors can be divided in two
different classes: effects which are due to surface damage and those which are due to bulk
damage, the latter being the greatest source of concern since they ultimately limit the detector
functionality.
3.1.1 Surface damage effects
The electron-hole pair generation in the silicon bulk, induced by ionizing radiation, is a com-
pletely reversible process without damaging effect. The behaviour is different in the insulating
oxide layers present on the detector surface since some holes become trapped in the oxide or
interact with atoms at the silicon-oxide interface to form interface states. Fixed positive charge
in the oxide layer modifies the electric field in the detector, while interface states give rise to
new energy levels in the forbidden gap which can modify the device behaviour. The net effect
is the forming of an electron layer in the silicon close to the oxide interface with the conse-
quent decrease in inter-strip isolation, causing unwanted signal charge sharing, and an increase
in inter-strip capacitance, which is the major factor in determining the electronic noise of the
system. However we will show that a careful choice of the fabrication technology and of the
detector design can minimize these damage effects to an acceptable level. In particular the cou-
pling between the strips is influenced by the oxide quality (process dependent); this effect is
reduced by substantially over-depleting the device. The tests performed on Florence detectors
have shown that our devices can operate at high bias voltages thus minimizing the radiation
surface effects.
3.1.2 Bulk damage effects
Bulk damages are generated when the incident particle transfers enough kinetic energy to a
silicon atom to move it from its lattice site. The displaced atom is called primary knock-on
atom (PKA) or recoil atom . The PKA and the lattice vacancy are known as point-like defects
and introduce allowed energy levels in the forbidden gap.
The energy threshold for this process is 185 eV for a neutron impinging on a silicon atom but
the particles involved in the tracker radiation damage have energy orders of magnitude greater.
60
This implies that not only the incident particle can produce further displacements but the PKA
itself can be emitted with an energy enough to produce more displacements and defects. Most of
the initial energy is lost by ionisation and only a small fraction contributes to the displacements.
This is not true at the end of the recoil atom path where the energy loss density increases and
a dense defect region is created. Quickly most of the point-like defects recombine and only
those which do not annihilate form more stable complex defects or migrate towards the surface.
These defects can be classified as acceptors or donors, depending on the electrical properties
and on the energy level position they occupy in the forbidden gap, and since the acceptor type
dominates, this results in an effective doping concentration change in silicon.
The two principal effects of this process are a change in the effective doping concentration
of the substrate material and an increase of the leakage current.
In spite of the large amount of studies performed on irradiated silicon detectors a complete
model to describe the changes in effective doping concentration as a function of absorbed flu-
ence, time and annealing temperature has not been proposed yet. In the following we will refer
to the empirical model known as Hamburg model , which agrees with most of the experimental
data [32] [34]. According to this model the change of effective doping concentration can be
parameterized as the sum of three contributions:
∆Neff = Neff,0 −Neff = Nc(φ) +Na(φ, t, T ) +NY (φ, t, T ) (3.1)
where Neff,0 and Neff are the effective doping concentrations, respectively before and after ir-
radiation, φ is the irradiation fluence, t is the time elapsed since irradiation and T is the absolute
temperature the detector has been maintained after exposure.
The first term Nc is the contribution due to defects that are stable in time; it depends on a
decrease of donors, exponentially saturated with fluence, and on an increase of acceptors, linear
with fluence:
Nc(φ) = NC,0(1− exp(−cφ)) + gcφ (3.2)
where NC,0 is closely related to the initial doping concentration and can be thought as the num-
ber of removable donors, c and gc are the donor removal and acceptor creation parameters. A fit
61
type inversion
p-typen-type
Figure 3.1: Change in the effective doping concentration, due to the stable contribution, as afunction of the irradiation fluence. The fit of Eq. 3.2 to the experimental points shows a goodagreement with data [35].
of equation 3.2 to experimental data measured on neutron and electron irradiated detectors [35]
is shown in Fig. 3.1.
With increasing fluence the doping density decreases until the residual donor and the newly
generated acceptor populations are equal and the silicon bulk becomes intrinsic; at higher flu-
ences the bulk is type inverted and the effective doping concentration is mainly due to radiation
induced defects. In any case the polarity of reverse biasing in initially p+-n devices does not
change with type inversion, simply the junction moves from the p+ implant strip side to the n+
contact on the back side of the detector. From the behaviour of Neff as a function of fluence
it is evident that the inversion point depends strongly on the initial resistivity (N eff,0) and this
characteristics reflects on the dependence of the full depletion voltage versus fluence. In fact
we can express the depletion voltage as a function of Neff (see Eq. 2.9) as:
Vdepl =ed2
2εNeff (3.3)
62
where d is the detector thickness. In Fig. 3.2 the predicted evolution of the depletion voltage ver-
sus LHC time operation for silicon detectors is shown, stressing on the dependence on substrate
resistivity. Low resistivity devices must be operated at higher voltages in the first period but
gain safer operating condition after heavy irradiation with respect to high resistivity substrates.
Time (years)
Dep
leti
on
Vo
ltag
e (V
)
0
50
100
150
200
250
300
350
0 1 2 3 4 5 6 7 8 9 10
= 1 k •cmρ Ω
= 4 k •cmρ Ω
Figure 3.2: Predicted evolution of the depletion voltage with respect to LHC irradiation time fortwo different initial resistivities. The detectors are supposed to be in the first layer of the barrel.For each initial resistivity two curves are shown, one assuming a total fluence of 1.6·1014 n/cm2,the other, more pessimistic, 2.4 · 1014 n/cm2 [3].
The change in the full depletion voltage is a fundamental process that has to be taken into
account when dealing with irradiated detectors since they should be operated overdepleted since
the first years of LHC running but, on the other hand, the breakdown threshold must never be
exceeded.
The second term in Eq. 3.1 describes the decay of the active acceptor defects, created during
the irradiation period, back to neutral inactive sites, hence the name of beneficial annealing.
This effect is produced by an arrangement of the defects in the short time after irradiation and
it shows an exponential decay with a time constant, strongly dependent on temperature, which
ranges from about 2 days at 20C to 250 days at -10C.
The last term is the reverse annealing effect and its behaviour is opposite to the beneficial
63
one. NY starts from zero at t=0 and saturates to a final value proportional to the fluence with
a time constant of two years at room temperature. Since reverse annealing is a cumulative
process it is necessary to cool the silicon detector (down to -10C) not only during beam periods,
when this procedure is necessary to reduce the leakage current too, but also during the stand-by
period. Otherwise, after years of operation, the increase of the defects would always lead to full
depletion voltages greater than the detectors breakdown threshold. The sharp edges in Fig. 3.2
are the beneficial and reverse annealing effects during the yearly scheduled maintenance period
when the tracker temperature is raised.
The resulting effect of the two annealing contributions produces a minimum in the effective
donor concentration after an annealing at 60C for 80 minutes. As the leakage current of the
irradiated detector is dependent on storage time and annealing temperature, the agreed standard
procedure is to measure the current at the reference temperature of 20C after a thermal treat-
ment of 80 minutes at 60C‘[36]. In our case only the diodes used to perform the dosimetry
measurement have undergone the beneficial annealing at 60C.
The other observable effect of bulk damage is the increase of the leakage current due to the
shorter lifetime of minority carriers (see Eq. 2.11), caused by the generation of deep impurities.
It has been shown [36] that the current density increase ∆I after irradiation is proportional to
the fluence φ:
∆I
V olume= αφ (3.4)
where the V olume is the one interested in the current generation, φ is the 1 MeV equivalent
neutron fluence and α is the damage constant, independent on the material and technology used
for manufacturing [36]. It is worth noting that in our case the leakage current after irradiation
is two orders of magnitude greater and so ∆I Iirr. In order to compare currents measured at
different temperatures (T1 and T2 in the following equation) we can use the relation:
I(T1) = I(T2) ·(T2
T1
)2
exp
(− Eg
2K·(1
T2− 1
T1
))(3.5)
where Eg = 1.12 eV and K is the Boltzmann constant [36].
The signal to noise ratio is also affected by the decrease in charge collection efficiency,
which is caused by the trapping of charge carriers at the defects in the silicon bulk. It has been
64
shown that the resulting signal loss is moderate and has a value lower than 10 % after a fluence
of 1.0 · 1014 n/cm2 [37].
3.1.3 The absorbed dose expressed as 1 MeV neutron equivalent fluence
The damage induced by non-ionizing energy loss of the incident particle in the silicon detector
depends on the particle type and energy. It is useful to refer to a normalized fluence which
doesn’t take into account the energy and particle type in order to be able to compare the results
obtained using different irradiation facilities and to predict the effects of new radiation environ-
ments. Usually the normalization is made in terms of 1 MeV neutron equivalent fluence in the
framework of the NIEL (Non Ionizing Energy Loss) scaling hypothesis [38]. This procedure as-
sumes that the lattice damage induced by particles of energy E depends only on the energy loss
in removing silicon atoms from their lattice sites, and neither on the spatial distribution of the
introduced displacement defects nor on the annealing sequences following the initial damage
event. Ionization energy loss and phonon production do not contribute to the lattice damages.
The NIEL effect can be expressed by the displacement damage cross section D(E) summing
over all the possible reaction channels for the initial particle and its energy. If we consider
that each PKA has a specific recoil energy ER and that only a fraction of the recoil energy is
deposited in form of displacement damage according to the ER dependent Lindhard partition
function P (ER), we can calculate D(E) as:
D(E) =∑k
σk(E)
∫fk(E,ER)P (ER)dER (3.6)
where σk(E) is the individual reaction cross section and fk(E,ER) is the energy distribution
of recoils in reaction k .
Starting from the displacement damage cross section D(E) it is possible to define an index
of the damage, called hardness factor k . Usually the hardness factor k is defined so to com-
pare the damage produced by a particular irradiation type to the damage that would have been
produced by mono-energetic neutrons of 1 MeV with the same fluence:
k =1
D(En = 1MeV )·∫D(E)φ(E)dE∫
φ(E)dE(3.7)
65
The equivalent 1 MeV neutron fluence Φeq which produces the same damage as an arbitrary
beam with a spectral distribution φ(E) and a fluence Φ is given by:
Φeq = kΦ = k
∫φ(E)dE (3.8)
In the following we will always refer to Φeq.
3.2 Irradiation of silicon detectors and dosimetry
In order to characterize the performances of radiation damaged detectors a set of devices, be-
longing to the same production batch of the sensors described in chapter 2.5, has been irradiated
using the neutron beam facility at the Louvain-la-Neuve cyclotron [39]. The specimens selected
allow to build two complete irradiated modules, a <100> high resistivity (HR) and a <100>
low resistivity (LR) one, so that a comparison with the performances before and after irradiation
can be fulfilled.
The sensors were exposed to a 20 MeV mean energy neutron beam together with some
silicon diodes that later allowed us to perform a dosimetry measurement. The cyclotron line
used to irradiate the detectors generates an intense fast neutron beam from the reaction 9Be +
d → n + X, obtained by impinging a 50 MeV deuterium beam on a 1 cm thick berillium
target. The other reaction products are stopped by a three layer filter, made of polystyrene (1
cm), cadmium (1 mm) and lead (1 mm), only a 10 % fraction of γ rays produced in the target
passes through the stop. The detectors were placed orthogonally to the beam at 40 cm distance
from the target, so to have an uniform irradiation over all the sensitive area. The irradiation
experimental setup scheme is shown in Fig. 3.3.
During the irradiation the detectors were kept at room temperature and unbiased. After
irradiation the sensors have been maintained at low temperature (-10C) to reduce reverse an-
nealing effects. A preliminary estimation of the time necessary to heavily irradiate the detectors
has been made starting from previous irradiation sessions performed with the same neutron
beam [40] (6 hours corresponding to 1.9 · 1014n/cm2 nominal fluence); in any case the equiva-
lent fluence has been measured later directly on our samples so to have an experimental check.
The 1 MeV neutron equivalent fluence is measured experimentally using two completely
independent methods.
66
1.2 cmCollimator
2 cm
Berillium target
DetectorsNeutron beam
1 cm40 cm
1 cm Filter
integrator
Deuterium beam
CurrentFlux meter
Figure 3.3: Final stage of the experimental setup used to irradiate the detectors.
In the first case we consider the relationship between the increase in the current density ∆I ,
related to the irradiation, and the equivalent fluence according to Eq. 3.4. A set of diodes, built
on the test structures surrounding the active area of the detector on the original silicon wafers,
have been irradiated together with the sensors. From the current behaviour before and after
irradiation the fluence has been calculated.
The damage constant α, measured at room temperature and for silicon detectors that have un-
dergone a beneficial annealing lasting 80 minutes at 60 C, is α = 4.0 · 10−17 A/cm, with an
accuracy of 5% [41]. This value is independent on the substrate and technology used so that is
the same for all test structures.
The current measurement must be performed on a properly defined diode geometry due to
the volume factor that appears in Eq. 3.4. Thus the current flowing in the device is measured
with the diode guard ring connected to ground; in this way the volume is well defined and is
about 0.440 mm3.
Since the α value refers to 20 C, the equation 3.5 has been used to obtain the correct current
value starting from the -10 C measured ones (at 500 V). The resulting mean 1 MeV equivalent
neutron fluence is (0.96 ± 0.12) · 1014 n/cm2. The uncertainty is mainly due to the fact that
diodes were positioned in three radial region around the neutron beam axis and have suffered
slightly different fluences.
In Fig. 3.4 the setup of the diode current measurement is sketched (a) and the current as a
function of the bias voltage is shown for several test structure (b).
67
H
Keithley 480Bias voltage Picoamperometersupply
Keithley 237
H L L
(a)
T=-10oC
Post-annealing
Vbias(V)I di
ode(
A)
0.05
0.075
0.1
0.125
0.15
0.175
0.2
0.225
0.25
x 10-6
0 100 200 300 400 500 600 700
(b)
Figure 3.4: (a) Setup used to measure diode test structure current. (b) Diode current vs. biasvoltage for several diodes. (Current values for fluence determination are taken at 500 V.)
In the second case the flux estimation was done using a reference detector taken from the
same production batch as a previously irradiated one (“old” in the following) and with the same
electrical and geometrical characteristics . The “old” detector was exposed to a neutron beam
at the ATOMKI Cyclotron, Hungary, with a known dose of 0.97 · 1014 1 MeV equivalent n/cm2
with an uncertainty about 15% [42]. The reference detector has been irradiated at Louvain-La-
Neuve together with the other detectors and diodes and has undergone the same treatment, after
exposure, than the “old” one so that the α value is the same. According to Eq. 3.4 the leakage
current is proportional to the fluence and from a comparison with the values obtained from
the “old” detector the fluence can be measured once we know the reference detector leakage
current, being α and the volume the same. In Fig. 3.5 the leakage currents corresponding to the
reference detector and to the “old” detector as a function of the bias voltage are compared. From
their values at 250 V the 1 MeV equivalent neutron fluence turns out to be (1.3±0.2)·1014n/cm2.
From these two measurements we can estimate a fluence of 1.1 ·1014 n/cm2 that corresponds
to 10 years of operation at LHC for our detectors and which is enough to produce type inversion
68
T = -10oCDose to be calculated (Ref.)
Φ = 9.7 x 1013 n/cm2 (Old)
Vbias(V)
I(A
)
0
0.05
0.1
0.15
0.2
0.25
0.3
x 10-3
0 50 100 150 200 250 300
Figure 3.5: I-V characteristics for the reference (Ref.) detector and the previously irradiatedone (Old).
(see chapter 3.3.3 and [42]).
3.3 Characterization of irradiated detectors
In the following sections the results of irradiated detectors characterization are reported with
particular attention paid to the implications on performances and operating conditions of the
final modules.
3.3.1 Leakage current
The leakage current measurement has shown an increase up to three orders of magnitude after
irradiation. For this reason (and in order to reduce the reverse annealing effect) all the measure-
ments have been done at -10C and this temperature has been adopted as the standard for the
subsequent irradiated module testing procedures.
69
3.3.2 Bulk capacitance and full depletion voltage
The bulk capacitance measurement allows to extract the full depletion voltage (see Eq. 2.13)
from a plot 1/C2bulk versus bias voltage. The results are shown in Table 3.1 and it is evident that
the LR devices have a lower full depletion voltage value (about 130 V) than the HR ones (250
V).
From the full depletion voltage value the effective doping density is obtained using Eq. 2.9.
For comparison purpose in Table 3.1 the full depletion voltage values for non irradiated detec-
tors are reported as well.
Detector Vdepl irr. |Neff | irr. Vdepl non-irr. |Neff | non-irr.(V) (cm−3) (V) (cm−3)
< 111 > HR 250 3.6 · 1012 54 7.9 · 1011
< 100 > LR 128 1.9 · 1012 240 35 ∗ 1011
Table 3.1: Full depletion voltage and effective doping density after heavy neutron irradiation(1.1 ·1014 1MeV equivalent neutron fluence).Values for non irradiated detectors are reported forcomparison purpose.
3.3.3 Bias resistor
From the bias resistor behaviour as a function of bias voltage it is evident that our irradiated
detectors have undergone bulk type inversion. As shown in Fig. 3.6 the resistance reaches
its maximum stable value only after full depletion, while before irradiation the Rbias value is
voltage independent.
In fact, after type inversion, the depletion starts at the n+-p interface on the ohmic back-side
and the not-fully depleted substrate contributes with a resistance in parallel to Rbias. Thus the
measured resistance is reduced until the bulk is completely depleted. Before irradiation this ef-
fect disappears even with a few volts of polarization since the depletion starts from the junction
side where the resistors are located.
From the behaviour of the bias resistance versus voltage the new full depletion voltage is ob-
tained and agrees with the value calculated using Cbulk measurement.
70
Vbias(V)
Rbi
as(M
Ω)
strip 116
strip 283
1.8
2
2.2
2.4
2.6
2.8
0 50 100 150 200 250 300 350 400 450
Figure 3.6: Bias resistor value vs. bias voltage. Two different resistors are shown, belonging todifferent regions of a F5 < 111 > HR sensor.
The metal strip resistance wasn’t measured due to setup problems but it is likely not to
have changed after irradiation and, in any case, its implication on detector performances are of
second order.
3.3.4 Coupling capacitance
The measurements performed on the irradiated detectors show that the coupling capacitors are
not damaged by the neutron heavy irradiation. In Fig. 3.7 the pinhole distribution before and
after irradiation for the same sensor is shown. No appreciable difference has emerged and this
makes us confident of the good behaviour of our detectors with respect to oxide layer defects
even after heavy irradiation. The measurement is made at room temperature with a 1 V direct
bias voltage across the coupling capacitor as explained in section 2.5.1.
The CAC value as a function of the bias voltage is influenced by the type inversion and
only with detectors overdepleted the coupling capacitance reaches its maximum. In order to
maximize the charge collection it is necessary to operate the irradiated devices in over depletion
71
strip number
I (A
)
f4-4080non irradiated
strip number
I (A
)
f4-4080irradiated
10-13
10-11
10-9
10-7
10-5
100 200 300 400 500
10-1210-1110-1010-910-810-710-610-5
100 200 300 400 500
Figure 3.7: Comparison of the pinhole defects before and after irradiation on the same detector.
regimes. In Table 3.2 the CAC values are shown.
3.3.5 Interstrip capacitance
The implant capacitance was measured, only for the <100> detectors, with the wafers kept at
-10C in a climatic chamber. The effect of the charge accumulation layer at the silicon-oxide
interface is clearly visible from the increase in the Cimp value after irradiation, compared with
the non irradiated device value, shown in Fig. 3.8 (a) as a function of the bias voltage.
The different behaviour can be explained in terms of the free electrons accumulation layer
present at the silicon oxide interface. The layer can be modelled as a bias voltage dependent
resistor Re which increases its value when the charges are removed by the increasing field in
the region under the oxide. Thus the implant impedance is the parallel between the implant
capacitance and Re (see Fig 3.8 (b)). When the depletion reaches the strip side, the capacitance
72
Vbias(V)
Cim
p(pF
)f = 400 Hz
irradiated
non irradiated
0
2
4
6
8
10
12
0 200 400 600 800 1000
(a)
+ + + + + + + + + + + + + + + + + - - - - - - - - - - - - - - - - - --- - - - - -
interface defects
R e
C imp
electrons
(b)
Figure 3.8: (a) Implant capacitance vs. bias voltage for a F5 high resistivity detector before andafter irradiation. (b) Model of the implant capacitance and of the free electron accumulatinglayer, due to surface radiation damage, present at the silicon-oxide interface.
decreases sharply until the bias voltage reaches a value two times higher than the full depletion
one. After that, the Cimp value decreases more slowly due to further electron confinement as
it happens for non irradiated detectors. The behaviour at bias voltage lower than 100 V can be
ascribed to the type inversion of the bulk and to the incomplete isolation of the strips from the
bulk.
In case of overdepleted device the implant capacitance turns out to be 1.71 pF and 1.27 pF
for the F4 and F5 detectors respectively, with an increase of about 10 % compared to the non
irradiated detector value.
In Table 3.2 the main electrical parameters are summarized. The measurement were per-
formed at -10C and the values are obtained with the detector over-depleted.
73
Detector Type Rbias Cb CAC Cimp
(MΩ) (pF/cm) (pF/cm) (pF/cm)
F4 < 111 > 2.49 0.41 33.3 -
F5 < 111 > 2.49 0.45 33.1 -
F4 < 100 > 3.23 0.41 38.9 0.259
F5 < 100 > 3.23 0.48 37.9 0.225
Table 3.2: Main electrical parameters after heavy neutron irradiation (1.1·1014 1MeV equivalentneutron fluence). All the measurement were made at -10C and, when present, with a biasvoltage greater than twice the full depletion voltage.
3.3.6 Total capacitance
After the irradiation the behaviour of the total capacitance seen by the front-end electronics,
with respect to the bias voltage, is changed according to the implant capacitance increase. In
this case the minimum value of the capacitance is obtained for bias voltage greater than twice
the full depletion voltage. This effect is depicted in Fig. 3.9 where the full depletion voltage for
the irradiated detector is 128 V. We notice that the detectors have to operate at high bias voltage
Vbias(V)
Cto
t(pF
)
Module <100>
non irradiated
irradiated
5
10
15
20
25
30
35
40
45
100 200 300 400 500 600 700 800
Figure 3.9: Capacitance seen by the pre-amplifier input vs. bias voltage for the <100> detector(F4+F5) in case of irradiated and non-irradiated devices.
74
to minimize the effects of irradiation which, in this case, increases the Ctot value of about 10 %.
It is evident the advantage of using low resistivity substrates which undergo type inversion
at higher fluences thus having lower depletion voltage than high resistivity ones at the end of
the detector lifetime. In this conditions heavily irradiated detectors can be easily overdepleted
decreasing the noise contribution due to the interstrip capacitive load to the front-end electron-
ics.
75
Chapter 4
The APV6 front-end chip
In this chapter the main features of the CMS Silicon Tracker front-end chip prototype (APV6)
will be described. The APV6 chips, sitting on a ceramic support (hybrid), have been used to
test the detectors described in this work and to perform a first “full system test” of the Silicon
Tracker. The deep understanding of the chip functionality is one of the main topics that I have
studied for this thesis. In this framework part of the work has concerned the design of a test
procedure for the hybrids and their APV6 chips. The setup described in chapter 5 has revealed
very flexible in measuring all the main chip parameters in a few minutes. A set of 8 hybrids,
for a total number of 30 APV6 chips, has been tested to select the ones to be used for the
Milestone 99 modules. The Milestone 1999 had the goal of evaluating the capability to build
several Silicon Microstrip Tracker modules and to test their performances.
4.1 The APV6 chip
The read out architecture of CMS Tracker takes advantage on Very Large Scale Integration
(VLSI) technique that allows amplifying the signal, released by a particle crossing the detector,
very close to the silicon sensor, reducing the noise pick up. Front-end electronic characteristics
are imposed by the hard requirements, in terms of signal time localization within a single bunch
crossing, high occupancy, radiation hardness and low noise level, necessary to achieve the de-
sired tracking performances in the environment where the Silicon Tracker will have to operate.
Research and development (R&D) program started in 1992 (RD20 collaboration [28]) has lead
to the construction of fast readout electronics fulfilling such requirements and whose main block
is the APV (Analogue pipeline Voltage mode) front-end chip series. The last prototype for
77
CMS Tracker front-end chips is the APV6, built using radiation-hardened Harris AVLSIRA
process [43] in 1.2 µm bulk CMOS technology from a Rutherford Appleton Laboratory and
CERN design [5]. The APV6 chip consists of a 128 channels analogue section and some sys-
tem features including a slow control communication interface, programmable on chip analogue
bias network and internal test pulse generation. Each channel contains a pre-amplifier and a
shaper stage, with a peaking time of nearly 50 ns, followed by a 160 location pipeline memory
in which samples (strip signals) are written at the 40 MHz LHC machine frequency. This ana-
logue memory is made of a 128 × 160 array of capacitor cells, whose dimensions are 35 µm
×30µm. Each cell contains two transistors, to perform read or write operations, and a 0.25 pF
storage capacitor.
The pipeline memory contains a record of the most recent data in a window of 160 × 25
ns= 4µs, in order to match the maximum CMS Level 1 trigger latency of 3.2 µs. Two pointers
control readout operations. A write pointer cyclically moves through the pipeline, one row per
bunch crossing, and decides in which location sampled data has to be written. A read pointer
follows the write one by an interval, referenced as trigger latency and measured in number of
pipeline clock cycles, that is the time between an analogue signal being applied to the APV6
input and the corresponding logical trigger time arrival to the chip. This data access mecha-
nism allows the marking and queuing of requested locations for output while embedded logic
ensures that samples waiting readout are not overwritten with new data. The pipeline buffering
is crucial in a high rate experiment like CMS in order to eliminate the dead time contribution
of the level-1 trigger. Following a trigger, a series of samples from the memory are processed
by the APSP (Analogue Pulse Signal Processor) section of the front-end chip. This part can be
operated in two modes: peak or deconvolution (see section 4.1.3).
After the APSP the processed data are held in a further memory buffer before switching
through an output analogue multiplexer. This additional buffer is required so that as one event
is multiplexed out another may be prepared for consecutive transmission reducing readout dead
time due to the statistical fluctuations of the “time-interval” distribution between two consecu-
tive level-1 triggers. The multiplexer operates at 20 MHz and uses a nested architecture to save
power since only the final 4:1 stage has to run at full speed. This has, as a consequence, that
the analogue data come out in a non-consecutive channel order but are interleaved [44] . A fifth
78
input to the final stage allows the insertion of digital data, containing error coding and pipeline
address information (see Fig. 4.1), at the beginning of the analogue levels.
Address
4 bit header 128 analogue levels
7.0 microseconds
0uA
100
200
300
400
500
600
Time
Ou
tpu
t C
urre
nt
(u
A)
Address
4 bit header 128 analogue levels
7.0 microseconds
0uA
100
200
300
400
500
600
Time
Ou
tpu
t C
urre
nt
(u
A)
Figure 4.1: APV6 Output frame. A signal, as it appears in deconvolution mode (blue upperframe) and in peak mode (green lower frame), is also shown.
When there is nothing to transfer the analogue output of the chip is at the logic 0 level with
single logic 1 states, called tick marks, every 1.75µs. The output from the APV6 is in current
form in the range from 0 to 600 µA and a MIP equivalent signal is represented by a current
value of the order of ∼ 50µA. A layout picture of APV6 chip is shown in Fig. 4.2, with all
major logic and analogue blocks.
The chip overall size is 12.0 × 6.25 mm2. On the left the 128 analogue inputs pads, grouped
into four section of 32 separated by large power supply pads, are visible. Each group of inputs
is arranged in two staggered rows; pads on the same row are spaced at 86 µm but the other row
is offset 43 µm, to allow microbonding, and this results in an effective bond pitch of 43 µm.
On the right side, from top to left, remaining power supply, test, bias reference, data, address,
clock, trigger and serial control pads are located [5].
The power supplies are nominally run at ±2 Volts and ground, with a power consumption
of 2.4 mW/channel.
79
Figure 4.2: Layout of APV6 readout chip. The real dimensions are 12.0× 6.25 mm2.
4.1.1 Analogue stages
Each APV6 channel is made of a pre-amplifier and a shaper stage, as shown in Fig. 4.3. The
software controlled parameters VSHA and VPRE allows to change the impedances that affects
the timing response of the analogue circuit. We have showed [47] that, contrary to what has
Vpbp
Vpcasc
Vpbn
M_p_pinp3000/1.4
M_p_pis150/10
Cfp .25p
M_n_pfb2.4/60
M_n_pcasc400/1.2
M_n_pis2330/10
M_n_psf400/1.2
M_n_pis3200/10
Vpsfb
Vsbp
Vscasc
Vsbn
M_p_sinp800/1.4
M_p_sis150/10
Cfs .25p
M_n_sfb2.4/60
M_n_scasc400/1.2
M_n_sis2100/10
M_n_ssf400/1.2
M_n_sis3200/10
Vssfb
VPRE
VSHA
Cc1.8p
Vsbp
Vscasc
Vsbn
M_p_sinp800/1.4
M_p_sis150/10
Cfs .25p
M_n_sfb2.4/60
M_n_scasc400/1.2
M_n_sis2100/10
M_n_ssf400/1.2
M_n_sis3200/10
Vssfb
PreamplifierShaper
Figure 4.3: APV6 front-end electronics scheme. The components interested by the VSHA andVPRE registers are shown.
been reported in literature [5], modifying the value of the VSHA register the APV6 output
80
doesn’t behave as a true two poles CR-RC filter.
4.1.2 Control interface
The configuration, bias settings and error states of the APV6 are handled by a two wire se-
rial interface which conforms the I2C standard, so that it may be controlled using commercial
components [45]. The APV6 chip can only act as slave device.
Every I2C transmission is composed of three bytes. The first byte contains the APV6 ad-
dress, the second one the command to be executed (register name, read or write operation) and
the last one the register value to be set. The APV6 binary address “1111” is reserved for broad-
cast addressing, so when it is used all connected chips will respond. Consequently a maximum
of 15 APV6 chips may share the same controller with different addresses.
Up to 13 variables are set or read from APV6. The meaning of the main (from an user point
of view) registers is reported in Table 4.1.
Name Description
Latency LAT Distance between write and read pipeline pointers.Value up to 160 (1 step=25 ns).
MODE Allows to switch between Peak and Deconvolution mode,to turn the power OnOff,to use the internal calibration.
Analogue bias Programmed values are converted by on-chip DACs.VSHA,VPRE VSHA and VPRE act on the feedback stages of the shaper
and pre-amplifier circuit respectively.VADJ VADJ allows to change the output frame level.CDRV Selects which group of 8 channels to pulse in calibration
mode.CLVL Selects the charge injected in calibration mode.CSKW Sets the delay between trigger and calibration pulse
8 steps of 3.125 nsError (read only) latency or FIFO error
Table 4.1: Principal APV6 internal registers.
The Latency register allows to select the separation between the write and read pointer of
the pipeline in units of 25 ns. This distance, expressed in time unit, is referred to as latency and
it is a fundamental parameter of the timing sequence that controls the entire DAQ chain (see
chapter 5.4).
81
4.1.3 Operation modes
The APV6 chip can be operated in two modes: peak mode in which the output sample cor-
responds to the peak amplitude of the amplifier output following a trigger, and deconvolution
mode, in which the output corresponds to the peak amplitude coming out from the APSP cir-
cuitry. In deconvolution mode three samples are sequentially read from the pipeline, as shown
in Fig. 4.4 in comparison with the peak sampling, and the output is a weighted sum of all three.
Time (5ns/division)
A.U
.
Ideal CR-RC output shape
Deconvolution ModePeak mode
-50 -25 0 25 50 75 100 125 150 175 200
Figure 4.4: Processed samples in peak and deconvolution mode after a trigger request.
This last operation effectively results in a re-shaping of the analogue pulse shape to one
confined within a bunch crossing time interval. The technique is referred to as deconvolution
since it retrieves the original current pulse from the amplifier shaped pulse [31]. By inverting
the transfer function of the shaper it is possible to calculate the set of weights which, applied to
three consecutive samples, perform the deconvolution operation. The weights are implemented
on the chip APSP circuitry by using three different capacitors.
The use of the deconvolution mode is mandatory in high luminosity LHC operations since
otherwise the effect of pile-up would result in a persistent background, for each triggered event,
82
due to signals generated in previous events thus spoiling the track finding algorithm perfor-
mances. By reducing the particle signals within a single bunch crossing, the deconvolution
obtains a faster pulse shape at the expense of an increase in both power consumption and, what
is worse, in the electronic noise. In Fig. 4.5 the reconstructed shapes for the peak and decon-
volution mode are compared. These curves are obtained changing the latency value by 25 ns
steps in order to reproduce the effect of a particle signal coming from different bunch crossings.
It is evident that in deconvolution mode the signal from the two bunch crossing closest to the
optimal one are much more suppressed with respect to the ones in peak mode.
Peak mode
CK 40MHz
0 25 50 75 100 125 150 175 200 225 250 275 300 (ns)
Deconvolution mode
0 25 50 75 100 125 150 175 200 225 250 275 300 (ns)
CK 40MHz
Figure 4.5: APV6 output shape in peak and deconvolution mode.
The reduced time shaping has been obtained at the expense of an increase of the chip noise.
Nevertheless this is the default operating mode for the CMS Tracker in high luminosity runs and
the detectors and electronics must satisfy the required performances in terms of signal to noise
ratio using the deconvolution mode. As we will see in chapter 7 the detectors tested during
83
this work have completely fulfilled the expected performances, even in the worst scenario. The
equivalent noise charge introduced by the APV6 chip has been measured as a function of the
input capacitance [5], both in peak and deconvolution mode, and has shown a linear dependence
on the detector input capacitance to the chip, and is given by:
ENC(e−) = 510 + 36 · Cinput(pF ) (4.1)
for the peak mode, and by:
ENC(e−) = 1000 + 46 · Cinput(pF ) (4.2)
for the deconvolution mode.
A further APV6 feature, fundamental for test purposes, is the internal calibration system. In
calibration mode a user adjustable charge level is injected in the input of a group of 16 channels
when a 50 ns trigger signal (lasting two clock cycles) reaches the APV6. The channels are
spaced with a module 8 pattern but they appear as a single block at the analogue output due
to the multiplexer architecture. In Fig. 4.6 the APV6 analogue frame acquired before digital
conversion is shown. The third group of 16 channels is pulsed with a charge corresponding to
1 MIP. It is visible the digital header before the analogue part and a tick mark following the
frame.
4.2 APV6 chip response
The complete output pulse shape produced by the APV6 analogue amplification section can-
not be measured directly but an image can be built up by sampling the calibration pulse at a
fixed time and progressively shifting its starting time by means of the CSKW register. In data
acquisition mode the same operation can be done by delaying the APV6 trigger with respect
to the physical trigger (see section 5.4). This measurement performed in peak mode allows to
obtain an image of the shaper output (Fig. 4.7(a)) while in deconvolution it gives the possibil-
ity to verify the effectiveness of the algorithm implementation on the APV6 chip (Fig. 4.7(b)).
Furthermore the time delay scan is a powerful method to find experimentally the optimum sam-
pling point in order to measure the signal at its peak value. All the measurement performed in
laboratory and during the beam tests have been preceded by an optimization of the sampling
point.
84
Figure 4.6: APV6 analogue frame (upper curve) acquired before the digital conversion. On topof it the group of 16 channels pulsed by the internal calibration operation mode is visible. Thelower curve is the output enable signal provided by the chip in correspondence of the analogueframe.
Delay (ns)
Nor
mal
ized
out
put
Peak Mode
0
0.2
0.4
0.6
0.8
1
0 20 40 60 80 100 120 140 160 180
(a)
Delay (ns)
Nor
mal
ized
out
put
Deconvolution mode
-0.4
-0.2
0
0.2
0.4
0.6
0.8
1
0 20 40 60 80 100 120
(b)
Figure 4.7: Normalized output shapes in peak (a) and deconvolution mode (b).
85
4.2.1 APV6 characterization
In order to study the chip behaviour as a function of the shaper feedback impedance the output
shape in peak mode has been measured, with the internal calibration mode, for different VSHA
values. A two poles semi-gaussian curve has been fitted to the experimental points obtaining
only a marginal agreement (see Fig. 4.8).
Delay (ns)
Char
ge (A
DC
coun
ts)
VSHA = 2.0 V calib. data
VSHA = -0.20 V calib. data
0
50
100
150
200
250
0 20 40 60 80 100 120 140 160 180
CR-RC fit
Figure 4.8: Semigaussian fit performed on the APV6 output for two different values of theVSHA register.
The function which better approximates the experimental curve is a four pole (two real and
two complex) transfer function (see Appendix A). This result agrees with the analytical study of
the front-end circuit [47]; in Fig. 4.9 the four poles transfer function inverse Fourier transform
is fitted to the experimental data.
Since the APV6 chip behaves as a true CR-RC filter with a 50 ns time constant only on a
first order approximation, the weights of the deconvolution algorithm are not properly adjusted
and the charge measured in deconvolution mode is slightly lower than the expected one. This
has been confirmed both by the laboratory measurements [7] and by the electronics response
86
Delay (ns)
Char
ge (A
DC
coun
ts)
VSHA = 2.0 V calib. data
VSHA = -0.20 V calib. data
0
50
100
150
200
250
0 20 40 60 80 100 120 140 160 180
4 poles (2 real+2 complex) fit
Figure 4.9: Four poles transfer function inverse Fourier transform fit to experimental data.
simulation [47] (see Fig. 4.10). Fig. 4.10(b) refers to the simulated response of an ideal CR-RC
filter and shows that the corresponding output shapes in peak and deconvolution mode to iden-
tical input signals are equal. On the other hand Fig. 4.10(a) shows the simulated behaviour of
the APV6 chip using its detailed description explained in [47]. We see that the deconvolution
maximum is about 8% lower than the peak one, in agreement with the experimental results.
Furthermore the detailed simulation is able to reproduce the undershoot present in the experi-
mentally determined curves (see Fig.4.7(b)).
For all the APV6 chip we tested the linearity response has been measured using the internal
calibration system. Typical calibration curves, for a set of four different APV6 chips housed on
the same hybrid, are shown in Fig. 4.11 and in Fig. 4.12 for the peak and deconvolution mode
respectively. The CLVL values ranges from 0.3 MIP to 4 MIP equivalent charge.
The effect of the VSHA register when it is set to the extreme values allowed is reported
in 4.13. It is clearly visible that with a longer time constant (full circles in Fig. 4.13(a)), in peak
mode the output shape is deeply broadened and the charge measured is increased by 30% with
87
-20
0
20
40
60
80
100
-50 0 50 100 150 200 250
APV6 chip reponse
Peak shapeDec. shape
Time (ns)
A.U
.
0 50-50 100 150 200 250
-20
0
20
40
60
80
(a)
-20
0
20
40
60
80
100
-50 0 50 100 150 200 250
Peak shape
Dec. shape
True CR-RC filter
Time (ns)
A.U
.
250200
80
150100500 -50
-20
0
20
40
60
(b)
Figure 4.10: (a) Simulated effect of the non ideal CR-RC behaviour on the output shape of theAPV6 chip. The signal measured in deconvolution mode is lower than the one measured inpeak mode. (b) APV6 output shapes for an ideal CR-RC 50 ns filter. The green lines are thedeconvolution shapes, the black one the peak shapes.
Clvl (MIP equivalent)
AD
C c
ount
s
chip 12chip 14chip 1achip 1c
0
20
40
60
80
100
120
140
160
180
0 0.5 1 1.5 2 2.5 3
Figure 4.11: Calibration register response linearity in peak mode for a set of four APV6 chipshoused on the same hybrid. The detector is not bonded to the hybrid.
88
Clvl (MIP equivalent)
AD
C c
ount
s
chip 12chip 14chip 1achip 1c
0
20
40
60
80
100
120
140
160
180
0 0.5 1 1.5 2 2.5 3
Figure 4.12: Calibration register response linearity in deconvolution mode for the same APV6chips tested in Fig. 4.11
respect to the normally used shaper time constant. This behaviour is less critical in deconvolu-
tion mode.
Delay (ns)
Cha
rge
(AD
C c
ount
s)
VSHA = 2 VVSHA = -0.2 V
-50
0
50
100
150
200
250
300
0 20 40 60 80 100 120 140 160 180
(a)
Delay (ns)
Cha
rge
(AD
C c
ount
s)
VSHA = 2 VVSHA = -0.2 V
-100
-50
0
50
100
150
200
250
0 20 40 60 80 100 120
(b)
Figure 4.13: VSHA register effect on the output shape in peak (a) and deconvolution mode (b).
89
The VSHA parameter can be used to increment the S/N ratio but the broadening of the
signal, especially in peak mode, deeply affects the detector response for the bunch crossing
following the particle crossing time, considerably increasing the occupancy.
4.3 The APV25 read-out chip
The new prototype of the front-end chip for the CMS silicon microstrip detectors is the APV25.
It can be considered the straigthforward translation of the APV6 chip in the “deep submicron”
0.25 µm IBM technology. The APV25 maintains all the APV6 features but it takes advantage
of the intrinsic radiation tolerance of the submicron process. The S/N ratio is considerably in-
creased while the power consumption is decreased. Furthermore the industrial scale of IBM
manufacturer guarantees more flexibility in handling the initial debugging runs and can be con-
sidered more reliable in the long term production phase with a large cost saving with respect to
the other specialized manufacturers. The use of the APV25 is one of the key issues, from both
technical and economical point of view, in the possibility to build the all-silicon solution for the
CMS Silicon Tracker.
The design has been slightly modified to take into account the advantages offered by the
smaller size process, for example increasing the pipeline depth from 160 to 192 locations. The
power supply lines for this new circuitry are ±1.25 V and ground. On the other hand the
APV25 can operate both in peak and deconvolution mode, has an internal calibration system
and an analogue storage pipeline similar to the APV6 chip. So all the measurement performed
on the APV6 are a very good starting point to quickly test this new prototype in order to enter
the final production phase.
90
Chapter 5
The laboratory setup
The APV6 and the full size module testing procedures require the setting up of a flexible and
reliable system, able to switch between the different experimental situations that arise in the
R&D phase. In particular the same setup should be able to test the hybrid equipped with the
front-end chip alone, to test the fullsize module with a β source and eventually to allow the use
of a laser beam in order to perform a fast check of the response of all the strips and electronic
channels of the device.
A Data Acquisition System (DAQ) performing all these different tasks has been built in
the framework of this thesis. We took advantage on the availability of some official CMS
electronic chain blocks in order to easily compare the results obtained in our laboratory with
the CERN Beam Test ones. Furthermore the setup allows the testing of irradiated detectors
inside a climatic chamber.
In this chapter a detailed description of the laboratory setup is presented, with particular
emphasis on the custom electronic card and solutions especially developed by the CMS Florence
group in this context. The laser test facility will be described in chapter 6.
5.1 The Florence laboratory setup
The DAQ system is based on a VME crate equipped with a RIO8062 CPU running the real time
operative system Lynx-OS. A schematic view of the laboratory setup is shown in Fig. 5.1.
The module under test, or simply the front-end hybrid, is housed inside a climatic cham-
ber, model Haræus VTM 04/500, that allows to keep the detector temperature stable at the
appropriate operative point (usually -10C). The detector electronics is connected by means of
91
I2C
CPU
Trigger
Kapton
Selected MIPsScintillatorPM
VME
FED
RS232
hp8131A
Interface card
Hybrid
β Source
Magnet
Logic Unit
SEQUENCER
Climatic chamber
Silicon detector
Delay generator
Trigger
Clo
ckT
rigg
er
Analogue output
Clock
Figure 5.1: Block scheme of the laboratory setup.
a “Kapton” cable to an interface card, which itself is placed inside the climatic chamber. The
interface card was developed by the CMS Silicon Microstrip Detector collaboration [48] and
is used for laboratory tests as well as in beam test environment. This card provides the power
supply, the I2C commands, the clock and trigger signals to all the APV6 chips located on the
same hybrid. The clock and trigger signals are buffered by a LVDS receiver prior to reach the
front-end chips. Furthermore the interface card receives the analogue output signals from the
APVs and, after an amplification stage, sends them to the FED ADC through up to four differ-
ential cables, one for each APV6 chip. The detector is biased with a high voltage power supply,
model EG&G Ortec 556H.
Detector response to MIPs is investigated using a β source. The sensor is installed on a box
containing a 90Sr source and a bending electromagnet. By adjusting the current flowing in the
magnet coil, only electrons with momentum close to the end point of the spectrum ( 2MeV)
are bent towards the detector, thus simulating minimum ionizing particles. A plastic scintillator,
coupled to a low noise fast photomultiplier and located on the opposite side of the silicon surface
with respect to the source, provides the trigger signal only for the particles that have completely
crossed the detector. The trigger signal, properly discriminated and shaped, is delayed in time
with an HP8131A pulse generator before being sent to the APV6. In this way we can adjust
92
the delay between the arrival of the particle and the APV6 trigger at about 1 ns steps, being
able to study the APV6 response at different sampling time. As we have seen in section 4.1.2
this is also necessary since in “data acquisition mode” the APV6 can adjust the latency between
the particle passage and the trigger, using the internal LATENCY register, only in 25 ns coarse
steps.
In case of “internal calibration mode” the trigger signal is generated by a pulser which feeds
directly the Sequencer card.
The clock and trigger signal levels needed to run the APV6 chips and to operate the ADC
are provided by a custom made Sequencer card (section 5.3) that is the main block of the DAQ
systems. In order to follow the physical flow of data in our system it is necessary to first describe
the interface card that connects the module to the outer world.
5.2 The Tracker Interface Card
The hybrid is connected to the Tracker Interface Card (TRICARD) with a kapton flat cable
ending in two ERNI connectors (1.27 mm pitch), a 26 pins one devoted to the power supply
(± 2 V and GND) and a 50 pins one for the remaining services. The power levels are stabilized
on board by two voltage regulators working with ± 6 V.
The analogue output from every single APV6 chip is pre-amplified and transmitted by a
differential twisted-pair cable towards the ADC. The hybrid houses up to eight front-end chips
but the interface card is able to manage the analogue outputs of only four of them. This is not
a severe limitation since all of the modules produced for the Milestone 99 have a maximum of
512 strips (corresponding to 4 APV6 chips).
The clock and trigger signals are received from the Sequencer board and transmitted in a
LVDS (Low Voltage Differential Signal) logic standard to match the APV6 requirements, as
shown in Table 5.1.
The I2C control signal is transmitted to the TRICARD by a 4 way “Lemo” cable connected
to a VME board that houses four independent I2C drivers.
In Fig. 5.2 the TRICARD connected to a fullsize module is shown.
93
Signal Logic state Voltage level
CLKP , TRGP 0 <-200 mV1 >+200 mV
CLKN , TRGN 0 >+200 mV1 <-200 mV
Table 5.1: Clock and trigger logic levels for the APV6 positive and negative lines.
Figure 5.2: The interface board connected to a fullsize module.
5.3 The Sequencer
The Sequencer card is described in details since it is a custom made device that has been com-
pletely developed in the framework of this thesis. The Sequencer board is the main block of the
electronic chain and is devoted to generate all the signals needed by the APV6 to work and by
the ADC to sample and store the data. Its main feature is the capability of perform a correct
94
timing of the clock and trigger signals and to adjust the delay between the particle crossing time
and the front-end electronic trigger.
It contains the 40 MHz oscillator which provides the clock signal to all the DAQ system,
simulating the LHC machine bunch crossing rate. The correct timing sequence, described in
detail in section 5.3.1, is realized using an FPGA (Field Programmable Gate Array) chip. An
additional degree of freedom in the delay has been recovered programming an FPGA section
so to obtain a coarse 25 ns delay. The delay value is adjusted on the Sequencer board through a
RS232 serial interface hosted on the same VME CPU running the DAQ software.
40 MHzClock
Trigin
JTAG
Calin
Reset
ClkpClkn
ClkpClkn
Trgp
Trgp
Trgn
Trgnsignals
Reset button
APV signals
FEDALTERA
MAX7160
Serin
Figure 5.3: The Sequencer board.
Fig. 5.3 shows a picture of the Sequencer and the scheme with the I/O signals and the
95
fundamental blocks.
The main components and signals of the Sequencer board are listed in the following:
• The clock is the 40 MHz system clock.
• The MAX7160 is the FPGA that generates the timing signals.
• The “Calin” and “Trigin” inputs are reserved to the trigger signal in case of internal
calibration or DAQ mode measurements respectively.
• The “Reset” input allows to perform a software reset of the APV6 chip in case of error
condition; the same functionality is exploited by a hardware button named “Reset button”.
• The “Clkp”,“Clkn”,“Trgp”,“Trgn” outputs are the LVDS signals that carry the clock and
trigger. A couple of LVDS transmitters, model DS90C031, drives these lines to the Inter-
face card and to the FED ADC using different cables.
• The “Serin” input receives from the serial interface RS232 the delay parameters to be
used by FPGA.
The FPGA is programmable through the “JTAG” connector. In Appendix C is reported the
complete layout of the custom Sequencer card, entirely designed by the CMS Florence group.
5.3.1 The timing circuit
The timing circuit is built using an FPGA Model MAX7160 manufactured by ALTERA Cor-
poration. This EEPROM contains 3200 programmable logic gates and provides a flexible way
to realize the timing of the signals which drive the APV6 chip and the ADC. The whole set of
signals in input and output from the FPGA is summarized in Fig. 5.4. The input signals have
been translated to TTL level to match the device requirements. The “Serin” line carries the
data to program the delay, the “Clock” line the master clock and the “Reset” line the request
for APV6 reset. Furthermore there are two inputs dedicated to the DAQ and the calibration
triggers. The output lines are reserved for the clocks (“Clockapv” and “Clockfed”) and final
triggers (“Trigapv” and “Trigfed”), properly synchronized.
As we have seen in section 4.1.3 the internal calibration pulse is generated when the APV6
chip receives a signal lasting two clock cycles, i.e. 50 ns, on the trigger line. In this case a
96
clock 40MHz
Res
et
MAX7160
Calin
TrigapvTrigfed
ClockapvClockfedTrigin
Serin
Figure 5.4: Block scheme of the FPGA MAX7160.
charge spike is generated at the pre-amplifier input of every channel after a time selected with
the CSKW chip register. The 50 ns signal is internally generated by the FPGA using a D-type
Flip-Flop chain. Three Flip-Flops sequentially connected as shown in Fig. 5.5(a), with the
Delay (D) input connected to the power line Vss=5 V, the “Calin” signal to the clock input of
the first gate and the master “Clock” to the remaining two, make the calibration pulse.
A similar chain is used for the reset signal, lasting 75 ns or more; in this case we have used
4 D-type FLIP-FLOPs (see Fig. 5.5(b)).
The output pulses “Calout” and “Resout” are sent to the “trigapv” line via an OR gate (see
Fig. 5.6) and correspond respectively to the sequence recognized by the APV6 as “Calibration
request” and “Reset”.
The “Trigger” signal circuit, sketched in Fig. 5.6, delays the trigger with 25 ns steps and
sends it on the “trigapv” line. This circuit works both in DAQ and calibration mode. Its main
component is a 9-bit asynchronous counter (“LPM-counter”) built with a J-K type Flip-Flop
chain. The number of 25 ns clock cycles corresponding to the time delay is loaded to the
counter when the “aload” input is high and it is determined by the RS232 serial data stream
content decoded by another section of the FPGA. The counter starts when the input “count-en”
is enabled by a “Trigin” or “Calin” signal. After the programmed delay, lasting a time referred
to as D2 in the following, a combination of two exits of the counter makes a 25 ns trigger pulse
on the “trigapv” line. The same two exits (“qout0” and “qout8”) allow to reset the FLIP-FLOP
state and the counter through the “aclear” input.
The logic circuit implemented on the FPGA has been designed and tested with the MAX-
97
clear
D Q
clear
D Q
clear
D Q
clear
D Qck40ck40
Vss
(b)ck40Reset
Resout
clear
D Q
clear
D Q
clear
D Qck40ck40
Vss
Calin
Calout
(a)
Figure 5.5: The internal calibration and reset FPGA section.
clear
D Q
clear
D Q
clear
D Q
clear
D Q
clear
D Q
clear
D Q
LPM-Counter
qout0
qout8
TrigapvResout
Calout
count-en
qout[0..8]ac
lear
aloa
d
qout8
qout0
ck40 ck40 ck40
VssTrigin
Calout
Vss
ck40
Figure 5.6: Trigger delay section and trigger final stage of the timing circuit.
98
PLUS II software, distributed by ALTERA Corporation. The same PC running the software
allows to program the FPGA on the fly through a parallel port directly on the Sequencer socket,
avoiding a dangerous and time consuming extraction of the chip . The FPGA MAX7160 has
been chosen due to its sufficient number of macrocells (160, each containing a programmable
register, a FLIP-FLOP and several elementary logic gates) and its small maximum transit time,
certified within 7 ns by the manufacturer.
In Fig. 5.7 the sequence of a calibration request followed by a trigger pulse, generated by
the MAX7160, is shown. Great attention has been paid to adjust their relative phase with the
clock phase, so that the clock rising edge always finds the trigger line in a well defined logical
state.
Figure 5.7: Typical calibration request pulse (lasting 50 ns) followed by a trigger (black trace)and 40 MHz clock signal (grey trace) acquired with a digital oscilloscope on the Interface card.In this case D2 has been fixed to 100 ns.
5.4 The timing sequence
The APV6 pipeline is an essential feature of the CMS Silicon Tracker electronics since al-
lows to sample continuously the signal under investigation, in our case the charge collected on
every channel, and to retrieve the useful information only when a first level trigger signal is
received. To correctly readout the pipeline a deep knowledge of the timing sequence involved
99
in the trigger pulse distribution is required. The main parameter related to this problem is the
elapsed time between the physical trigger pulse, connected to the particle crossing time, and the
front-end triggerarrival time at the APV6 chip input. In internal calibration mode the physical
trigger is replaced by the calibration pulse. In the following sections the timing sequence of the
calibration and DAQ mode will be reviewed.
5.4.1 Internal Calibration Mode
The entire sequence is started by the “Calin” TTL input on the Sequencer. The MAX7160 cir-
cuit produces, as described in the previous section, a 50 ns pulse, corresponding to a calibration
request, and, after a time delay D2, a second 25 ns pulse that plays the role of trigger. The delay
D2 can be adjusted at 25 ns steps.
The APV6 chip has an internal calibration chain made of T-type Flip-Flops. When the
chip receives a calibration request signal, a clock pulse is sent to the Flip-Flop chain after a
programmable delay CSKW (see section 4.1.2) The consequent transition between the logic
states “1” and “0” of the Flip-Flop output releases a known charge to be injected in the pre-
amplifier input capacitances.
The timing diagram of this process is sketched in Fig. 5.8 and is described in the following.
A trigger pulse enters the “Calin” Sequencer input (1) and a 50 ns calibration request signal
reaches the APV6 chip. The released charge produces, at the shaper output, a signal that reaches
its maximum after a time T1 from the falling edge of the 50 ns calibration request (3). T1
depends on the delay adjustable at 3 ns steps through the CSKW APV6 internal register and
on the time the output signal needs to reach its maximum (typically 50 ns). The output signal
is continuously sampled in correspondence of the rising edge of the 40 MHz clock (4), and is
stored in the pipeline. Finally, after a time D2 from the rising edge of the calibration request, the
trigger pulse reaches the APV6 (2) and the pipeline cell addressed by the value of LATENCY
register (corresponding to the time Tlat in Fig. 5.8) is read out (3). It is worth noticing that the
CSKW register allows to perform a fine scan around the signal maximum trimming the proper
time sequence.
The delay related to the APV6 chip internal LATENCY register and the delay D2 provided
by the Sequencer carry out the same function and are completely inter-exchangeable.
100
1
2
4
3
D2
50ns
6ns
T1
25ns
latT
Figure 5.8: The Internal calibration timing sequence. The dotted curve in (3) shows a calibrationpulse optimized, by means of the CSKW register, so to have its maximum in correspondence ofthe clock rising edge. ( Time is not in scale).
5.4.2 DAQ mode
In data acquisition mode the trigger is generated by a particle crossing a plastic scintillator or
by a pulse generator in case the system is used as laser test station (see chapter 6). In Fig. 5.9
the timing diagram is shown, together with the time values measured with our laboratory setup
used to better explain the event sequence.
The photomultiplier signal (2) is processed by a constant fraction discriminator that gener-
ates an output signal lasting 50 µs (3). Since it is very dangerous to rely only on the 25 ns step
timing provided by the LATENCY register or by the Sequencer delay to correctly sample the
output analogue signal from the APV6 on its maximum, it is necessary to add in this point a
further delay stage. A pulse generator, model HP8131A, has been used to generate a 6 ns pulse
delayed in time D1 with respect to the discriminator output (plus 120 ns of cables and internal
delays), at 1 ns steps (4). From this point onward the trigger chain is identical to the one de-
scribed for the internal calibration acquisition mode. In particular a trigger signal is generated
by the MAX7160 after a delay D2 with respect to the pulse signal (5) and reaches the APV6
with a further delay due to cables (6).
101
µs
1
2
3
4
5
6
50
6ns
25ns10ns
Latency
5ns
10ns 120ns+D1
D225ns
Figure 5.9: The DAQ timing sequence. (Time is not in scale). The sequence starts when aparticle crosses the scintillator (1). The photomultiplier signal (2) undergoes some electronicsprocessing and reaches the APV6 as a 25 ns trigger signal (6) after a delay due to the cables andto user adjustable registers (D1 and D2).
The time difference between the particle crossing the scintillator (1) and the trigger arrival
to the APV6 (6) has been measured with a digital oscilloscope in order to obtain a coarse
evaluation of the LATENCY register value. The final relationship between the LATENCY
register, the D1 and D2 times and the trigger delay (in Fig. 5.9 145 ns are introduced by the
cables and other electronic components), is given by:
LATENCY = Mod25(145ns + D1 + D2) (5.1)
It should be noted that only triggers arriving within a ±3ns time window around the clock
rising edge are used as APV6 final triggers (see Fig. 5.10). In fact the FLIP-FLOP chain that
starts the counter, shown in Fig. 5.6, is activated only in this case. This is unavoidable since we
are using a synchronous system (APV6+40 MHz clock) designed to be used in a synchronous
environment (LHC+CMS) on an asynchronous test bench (β source).
102
Non accepted triggers
Accepted triggers
clock 40Mhz
6ns
6ns6ns
25ns
Figure 5.10: Relationship between accepted triggers and rising clock edge.
In this way only a fraction of the particles generating the triggers are processed, assuring
that their signals are properly sampled. Our setup, in the experimental conditions described
above and with the 90Sr source filtered through a window of approx 2 cm2, allows a final data
acquisition rate of about 50 Hz.
5.5 The Data Storage
The analogue output from each APV6 chip is sent to the TRICARD where it receives a first
amplification and an offset adjustment to match the levels and fully exploit the dynamic range
of the ADC circuit.
The ADC board is a PCI mezzanine card (PMC) inserted in the PMC slot of the RIO8062
CPU that controls the acquisition. The characteristics of this card, referred to as FED (Front
End Driver) in the following, will be described more in detail in the next section.
5.5.1 The FED ADC
The FED is a prototype of the ADC card that will be used in the experiment [6]. It contains 8
ADC channels and a Xilinx array that is programmable through the PCI connector and allows to
perform some preliminary operations on the acquired data. In particular it is possible to decide
the number of sample acquired for every trigger, the number of ADC channels to be used and
the sampling point with respect to the clock phase.
The FED ADC is a 9 bit converter running at 40 MHz. Since the output rate of the APV6
103
multiplexer is only 20 MHz each channel is sampled and stored twice during the acquisition. In
CMS two APV6 chips will be further multiplexed thus obtaining a 40 MHz analogue output to
be digitized.
The sampling clock is provided to the FED by the Sequencer together with the trigger signal
that is necessary to start the acquisition. Due to the fact that there is a fixed delay and a jitter,
of the order of few µs but not predictable, between the trigger arrival time to the APV6 and
the output of the analogue frame, it is necessary to acquire a number of samples larger than
the 140 strictly necessary to get information about the 128 channels and the header. In our
setup we acquired up to 1024 samples for every trigger, covering a time window of 25.6 µs.
The string of conversions performed on the APV6 analogue output, connected to the FED, in
correspondence of a trigger is considered as a single event. Data are written in storage devices
as ASCII files containing the ADC values for all the APV6 chips in the readout chain, together
with some global information related to the software and type of acquisition performed, and are
immediately available for the offline analysis.
The maximum data acquisition rate in our system is of the order of 100 Hz for a single APV,
completely limited by the data storage rate on disk.
104
Chapter 6
The laser test station
To be ready for the production phase of the final CMS silicon strip detectors a set of procedures
have to be defined in order to check the quality of the modules. One of the key steps that
have to be followed is the implementation of a flexible and affordable system that allows the
full functionality test of a complete detector, with respect both to the sensors and electronics
quality. In the context of this thesis a laser test station has been built, based on the same DAQ
system used for the MIPs measurements (see chapter 5).
A laser beam is the most suitable solution with respect to compactness, costs and measure-
ment rapidity to fully test all components of a silicon detector. Furthermore the readout signal
is easily detectable since the illuminated area is well known and stable; with this apparatus the
signal can be observed on-line even with an oscilloscope.
The laser radiation must excite electrons from the valence to the conduction band in order
to release charge in the detector, but at the same time must cross completely the detector to
simulate the passage of a particle through the entire silicon thickness. Since the crystalline
silicon wafers become transparent in the near infrared a laser radiation at λ=1064 nm can be
used [23] (see Fig. 6.1).
At this wavelength the single photon energy is 1.16 eV, compared with 1.12 eV band energy
gap. In a semiconductor with an energy gap ∆E between the valence band and the conduction
band the absorption of one photon of energy hν ≥ ∆E causes the creation of an electron-hole
pair. The absorption coefficient scales as√hν −∆E. The sensitivity of the sensors is still about
0.1 A/W and in literature we found a total transmission rate of 71% for 300 µm thick sensors
with oxide coatings on both surfaces at room temperature [49]. With these characteristics the
105
Figure 6.1: Absorption coefficients for pure Ge, Si and GaAs as a function of the photon energy.Figure taken from Ref. [23].
laser beam is able to uniformly produce electron-hole pairs along its path in a silicon detector.
One of the main tasks of the job developed in this thesis has been the design of a driver
for the pulsed laser diode, the choice of an optical focusing system and the integration of two
remote controlled translation stages in the system.
6.1 The laser source
The choice of the laser source is mainly dictated by compactness and operational easiness.
The laser diodes have such characteristics and in addition are low power devices. Progresses
obtained in the last decade in semiconductor engineering, and consequent enlarged spectral
emission ranges, have made laser diodes the best candidates for applications in spectroscopy
field as well as in telecommunication, office devices, CD player etc, favouring their diffusion.
The device is a broad area high power pulsed laser operating at 1064 nm wavelength, model
C86119E manufactured by EG&G [50]. It employs MOCVD grown strained InGaAs/AlGaAs
layers offering high efficiency, low threshold and continuous wavelength tuning at approxi-
mately 0.3 nm/C. This last feature is not fundamental in applications that require only the
production of donor-acceptor pair in a silicon wafer.
106
The basic principle of semiconductor lasers may be summarized as follows. When an elec-
tric current is sent in the forward direction through a p-n semiconductor diode, the electrons and
holes can recombine within the p-n junction and may emit the recombination energy in the form
of electromagnetic radiation. The wavelength is determined by the energy difference between
the energy levels of electrons and holes, which is essentially defined by the band gap. The
spectral range of spontaneous emission can therefore be varied within wide limits by the proper
selection of the semiconductor material and its composition in binary compounds. Above a cer-
tain threshold current the radiation field at the junction becomes sufficiently intense to make the
induced-emission rate exceed the spontaneous or radiationless recombination processes. The
radiation can be amplified by multiple reflections from the plane end faces (orthogonal to the
junction plane and optically treated) of semiconducting medium and may become strong enough
that induced emission occurs at the p-n junction before other relaxation processes deactivate the
population inversion. The wavelengths of the laser radiation are mostly determined by the spec-
tral gain profile and by the eigenresonances of the laser resonator. Usually the cavity face with
the larger transmission coefficient is devoted to radiation output while the light exiting the other
face is collimated on a monitor photodiode that allows the check of device functionality.
A rugged 14 pin, flanged, dual-in-line package encloses the laser diode, the silicon monitor
photodiode, the thermoelectric cooler and the thermistor used for the test station. The laser
output face is optically coupled, internally to the package, to a multimode 100 µm fiber.
The laser must be operated by pulsing it in the forward bias direction. To this end a custom
driver circuit has been designed and will be described in section 6.2. The maximum rated pulse
duration (200 ns) and duty factor (0.1 %) must never exceeded. If the specified pulse duration
or duty cycle is exceeded, the lasing action may be quenched because of the heat generated in
the junction and the device may be eventually destroyed. However the repetition rate may be
increased if the pulse duration is reduced provided the maximum duty factor is not exceeded.
The peak forward current is 4 A and the peak reverse voltage is 2 V, providing a 100 mW peak
output power from the fiber.
For our application the spectral purity of the emission is not relevant so we haven’t stabilized
the device temperature. Nevertheless the calibration of the thermistor value vs. temperature has
been measured in a climatic chamber since the data sheet provided by the manufacturer doesn’t
107
report this relation. The results are shown in Fig. 6.2, where the uncertainty on the resistance is
about ±0.5 kΩ due to the thermal drift.
Temperature (oC)
The
rmis
tor
(KΩ
)
0
2.5
5
7.5
10
12.5
15
17.5
20
22.5
25
0 10 20 30 40 50 60
Figure 6.2: Thermistor calibration.
6.2 The laser driver
The laser driver has been designed to obtain a sequence of radiation pulses with the desired
duration and intensity. If the amplitude of the pulse is not a problem, more attention requires the
short duration of the laser pulse that must be of the order of few nanoseconds to be comparable
to the collection time of the charge released by a relativistic particle crossing the detector. In
addition, the circuit must maintain the laser operational conditions within the maximum ratings
provided by the manufacturer even in case of malfunctioning of some of its components.
The laser diode emits a radiation pulse in correspondence of a trigger TTL signal (T1 in
Fig. 6.3), which acts as trigger also for the DAQ system described in chapter 5. The circuit
schematic is reported in Fig. 6.3 with a set of electrical component values used during the tests.
The main block of the driver circuit is a differential pair made with two high bandwidth
transistor (model NPN-BLF80). The differential pair is switched by a fast trigger signal named
T2 derived by T1. The diode is placed on a branch of the pair and is kept in forward conduction
108
T1
A A
B B
C C
D D
E E
44
33
22
11
Vdd
= -2
V
Vss
= -5
V
Vcc
= 5
V
Vdd
Vdd
Vdd
Vss
Vss
Vcc
Sign
al
U1A
7414
12
U1A
7414
12
U2A
1012
4
12
C2
270
pF
U1A
7414
12
D2
DIO
DE
R3
100
U2A
1012
4
12
R2
51
D1
DIO
DE
C3
1.3
nF
U2A
1012
4
12
Q2
NPN
-BLT
80
Q3
NPN
-BLT
80
U1A
7414
12
R6
5.6
U1A
7414
12
R1
52
R7
5.6
R8
47
R9
47
C4
1 nF
Q1
NPN
-BLT
80R
4
50
R5
1K R11
1K5
R12
33
C4
100
nF
R13
1.4
R15
27
D3
DIO
DE
D4
lase
r
C5
100
nF
U1A
7414
12
R14 27
T3
T2
T1
Figure 6.3: Laser driver schematic.
when the corresponding transistor base is low. A pair of resistor networks correctly bias the
transistor bases while a normal diode on the branch opposite to the laser one symmetrizes the
109
response of the device. The current flowing through the laser diode is regulated by a resistor
switch (R13) after the collector of the differential pair output transistor (Q1). In order to reduce
the device working time this transistor conducts only in correspondence of a trigger signal T3,
derived from T1 and synchronous to T2 but longer. Furthermore a RC filter allows the current
to flow through the resistor switch only when the current flows through the laser (D4).
The trigger signal (T1) from a local oscillator reaches the differential pair after a series of
logic gates (NOT gates) and a CR shaping stage which allows to adjust the duration of the TTL
pulse, properly selecting the values of a capacitor (C2) and a resistor (R2). A diode suppresses
the undesired signal polarity. The trigger T1 is sent in a second branch and is shaped with a
larger time constant in order to obtain the signal T3. The circuit is biased between +5V (Vcc)
and -5V (Vss), with a further bias reference Vdd=-2V.
The design has been tested with a SPICE simulation and the results are in agreement with the
pulses obtained experimentally (see Fig. 6.4). The regulation of the collector resistance (R13)
allows a dynamic arrangement of the current flowing into the diode and of the output power,
while the RC (R2-C2) time constant responsible of the pulse signal duration can be adjusted
to obtain pulse width down to 20 ns with good shapes. The voltage drop, corresponding to
the current pulse in the laser, measured with a digital oscilloscope connected to the laser diode
cathode is shown in Fig 6.4.
Figure 6.4: Laser pulse duration for R=51 Ω and C=300 pF.
110
6.3 The optical and the positioning systems
The laser beam is strongly divergent at the multimode fiber output. A measurement performed
with a photodiode array over several distances from the fiber end leads to a divergence of about
25 degrees. In order to obtain a smaller spot size, to avoid keeping the fiber in a dangerous way
close to the detector surface, it is necessary to use an optical collimation system between the
fiber and the sensor. Furthermore the system must be compact and must be positioned vertically
to scan the detector surface that is placed on a horizontal plane over a pair of translation axis.
A system based on a microscope objective as a collimator and a convergent lens, coaxial to
the fiber, has been chosen (Fig. 6.5). The fiber, kept safe in its final stage inside a fiber holder, is
Microscope objective
Laser beam
Fiber
Fiber holder
Fiber coupler
LensSilicon detector
(a) (b)
Figure 6.5: Laser beam collimation system.
housed inside a fiber coupler (model Newport-M-F1015). On the same optical axis an objective
lens is placed in order to collimate the strongly divergent beam coming out from the fiber. The
fiber coupler is devoted to easily adjust the distances between the fiber and the objective and
their relative position by means of a set of three dimension micrometric positioning system.
The collimated beam is directed on a convergent lens and is focused on the detector surface.
The lens is placed on a micrometric translation stage (model Newport-UMR8.25) that allows
to change the distance between the detector and the lens, so to have an additional degree of
111
freedom in the overall geometry. The fiber coupler and the translation stage are fixed on an
aluminium bridge standing over the detector.
The detector is placed on a pair of orthogonally coupled translation axes in order to make the
system able to perform a scan of the entire sensor surface. The longer axis is fixed on the laser
station reference surface and holds, orthogonally to its displacement direction a second shorter
axis that carries the detector. The two axes, manufactured by Newport (Model MT70 300 and
M UTM150 PP 1HL respectively), are step motorized with a position resolution of 1µm. They
are driven by a controller model MM2500 that allows manual and software displacement. A
software control code, based on a Macintosh/OS, has been developed to move the axes via
a parallel IEEE-488 interface. Position and translation velocity are set by the user so that the
detector scan can be localized and optimized. A picture of the whole system is shown in Fig. 6.6
Translation axis
Laser Driver
Optical system
Detector
TRIcard
Laser source
Figure 6.6: The laser test station.
112
6.4 System performances
The laser system setup structure is summarized in Fig. 6.7. Several configuration of the optical
system have been tested in order to obtain a spot size as small as possible. It has been found
that, due to the fiber coupling, the laser beam doesn’t behave as a true gaussian beam and it is
more difficult to focus.
Translation axis
LASERPulse generator
Analog signal( to VME )
Pc/Mac
GPIB
Axis Driver
DetectorSequencer
(TRIGGER)
Figure 6.7: The laser test station structure.
The spot size has been measured with a photodiode linear array made of 128 pixels each
with a 25 µm width. The array scanning is carried out at a multiplexer frequency of 2.725 KHz,
corresponding to 2.75 µs/pixel. Beam waist measurements performed with the photodiode array
agree with data collected with the silicon detectors. The narrower beam waist is obtained with a
19 mm focal-length plano convex lens and a 40 mm focal-length 0.10 N.A. objective lens, and
corresponds to a FWHM about 80 µm, as shown in Fig. 6.8.
The second adjustment concerns the laser intensity that is regulated by changing the resistor
R13 in Fig. 6.3. The high intensity configuration allows to detect easily the laser beam position,
mainly for debugging purposes, since the signal on the lightened strips is visible directly on
the oscilloscope, while a lower intensity configuration is used to emulate MIPs charge release.
The laser intensity calibration has been performed by changing the R13 resistor value and by
measuring the charge released. In Fig. 6.9 the measured charge (in peak mode) as a function of
the R13 resistor value is shown.
The setup allows to perform a scanning of the detector response over all the strips in a couple
113
Figure 6.8: Beam waist measurement performed with a photodiode array. (1µs=9.1 µm);FWHM ∼ 80µm.
R13 resistor (Ω)
Cha
rge
(AD
C c
ount
s)
10
20
30
40
50
60
70
5 6 7 8 9 10 11 12
Figure 6.9: Laser intensity calibration. The charge measured in peak mode vs. the R13 resistorvalue is shown.
of minutes if we take into account a 50 Hz event acquisition rate and we want to collect enough
statistic on every channel (512 strips in total). This procedure has revealed itself to be very
effective in finding unbonded or broken strips/channels. Furthermore broken bondings can be
easily traced to the crystal-hybrid or the crystal-crystal interface. A laser scan, together with
114
a pedestal and noise measurement, can give all major information about every single detector
channel and is a good candidate to setup a module test procedure. The uniformity response
of all the detector channels is shown in Fig. 6.10 (a); previously unbonded strips are clearly
visible in correspondence where no clusters have been reconstructed . The cluster charge (laser
induced) distribution is shown in Fig. 6.10(b), where the fit to a gaussian curve is superimposed.
Strip number
Rec
onst
ruct
ed c
lust
ers
0
5
10
15
20
25
30
0 100 200 300 400 500
(a)
67.31 / 60Constant 40.33Mean 27.84Sigma 4.805
Entries
AD
C c
ount
s
0
10
20
30
40
50
60
5 10 15 20 25 30 35 40 45 50 55
ADC counts
Ent
ries
ADC counts
(b)
Figure 6.10: Detector channel response to a laser beam continuously scanning the detectorsurface (a) and cluster charge distribution for a MIP-like laser generated charge (peak mode)(b).
The mean value of the distribution in Fig. 6.10(b) is close to the most probable signal re-
leased by a MIP.
115
Chapter 7
Performances of the detector prototypes
In this chapter the performances of the detector prototypes described in section 2.5 and 3.3 are
summarized. This study is important since the size, strip design and front-end electronics of
these detectors are very similar to the ones that will be installed in the inner forward part of the
CMS silicon tracker.
All the detectors have been carefully characterized in laboratory measuring the electrical
parameters, the noise and MIPs response. Later they were tested at CERN with a muon beam
in a more complex experimental setup. The performances in terms of signal to noise ratio have
been compared with the values estimated from the detectors electrical parameters. The complete
set of modules available, which covers several of the main possible choice for the substrate type,
and the presence of irradiated detectors has allowed a wide analysis of the performances of this
kind of prototypes and has showed that satisfactory values of S/N ratio have been obtained even
in the worse experimental conditions. Finally, the first results on the MIP response of a 500 µm
thick detector bonded to the APV6 chip are presented.
All these measurements, which are the main object of the work performed during the thesis
and which I have done personally, have been submitted for publication [8] [53] [54].
7.1 Off-line analysis
The information related to the analogue output frame of each APV6 chip are converted by the 40
MHz CMS ADC (FED [6]) and stored on disk for subsequent off-line analysis. The data taking
is done on individual runs, each one consisting of approximately 1000 events for the laboratory
measurements and 10000 events for the beam test acquisition. For each front-end chip an event
117
contains a redundant number of consecutive digitized samples of the analogue output and is
defined by an external trigger generated by scintillation counters. Only a fraction of the data
stored in every events consists of the 12 bit logical header and the 128 samples related to the
detector channels read out by the APV6 chip. The first operation performed by the analysis
code is to extract the 128 voltage levels, converted by the ADC, by means of a robust threshold
recognition algorithm. Since the APV6, due to the multiplexer design, presents at its output the
channel voltage levels in a non sequential way with respect to the strip number on the detector
a reorder is needed. The 128 ADC values thus obtained, referred to as ADCni in the following,
with i channel number and n event number, are the starting point for every subsequent off-line
analysis. The signal Sni of each channel and its noise σi are calculated from the raw data after
a subtraction of the common mode and pedestals. Only at this stage the presence of a charge
signal (cluster) due to particle crossing is searched. Since we are still in a debugging phase for
both the detectors and software no zero suppression has been performed at DAQ level. This will
be a fundamental component of the final CMS DAQ software.
The pedestal of each channel PEDi is defined as the average strip output level when no
signal (charge or calibration) is present. Its calculation is performed by an iterative process. A
first raw estimation PEDrawi is given by the average on N events of the ADC counts for each
individual channel:
PEDrawi =
∑Nn=1 ADCn
i
N(7.1)
A correction has to be applied, for each event, to the raw data in order to take into account
the shift of the baseline for all the strips belonging to the same electronic section of the APV6
chip. This effect is known as common mode noise and it is event dependent since it is mainly
due to electronic noise pickup at the pre-amplifier inputs. The common mode noise CMN n is
calculated, event by event, as the average of the pulse heights for contiguous strips removing
the channels that may carry a particle signal or that are too noisy:
CMNn =1
Nch
Nch∑i=1
(ADCni − PEDraw
i ) (7.2)
where Nch is the number of channels that are considered to be affected by the same common
noise shift. In the case of the APV6 chip, since the analogue inputs are grouped in 4 blocks
118
of 32 channels, we calculated four different common noise values on a 32 channel basis. The
common mode algorithm has revealed to be robust and all the common mode distributions for
our detectors are well fitted with a gaussian curve.
The final value of the pedestals is then recalculated after the common mode noise subtraction
as shown in Eq. 7.3. From each channel the corresponding common noise is subtracted and the
pedestal for the ith channel is recalculated:
PEDi =1
N
N∑n=1
(ADCni − CMNn) (7.3)
The reordered pedestals profile, common mode subtracted, is shown in Fig. 7.1 for a set of four
APV6 chips housed on the same hybrid; the 32 strips pattern is clearly visible. The common
Channels
PE
Di (
AD
C c
ount
s)
APV6 chip 12
Channels
PE
Di (
AD
C c
ount
s)
APV6 chip 14
Channels
PE
Di (
AD
C c
ount
s)
APV6 chip 1a
Channels
PE
Di (
AD
C c
ount
s)
APV6 chip 1c
0
50
100
150
200
250
300
50 1000
50
100
150
200
250
50 100
0
50
100
150
200
250
50 1000
50
100
150
200
250
300
50 100
Figure 7.1: Pedestal profiles of the four chips housed on the hybrid after common mode sub-traction. The APV6 chips are bonded to the detector.
mode noise is calculated a last time using the new value of the pedestals obtaining the final
119
CMNn. In Fig. 7.2 the typical distributions for the common mode noise corresponding to the
4 groups of 32 contiguous strips are shown. The signal Sni present during the event n on the ith
common noise chips =1
IDEntriesMeanRMS
91 4000 -.1348E-02
4.878 92.68 / 58
Constant 199.9Mean -.6556E-01Sigma 4.681
common noise chips =2
IDEntriesMeanRMS
92 4000 -.4798E-02
5.136 86.20 / 67
Constant 195.7Mean -.6366E-01Sigma 4.786
common noise chips =3
IDEntriesMeanRMS
93 4000 -.1498E-02
5.061 116.9 / 65
Constant 199.8Mean -.2029E-01Sigma 4.652
common noise chips =4
IDEntriesMeanRMS
94 4000 -.5980E-03
4.870 69.54 / 59
Constant 201.0Mean -.4564E-01Sigma 4.682
0
50
100
150
200
250
-20 0 200
25
50
75
100
125
150
175
200
225
-20 0 20
0
50
100
150
200
250
-20 0 200
25
50
75
100
125
150
175
200
225
-20 0 20CMN (ADC counts)
CMN (ADC counts) CMN (ADC counts)
Ent
ries
Ent
ries
chip 1C
chip 12
chip 1A
chip 14
CMN (ADC counts)
Figure 7.2: Distribution of the common mode noise of 4 APV6 chips housed on the samehybrid. The APV6 chips are bonded to a complete detector.
strip is calculated subtracting from the sampled ADC value ADCni the strip pedestal and the
common mode noise effect:
Sni = ADCn
i − PEDi − CMNn (7.4)
The statistical fluctuation of the signal around its pedestal value defines the noise σi associated
with a channel:
σ2i =
1
N − 1
(N∑
n=1
(ADCni − CMNn)2 −N · PED2
i
)(7.5)
The noise is calculated on the same event sample in the case of beam test runs or on a
separate file acquired in absence of particle beam during laboratory measurement. In both cases
120
the whole process is performed with several iterations; in a first step all the strips and events are
included so to have a preliminary estimation of the strip signal and noise. These quantities are
later used to identify the strips which have collected some charge and that have to be removed
from the pedestal, common and strip noise calculation. Furthermore the analysis code identifies
and tags the detector channels that show an anomalous behaviour with respect to noise. More
in detail this happens when a strip is not connected or it is noisy. The latter case, if not related
to the front-end electronics, is due to an increase of the leakage current, of the capacitive load
subsequent to a short between adjacent strips or to a pinhole in the coupling oxide layer. The
overall effect is the removal from the off-line analysis of the strips with a noise exceeding a
given range around the detector noise mean value. In Fig. 7.3 the noise profile of 128 strips (a)
and their distribution (b) is shown for a non irradiated module.
profilo sigma (cmn sub) chips =1
IDEntries
21 128
0
0.5
1
1.5
2
2.5
3
3.5
20 40 60 80 100 120
1
2
3
Noi
se (
AD
C c
ount
s)
0 20 40 60 100 12080Strips
(a)
sigma (cmn sub) chip =1
IDEntriesMeanRMS
31 128
1.545 .8381E-01
19.69 / 9Constant 28.55Mean 1.536Sigma .6004E-01
Noise (ADC counts)
Ent
ries
0
5
10
15
20
25
30
35
40
0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2
Entries
0
40
1 2
10
20
30
Peak mode
0Noise (ADC counts)
Ent
ries
(b)
Figure 7.3: Measured noise profile for a single chip of a non irradiated detector operating in peakmode (a). Distribution of the noise for the same APV6 chip. The fit to a gaussian distribution issuperimposed (b).
The noise value that will be used to estimate the detector performances in the following is
obtained as the most probable value of the gaussian fitted to the noise distribution.
121
7.1.1 Cluster and total charge reconstruction
The experimental evidence of the passage of a particle through the detector is obtained searching
for a group of contiguous strips (cluster) with a total signal value compatible with the charge
released by a MIP. The cluster finding algorithm is based on the previously defined quantities
“strip signal” and “strip noise”. Three parameters allow to select only the strips and clusters
with a signal which exceeds given thresholds. For each event the analysis code loops over the
strips to find the ones with a signal to noise ratio greater than a given threshold Ts:
Sni > Ts × σi (7.6)
These strips assume the role of cluster seeds. Then the neighbour strips, up to three for each
side of the seed, are checked to see if they have a bigger S/N and in that case they are promoted
to cluster seed. When this search is terminated the seed is established and the nc neighbour
strips satisfying the condition:
Sni > Tn × σi (7.7)
are added to the seed to form a cluster. The maximum number of strips allowed in a cluster is
limited by the fact that in our tests particles cross the detector orthogonally and we have always
measured a cluster multiplicity below 7 strips. On the other hand, setting a loose limit to the
cluster size we risk to merge into a single cluster the strips affected by the close passage of two
different particles. Anyway the lateral strips search on each side of the seed ends when the first
strip not satisfying condition 7.7 is found.
Finally, the total cluster charge is computed as the sum of the signal over the accepted strips:
Sncluster =
∑nc
Si (7.8)
The last threshold is given on the total charge; the cluster is retained if:
Sncluster > Tc × σi(seed) (7.9)
The threshold values used in this analysis are:
Ts = 4 Tn = 2 Tc = 5 (7.10)
and are a compromise between the need of maintaining a good reconstruction efficiency and the
risk of introducing noise cluster (ghost hits).
122
7.2 The 300 µµµm detectors
All the modules built in the framework of this thesis have been carefully characterized with
respect to S/N ratio performances in laboratory with MIPs extracted from a β source and in
three beam test sessions at CERN during summer 2000. I have taken care of all the measurement
done in laboratory and I have participated to the beam test activities together with other groups
of the CMS Tracker collaboration.
The APV6 registers have been set out, during all the measurements, at their nominal value;
in particular the VSHA register is adjusted to 2 V (see section 4.2).
7.2.1 The βββ source measurements
The detectors have been exposed to MIPs in the experimental setup described in 5.1. For the
whole set of detectors we have computed the cluster charge distributions and fitted them to a
Moyal function (see Eq. 2.14 for the meaning of the plot parameters) approximating the energy
loss distribution in a thin silicon layer. The measurements have been performed at -10C for the
irradiated devices and at room temperature for the non irradiated ones.
The cluster charge distribution with the APV6 chip operating in peak mode is shown in
Fig. 7.4; the result of the fit is superimposed. The cluster charge value to compute the S/N
ratio will be referred to as the peak value of the fitted function to such distribution. The cluster
charge measurements have shown a good reproducibility over long term periods (about 5% for
the cluster signal).
The strip noise profile for all the detectors is quite flat and follows a gaussian distribution
around the mean value. For this reason we have always used the strip noise mean value of the
fitted gaussian to calculate the signal to noise ratio without applying a weighted sum to the strip
noises involved in the cluster.
The S/N ratio
The main target of this analysis is to measure the signal to noise ratios for the two kinds of
substrates, both irradiated and non irradiated. The signal to noise ratio, as stated before, is
defined as the ratio of the most probable value of the “Moyal function” fit to the cluster signal
distribution to the most probable value of the gaussian fit to the strip noise distribution. The
123
EntriesMean
871 42.65
63.32 / 41P1 309.6P2 35.09P3 5.230
ADC counts
Ent
ries
0
10
20
30
40
50
60
70
80
90
0 20 40 60 80 100 120
(a)
EntriesMean
528 44.88
59.57 / 51P1 147.3P2 34.22P3 6.422
ADC countsE
ntri
es
0
5
10
15
20
25
30
35
40
45
0 20 40 60 80 100 120 140 160
(b)
Figure 7.4: Cluster charge distribution for the non irradiated (a) and irradiated (b) <111>detectors. The measurement are performed in peak mode at a bias voltage of 200 V (a) and 500V (b). See Eq. 2.14 for the fit parameter meaning.
results are summarized in Table 7.1. Due to the uncertainties related to the fit procedure and
mainly to the spread observed on measurement performed with the same APV6 and detector
conditions, the S/N values are affected by an uncertainty of the order of ±1mainly of systematic
origin. In particular we have observed a strong dependence on the detector temperature, which
needs to be stabilized. In the worst cases the irradiation procedure has reduced the signal to
noise ratio of about 15 % but even in deconvolution mode the S/N value is far above the limit
of 10. The tests performed have shown no experimental evidence, in the uncertainty limits
mentioned above, of a different behaviour between the <111> HR and the <100> LR devices,
once they are operated overdepleted. This is true for both non irradiated and irradiated detectors
after taking into account the different depletion voltages.
On all the devices we performed a voltage scan to study the effect of different bias voltages
on the charge collection mechanism. In Fig. 7.5 the results are shown for the non irradiated and
irradiated <111> HR detectors, in terms of signal to noise ratio, both in peak and deconvolution
124
mode.
Vbias (V)
S/N
Peak mode
Deconvolution mode
Non irradiated HR detector
0
2.5
5
7.5
10
12.5
15
17.5
20
22.5
0 50 100 150 200 250 300
(a)
Vbias (V)
S/N
Irradiated HR detector
Peak modeDeconvolution mode
8
10
12
14
16
18
100 150 200 250 300 350 400 450 500
(b)
Figure 7.5: Bias voltage scan for the non irradiated (a) and irradiated (b) high resistivity detec-tor. Measurements are performed both in peak and deconvolution mode with a β source.
The signal to noise increases with the applied voltage even after the silicon bulk is fully
depleted due to the increase in charge collection and a reduction in noise. A possible expla-
nation of the signal increase after the depletion voltage has been reached is that our LHC type
electronics suffers of a ballistic defect due to the non negligible charge signal formation time.
This effect is particularly evident when dealing with irradiated devices. The measurements per-
formed have shown that we can safely over deplete the irradiated devices but an efficient cooling
system is crucial to operate the detector. We can observe the increase, due to irradiation effects,
of the bias voltage that optimize the S/N ratio, as expected from the considerations presented in
chapter 3. From the S/N ratio versus bias voltage plot we determine the operating voltage for
the whole set of detectors.
A careful characterization has been performed on the modules to test their response to a
change of the main APV6 registers. In particular the cluster charge measurement as a function of
the trigger delay with different VSHA values has proved that the APV6 response doesn’t change
125
with an external load, as shown in Fig 7.6 for peak mode measurements. In this plot the APV6
output shape obtained in calibration mode with the hybrid not connected to the detector (open
circles) is superimposed to the shape obtained with β source MIPs on the complete module
(full triangles). The signals are normalized to the peak value and their shapes are completely
overlapped.
Delay (ns)
Sign
al (
AD
C c
ount
s)
VSHA = 0.56 V β source
VSHA = 2.0 V β source
VSHA = -0.20 V cal. mode
VSHA = 2.0 V cal. mode
Non irradiated detector0
50
100
150
200
250
0 20 40 60 80 100 120 140 160 180
Figure 7.6: Effect of the VSHA register in calibration mode (with the APV6 not bonded to thedetector) and with a β source. Data are normalized to the shape peak value.
The decrease of the VSHA value (in Volts) correspond to a larger shaper time constant (see
Fig. 7.6, full circles and open squares) so the charge collected increases; this is shown also in
Fig 7.7 for the non irradiated device.
Anyway the VSHA value that has been used during beam tests and laboratory performance
measurements is the one corresponding to the fastest response (VSHA=2 V).
7.2.2 The Beam Test measurements
The detectors have been exposed to a muon beam during two beam tests in the area called X5
at the CERN SPS during summer 2000. The particles had a momentum of about 100 GeV/c
and an intensity of 104 muons per spill (extraction cycle). The experimental setup used during
the beam tests is more complex than the one described in chapter 5.1 since it is based on a
126
Vsha (Volts)
Cha
rge
(AD
C c
ount
s)
non irradiated detector
β source
105
110
115
120
125
130
135
140
0 0.5 1 1.5 2
Figure 7.7: Effect of VSHA on the charge collected in peak mode with the β source.
DAQ system similar to the final one foreseen for the CMS tracker. In particular the trigger and
clock signals are delivered through a Front End Controller (FEC) and are processed by a Central
Control unit (CCU) before reaching the detectors [3]. The analogue APV6 output signals are
transmitted to the FED by means of analogue laser drivers and optical fibers [51]. Furthermore
this system has been successfully operated for the first time under LHC like beam conditions in
a 25 ns structured beam provided by the SPS accelerator complex [52].
All the detectors were installed in a temperature controlled box. The non irradiated devices
are operated at 10 C while the irradiated ones at -15C.
The muon beam was centred on the central region of the detector as shown in Fig. 7.8 where
the cluster seed is plotted for the two central APV6 chips only.
In order to reduce the noisy clusters we select only the events in which the clusters show
a strong position correlation with two other reference detectors (straight tracks). In Fig. 7.9
the cluster seed position is plotted for the <111> HR non irradiated detector and two other
detectors, showing a strong correlation.
From the track clusters we have measured the cluster charge distribution and the corre-
sponding noise. A typical charge distribution is showed in Fig. 7.10 for the < 111 > HR non
irradiated detector operating in peak (a) and deconvolution mode (b).
127
EntriesMean
2157 126.3
0
20
40
60
80
100
Ent
ries
Strip number
0 50 100 150 200 250 300
Rec
onst
ruct
ed c
lust
ers
Strip number
Figure 7.8: Beam profile measured during the beam test.
050
100150
200250
300
050
100150
200250
Stri
p nu
mbe
r (F
lore
nce)
Strip number (Module 11) Strip number (Module 10)
0
25
50
75
100
125
150
175
200
225
250
Figure 7.9: Cluster position correlation between 3 detectors.
128
90.54 / 67P1 262.3P2 46.13P3 6.675
0
10
20
30
40
50
60
70
80
0 20 40 60 80 100 120 140 160 180 20020 40 60
Non irradiated
80 100 120 140 160 180
20
40
60
80E
ntri
es
Charge (ADC counts)
S/N=20
(a)
112.9 / 72P1 273.9P2 38.77P3 7.350
0
10
20
30
40
50
60
70
80
90
0 20 40 60 80 100 120 140 160 180 2000 20
20
40
40
60Charge (ADC counts)
60
80
80
100 120 140 160 180
S/N=13
Non Irradiated
Ent
ries
(b)
Figure 7.10: Cluster charge distribution for the < 111 > HR detector measured in a muon beamin peak mode (a) and deconvolution mode (b).
The noise profile and the noise distribution measured for an APV6 of the same detector in
peak mode with the X5 beam test experimental setup are shown in Fig. 7.11.
The performances obtained in terms of signal to noise ratio agree with the values obtained
in laboratory with the β source (see Tab. 7.1). Furthermore, from the comparison of the APV6
digital header amplitudes (see chapter 4.1) acquired in the two different experimental setups,
that are closely related to the relative analogue amplification gain, we have confirmed that the
beam test and Florence setup charge and noise measurements are consistent.
The signal to noise ratio as a function of the bias voltage has been measured for all the four
detectors showing a good agreement with the data obtained in laboratory. The S/N versus bias
voltage for the non irradiated devices is shown in Fig. 7.12
7.2.3 Results summary
The performances obtained during the laboratory and beam test measurement have shown a
very good agreement in terms of S/N ratio. This makes us confident that our setup can perform
129
Channel
Noi
se(A
DC
cou
nts)
7.569 / 4Constant 43.47Mean 2.288Sigma .1406
ADC counts
Ent
ries
0
0.5
1
1.5
2
2.5
3
20 40 60 80 100 120
0
10
20
30
40
50
0 1 2 3 4 5 6 7 8 9 10
Non irradiated detector
Peak mode
Non irradiated detector
Figure 7.11: Noise distribution for the non irradiated <111> HR detector.
Vbias (V)
S/N
Detector 111 non irr.Detector 100 non irr.
Non irradiated detector
0
2.5
5
7.5
10
12.5
15
17.5
20
22.5
0 50 100 150 200 250 300 350 400
Figure 7.12: Bias voltage scan for the non irradiated detectors.
130
the test of the detectors functionality in all the situations where spatial resolution measurements
and full electronics integration are not required. From the other side our detectors have provided
a reference benchmark to test the noise performance of the more complex beam test setup at
CERN. In order to fully validate the measured signal to noise ratio a comparison is fulfilled with
the expected values obtained starting from the relations derived in chapter 2.6 after the sensors
electrical characterization.
In Table 7.1 the main results in terms of S/N ratio are summarized. The measured S/N ratios
are slightly lower than the expected ones, in particular for the non irradiated devices, but this
discrepancy has almost entirely disappeared for the irradiated detectors.
The experimental conditions with respect to temperature and leakage current are taken into
account to calculate the expected S/N values. The accuracy on the S/N ratio is of the order
of ±1 as emerged from measurement reproducibility over long term operation and different
environment condition.
Detector <111> HR <100> LR
non irradiated irradiated non irradiated irradiated
LABORATORY
S/N Pk 20.1 @200 V 16.9 @500 V 18.5 @400 V 17.0 @450 V
S/N De 12.0 @200 V 11.5 @500 V 11.5 @400 V 11.2@450 V
BEAM TEST
S/N Pk 20.0 @200 V – 18.5 @350 V –
S/N De 12.7 @200 V 10.6 @500 V 11.8 @600 V
Expected values
S/N Pk 22.5 – 22.5 16.7
S/N De 14.3 – 14.3 12.7
Table 7.1: 300 µm Milestone 99 detectors performances. The S/N ratio is reported both in peak(Pk) and deconvolution (De) mode.
131
7.3 The 500 µµµm detector
The outer part of the silicon tracker is based on large area detectors in order to take advantage of
the new 6” production lines. Since the increase in strip length will worsen the noise contribution
it is necessary to employ thicker detectors so to collect more charge and maintain signal to noise
well above the required threshold of 10.
In the framework of this thesis the first results on the performances of 500 µm thick silicon
detectors, before and after irradiation, bonded to the CMS front-end electronics have been ob-
tained [54] [53]. Is is worth stressing that a wafer thickness of 500 µm is currently an industry
standard for 6” lines and this would result in significant cost savings for the sensors and a further
simplification in the production phase.
7.3.1 The modules
The prototypes under study were produced by ST Microelectronics (Catania, Italy) starting from
a design of the CMS Pisa group that has looked after to the preliminary electrical characteriza-
tion too. They are manufactured on 500 µm thick n-type silicon bulk from 6” wafers and have
an active area of 31 × 50 mm2 with a “barrel type” geometry. The bulk resistivity is greater
than 4 KΩ·cm and the crystal lattice orientation is <100>. The sensors are single sided with
256 p+ strips with an implant width of 24 µm and a pitch of 122 µm. The implant strips are AC
coupled to the APV6 read-out chips through integrated capacitors obtained with multi-layers of
SiO2. The bias voltage is provided through polysilicon resistors in the same way as described
for Florence detectors (see chapter 2.4.1). The active area is surrounded by six guard rings in
order to increase the breakdown voltage and a n+ implantation along the edge of the device is
added to reduce the cut line contribution to the leakage current.
In our analysis we used two identical sensors, one of which has been irradiated, bonded to
hybrids equipped with APV6 chips. For a later comparison of the performance of the 500 µm
modules with respect to the 300 µm ones, a 300 µm thick sensor with 120 mm long strips and
a pitch of 122 µm has been bonded to the same hybrid of the irradiated device. This reference
detector has 128 strips.
The detectors have been uniformely irradiated, using the Louvain-la-Neuve cyclotron facil-
ity described in section 3.2, up to a flunce of 1.6 · 1013 1 MeV equivalent n/cm2, similar to the
132
expected value for 10 years of LHC operation in the outer silicon tracker region [4]. During
irradiation the detectors were fully depleted by applying a bias voltage of 150 V and they were
kept at a temperature of -10C. After irradiation they were stored at -25C in order to reduce
reverse annealing effects.
The characterizations performed on detectors before and after irradiation has shown that
the devices can be operated in stable conditions up to 400 V in both cases [53]. From a bulk
capacitance measurement as a function of the bias voltage the full depletion voltage turns out
to be 53 V and 42 V, for the non irradiated and irradiated detector respectively. Due to the low
neutron fluence the irradiated detector is not type inverted.
7.3.2 Florence laboratory results
The detectors have been exposed to minimum ionizing particle beam obtained selecting elec-
trons from a β source with the same experimental setup described in chapter 5.1 The mea-
surement have been performed at -10C. To study the effect of temperature we operated the
non irradiated detector both at -10C and at 10C but the data collected, in this case, show no
significant variations.
Data were taken both in peak and in deconvolution mode and the detectors were operated
within a wide range of bias voltages. In Fig. 7.13 the cluster charge distributions in peak mode
for the 300 µm and the 500 µm thick detectors in the same operating conditions are shown; in
this particular case the bias voltage is 150 V.
Comparing the total cluster charge for the 300 µm and 500 µm detectors we observe a direct
proportionality between charge and thickness (see Table 7.2). The charge collected as a function
of the bias voltage is plotted in Fig. 7.14 for the 500 µm irradiated and non irradiated detectors.
Main results are summarized in Table 7.2, where the most probable signal values and the
noise are reported for data taken at a bias voltage of 150 V and a temperature of -10C for
all the detectors. The noise increases only slightly after irradiation ( 5% both in peak and
deconvolution) and the cluster charge loss is below 10 %.
It should be stressed here that the performances in term of signal to noise ratio for these
prototypes with reduced strip length are not important. It is of main importance the demonstra-
tion that the collected charge for the different devices scales with the thickness and the effect of
133
23.49 / 22P1 259.6P2 35.74P3 5.418
ADC counts
Ent
ries
/bin
0
10
20
30
40
50
60
70
0 50 100 150 200 250 300
(a)
83.86 / 37P1 796.6P2 59.90P3 8.304
ADC countsE
ntri
es/b
in
0
25
50
75
100
125
150
175
200
225
0 50 100 150 200 250 300
(b)
Figure 7.13: Cluster charge distribution in peak mode for the 300µm (a) and 500µm (b) thickdetectors.
Detector Signal Noise Signal Noise Signal/µm
(Peak) (Peak) (Dec.) (Dec.) Peak Dec
500 µm irradiated 55.0 1.40 45.0 2.26 0.11 0.09
500 µm non irradiated 59.9 1.32 49.2 2.14 0.12 0.10
300 µm Ref. non irr. 35.8 1.79 29.3 2.64 0.12 0.10
Table 7.2: Signal and noise results for 500µm thick detectors compared to a 300 µm referencedetector. The values reported are in ADC counts. The strip lengths are different in the two casesso that noise cannot be directly compared.
irradiation on 500 µm samples does not alter significantly the noise and charge values.
134
Vbias (Volts)
Cha
rge
(AD
C c
ount
s)
Non irradiated detector - peak modeIrradiated detector - peak modeIrradiated detector - deconvolution mode
10
20
30
40
50
60
70
80
0 50 100 150 200 250 300
Figure 7.14: Collected charge versus bias voltage for the 500µm irradiated and non irradiateddetector.
135
ConclusionsThe work performed in the context of this thesis has concerned the study of the performances
of the silicon microstrip detector prototypes for the forward part of the CMS tracker at LHC,
with particular emphasis on the radiation effects and their influence on the signal to noise ratio.
A set of modules, with the same geometrical characteristics, has been built by the CMS Flo-
rence group starting from two different substrates: a high resistivity <111> crystal orientation
and a low resistivity <100>. Two of them, one for each kind, have been heavily irradiated with
a neutron beam (up to a fluence of 1.1 · 1014 1 MeV equivalent n/cm2, that is the value expected
after 10 years of LHC operation in the region where these detectors are going to be installed) to
study the radiation damage effects.
In order to measure the signal and noise performances, a β source test and DAQ system,
which has revealed to be very flexible in performing the modules and front-end electronics
characterization, has been built in the framework of this thesis in Florence laboratory. On the
same DAQ system I have implemented a laser station with the primary goal of providing a fast
check of the detector channels functionality.
All the tested detectors were bonded to the APV6 chip, a prototype of the Silicon microstrip
tracker front-end electronics very similar to the final version that will be used in the experiment.
A deep investigation has been done in order to understand the operating principle of the APV6
and its main features. The flexibility of the laboratory DAQ setup has allowed to plan a fast test
procedure of all the main APV6 functions (logic, channel response, noise). The main object
of the work I performed during the thesis has been the testing of the modules during their
construction stages, from the APV6 chip alone to the irradiated modules.
The whole set of detectors has been exposed to a high energy particle beam in three different
test periods at CERN. In this case a complete prototype of the CMS tracker read-out and control
system has been used, with components as close as possible to the final design. This system
has been successfully operated for the first time under LHC like conditions in a 25 ns structured
beam.
Main results obtained from the laboratory and beam test measurements, which fully agree
137
among them, may be summarized as follows:
• The HR and LR devices are both type inverted with full depletion voltages values re-
spectively of 250 and 130 V after an irradiation with 1.1 · 1014 1MeV equivalent n/cm2.
From the bias voltage scan it appears that the detectors have to be operated overdepleted
in order to optimize the S/N ratio.
• No experimental evidence on a difference between the <111> HR and <100> LR mod-
ules in terms of signal to noise ratio has been obtained, provided that they are used overde-
pleted.
• The signal to noise ratio changes, before and after irradiation, from 20 to 17 in peak mode
and from 12 to 11 in deconvolution mode.
• An efficient cooling system is crucial to operate the irradiated detectors, avoiding the
potentially dangerous thermal runaway effect.
The measured S/N ratios show only a slight discrepancy with the values expected from a
calculation based on the detector electrical parameters measurement and the APV6 chip perfor-
mances.
From previous experience coming from test beam measurements it is known that a S/N≥10,
using our clustering algorithm, assures an efficiency close to 100 % while reduces under a
negligible level ghost hits related to statistical noise fluctuations.
The collected results have shown that this threshold is guaranteed, with the APV6 front-end
chip, even in the presence of irradiated detector working in the high-luminosity LHC condi-
tions. In particular the detectors were irradiated up to the foreseen fluence for 10 years of LHC
operations and the main goal of a S/N∼ 11 in deconvolution mode, while keeping the full de-
pletion voltage below an acceptable limit, has been obtained. Initial low resisitivity substrates
help in reducing the optimal detector operating bias voltage in the crucial high luminosity phase
of LHC once the tracker will be heavily irradiated. This makes us confident that this kind of
detectors will ensure satisfactory performances for the whole lifetime of the experiment.
138
Finally, the total cluster charge and noise have been measured, for the first time, for a 500µm
thick silicon detector bonded to CMS electronics, comparing these values before and after irra-
diation. The results are in good agreement with expectation in terms of linearity of the charge
collected with respect to the thickness and in term of measured noise, both in peak and in de-
convolution mode. These data show that 500µm devices are a promising technology for large
area silicon trackers.
139
Appendix A
APV6 response parametrization
The transfer function for the circuit shown in Fig. 4.3 can be derived in two independent ways.
The first possibility is to find the best transfer function F (ω) by fitting experimental data with
the inverse Fourier transforms of a multipole functions plausible set. In a second case an an-
alytical approach can be used and from a comparison with the experimental points the main
parameters of a simplified equivalent circuit can be calculated.
I Method
The experimental data are well fitted with the time function F (t) obtained as the inverse
Fourier transform of a four poles (two real and two complex) function F (ω) given by:
F (ω) =1
(1 + jωτ1r)(1 + jωτ2r)(1 + jωθ)(1 + jωθ∗)(A.1)
The time domain corresponding function F (t) is given by:
F (t) = QR·[A · e−
t−t0τ1r +B · e−
t−t0τ2r + (C · sin (ω0(t− t0)) +D · cos (ω0(t− t0))) · e−
t−t0τs
](A.2)
where: τs = |θ|2
(θ)
ω0 = (θ)|θ|2
while the expressions for A,B,C,D are:
A =(1+τ2
s ω20)τ2
1r
(τ1r−τ2r)·[(τs−τ1r)2+(ω0τsτ1r)2]
B =(1+τ2
s ω20)τ2
2r
(τ2r−τ1r)·[(τs−τ2r)2+(ω0τsτ2r)2]
C =(1+τ2
s ω20)·[τ1rτ2r(1−τ2
s ω20)+τs(τ1r+τ2r−τs)]
ω0·[(τs−τ1r)2+(ω0τsτ1r)2]·[(τs−τ2r)2+(ω0τsτ2r)2]
D =(1+τ2
s ω20)τs(2τ1rτ2r−τ2rτs−τ1rτs)
[(τs−τ1r)2+(ω0τsτ1r)2]·[(τs−τ2r)2+(ω0τsτ2r)2]
141
VSHA (V) τ1r (ns) τ2r (ns) τs (ns) 1/ω0 (ns) t0 (ns) QR
2.00 31.8± 1.2 31.8± 1.2 21.0± 0.6 14.9± 0.3 3.3± 0.3 9349
-0.20 23.4± 2.5 871.1± 155.2 21.8± 2.0 13.6± 0.4 2.9± 0.6 144760
Table A.1: Numerical results for the free parameters of Eq. A.2 fitted to Florence data.
The parameters τ1r, τ2r, τs, ω0, the amplitude coefficient QR and a delay time t0 are let as free
parameters during the fit procedure. The fit results are shown in Fig. 4.9 for two different shaper
time constants. The numerical results for the correspondent free parameters are summarized in
Table A.1.
It is worth noting that the VSHA register mainly affects just one pole (the real τ2). Thus
the shaper feedback resistance change, operated by VSHA, is responsible of the pulse shape by
moving one real pole of the transfer function.
II method
The single channel APV6 simplified equivalent circuit is is shown in Fig. A.1.
The pre-amplifier and the shaper circuit are both based on a “cascode” (Fig. A.1(b)) with
the feedback capacitor Cf = 0.25 pF. The “cascode” output is followed by a source follower
and fed back to the input through the transistor resistance adjustable by means of the VPRE and
VSHA registers. Cx is the first transistor input gate capacitance, plus, for the pre-amplifier, the
detector capacitance if present; G,Cd and Rd are the transconductance, drain capacitance and
resistance of the common-source input transistor of the cascode [47].
The cascode with feedback can be sketched as an ideal amplifier A with input impedance
ZA:
A =
(−G +
1
Zf
)· ZdZf
Zd + Zf
ZA =Zx · (Zf + Zd)
Zf + Zd + Zx +GZxZd
where Zx =1
jωCx, Zf =
1jωCf
, Zd =Rd
1+jωCdRd.
The final model is reported in Fig. A.1(c) where RB is the source follower resistance, Rfpa
142
Cf
Cx Cd Rd
VoVi
ViG
Vi Vo
ZA
A
ZAshZApa Cspa
Cssh
Rfpa RfshCc
V
Vpbp
Vpcasc
Vpbn
M_p_pinp3000/1.4
M_p_pis150/10
Cfp .25p
M_n_pfb2.4/60
M_n_pcasc400/1.2
M_n_pis2330/10
M_n_psf400/1.2
M_n_pis3200/10
Vpsfb
Vsbp
Vscasc
Vsbn
M_p_sinp800/1.4
M_p_sis150/10
Cfs .25p
M_n_sfb2.4/60
M_n_scasc400/1.2
M_n_sis2100/10
M_n_ssf400/1.2
M_n_sis3200/10
Vssfb
VPRE
VSHA
Cc1.8p
Vsbp
Vscasc
Vsbn
M_p_sinp800/1.4
M_p_sis150/10
Cfs .25p
M_n_sfb2.4/60
M_n_scasc400/1.2
M_n_sis2100/10
M_n_ssf400/1.2
M_n_sis3200/10
Vssfb
i
(a)
paApa Vi shAsh
Vout
δ(t)QIs = RBRB
Voutpa
Vi pa Vi sh(c)
(b)
Figure A.1: Single channel APV6 circuit model.
and Rfsh are the feedback resistances driven by VPRE and VSHA registers, CSpa and CSsh
are the capacitances on pre-amplifier and shaper outputs. The transfer function is analytically
calculated as in the following:
F (ω) =Vout
Is(A.3)
=ZAshZLsh(RB + AshRfsh)(Rfsh + ZAsh) · T 2
pa
ZApaZLpa(ZcZAsh + ZcRfsh + ZAshRfsh)(RB + ApaRfpa) + Z2Ash
ZLsh(RB + AshRfsh)Tpa
(A.4)
143
ZINsh = Zc +ZAsh
ZAsh + Rfsh
·(Rfsh +
ZAshZLsh(RB + AshRfsh)
(Rfsh + ZAsh)(RB + ZLsh) + ZLsh(RB −AshZAsh)
)
ZLpa = ZINsh//CSpa =ZINsh
1 + jωZINshCSpa
Tpa =Vpa
out
Is=
ZApaZLpa(RB + ApaRfpa)
(Rfpa + ZApa)(RB + ZLpa) + ZLpa(RB −ApaZApa)
Zc =1
jωCc
ZLsh =1
jωCSsh
In a more compact form F (ω) can be expressed as:
F (ω) Rfsh · jωRfpaCc · (1− jωτ0)
1 + jωa− ω2b− jω3c+ ω4d+ jω5e+ ω6f(A.5)
where a, b, c, d, e, f are function of the circuit elements.
The transfer function has an amplitude proportional to Rfsh and this means that is strictly
dependent on the VSHA register, as we expected. The transfer function shows six poles and
two zeros which, after a double pole-zero cancellation, even if not exact, is compatible with the
four poles transfer function obtained with the phenomenological approach.
144
Appendix B
The effect of deconvolution on noise
The evaluation of noise after the deconvolution operation is important for detectors equipped
with APV6 chips since they will run, during the LHC high luminosity phase, with this oper-
ational mode. The noise treatment must, in our case, deal with a system that is based on the
sampling of a continuous waveform.
In general the deconvoluted signal at some arbitrary time, in the presence of noise alone, is
given by:
sk =∑i
wivk−i+1 (B.1)
where vi are the noise output voltages and wi the weights. Both sk and vi have a mean
expectation value equal to zero.
The noise is defined as the variance of sk, which is:
σ2(sk) =<∑ij
wivk−i+1wjvk−j+1 > (B.2)
The rms noise after shaping can be considered as the sum of two components, σp and σs, repre-
senting parallel and series noise at the amplifier input [55] [56]. This can be written as:
σ2tot = α
∫ ∞
−∞[h(t)]2dt+ β
∫ ∞
−∞[h′(t)]2dt = σ2
p + σ2s (B.3)
where h(t) is the response to a charge pulse and α and β are coefficients which depend on the
details of the noise sources. Since they are uncorrelated we can treat the two contributions to
the noise separately; for the CR-RC filter they can be written as:
σ2(s) = (w21 + w2
2 + w23)σ
2 + 2w1w2C(∆T ) + 2w2w3C(∆T ) + 2w1w3C(2∆T ) (B.4)
145
where ∆T is the sampling interval and the functions C(∆T ) give the average correlation be-
tween a measurement at some time and another delayed by ∆T ; C(0) = σ2. For the CR-RC
shaper these functions can be calculated and turn out to be:
C(x) = σ2p(1 + x)e−x (B.5)
for the parallel noise and
C(x) = σ2s(1− x)e−x (B.6)
for the series noise, with x = ∆Tτ
ratio of the sampling interval to amplifier time constant. For
large values of x, or samples well separated in time, C(x) → 0, meaning no correlation present,
while for x→ 0, C(x) → σ2, meaning complete correlation.
The two components of the noise after deconvolution can be expressed, after some algebra,
as:
σ2p(s) = (σ2
pe−2/x2)[e2x − 4x− e−2x] (B.7)
σ2s(s) = (σ2
se−2/x2)[e2x + 4x− e−2x]
In our case, in which x=0.5, it turns out:
σp(s) ≈ 0.45σp (B.8)
σs(s) ≈ 1.45σs
We can observe that the deconvolution method decrease the parallel noise while series noise
increases, according to the fact that the operation is equivalent to using a filter with a shorter
time constant. It can be demonstrated that using a CR-RC filter with a shorter time constant τ ′
the consequences for the parallel and series noise are:
σ2p(s) = σ2
py (B.9)
σ2s(s) = σ2
s/y (B.10)
where y = τ ′τ
. In our case the deconvolution actually reduces parallel noise more than re-
shaping with a similar ratio of time constant, while the series noise is increased a slightly more.
146
Thus provided that the series noise of the system is sufficiently low, the deconvolution method
offers greater immunity to increased parallel noise from radiation induced leakage currents dur-
ing LHC operations.
147
Appendix C
The Sequencer schematic
The layout of the Sequencer described in section 5.3 and developped in the framework of this
thesis is shown in Fig. ??.
149
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AcknowledgementsSincere thanks to all members of the CMS Florence group for their advice, support and spur
to carry on with this work. In particular I’m in debt with Carlo, who has spent several Christ-
mastime days in helping me to write the thesis, and with Dr.Marco Meschini, Dr.Raffaello
D’Alessandro, Dr.Alessandro Buffini, for the encouragement and suggestions. I’m also very
grateful to Prof. Anna Cartacci for the continuously shown kindness and to Prof. Giuliano
Parrini for the helpful comments and for the constant presence. Last, but not the least, I wish to
thank the BAR managers and Mrs Gabriella and Mrs Guglielma that make people to be like one
of the family.
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