IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 62, NO. 2, FEBRUARY 2013 503
Power Consumption in Direct Interface CircuitsFerran Reverter
Abstract—This paper analyzes theoretically and experimentallythe current consumption in direct interface circuits, i.e., circuitsin which the sensor is directly connected to a microcontroller(μC) without using either a signal conditioning circuit or ananalog-to-digital converter. The theoretical analysis, which takesinto account the current consumed by both the internal electronicsof the μC and the external components, proposes formulas toestimate the average current consumption in active mode. Theestimated values fairly agree with those obtained in the experi-mental tests carried out by an AVR ATtiny2313 μC at differentoperating conditions. For example, the current consumption inactive mode at 3 V–4 MHz was about 1.5 mA for the measurementof a 1-kΩ resistive sensor and 0.6 mA for the measurement of a177-pF capacitive sensor. The results reported herein are expectedto be useful for the design of direct interface circuits intended forbattery-powered measurement systems.
S ENSOR electronic interfaces are generally based on theblock diagram shown in Fig. 1(a) [1]. First of all, the
sensor transforms a signal from a given energy domain (suchas thermal, magnetic, mechanical, chemical, or radiant) to theelectrical domain [2] by changing, for example, its electricalresistance or capacitance. Afterwards, the signal conditioningcircuit, which generally relies on operational amplifiers, per-forms some or all of the following tasks: sensor-output-to-voltage conversion, amplification, filtering, linearization, anddemodulation. The resulting analog signal is then digitizedvia an analog-to-digital converter (ADC). Finally, a digitalsystem [e.g., microcontroller (μC)] acquires, stores, processes,controls, communicates (to other devices), and/or displays thedigital value with information about the measurand. Owing tothe rapid advances in integrated circuit (IC) technologies, nowa-days, there are commercially available ICs that integrate thecircuitry to perform several of the functions shown in Fig. 1(a).
The block diagram in Fig. 1(a) can be simplified, for somesensors, to that shown in Fig. 1(b), where the sensor is di-rectly connected to the μC without using either the signalconditioning circuit or the ADC, thus reducing the cost, com-plexity, physical space, and power consumption of the circuit.These direct interface circuits rely on appropriately exciting
Manuscript received April 3, 2012; revised July 26, 2012; acceptedAugust 15, 2012. Date of publication September 17, 2012; date of currentversion December 29, 2012. The Associate Editor coordinating the reviewprocess for this paper was Dr. Serge Demidenko.
The author is with the Castelldefels School of Technology (EETAC), Uni-versitat Politècnica de Catalunya, 08860 Castelldefels, Spain (e-mail: [email protected]).
Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TIM.2012.2216473
Fig. 1. (a) Classical block diagram of a sensor electronic interface. (b) Direct-sensor-to-μC interface circuit.
the sensor to get a signal (usually, a time-modulated or quasi-digital signal) that can be directly measured by the μC usingan embedded digital timer/counter. Such circuits were initiallyproposed in application notes of μC’s manufacturers [3]–[5],but recently, they have been carefully analyzed and appliedto measure different types of resistive [6]–[10] and capacitive[11]–[14] sensors. For example, a nonlinearity error of 0.01%full-scale span (FSS) and a resolution of 13 b when measuringresistive sensors [6], [8] and a nonlinearity error of 0.1% FSSand a resolution of 9 b when measuring capacitive sensors[11] have been reported. These results are quite remarkable tak-ing into account that the circuit just requires a low-cost general-purpose μC. As indicated before, direct interface circuits areexpected to be low power since they require a few components.However, the current consumption of these circuits has not beenanalyzed, and just some numerical values have been provided.For instance, the current consumption in active mode was9.5 mA at 4.5 V–20 MHz [10] and 2.2 mA at 3 V–8 MHz [15].
This paper analyzes theoretically and experimentally the cur-rent consumption of direct interface circuits when measuringresistive and capacitive sensors. It also proposes an appropriateconfiguration of the resources embedded into the μC in orderto reduce the current consumption of the circuit in activemode. The analyses and proposals reported herein are expectedto be useful for those interested in applying direct interfacecircuits to battery-powered measurement systems, such asautonomous sensors.
II. OPERATING PRINCIPLE
Before analyzing the current consumption, this section re-views the operating principle of direct interfaces based on anRC circuit, in which the μC measures the time interval neededto discharge a capacitance C to a given threshold voltagethrough a resistance R [16].
A. Direct Interface Circuit for Resistive Sensors
Fig. 2(a) shows a direct interface circuit to measure a resis-tive sensor (Rx). Its operating principle involves two stages:charging stage and discharging stage. During the chargingstage [see Fig. 2(c)], Pin 1 is set as an output providing a
504 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 62, NO. 2, FEBRUARY 2013
Fig. 2. (a) Direct interface circuit for a resistive sensor (Rx). (b) Waveform ofthe voltage at node A during the charge–discharge process. (c) Pin configurationduring the charging stage. (d) Pin configuration during the discharging stage.
digital “1” (with an analog output voltage equal to the supplyvoltage VDD), whereas Pin P is set as an input offering highimpedance (HZ). Therefore, the capacitor Cd is charged towardVDD through the resistor Ri for a time interval equal to kc timesthe time constant (RiCd), kc being higher than five to ensurean appropriate charging process. The resistor Ri is necessary todecrease the cutoff frequency of the low-pass filter during thecharging stage and, hence, to improve the rejection of powersupply noise/interference [17]. During the discharging stage[see Fig. 2(d)], Pin 1 is set as an HZ input and Pin P is setas an output providing a digital “0,” and consequently, Cd
is discharged toward ground through Rx while the embeddedtimer measures the time interval required to do so. When theexponential discharging voltage crosses the lower thresholdvoltage (VTL) of the Schmitt trigger (ST) buffer embeddedinto Pin 1, the timer is read and a digital number proportionalto Rx is achieved. The resulting waveform of the voltage atnode A [Fig. 2(a)] during the charge–discharge process isshown in Fig. 2(b), where the discharging time is
Td = RxCd ln
(VDD
VTL
). (1)
In order to have a measurement result insensitive to bothmultiplicative and additive parameters, the circuit in Fig. 2(a)(with an additional reference resistor Rref ) usually measuresthree discharging times [6] and then applies the three-signal au-tocalibration technique [18]. Accordingly, the sensor resistancecan be then estimated by
R∗x =
Td1 − Td3
Td2 − Td3Rref (2)
where Td1, Td2, and Td3 are the discharging times of the sensor,reference, and offset measurements, respectively.
Fig. 3. (a) Direct interface circuit for a capacitive sensor (Cx). (b) Waveformof the voltage at node A during the charge–discharge process. (c) Pin config-uration during the charging stage. (d) Pin configuration during the dischargingstage.
B. Direct Interface Circuit for Capacitive Sensors
For the measurement of a capacitive sensor (Cx), it isrecommended to swap the position of the resistance and thecapacitance, as shown in Fig. 3(a). The operating principle ofsuch a circuit also requires two stages. During the chargingstage [see Fig. 3(c)], both pins are set as an output: Pin 1provides a digital “1,” whereas Pin P provides a digital “0.”Therefore, taking into account that the resistor Rd is muchhigher than Ri, Cx is charged toward VDD through Ri for a timeinterval equal to kcRiCx. During the discharging stage [seeFig. 3(d)], Pin 1 is set as an HZ input and Pin P does not changeits state, and consequently, Cx is discharged toward groundthrough Rd. When the voltage-threshold crossing is detected,the timer is read and a digital number proportional to Cx isachieved. Here, the discharging time [see Fig. 3(b)] equals
Td = RdCx ln
(VDD
VTL
). (3)
The circuit in Fig. 3(a) (with an additional reference ca-pacitor Cref ) generally measures three discharging times tocompensate for multiplicative and additive parameters [11].Then, the sensor capacitance can be estimated by
C∗x =
Td1 − Td3
Td2 − Td3Cref (4)
where Td1, Td2, and Td3 are again the discharging times of thesensor, reference, and offset measurements, respectively.
III. THEORETICAL ANALYSIS
The current consumption of direct interface circuits in activemode (i.e., when the μC is working to perform the measure-ment) is analyzed using the waveforms in the time domain
REVERTER: POWER CONSUMPTION IN DIRECT INTERFACE CIRCUITS 505
Fig. 4. (a) Waveform of the voltage at node A [Figs. 2(a) and 3(a)] when the circuit carries out the three measurements involved in the autocalibration technique.(b) Current consumed by the internal electronics of the μC. (c) Current flowing through the external components of the μC. (d) Overall current consumed.
shown in Fig. 4. Fig. 4(a) shows the voltage at node A [seeFigs. 2(a) and 3(a)] when the circuit carries out the threemeasurements involved in the autocalibration technique, whereTc1, Tc2, and Tc3 are the time intervals required to charge thecapacitance, Td1, Td2, and Td3 are the discharging times to bemeasured, Tp is the time interval required to process the threemeasurements by means of (2) or (4), and TT is the time intervalrequired to complete one measurement.
Fig. 4(b) shows the current (iint) consumed by the internalelectronics of the μC during the measurement; note that thecurrent of the input/output ports of the μC is not consideredin Fig. 4(b), but it is considered in Fig. 4(c). The charging,
discharging, and processing stages require average internal cur-rents equal to Iint1, Iint2, and Iint3, respectively, whose valuesdepend on the configuration of the resources embedded into theμC. To reduce the current consumption, we propose to set thecentral processing unit (CPU) and the timer as indicated in Ta-ble I; the rest of peripherals are assumed to be always switchedoff. In the charging stage, the charging time is to be controlledby the CPU running at low frequency (e.g., tens or hundreds ofkilohertz); two remarks are as follows: 1) Most of the currentμCs have a prescaler to divide the master clock frequency, and2) a low-accuracy charging time (due to a low-frequency clock)is not a problem at all since the information is in the discharging
506 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 62, NO. 2, FEBRUARY 2013
TABLE IPROPOSED STATE (ON/OFF) AND RUNNING FREQUENCY OF THE CPU AND THE EMBEDDED TIMER FOR EACH STAGE OF THE MEASUREMENT
time. In the discharging stage, the discharging time is to bemeasured by the timer running at high frequency (e.g., units ortens of megahertz) to have a good resolution, whereas the CPUis off whenever this does not stop the operation of the interruptsystem and the timer. In the processing stage, the CPU runs athigh frequency (e.g., units or tens of megahertz) to compute asfast as possible the resistance or capacitance of the sensor bymeans of (2) or (4). The following relation between currents isexpected: Iint1 < Iint2 < Iint3.
Fig. 4(c) shows the current (iext) flowing through the exter-nal components of the μC during the measurement. A positivepolarity of current [see the charging stages in Fig. 4(c)] meansthat the current flows from the supply voltage of the μC tothe capacitance of the RC circuit, whereas a negative polarity[see the discharging and processing stages in Fig. 4(c)] meansthat it flows from the capacitance to ground and, hence, noelectric charge is required from the supply voltage. Therefore,the external current consumption is zero during the discharg-ing and processing stages, but not during the charging stage.If the charging time is much longer than the time constant(i.e., kc > 5), then the average external currents during Tc1, Tc2,and Tc3 are
Iext1 =VDDC
Tc1(5a)
Iext2 =(VDD − VTL)C
Tc2(5b)
Iext3 =(VDD − VTL)C
Tc3(5c)
respectively, where C is the capacitance being charged (i.e., Cd
in Fig. 2 and Cx in Fig. 3).Fig. 4(d) shows the resulting current consumed by the direct
interface circuit when both Fig. 4(b) and (c) (just positivepolarities of current) are considered. Accordingly, the averagecurrent consumption in active mode is
IT = (Iext1 + Iint1)Tc1
TT+ Iint2
Td1
TT+ (Iext2 + Iint1)
Tc2
TT
+Iint2Td2
TT+ (Iext3 + Iint1)
Tc3
TT+ Iint2
Td3
TT+ Iint3
Tp
TT. (6)
In terms of current consumption analysis, we propose to assumethat Tc1 ≈ Tc2 ≈ Tc3 ≈ Tc and Td1 ≈ Td2 ≈ Td3 ≈ Td. Then,
using (5), (6) can be simplified to
IT ≈ 1
TT[C(3VDD−2VTL) + 3Iint1Tc + 3Iint2Td+Iint3Tp] .
(7)
When measuring resistive sensors, the contributions ofthe following components in (7) are expected to be minor:1) Iint1, since it is much smaller than Iext when using a low-value Ri (e.g., hundreds of ohms) [17], and 2) Iint3, because3Tc + 3Td � Tp when using a high-value Cd. Accordingly, (7)can be simplified to
IT ≈ VDD − 0.67VTL + Iint2Rx ln(VDD/VTL)
kcRi +Rx ln(VDD/VTL)(8)
which shows that IT is independent of Cd. On the other hand,when measuring capacitive sensors, the contributions that areexpected to be minor are the following: 1) Iext and Iint1, sincethe charging stage is very short when measuring sensors in thepicofarad range, and 2) Iint3, because 3Td � Tp when using ahigh-value Rd. Consequently, (7) can be simplified to
IT ≈ Iint2 (9)
which shows that IT is equal to the current consumed by theinternal electronics of the μC during the discharging stage.
For those applications in which the interface circuit doesnot read the sensor continuously but every T0 second (aswhat happens, for instance, in autonomous sensors), the overallaverage current consumption could be estimated by
I0 = ITkTT
T0+ Isleep
Tsleep
T0(10)
where k is the number of times that the sequence in Fig. 4(a)is repeated in active mode to average k measurements, Tsleep isthe time interval in which the μC is in sleep mode, Isleep isthe current consumption of the μC in such a mode, andT0 = kTT + Tsleep.
IV. EXPERIMENTAL RESULTS AND DISCUSSION
The experimental tests were carried out by means of anAVR ATtiny2313 (Atmel) 8-b μC at two operating conditionsof supply voltage (VDD) and frequency (fclk): 3 V–4 MHzand 5 V–20 MHz; note that fclk is the running frequency ofthe digital electronics (e.g., timer) embedded into the μC, butnot the frequency at which the sensor is measured. Table IIshows some experimental data of the AVR μC at such operatingconditions. The currents Iint1, Iint2, and Iint3 were measured
REVERTER: POWER CONSUMPTION IN DIRECT INTERFACE CIRCUITS 507
TABLE IIEXPERIMENTAL DATA OF THE AVR μC AT
DIFFERENT OPERATING CONDITIONS
by a digital multimeter (Agilent 34401A) set in dc currentmode and connected in series with the supply pins while theμC repeated the task of interest continuously (i.e., in a loop).For example, Iint2 was measured while the embedded timer rancontinuously at high frequency (see Table I), but the CPU andthe rest of peripherals were off. In such conditions, the currenthad a period that was much smaller than the integration time(200 ms) of the integrating ADC embedded into the multimeter,and hence, the readings were stable [19], [20]. From Table II,both operating conditions show that Iint1 < Iint2 < Iint3, asexpected. Moreover, the three currents clearly increase withfclk and VDD. The ratio of currents between the two operatingconditions is quite similar to the ratio of fclk (i.e., 20 MHz/4 MHz) multiplied by the ratio of VDD (i.e., 5/3).
After characterizing the μC, the direct interface circuitsshown in Figs. 2(a) and 3(a) were built. To have a performancesimilar to that shown in Fig. 4, the circuits in Figs. 2(a) and3(a) carried out three times the charge–discharge process [seeFigs. 2(b) or 3(b)] through the same components and thenapplied (2) or (4); such a simple measurement setup withoutany reference component does not make sense in terms of theresult obtained from (2) or (4), but it does in terms of currentconsumption analysis. The current IT was measured using thesame digital multimeter while the circuit repeated continuouslythe sequence shown in Fig. 4(a); note that the period (TT ) ofthe resulting current was also much smaller than the integrationtime of the multimeter. Other general remarks about the imple-mentation of the circuits in Figs. 2(a) and 3(a) are as follows:1) The charging time was long enough (kc ≈ 6) to have theright voltage across the capacitance after the charging stage;2) the discharging time was measured by the embedded 16-bTimer 1; and 3) the tasks of Pin 1 were implemented by pinPD6/input capture pin, which is associated to a capture moduleand includes an ST buffer.
The circuit in Fig. 2(a) was built with Rx = 1000 Ω (whichis a common value for resistive temperature sensors [6]), Ri =100 Ω, and different values of Cd (220 nF, 470 nF, and 1 μF).Tables III and IV show the resulting time intervals (measuredwith a Tektronix TDS2002B digital oscilloscope) at 3 V–4 MHz and 5 V–20 MHz, respectively. The overall time interval(TT ) is in the range of units of milliseconds (between 1 and6 ms), which seems acceptable for measurements carried outby battery-powered circuits; note that TT is slightly shorterat 5 V–20 MHz since the processing of (2) is faster and thevalues VDD/VTL and, hence, Td are smaller. Figs. 5 and 6 show
TABLE IIIEXPERIMENTAL VALUES OF THE TIME INTERVALS INVOLVED
IN FIG. 4(a) FOR THE CIRCUIT IN FIG. 2(a) AT 3 V–4 MHz
TABLE IVEXPERIMENTAL VALUES OF THE TIME INTERVALS INVOLVED
IN FIG. 4(a) FOR THE CIRCUIT IN FIG. 2(a) AT 5 V–20 MHz
the current consumption at 3 V–4 MHz and 5 V–20 MHz,respectively, including the theoretical values calculated by (7),the approximated theoretical values calculated by (8), andthe experimental values. The experimental values are slightlyhigher (less than 5%) than the theoretical values calculated by(7), which means that the models developed in Section III arequite accurate to estimate the current consumption. The approx-imated theoretical values are smaller than those calculated by(7) since some contributions have been neglected in (8); note,however, that the values resulting from (8) are more than 95%and 85% of the overall current calculated by (7) in Figs. 5and 6, respectively. The current consumption at 3 V–4 MHz(approximately 1.5 mA) is about three times smaller than thatat 5 V–20 MHz (approximately 5 mA) basically because thecurrent consumption during the discharging stages (which lastabout 50% of TT ) is almost ten times smaller (see Table II).
The circuit in Fig. 3(a) was built with Cx = 177 pF (whichis a usual value for capacitive humidity sensors [11]), Ri =100 Ω, and different values of Rd (2.2, 4.7, and 10 MΩ). Fig. 7shows the current consumption at 3 V–4 MHz including thetheoretical values calculated by (7), the approximated theo-retical values calculated by (9), and the experimental values.As in Figs. 5 and 6, the experimental values in Fig. 7 areslightly higher (less than 10%) than the theoretical values.The approximated theoretical values are smaller than thosecalculated by (7), as expected, but they represent more than80% of the nonapproximated value when using a high-valueresistor. The current consumption in Fig. 7 is about two to threetimes smaller than that in Fig. 5 since the current consumptionduring the charging stage is almost negligible when measuringlow-value capacitive sensors.
Fig. 8 shows the charge required to carry out one completemeasurement (i.e., QT = IT · TT ) for the circuit in Fig. 2(a)when measuring a 1-kΩ resistive sensor at both operating con-ditions. Although IT is quite independent of Cd (about 1.5 mAin Fig. 5 and 5.0 mA in Fig. 6), the required QT clearlyincreases with Cd: from 2.5 to 9.0 mA · ms at 3 V–4 MHzand from 6.3 to 25.0 mA · ms at 5 V–20 MHz. This increase of
508 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 62, NO. 2, FEBRUARY 2013
Fig. 5. Theoretical and experimental values of current consumption for thecircuit in Fig. 2(a) at 3 V–4 MHz.
Fig. 6. Theoretical and experimental values of current consumption for thecircuit in Fig. 2(a) at 5 V–20 MHz.
charge is due to the fact that TT is not constant, but it increaseswith Cd, as shown in Tables III and IV. Consequently, increas-ing Cd involves approximately the same current consumptionbut for a longer measuring time, and hence, more charge isrequired from the battery. On the other hand, increasing Cd
involves a better measurement resolution [8], and hence, there isa tradeoff between resolution and battery lifetime. Furthermore,if the average of k measurements is used as an estimate to havea higher resolution (for example, k = 100 to have a resolutionof 13 b when measuring resistive sensors [8]), then the chargeshown in Fig. 8 must be multiplied by k. However, autonomoussensors generally require a low resolution (for example,8–10 b), and hence, no averaging or a low value of k (forexample, k < 10) is to be expected.
The AVR ATtiny2313 used in the previous experimental testsis not a low-power μC. However, no significant differencesin the current consumption are expected if a low-power μCis used, at least in active mode. This is because the currentconsumption of low-power μCs in active mode at high frequen-cies is quite similar to that specified in Table II. For example,Table V shows the current consumption of a direct interface cir-cuit implemented with a low-power μC (MSP430F2274 fromTexas Instruments Incorporated) at 3 V–8 MHz for the differentoperating stages [15]. The current during the discharging stageis 0.4 mA, which is similar to the value of Iint2 specified in
Fig. 7. Theoretical and experimental values of current consumption for thecircuit in Fig. 3(a) at 3 V–4 MHz.
Fig. 8. Charge required to carry out one complete measurement for the circuitin Fig. 2(a) at 3 V–4 MHz and 5 V–20 MHz.
Table II for the AVR at 3 V–4 MHz, and the current during theprocessing stage is 2.8 mA, which is almost two times (sincethe frequency is two times higher) the value of Iint3 specifiedin Table II. The main difference between the results reportedherein and those from [15] is in the charging stage. Accordingto Section III, the current during the charging stage should beas follows: Iext = (3 V · 1 μF)/0.83 ms = 3.6 mA, which issignificantly smaller than the value reported in [15] (5.8 mA).This is due to the fact that the timer was set to run at 8 MHzduring the charging stage, which is completely unnecessary.Using the configuration proposed in Table I, the average currentconsumption in active mode in [15] could be 1.1 mA [obtainedfrom (8)] instead of 2.16 mA (Table V).
V. CONCLUSION
This paper has analyzed using both theoretical and exper-imental methods the current consumption of direct interfacecircuits in active mode. The models developed in Section IIIenable us to estimate values of current consumption that fairlyagree with the experimental results obtained using a commer-cial μC (AVR ATtiny2313). The current consumption clearlyincreases with the supply voltage and the operating frequencyof the μC, for example, from about 1.5 mA at 3 V–4 MHz
REVERTER: POWER CONSUMPTION IN DIRECT INTERFACE CIRCUITS 509
TABLE VTIME INTERVALS AND CURRENT CONSUMPTION FOR EACH OF THE OPERATING STAGES IN A DIRECT INTERFACE CIRCUIT
IMPLEMENTED WITH AN MSP430 μC AT 3 V–8 MHz. SUCH A CIRCUIT WAS INTENDED FOR THE MEASUREMENT
OF MAGNETORESISTIVE SENSOR (R0 = 2 kΩ) USING Ri = 120 Ω AND Cd = 1 μF [15]
to 5.0 mA at 5 V–20 MHz when measuring a 1-kΩ resistivesensor. Furthermore, the current consumption is expected to belower for the measurement of capacitive sensors than for themeasurement of resistive sensors since the charging stage isvery short in the former case; the average current consumptionwhen measuring capacitive sensors can be approximated to theinternal current (Iint2) consumed by the μC during the dis-charging stage. This paper has also proposed some guidelinesfor the configuration of the embedded resources of the μC inorder to have a low current consumption in active mode. Webelieve that such analyses and proposals can be useful for thoseinterested in applying direct interface circuits to low-powermeasurement systems, such as autonomous sensors.
REFERENCES
[1] R. Pallàs-Areny and J. G. Webster, Sensors and Signal Conditioning,2nd ed. New York: Wiley, 2001.
[2] J. H. Huising, “Smart sensor systems: Why? Where? How?” in SmartSensor Systems, G. C. M. Meijer, Ed. Chichester, U.K.: Wiley, 2008,pp. 1–21.
[3] D. Cox, “Implementing ohmmeter/temperature sensor,” Microchip Tech-nol. Inc., Chandler, AZ, AN512, 1994.
[4] L. Bierl, “Precise measurements with the MSP430,” Texas Instrum.,Dallas, TX, 1996.
[5] R. Richey, “Resistance and capacitance meter using a PIC16C622,” Mi-crochip Technol. Inc., Chandler, AZ, AN611, 1997.
[6] F. Reverter, J. Jordana, M. Gasulla, and R. Pallàs-Areny, “Accuracy andresolution of direct resistive sensor-to-microcontroller interfaces,” Sens.Actuators A, Phys., vol. 121, no. 1, pp. 78–87, May 2005.
[7] E. Sifuentes, O. Casas, F. Reverter, and R. Pallàs-Areny, “Direct interfacecircuit to linearise resistive sensor bridges,” Sens. Actuators A, Phys.,vol. 147, no. 1, pp. 210–215, Sep. 2008.
[8] F. Reverter and O. Casas, “Interfacing differential resistive sensors to mi-crocontrollers: A direct approach,” IEEE Trans. Instrum. Meas., vol. 58,no. 10, pp. 3405–3410, Oct. 2009.
[9] J. Courbat, D. Briand, L. Yue, S. Raible, and N. F. Rooij, “Drop-coatedmetal–oxide gas sensor on polyimide foil with reduced power con-sumption for wireless applications,” Sens. Actuators B, Chem., vol. 161,pp. 862–868, 2012.
[10] S. Y. Yurish, “A simple and universal resistive-bridge sensors interface,”Sens. Transducers J., vol. 10, pp. 46–59, Feb. 2011.
[11] F. Reverter and O. Casas, “Direct interface circuit for capacitive humiditysensors,” Sens. Actuators A, Phys., vol. 143, pp. 315–322, 2008.
[12] J. E. Gaitán-Pitre, M. Gasulla, and R. Pallàs-Areny, “Analysis of a directinterface circuit for capacitive sensors,” IEEE Trans. Instrum. Meas.,vol. 58, no. 9, pp. 2931–2937, Sep. 2009.
[13] F. Reverter and O. Casas, “Interfacing differential capacitive sensorsto microcontrollers: A direct approach,” IEEE Trans. Instrum. Meas.,vol. 59, no. 10, pp. 2763–2769, Oct. 2010.
[14] J. Pelegrí-Sebastiá, E. García-Breijo, J. Ibáñez, T. Sogorb, N. Laguarda-Miro, and J. Garrigues, “Low-cost capacitive humidity sensor for applica-tion within flexible RFID labels based on microcontroller systems,” IEEETrans. Instrum. Meas., vol. 61, no. 2, pp. 545–553, Feb. 2012.
[15] E. Sifuentes, O. Casas, and R. Pallàs-Areny, “Wireless magnetic sensornode for vehicle detection with optical wake-up,” IEEE Sens. J., vol. 11,no. 8, pp. 1669–1676, Aug. 2011.
[16] F. Reverter and R. Pallàs-Areny, Direct Sensor-to-MicrocontrollerInterface Circuits: Design and Characterisation. Barcelona, Spain:Marcombo, 2005.
[17] F. Reverter, M. Gasulla, and R. Pallàs-Areny, “Analysis of power-supplyinterference effects on direct sensor-to-microcontroller interfaces,” IEEETrans. Instrum. Meas., vol. 56, no. 1, pp. 171–177, Feb. 2007.
[18] G. C. M. Meijer, “Interface electronics and measurement techniques forsmart sensor systems,” in Smart Sensor Systems, G. C. M. Meijer, Ed.Chichester, U.K.: Wiley, 2008, pp. 23–54.
[19] V. Tiwari, S. Malik, and A. Wolfe, “Power analysis of embedded software:A first step towards software power minimization,” IEEE Trans. VeryLarge Scale Integr. (VLSI) Syst., vol. 2, no. 4, pp. 437–445, Dec. 1994.
[20] J. Rius, A. Peidro, S. Manich, and R. Rodriguez, “Power andenergy consumption of CMOS circuits: Measurement methods and ex-perimental results,” in Proc. 13th Int. Workshop PATMOS, Turin, Italy,2003, pp. 80–89.
Ferran Reverter was born in Llagostera, Spain, onJanuary 4, 1976. He received the B.Sc. degree inindustrial electronic engineering from the Universityof Girona, Girona, Spain, in 1998, the M.Sc. de-gree in electronic engineering from the University ofBarcelona, Barcelona, Spain, in 2001, and the Ph.D.degree in electronic engineering from the UniversitatPolitècnica de Catalunya (UPC), Barcelona, in 2004.
He was a Visiting Postdoctoral Fellow with DelftUniversity of Technology, Delft, The Netherlands,from 2005 to 2007, and with the Imperial College
London, London, U.K., in 2012. Since 2001, he has been with CastelldefelsSchool of Technology (EETAC), UPC, Castelldefels, Spain, where he is an As-sociate Professor in analogue electronics and digital systems. He is a coauthorof the book Direct Sensor-to-Microcontroller Interface Circuits (Marcombo,2005). His research interests are in the field of electronic instrumentation, inparticular, the design of interface circuits for smart sensors.
