Reconfigurable-bandwidth bandpass filter within 10–50%

Published in IET Microwaves, Antennas & PropagationReceived on 18th May 2012Revised on 20th September 2012Accepted on 5th November 2012doi: 10.1049/iet-map.2012.0274

Special Section on Advanced Tuneable/Reconfigurableand Multi-Function RF/Microwave Filtering Devices

ISSN 1751-8725

Reconfigurable-bandwidth bandpass filter within10–50%Miguel Á. Sánchez-Soriano1, Roberto Gómez-García2, Germán Torregrosa-Penalva1,

Enrique Bronchalo1

1Signal Theory and Communications Division, Miguel Hernández University, 03202 Elche, Spain2Department of Signal Theory and Communications, University of Alcalá, Alcalá de Henares 28871, Madrid, Spain

E-mail: [email protected]

Abstract: An original and simple approach to design electronically reconfigurable-bandwidth microwave planar bandpass filtersis reported. It exploits the use of transversal signal-interference filtering sections shaped by a branch-line directional couplerarranged in reflection mode with switchable stubs. Thus, by switching on/off these stubs through p–i–n diodes, sharp-rejection filtering functions with different bandwidths and multiple out-of-band transmission zeros are produced. Analyticalequations and guidelines for the synthesis of the basic switchable filter building block are detailed. Moreover, a 2 GHz two-stage proof-of-concept microstrip prototype, with five distinct bandwidth states ranging from ∼10 to 50%, is built andcharacterised for experimental verification.

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1 Introduction

The latest trends towards the development of advancedfrequency-agile multi-mode radiofrequency (RF)transceivers for next generation telecommunications andremote-sensing applications, such as software-defined radiosand radars, have relaunched the research into reconfigurablehigh-frequency electronics [1, 2]. Microwave tunable filtersare among the most critical and challenging devices in thisframework, to carry out the adaptive frequency-bandselection needed by these systems. Indeed, fullycontrollable bandpass filters in terms of centre frequency,bandwidth and even selectivity are strongly demanded [3].Centre-frequency tunability has been traditionally

accomplished in coupled-resonator bandpass filter networksby modifying the natural frequencies of their resonators. Inplanar technologies, this has been done by insertingvariable-reactance elements in the resonating lines, such asp–i–n, Schottky, varactor diodes or micro-electromechanicalsystems (MEMS) when power-handling capability andlinearity are an issue [4, 5].On the other hand, much less effort has been devoted to

design reconfigurable-bandwidth bandpass planar filters.This is owing to the difficulty involved in finding efficientmechanisms to control the inter-resonator couplings, asa main goal to attain passband-width variation. In [6], atechnique to continuously adjust the bandwidth ina combline filter by placing coupling reducers between itsconstitutive resonators was proposed. However, this circuitis only well suited to process narrow-to-moderate-bandwidth signals and shows low-order selectivity, becauseof the lack of transmission zeros in its transfer function.More recently, some contributions in this field from both

502& The Institution of Engineering and Technology 2013

the conceptual and technological perspectives have beenreported. For example, in [7], a discretely reconfigurable-bandwidth bandpass filter that uses switchable stubs in anin-parallel coupled-line arrangement was presented.Although it exhibits a relative-bandwidth tunability rangingfrom 20 to 50%, the absence of stopband transmission nullsremains as a drawback. A quasi-elliptical multi-mode-resonator tunable filter with a separate control for eachpassband edge was described in [8]. In this case,close-to-passband selectivity is enhanced but for a smallerrelative-bandwidth variation between 22 and 34%.Bandwidth-controllable filters based on periodic structures,ferroelectrics and left-handed materials have also beendevised, albeit with limitations regarding in-band power-insertion losses, linearity and reconfigurable-bandwidth ratio[9–11].In the very last few years, signal-interference circuits have

become a suitable choice for high-frequency filter realisation.In these circuits, the filtering action is derived from theinteractions at the output node between the signalcomponents coming from different signal-propagation pathsof the filter. Signal-interference filters have been applied tobandpass, bandstop and lowpass designs with single/multi-band characteristics. For planar substrates, untypicalfiltering networks, such as in-parallel transmission-linestages, power couplers/dividers and baluns arranged inreflection mode, have been proposed for microwave-to-millimetre-wave bands [12–22]. Signal-interferencestructures have also shown their suitability for activefiltering in recursive, transversal and channelised schemesfor hybrid- and integrated-circuit technologies [23–26].Moreover, electronic reconfigurability has also been addedto them, enabling frequency-agile filter functions to be

IET Microw. Antennas Propag., 2013, Vol. 7, Iss. 7, pp. 502–509doi: 10.1049/iet-map.2012.0274

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performed [26–31]. Modern advances in the development ofefficient tunable notch filters, aimed at the mitigation ofinterfering signals in ultra-wideband systems, are the bestexponent of this trend [31].In this work, a new class of microwave planar bandpass
filter with electronically reconfigurable bandwidth ispresented. It consists of the cascade of two transversalsignal-interference filtering sections shaped by a branch-linecoupler with switchable stubs. Among its main features, itsextremely large passband-width tuning range, from ∼10 to50% in relative terms, must be emphasised. Also, multipleout-of-band transmission zeros are generated at both sidesof the filter passband for all the bandwidth states, thusconferring high-selectivity performances to each tunedtransfer function. Furthermore, unlike some previoustunable-bandwidth filter solutions, its circuit simplicity andsmall in-band power-insertion losses driven by theavoidance of coupled lines should be highlighted.Specifically, it should be mentioned that, in [32], a

frequency-reconfigurable bandpass filter also exploitingsignal-interference principles was described. Nevertheless, itutilises the switching of different-length transmission-linesegments arranged in parallel as a tunable version of thefixed-frequency transversal filter scheme suggested in [13].In the present work, by using branch-line couplers withswitchable stubs, better performances in terms of in-bandamplitude flatness, sharpness of passband cut-off slopes,and bandwidth-tuning ratio are attained. Also, hardwarecomplexity is reduced in this new approach, sincewide-band power dividers and combiners as those utilisedin [32] are fully circumvented.The rest of the paper is organised as follows: the devised

reconfigurable filter-building block and its operatingprinciple are described in Section 2. Here, analyticalequations and guidelines for its theoretical design are alsoprovided. In Section 3, a two-stage microstrip filterprototype at 2 GHz with five electronically switchablepassband-width states is manufactured and tested forexperimental validation. Finally, a summary and the mostrelevant conclusions of this work are set out in Section 4.

2 Reconfigurable filter-building block

Fig. 1 shows the proposed reconfigurable filter-buildingblock. It consists of a branch-line coupler made up bytransmission lines with characteristic impedances Z1 and Z2and electrical lengths of π/2 at the design centre frequencyf0. The branch-line coupler is loaded at its direct andcoupled ports with shunt open ended stubs (#1 and #2) ofcharacteristic impedances Zs1 and Zs2 and electrical lengthsθs1 and θs2, respectively. The basic idea is to switch on/offthe stubs #1 and #2 through the use of p–i–n diodes in

Fig. 1 Schematic diagram of the proposed filter-building block forreconfigurable bandwidth


order to provide two different bandpass filteringconfigurations: the narrowband and the wideband states.

2.1 Narrowband state

When both stubs are on (p–i–n diodes are off) in Fig. 1, theresulting filtering configuration corresponds to a transversalfiltering section as studied in [12] (Fig. 2). This narrowbandstate filter configuration can be designed for a wide range ofpassband bandwidths (from ∼10 to 40%) with feasibledimensions, as seen in Fig. 3. The electrical lengths of thestubs are synthesised to provide a maximum power transferat f0 along with a spectrally symmetrical response. For that,the following relations must be met at f0

us1 = mp

2, us2 = m+ 2n( )p

2m, n [ N (1)

By selecting m = 1 and n = 2, a very selective response withseveral transmission zeros in the stopband is achieved witha moderate size [12]. These numbers lead to θs1 = π/2 andθs2 = 5π/2 at f0. In this state, for Z2 = Z0 (this condition mustbe fulfilled in order to obtain a maximum power transfer atf0), the 3 dB relative bandwidth (BW3 dB) and the separationbetween passband adjacent transmission zeros (Δfz), whichdefines the filter selectivity, are synthesised by adjusting Z1,Zs1 and Zs2, as shown in Fig. 3. The filter performance ofthis state is illustrated in Fig. 4 for some design values. Ascan be seen, the filter response is very selective with highpower rejection levels, although the lower and upperstopbands are not very broad. This latter aspect is addressedin next subsections.

Fig. 2 Schematic diagram of the proposed filter-building block innarrowband configuration

Fig. 3 BW3 dB (full lines) and Δfz (dashed lines) of the narrowbandstate configuration as a function of Zs2, under varied Z1In these design curves, Z2 = Zs1 = Z0


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2.2 Wideband state

In the wideband state, stubs #1 and #2 are off (p–i–n diodeson), forming the filtering structure shown in Fig. 5a. Thiscircuit network is electrically equivalent to the onedisplayed in Fig. 5b, since there is no current circulationthrough the coupler’s branch, which is between the twogrounds. The S-parameters of this configuration can beanalytically calculated as

S11=jsin2 Z2/Z0

[ ]− Z0/Z2[ ]( )+ j Z0/Z1

( )cos2u 2+ Z2/Z1

[ ]( )D

(2)

S21=2sinu

D(3)

where

D=2cosusinu 1+Z2Z1

( )+ j sin2u

Z2Z0

+Z0Z2

( )

− jZ0Z1

cos2u 2+Z2Z1

( )

and θ = π/2( f/f0). From (3), it can be seen that this filteringconfiguration presents transmission zeros (S21 = 0) at fz =2pf0, where p is an integer. Moreover, as previouslymentioned, if Z2 = Z0, a maximum power transfer (S11 = 0)is also obtained in this state at f0 (as well as at odd

Fig. 4 Theoretical response for the narrowband state filterconfiguration, for some values of Zs2In this example, Z1 = 0.6Z0, Z2 = Zs1 = Z0

Fig. 5 Schematic diagram of the proposed filter-building block inwideband configuration

a Full circuitb Simplified equivalent circuit


multiples of f0), as deduced from (2). Regarding thebandwidth features, this state can be designed to presentBW3 dB ranging from ∼30% to ∼140%, as seen in thedesign curves of Fig. 6. Fig. 7 shows the filter responses ofthe wideband state configuration for some design values.The design procedure of the reconfigurable building block

is, therefore, as follows: for a fixed Z2 = Z0, it is found the Z1value that gives the required bandwidth in the wideband state.Once Z1 is known, Zs1 and Zs2 are synthesised in order to meetthe bandwidth specifications of the narrowband state. As bothstates can be designed almost independently, a broad tuningrange can be obtained. For instance, a tuning range of 6:1 isreadily achieved between states with this reconfigurablefiltering structure.

2.3 Additional states for two cascadedreconfigurable building blocks

More states can be obtained by connecting in cascade two ormore reconfigurable building blocks. If two building blocksare used for the design (as seen in Fig. 8), the powertransmission parameter of the whole structure is

S21F = S21AS21B1− S22AS11B

(4)

where A and B refer to the first and second stages,

Fig. 6 3 dB relative bandwidth of the wideband state configurationas a function of Z1, under varied Z2

Fig. 7 Theoretical response for the wideband state filterconfiguration, for some values of Z1In this example, Z2 = Z0


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respectively. For simplicity, it is assumed that both stages aredirectly connected, that is, without the connecting line.As each stage provides two states, the whole filter can

exhibit four different states:

† First state: first stage in narrowband configuration andsecond stage in wideband configuration. In this state,the resulting bandpass filter has the bandwidth (and thepassband performance) defined by the first stage, sincethe second stage has a much wider passband bandwidth(see (4)). Moreover, the second stage rejects the spuriousbands of the first stage, extending the stopband bandwidthof the whole filter.† Second state: first stage in wideband configuration andsecond stage in narrowband configuration. In this state, forthe same reasons as before, the second stage performancepredominates.† Third state: both stages in wideband configuration. Thisstate provides the widest passband bandwidth, with a betterselectivity than a single stage.† Fourth state: both stages in narrowband configuration. Inthis state, the resulting passaband bandwidth is defined bythe stage with a narrower passband bandwidth. The inherentspurious bands of each stage are not rejected; thus, thestopband bandwidth in this state is worse than in the firstand second states, although the filter selectivity is higher.

Thus, in the design procedure, the first stage is designed (innarrowband configuration) to provide the desired narrowestpassband bandwidth, and the second stage (also innarrowband configuration) to produce a passbandbandwidth halfway between the narrowest and the widest.The branch-line coupler of each stage (which defines thewidest bandwidth state, i.e. third state) can be designed tobe identical without hardly losing design flexibility.Following this procedure, the narrowest bandwidth statewill be the first state, the intermediate bandwidth state willbe the second state and the widest one will be the thirdstate. In the proposed two-stage filter, all states can also be

Fig. 8 Schematic diagram of the proposed two-stagereconfigurable filter


designed in an almost independent way (apart from thefourth state).An additional fifth state, with a passband bandwidth much

wider than in the narrower states (first and second) but withless bandwidth than in the third state can be achievedtaking advantage of the two-cascaded-stage filter. To obtainthis additional state, one stage must be set in widebandconfiguration, whereas the other stage is configured as atransversal filtering section but just switching off the largeststub, resulting in the circuit shown in Fig. 9. The newfiltering section, by itself, features a selective response witha wide passband bandwidth, but with spurious bands at twosides of the passband. However, in the whole circuit, thesespurious bands are rejected by the other stage working inwideband configuration, giving an adequate filtering profile.The passband bandwidth in this state, for fixed branch-linecoupler impedances, is controlled by the shortest loadingstub impedance (Zs1A in Fig. 9), as seen in the designexamples of Fig. 10. In this figure, for the design examplewith Zs1 = 0.6Z0, the fractional bandwidth is BW3dB = 52%,whereas for the case with Zs1 = Z0, is BW3dB = 44%.

Fig. 9 Schematic diagram of the additional state in the proposedtwo-stage reconfigurable filter

Fig. 10 Theoretical response for the additional state, under variedZs1In this example, Z1 = 0.6Z0, Z2 = Z0, Zcon = 1.3Z0 and θcon = π/2 at f0


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Fig. 11 Photograph of the implemented two-stage reconfigurable filter

3 Implementation and experimental results

To validate the proposed reconfigurable filter-building block,a two-stage reconfigurable-bandwidth bandpass filter withfive states is implemented and tested. The filter is designedto be centred at 2 GHz on Taconic TLX-8 substrate (relativepermittivity εr = 2.55, thickness h = 382 μm, metal thicknesst = 32 μm and loss tangent tanδ = 0.002.) The p–i–n diodesused for switching on/off the stubs are from NXPSemiconductors (BAP55LX). They have a forward biasedresistance of 1 Ω (at 10 mA), a diode capacitance of 0.25pF and a series inductance of 0.4 nH. The filter is designedin accordance with the guidelines given in Section 2.3.The branch-line coupler in both stages is synthesised withZ1 = 30 Ω and Z2 = 50 Ω. The loading stubs impedances ofthe first stage are chosen as follows: Zs1A = Zs2A = 50 Ω,whereas those of the second stage are: Zs1B = 50 Ω and Zs2B =30 Ω. The connecting line is designed to provide a goodmatching between stages. Its characteristic impedance isselected to be Zcon = 62 Ω and its electrical length θcon = π/2at 2 GHz. The full-wave simulator Sonnet has been used inthe design process in order to take into accountdiscontinuity and diode parasitic effects. Fig. 11 shows thephotograph of the implemented filter. It can be seen thattwo/three diodes are used to switch off the shortest andlargest stubs, respectively. This is to completely neutralisethe stubs when they are off and so, to avoid that they canresonate in the band of interest, degrading the passbandfilter response. The filter operates as follows: the biascircuit #3 controls the second stage (switching betweennarrowband and wideband configurations), whereas the biascircuits #1 and #2 control the first stage (switching amongnarrowband, wideband and the new filtering section


configurations). In this way, the five distinct statescommented in the previous section can be set.Table 1 summarises the simulated and measured

performances of the implemented reconfigurable filter. The‘additional’ state of Section 2.3 is herein called fifth state.The full wave and measured responses are shown inFigs. 12 and 13. The measured results agree very well withthe simulation ones. The implemented filter presents a verywide tuning range of 5:1 (between first and third states), inaddition to intermediate bandwidth states (second and fifthstates). A frequency shift down to lower frequencies can beobserved in the third state. This is due to the fact that inthis state all diodes are on, which slightly ‘enlarges’ thecircuit owing to the parasitic inductance associated tothe diodes. Note also from Fig. 12, for all the tunedcases and just unlike most of currently availablereconfigurable-bandwidth filter solutions, the plurality ofout-of-band transmission nulls produced at both sides of thefilter passband owing to signal-energy cancellation effects,

Table 1 Filter performance

State BW3 dB (%) f0 , GHz IL, dBSim./Meas. Sim./Meas. Sim./Meas.a

first 11/10 1.95/1.95 1.86/2.25second 22/19 1.95/1.95 1.52/1.77third 51/52 1.80/1.82 1.61/1.64fourth 14/11 1.94/1.95 1.36/2.09fifth 38/37 1.90/1.92 1.59/1.66

IL denotes insertion loss at f0.aSMA connector losses are included.


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Fig. 12 Simulated and measured results of the implemented reconfigurable filter

a First stateb Second statec Third stated Fourth statee Fifth state

leading to a sharp-rejection filtering response for eachreconfigured state.For a full characterisation of the reconfigurable filter,

linearity measurements have been also performed. Table 2gives the input third-order intercept point (IP3in) of thedifferent states measured using two tones 1 MHz apart


within the passband with an input power of 0 dBm. As seenin this table, the states with more p–i–n diodes in on-state(third and fifth states, which are also the widest bandwidthstates) present greater IP3in. It seems to be an expectedresult since in filter with active components, the wider thepassband bandwidth, the lower the distortion is [33].


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Moreover, p–i–n diodes, at microwave frequencies, usuallypresent less distortion when they are on [34–36]. Regardingthe measured noise figure, in each state the noise figure isslightly higher than the measured insertion loss (thisincrement of the noise figure with respect to the insertionloss is probably because of the shot noise of the p–i–ndiodes).

4 Conclusion

This paper has shown the potential of signal-interferencecircuits to design electronically controllable-bandwidthmicrowave bandpass filters with a broad tuning range.To this aim, a novel reconfigurable filter block consistingof a branch-line-hybrid-based transversal filtering sectionwith switchable stubs has been devised. This basic filternetwork, by switching the aforementioned stubs by meansof p–i–n diodes, enables filtering transfer functions withvery different bandwidth states to be obtained. Moreover,sharp-rejection performance is attained for all the tunedresponses, as a result of the large number of generatedout-of-band signal transmission nulls. Thus, this solutionoutperforms most of the state-of-the-art in tunablebandwidth filters in terms of selectivity andpassband-width tuning ratio. Design equation and graphsfor the theoretical design of this reconfigurable filterapproach have been expounded. Furthermore, to proveexperimental viability, a 2 GHz two-stage prototype,exhibiting five switchable bandwidths between a moderate( ∼10%) and an ultra-wideband (∼50%) value has beensuccessfully constructed in microstrip technology andtested. Apart from its circuit simplicity, its relativelycompact size, good linearity behaviour and low powerinsertion losses for all the tuned states must be emphasisedas major advantages.

Table 2 Filter linearity performance

State IP3in, dBm

first 24second 30third 32fourth 23fifth 35

Fig. 13 Zoom-in of the measurd results for the five states


5 Acknowledgments

This work was supported in part by the ‘Centro deInvestigación Científica y Tecnològica (CICYT)’ under theSpanish National Projects TEC2010-21520-C04-02 andTEC2010-21520-C04-04, in part by the Regional ProjectUAH2011/EXP-025 and by a research grant from theGeneralitat Valenciana.

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Reconfigurable-bandwidth bandpass filter within 10–50%

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