Signal Processing Requirements for Wimax (802.16E) Base Station

8/10/2019 Signal Processing Requirements for Wimax (802.16E) Base Station

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Signal Processing Requirements forWiMAX (802.16e) Base Station

M SHAKEEL BAIG

Signal Processing Group

Department of Signals and Systems

Chalmers University of TechnologyGteborg, Sweden, 2005 EX018/2005


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Signal Processing Requirements for WiMAX

(802.16e) Base Station(Master thesis)

M Shakeel Baig

Supervisors:Yusuf Jamal; Klas Brink; Rickard FahlqvistAnalog Devices Inc.

Stockholm, Sweden

Examiner:Prof. Mats VibergSignal processing group

Department of Signals and systems

Chalmers University of Technology

Gothenburg, Sweden

2005

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Abstract

802.16e provides specifications for non line of sight, mobile wireless communications in the

frequency range of 2-6 GHz. It is well implemented by using OFDMA as its physical layer

scheme. The OFDM symbol time (s

T ) is to be selected depending on the channel conditions,

available bandwidth and, simulations provide a means of selecting right values of sT in

different channel conditions. Additionally it has been shown that certain values of sT

outperform others in all conditions, thus invalidating their use. Moreover, a solution proposed

by INTEL is also analyzed.

One of the major requirements of OFDM is high synchronization. Detecting the timing offset

of a new mobile user, entering the network, which is not time aligned using cross-correlation

and auto-correlation in time domain and cross-correlation in frequency domain at the base

station has been simulated. Results point that the processing load can be significantly reduced

by using frequency domain correlation of the received data or by using auto-correlationfollowed by cross-correlation on localized data.

The use of adaptive antenna system in 802.16e improves the system performance, where

beamforming is implemented in the direction of desired user. Capons method and MUSIC

method have been simulated to compute the direction of arrival for OFDMA uplink. A new

user, while in the ranging process, transmits data with unknown time offset and unknown

direction. The thesis describes the procedure to find the two unknown one after another.

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Acknowledgements

This thesis was possible because of the will and wish of ALMIGHTY and I am grateful to

Him. Next, I express thanks to my Parents for their support during all these years of my

education.

I am grateful to Analog Devices Inc, Sweden for allowing me to work at their office for the

thesis, and my thanks to its employees for providing an encouraging environment, during my

stay at the office.

I express my sincere thanks to my supervisors, Yusuf Jamal, Klas Brink and Rickard

Fahlqvist, for their guidance and invaluable time spent on those numerous discussions we had

at the office. It is in those fruitful discussions that I learned many more things apart from

thesis itself and I am sure those will be helpful in future. Thanks for everything.

Whenever we did not find any solution, we had one man to call to, Michel Lopez, USA. My

special thanks to him for helping me during the thesis and for the careful reviewing of my

thesis report.

My thanks to Professor Mats Viberg, my thesis examiner, for his valuable ideas regarding

beamforming, the channel model and comments on the thesis report.

Lastly, I thank all my friends who helped me during my stay in Stockholm.

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Contents

ABSTRACT............................................................................................................................IV

ACKNOWLEDGEMENTS...................................................................................................VI

CONTENTS.........................................................................................................................VIII

LIST OF FIGURES ................................................................................................................ X

PART 0: INTRODUCTION.................................................................................................... 1

0.1GOAL OF THE THESIS ......................................................................................................... 30.2REPORT OUTLINE ...............................................................................................................30.3INTRODUCTION TO WIMAX.............................................................................................. 3

0.3.1 802.16c .................................................................................................................................. 3

0.3.2 802.16a .................................................................................................................................. 4

0.3.3 802.16d .................................................................................................................................. 4

0.3.4 802.16e .................................................................................................................................. 40.4OFDMSYSTEM DESCRIPTION ............................................................................................ 5

0.4.1 Effects of Receiver performance............................................................................................ 7

PART 1: CHANNEL ESTIMATION..................................................................................... 9

1.1INTRODUCTION .................................................................................................................. 91.2UP LINK TRANSMISSION .................................................................................................... 91.3THE BLOCK DIAGRAM ..................................................................................................... 10

1.3.1 Transmitter .......................................................................................................................... 11

1.3.2 Channel................................................................................................................................ 12

1.3.3 Receiver ............................................................................................................................... 15

1.4P

ROBLEM DESCRIPTION................................................................................................... 171.5SIMULATION RESULTS ..................................................................................................... 18

1.5.1 Performance in AWGN channel .......................................................................................... 18

1.5.2 Performance in a Rayleigh Fading Channel....................................................................... 20

1.5.3 Effect of mobile speed.......................................................................................................... 23

1.5.4 Performance in presence of Multipath ................................................................................ 24

1.5.5 Effect of multipath delay spread .......................................................................................... 28

1.5.6 Effect of Channel coding ..................................................................................................... 30

1.6DISCUSSION ..................................................................................................................... 32

PART 2: RANGING ..............................................................................................................34

2.1INTRODUCTION ................................................................................................................ 34

2.2PROBLEM DESCRIPTION................................................................................................... 352.3TIMING OFFSET CALCULATION......................................................................................... 35

2.3.1: Cross-correlation in time domain ...................................................................................... 35

2.3.2: Auto-correlation in time domain ........................................................................................ 36

2.3.3: Correlation in frequency domain ....................................................................................... 37

2.4FREQUENCY AND POWER OFFSET CALCULATION ............................................................. 392.5SIMULATION RESULTS ..................................................................................................... 392.6PROCESSING LOAD CALCULATION................................................................................... 42

PART 3: ANTENNA BEAMFORMING ............................................................................. 46

3.1INTRODUCTION ................................................................................................................ 46

3.2BEAMFORMING BASICS .................................................................................................... 473.3TYPES OF BEAMFORMING................................................................................................. 48

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3.4PARAMETERS EFFECTING BEAMFORMING ........................................................................ 503.5ADAPTIVE ANTENNA SYSTEM FOR OFDMAIN 802.16E .................................................. 503.6DIRECTION OF ARRIVAL COMPUTATION (DOA)............................................................... 53

3.6.1 Capons method................................................................................................................... 53

3.6.2 MUSIC algorithm ................................................................................................................ 54

3.7SIMULATION RESULTS ..................................................................................................... 54

3.8CONCLUSION ................................................................................................................... 57A. REFERENCES.................................................................................................................. 60

B. ACRONYMS...................................................................................................................... 62

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List of figures

Figure 0.1: Three orthogonal subcarriers ................................................................................. 1

Figure 0.2: Comparison between FDMA and OFDM [5] ......................................................... 6Figure 0.3: 3 subcarriers and multipath component.................................................................7Figure 0.4: Perfect synchronization, no ICI.............................................................................. 7

Figure 1.1: Time plan from [1] ................................................................................................ 10Figure 1.2: Tile Structure Uplink............................................................................................. 10Figure 1.3: Simulated Doppler spectrum................................................................................. 13Figure 1.4: A typical Rayleigh fading channel ........................................................................ 14Figure: 1.5. The Block Diagram .............................................................................................. 16Figure 1.6: SNR v/s BER for AWGN channel with QPSK modulation .................................... 19Figure 1.7: Response of tail biting convolutional code in AWGN channel ............................. 19Figure 1.8: Performance of 16 QAM modulation in AWGN channel...................................... 20Figure 1.9: Performance in fading channel at 3GHz (without multipath) .............................. 21Figure 1.10: Typical channel at different Doppler frequency (350, 680, 70, 135 Hz) ............ 21Figure 1.11: Performance in fading channel at 5.9GHz (without multipath) ......................... 22Figure 1.12: Performance of OFDM at 2 carrier frequencies ................................................ 23

Figure 1.13: Performance of 128 Sfor different speed of mobile .........................................24

Figure 1.14: Performance of 256 Sfor different speed of mobile .........................................24

Figure 1.15: Channel model .................................................................................................... 25Figure 1.16: Figure showing subcarriers and multipath......................................................... 26Figure 1.17: Performance at 3GHz in fading channel with multi path (guard time Ts/8)....... 26

Figure 1.18: Performance at 5.9GHz in fading channel with multi path (guard time Ts/8).... 26Figure 1.19: Performance for different amplitudes of multipath............................................. 27Figure 1.20: 16QAM in fading channel ................................................................................... 28

Figure 1.21: Performance when multipath delay is limited with in 16 S(guard time Ts/4) .. 29

Figure 1.22: Performance when multipath delay is limited with in 100 S(guard time Ts/4) 29

Figure 1.23: Convolutional encoder [1] .................................................................................. 30Figure 1.24: Convolutional coding performance .................................................................... 31

Figure 2.1: Initial ranging transmission symbol structure ...................................................... 36Figure 2.2: Process of finding peak corresponding to time offset..........................................37

Figure 2.3 frequency domain correlations............................................................................... 38Figure 2.4: Frequency domain correlation using IFFT.......................................................... 39Figure 2.5: Comparison of various correlation techniques..................................................... 40Figure 2.6: Cross-correlation with double precision code and with 2 bit quantized code......41Figure 2.7: Effect of ranging signal amplitude on system performance.................................. 42Figure 2.8: cross-correlation ................................................................................................... 43

Figure 3.1: 2 element linear array. .......................................................................................... 47Figure 3.2: 2 element linear array far field geometry ............................................................. 47Figure 3.3: Plot of Array factor (AF) ......................................................................................48Figure 3.4: Example (non-real) of switched beamforming...................................................... 49

Figure 3.5: Switched beamforming and Adaptive beamforming ............................................. 49Figure 3.6: Downlink part of AAS frame structure.................................................................. 51

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Figure 3.7: AAS diversity map zone [1]................................................................................... 52Figure 3.8: Capons method for DOA computation (DOA of signals from two MSS is not

distinguished). .......................................................................................................................... 55Figure 3.9: MUSIC method for DOA computation, DOA of signals from two MSS is seen.... 55Figure 3.10: Denominator of the MUSIC spectrum (searching for dip) ................................. 56Figure 3.11: Denominator of the MUSIC method. Y-axis limit set to 0 to 1. .......................... 57

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PART 0: Introduction

Wireless communication systems have been in use for quite a long time. Many standards (to

name 802.11, Bluetooth) are available based on which these devices communicate, but the

present standards fail to provide sufficient data rate, when the user is moving at high speed. In

view of this requirement for future mobile wireless communication systems, the present

standard, 802.16e has been proposed by Institute of Electrical and Electronic Engineers

(IEEE). 802.16 (WiMAX) provides specifications for both fixed Line of sight (LOS)

communication in the range of 10-66GHz (802.16c), and fixed, portable, Non-LOS

communication in the range of 2-11GHz (802.16a, 802.16d). Also it defines wireless

communication for mobiles, moving at speed of 125 KMPH, in the range of 2-6 GHz

(802.16e). 802.16e is well implemented with OFDMA as its physical layer scheme, hence

OFDMA is discussed here.

One of the limiting factors in the performance of mobile wireless communication systems is

the Inter symbol interference (ISI), caused by the multipath. In single carrier systems the

symbol duration (for large system capacity) is very small and spans a wide bandwidth in

frequency domain and the multipath arriving at different time instants is spread over multiple

symbols leading to ISI. The complex solution is to implement an equalizer at the receiver to

mitigate the effect of the channel. A much simpler solution is to opt for multicarrier systems,

like OFDM, which transmit low rate data (large symbol time) on several overlapping

orthogonal subcarriers. In addition a guard time is provided (Figure 0.1) at the start of each

symbol. By doing so, the symbol time is made large enough so that the system becomes less

sensitive to multipath.

Figure 0.1: Three orthogonal subcarriers shown separately (in practice a sum of 3 is

transmitted)

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Signal Processing requirements for WiMAX (802.16e) base station.

Multi carrier systems have the problem of inter carrier interference (ICI), due to the loss of

orthogonality between subcarriers. The use of cyclic prefix [5] in OFDM ensures

orthogonality over the receiver window thus avoiding ICI. In a fading channel, an OFDM

system performance is highly degraded and hence channel estimation is done to overcome the

effect of fading. For this, an OFDM system has pilot symbols (on pilot subcarriers) embedded

in between the data symbols (on data subcarriers), which provides the channel information atthe receiver. This channel estimation values at the receiver are interpolated over the data

subcarriers and data symbols are decoded. Much depends on the symbol time, subcarrier

spacing and pilot location in both time and frequency domain as the channel characteristics

should not change significantly between pilot subcarriers, else the interpolation would not be

accurate. The first part of the present thesis investigates the performance of channel

estimation for different symbol times and subcarrier spacing

In any OFDM system, the performance highly depends on synchronization between the

transmitter and the receiver. Loss of timing accuracy leads to ISI and ICI is the result when

there is loss of frequency accuracy. During start of initial ranging (process of establishingsynchronization), the timing offset between the mobile subscriber station (MSS) and the base

station (BS) is more than the round trip delay (RTD), which is quite high and additionally, the

system may have frequency offsets. Another challenge is that the MSS does not know what

power level is to be used for transmission. It starts transmitting with the least power and waits

for a response from the BS; if the BS has received the transmission from the MSS then it

transmits back a ranging response to the MSS. If the transmission is lost then the MSS restarts

the ranging process at a higher power level, which increases interference. In OFDMA system

[7], we use code division multiple access (CDMA) codes to improve the system efficiency in

detecting the new user. A new MSS will transmit this CDMA code, which the BS should

detect. Part two of the thesis looks into the above situation and the amount of interferencecaused by the unsynchronized new user entering the network.

The present demand in the field of wireless communication is not only to provide data

communication when the user is mobile but also to provide high data rate by consuming less

bandwidth (achieve good spectral efficiency). WiMAX, the IEEE standard provides

specification for efficient forward error correction techniques and optional schemes like

adaptive antenna system (AAS), space time coding (STC) and multi input multi output

(MIMO) systems. Of these AAS achieves high system capacity with implementation cost

mainly concentrated at the base station (BS), which can be easily tolerated. Hence is a good

solution for increasing system capacity with least cost. This gives many advantages asreduced interference, increased range and SNR and Space division multiple access (SDMA) at

the cost of some complexity at the transmitter (BS) which is usually acceptable.

A BS, in an AAS network, faces two problems when a new MSS tries to enter the network;

the timing offset is unknown and the direction of arrival is unknown. Part 3 of the thesis

explains the AAS scheme in 802.16e and discusses solution to the above problems, where the

two unknown are found one after another. Additionally, directional of arrival (DOA)

algorithm has been simulated for 802.16e uplink.

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0.1 Goal of the thesis

The aim of the thesis is to get an understanding of the IEEE WirelessMAN standard and

analyze the receiver signal processing requirements for 802.16e BS. The idea is to makecoarse calculation for signal processing load on the DSP TigerSHARC (but this report does

not include any processor specific information) when implementing certain components of the

receiver signal chain.

0.2 Report out line

As the thesis analyzes three different aspects of WiMAX; the report is mainly divided into 3

parts namely Channel estimation, Ranging and Antenna Beamforming. The report start with

brief explanation about the IEEE standard, WiMAX, a general OFDM system and some basicproblems faced when implementing the system. Next in the report, the problem of channel

estimation in a mobile environment and the response of various symbol times are explained.

Next is the description of the total signal chain of OFDMA for 802.16e standard, followed by

simulation results. A small discussion about the symbol times conclude the session.

Next, the report explains the challenges faced by the system when a new user tries to enter the

network. Some methods of overcoming these in 802.16e are discussed, followed by

simulations, processing load calculations and discussions. The last part deals with Antenna

beamforming and the requirements to implement it in 802.16e.

0.3 Introduction to WiMAX

Worldwide Interoperability for Microwave Access (WiMAX) provides specifications for both

fixed Line of sight (LOS) communication in the range of 10-66GHz (802.16c), and fixed,

portable, Non-LOS communication in the range of 2-11GHz (802.16a & 802.16d). In

addition, it defines wireless communication for mobiles, moving at a speed of 125 KMPH, in

the range of 2-6 GHz (802.16e). Support for both time division duplex (TDD) and frequency

division duplex (FDD) SS is provided, both using a burst transmission format whose framing

mechanism supports adaptive burst profiling in which transmission parameters, including themodulation and coding schemes, may be adjusted individually to each SS on a frame-by-

frame basis, thus providing high data rates.

0.3.1 802.16c

Wireless metropolitan area network- single carrier physical layer (WirelessMAN-SC PHY)

specification is targeted for operation in the 1066 GHz frequency band. The BS is essentially

an isotropic radiator, which transmits data (downlink) to all the users designated by their

connection identifier (CID). The subscriber station (SS) shall use highly directional antennas

directed towards the BS. The signal chain for this physical layer, at the transmitter is definedas, randomization, forward error correction (FEC) encoder, symbol mapping followed by

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pulse shaping and transmission. Randomization is done in order to ensure the carrier recovery

at the receiver. Mandatory FEC scheme comprises of Reed Solomon (RS) encoder and Block

convolutional coder (BTC), optional scheme includes parity check codes and convolutional

turbo codes (CTC). Moreover the encoding rate depends on channel conditions and required

bit error rate (BER). Adaptive modulation schemes (Quadrature phase shift keying (QPSK),

16 Quadrature amplitude modulation (16 QAM)) are used for symbol mapping, additionally,64 QAM is provided as an optional modulation scheme.

Application of this standard includes point to point (PPP) and point to multi point (PMP)

microwave communication, interconnection between remote locations and backhaul services.

Implementation cost and time is saved when compared with laying of cables.

0.3.2 802.16a

This part of the WiMAX standard uses single carrier (WirelessMAN-SCa PHY) physical

layer specification, similar to that of 802.16c, except that it is targeted for the frequencybelow 11 GHz and at NLOS. The SS can be personal computers with an external box connect

to an outdoor isotropic antenna [14]. Hence this is fixed NLOS wireless communication.

Support for both TDD and FDD is provided, similar to 802.16c. Since single carrier in

multipath environment is used, a receiver needs to perform efficient channel estimation and

equalization techniques to overcome the multipath effects. Another difference is the

concatenated FEC using RS and pragmatic trellis coded modulation (TCM) (rate

convolutional coding (CC)) with optional interleaving. Optionally to improve the

performance support is provided for BTC, CTC, Adaptive antenna systems (AAS) and space

time coding (STC) are provided.

802.16a devices can be used to provide with T1/E1 level services to enterprises, thus

eliminating wire lines and saving the implementation cost and time. Additionally it can be

used to provide backhaul for hotspots being served by 802.11. Also it can be used in

residential locations to provide broadband internet connections.

0.3.3 802.16d

This is targeted to provide a broadband internet connection to indoor users. The SS operating

on this standard use indoor antenna and a limited mobility (portable devices) is allowed [14].

802.16d uses orthogonal frequency division multiplexing (OFDM) as its physical layerspecification to enable NLOS communication below 11 GHz. Since OFDM is used, the

receiver is made simple by elimination of bulky equalizer. The other features have nearly

been kept similar in all the physical profiles of the standards. FEC includes concatenated RS-

CC followed by interleaving. Similar to 802.16a, AAS, STC schemes are provided but are

kept optional. Variable FFT size and symbol time is specified, which could be fixed

depending on type of environment and allocated bandwidth.

0.3.4 802.16e

Specifications are provided such that mobility of the SS at 125 KMPH is allowed. Orthogonalfrequency division multiple access (OFDMA) is used as the physical layer scheme. Channel

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coding is provided by use of mandatory CC and optional BTC, CTC and low density parity

check codes (LDPC). Data is randomized and interleaved to avoid loss of carrier recovery and

burst errors. In addition to AAS, STC, optional multi input multi output (MIMO) scheme has

been specified. Code division multiple access (CDMA) codes are used along with the random

window length based contention control algorithm for initial ranging, periodic ranging,

bandwidth request and handoff. The inter BS communications have been defined, which willbe used as a backbone network between the BSs to aid the inter-cell mobile subscriber station

(MSS) handoff. This ensures fast and accurate synchronization at the cost of slightly

increased complexity. Similar to 802.16d, variable FFT size and symbol time is provided

which could be set depending on the environment and allocated bandwidth.

Put together, the 802.16 technology would enable the SS to get broadband wireless access

(BWA) at all times in all locations, either when stationary, or at pedestrian speed or when

traveling at 125 KMPH.

Few of the difference between 802.16d and 802.16e are presented here. In OFDM, SS uses allthe available subcarriers for the allocated time, but in OFDMA, user is allocated region

having definition in both time and frequency. The subcarrier mapping is different in both the

standards, resulting in channel estimation done in 802.16d being complex, but done less

number of times. In 802.16e the channel estimation is simple, but more frequently done

(because data considered, per iteration is less Channel is flat only over limited subcarriers).

Another difference is use of CDMA codes for ranging in 802.16e, the receiver performs

correlation to detect the user (read part 2 of the thesis), and hence more processing is

involved.

0.4 OFDM system descr ipt ion

OFDM is a multi carrier transmission scheme where the information is transmitted on

multiple subcarriers, with a lower data rate, instead of one high data rate carrier (Figure 0.1)

and moreover, the subcarriers are orthogonal to each other, leading to saving of bandwidth

(Figure 0.2). The major disadvantage of an OFDM system is its requirement of perfect

synchronization in time and frequency. But the advantages of using OFDM are far more and

provide enough reasons for the popularity of the OFDM systems. A typical channel fade will

degrade only a few of the subcarriers, which in most cases can be compensated by use of

efficient interleaving and channel coding [8]. OFDM systems can be implemented very

efficiently by using the Inverse Fast Fourier transform (IFFT) at the transmitter and FastFourier transform (FFT) at the receiver. The overall complexity and its increase with data rate

in OFDM systems is far less than the single carrier systems [5], hence OFDM is becoming a

widely accepted technology and more prominent to be used in future mobile wireless

communication standards.

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Saving of bandwidth

Figure 0.2: Comparison between FDMA and OFDM [5]

For successful operation of OFDM system, it is required that the subcarriers should neverloose orthogonality between each other at any time. The advantage of an OFDM system is

lost when the subcarriers are no longer orthogonal to each other. This puts forward quite

stringent requirements to be fulfilled by the transmitter and the receiver.

where Tis multiple of =T

dttfft0

0.)2(2sin.2sin f

1 -- (eq. 0.1)

Ideally, to maintain orthogonality we need that the symbol duration be exactly inverse of the

subcarrier spacing and the FFT be considered over symbol duration such that it covers integer

number of cycles. Moreover, the consecutive subcarriers differ by 1 full cycle only (Figure

0.1). If the system is to operate in a multipath environment, then each subcarrier shouldexperience a flat fading, hence the subcarrier spacing should be less than the coherence

bandwidth and each symbol should experience a time-invariant channel, hence the symbol

time should be less than the coherence time else the complexity of receiver increases when

overcoming the fading effect.

Reduction of inter symbol interference, which would require bulky equalizer to be constructed

at the receiver in a single carrier system, is overcome by the use of guard time in an OFDM

system. A guard time is added in time domain between two OFDM symbols and the FFT is

considered over duration such that there is no component from the previous or next symbol,

(Figure 0.3) which nulls the ISI and thus avoiding the bulky equalizer. ISI is completelyeliminated when the multipath signal delay is within the guard time. When designing an

OFDM system proper values are selected depending on the environment so as to satisfy the

above condition. Multi carrier systems have the problem of inter carrier interference (ICI),

which results from loss of orthogonality between the subcarriers. This happens when the FFT

is considered over duration where the subcarrier is not present (non-integer number of cycles),

which would be the case when multipath is present and the guard time has amplitude zero.

This is reduced by use of cyclic prefix [5], where we transmit a copy the last part of the

symbol followed by the symbol itself. This ensures orthogonality over the FFT period in case

of delayed multipath (Figure 0.3).

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Figure 0.3: 3 subcarriers and multipath component shown separately, in practice the signal is

a sum of all subcarriers [5].

0.4.1 Effects of Receiver performance

An unstable and non synchronized local oscillator can cause frequency drift, resulting in FFT

bins being placed such that it samples component from other subcarriers along with the

required, leading to ICI (Figure 0.4 & Figure 0.5).

Figure 0.4: Perfect synchronization, no ICI Figure 0.5: Synchronization loss, result: ICI

OFDM spectrum of 5 subcarriers, vertical line representing FFT bins.

The phase noise from oscillator will cause the subcarrier spectrum to change and even though

FFT bins are placed at right place in frequency domain, with phase noise, we get non-zero

component of other subcarriers, which also results in ICI. Hence the stability of the oscillator

is very much required.

In a mobile fading channel, where the channel varies fast, the performance is highly degraded

and hence channel estimation is to be done to overcome the effect of fading. For this, an

OFDM system has pilot symbols (on pilot subcarriers) embedded in between the data symbols

(on data subcarriers), which provides the channel information at the receiver. This channel

estimation values at the receiver, are interpolated over the data subcarriers and the data

symbols are decoded. Much depends on the pilot spacing in both time and frequency domain

as the channel characteristics should not change significantly between pilot subcarriers, else

the interpolation would not be accurate.

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PART 1: Channel Estimation

1.1 Introduction

A general communication system consists of two blocks, a transmitter and receiver, connected

by a channel. The information transmitted by the transmitter passes through the channel and

then reaches the receiver. If the channel does not distort the transmitted signal, then the

receiver can retrieve the transmitted information successfully, but in practice the channel

alters the transmitted information making the task difficult for the receiver. The main aim of

the designer is to reduce the number of errors made at the receiver. To achieve this,

information is required at the receiver, as to how the channel alters the information, so that the

channel impairments can be mitigated.

When the user is mobile, the channel characteristics do not remain constant for a very long

time. Hence the channel parameters need to be tracked, so that the effect can be mitigated and

reconstruct the transmitted data. This part of the thesis deals with the requirements of Channel

estimation at the Base station (BS) for an 802.16e uplink. Symbol time has an effect on

system performance depending on the channel conditions. Different symbol times are

proposed in [1] and each one has been simulated and compared for various channel condition.

In addition a solution proposed by Intel coop. has also been analyzed. It is concluded that the

performance of the system, for few proposed symbol times, is relatively good in all

conditions.

1.2 Up link Transmission

Any practical standard provides details about receiver and transmitter requirements, operating

details but only the transmitter construction details are provided. Receiver construction and its

performance depends on the algorithms used in implementation and are often left open for

vendors to compete each other. Since this thesis is analysis BS receiver requirements the

uplink part of the system is being simulated a brief description about it is presented here.

The Uplink transmissions (Transmission from the Mobile Subscriber station (MSS) to the BS)

have definition in both frequency and time i.e. the bandwidth allocated to a MSS is defined by

a number of subchannels in frequency domain and a number of slots in time domain (figure

1.1). A subchannel is a combination (non sequential) of subcarriers, and a slot in OFDMA

uplink is defined as 3 OFDM symbols. Another way of representing subchannel is a

combination of 6 tiles. The tile (the smallest data unit (Figure 1.2)) spans for 3 OFDM

symbols in time and 4 subcarriers in the frequency domain. The data is mapped into a tile

structure as shown below.

Horizontal axis represents the frequency domain and the time axis is in vertical direction. 6(or 8 in certain special case) of these tiles form a subchannel, which is the minimum allocated

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transmission region for any MSS, spanning at least a total of 72 subcarriers (6*4 subcarriers *

3 (at least) OFDM symbols). The six tiles in a subchannel are mapped far apart on the total

spectrum (2048 subcarriers), for example tiles use subcarriers 448 to 451; 512 to 515; 984 to

987; 1189 to1192; 1505 to 1508; 1753 to 1756. Moreover the location of the tile structure

changes for every 3 OFDM symbols (due to rotation scheme).

Figure 1.1: Time plan from [1]

Symbol 1

Symbol 2

Symbol 3

Data subcarrier Pilot subcarrier

1 2 3 4

Figure 1.2: Tile Structure Uplink (Mandatory)

Since the subcarriers are far apart in both time and frequency domain except for with in a tile,

the channel estimation is to be done on each tile separately and hence any knowledge or priorestimate about the channel response which could improve the system performance is not

available.

1.3 The Block Diagram

The Block diagram (Figure 1.5) represents the whole system model or the signal chain at base

band. The block system is divided into 3 main sections namely the transmitter, receiver and

the channel. The model has been tested with and without the channel coding (part in doted

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box representing the channel coding and decoding). The bit error rate (BER) plots have been

obtained for at least 2000 errors to get a good confidence limit.

1.3.1 Transmitter

Data Generation:

The data is generated from a random source, consists of a series of ones and zeros. Since the

transmission is done block wise, when forward error correction (FEC) is used, the size of the

data generated depends on the block size used, modulation scheme used to map the bits to

symbols (QPSK, 16QAM), and whether FEC is used or not [1]. The generated data is passed

on to the next stage, either to the FEC block or directly to the symbol mapping if FEC is not

used.

Forward error correction:

In case error correcting codes are used, the data generated is randomized so as to avoid longrun of zeros or ones, the result is ease in carrier recovery at the receiver. The randomized data

is encoded using tail biting convolutional codes (CC) with a coding rate of (puncturing of

codes is provided in the standard, but not simulated here). Finally interleaving is done by two

stage permutation, first to avoid mapping of adjacent coded bits on adjacent subcarriers and

the second permutation insures that adjacent coded bits are mapped alternately onto less or

more significant bits of the constellation, thus avoiding long runs of lowly reliable bits.

Symbol mapping:

The coded bits (uncoded, if FEC not used) are then mapped to form symbols. Modulation

scheme used is QPSK or 16QAM (QPSK unless otherwise specified) with gray coding in theconstellation map. In any case the symbol is normalized so that the average power is unity,

irrespective of the modulation scheme used [1].

Subcarrier allocation:

The subcarrier allocation is mentioned in the section 1.2(Uplink transmission). This separates

data into set of 4 subcarriers for 3 time symbols, named as the tile structure. Symbols are

allocated indices representing the subcarriers and OFDM time symbol, and then passed onto

the next stage, the IFFT, to convert into time domain.

IFFT and cyclic prefix:An N point inverse discrete fourier transform (IDFT) of X(k)is defined as

( ) ( )

=

=1

0

21 N

n

N

knj

ekXN

nx

for n =0,1,.N-1. (eq. 1.1)

From the equation we can infer that this is equivalent to generation of OFDM symbol. An

efficient way of implementing IDFT is by inverse fast fourier transform (IFFT). Hence IFFT

is used in generation of OFDM symbol. The addition of cyclic prefix is done on the time

domain symbol obtained after IFFT. The IFFT size (N value) is considered as 2048 in

11


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simulations. This data is fed to the channel which represents Rayleigh fading channel model

and also implements multipath as shown in block diagram.

1.3.2 Channel

In NLOS wireless communication, the received signal is a combination of many multipath

signals, which are result of reflections from surrounding objects. These multipaths have

different amplitude and phase and may add either constructively or destructively leading to a

complex envelope, i.e. fading. Fading characteristics depend on the channel parameters (rms

delay spread and Doppler spread) and signal parameters (symbol period and bandwidth).

Multipath delay spread leads to time dispersion and frequency selective fading and Doppler

spread leads to frequency dispersion and time selective fading. Any mobile channel is one of

the four mentioned below [2] based on

Based on multipath time delay spread

Flat fading Freq selective fading

BW of Signal < BW of channel [ cs BB >

Delay spread < symbol period [ >>sT ] [


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Two independent Gaussian random sources (a & b) are used to generate the complex

Gaussian random variable (G = a+jb). A filter generated by eq. 1.2 is used to shape it in the

frequency domain. By using an IFFT (r (t) = IFFT (S (f).*G)), we get an accurate time

domain waveform of Doppler fading [2].

Figure 1.3: Simulated Doppler spectrum

Using Smiths method, the system generates time samples of the fading channel. The data is

multiplied in time domain with the fading channel output.

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Figure 1.4: A typical Rayleigh fading channel

Figure 1.5 shows simulated Rayleigh fading channel for the speed of 125 KMPH, and

frequency of 5.9GHz.

Output = fading * input

--(eq. 1.3))()()()(

tsettrtj

= is the transmitted signal)(ts )(t is the amplitude of the fading channel (Rayleigh distributed)

)(t is the phase of the fading channel (uniformly distributed)

According to the standard the maximum supported speed of mobile is 125 KMPH and the

operating frequency range is between 2 6 GHz. The system has been simulated for speeds

30, 80, 125 KMPH and frequency band of 3 GHz and 5.9GHz. Three multipaths were

simulated with uniformly distributed phase. For multipath the amplitude and delay has been

chosen as a random parameter, the first path does not have any excess delay and the amplitude

is scaled by a uniformly distributed number in the range of 0 to 1. The other 2 paths have theiramplitude scaled by uniformly distributed number between 0 to 0.9 and 0 to 0.7. The excess

delay is selected as a uniformly distributed random parameter. Finally additive white

Gaussian noise (AWGN) is added as a last component in the channel.

)/(__

)(*)/()(_

smlightofspeed

HzfrequencysmvelocyffrequencyDoppler d = --- (eq. 1.4)

--- (eq. 1.5)dc fTtimeCoherence /423.0)(_ =

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15

1.3.3 Receiver

The first thing done at receiver (in simulation) is removal of cyclic prefix, thus eliminating the

inter symbol interference (ISI). Data is then passed through the serial to parallel converter of

size 2048 and then fed to the FFT for frequency domain transformation. The signal was

distorted by the channel, to reconstruct the original signal we need information as to how thechannel acted on the transmitted signal so that we can mitigate its effect. This is called

equalization. In an OFDM system, this is done by channel estimation and interpolation. As

we need at least one tile structure (3 OFDM symbols) to detect the data, storage of 3 OFDM

symbols is provided followed by the subcarrier de mapping. The pilot subcarriers are used for

channel estimation and synchronization at the receiver. In the simulation least squares (LS)

estimate has been used for channel estimation at the pilot subcarriers. If is the

transmitted data (known if pilot), Y is the received data, and C is the unknown channel

response, then

)(tD

)(t )(t

)()(*)()( tNtCtDtY +=)t

-- (eq. 1.6)where (N represents the AWGN noise.

The channel can be estimated for known data symbols, i.e. pilot subcarriers as,

)(

)()(

tD

tYtC =

-- (eq. 1.7)

The estimate is simple but is highly affected by SNR or the noise power, as the assumption

made is absence of noise from the receiver power.

This information about channel at pilot subcarriers is interpolated over the whole tile

structure, to recover the data on each data subcarrier (Figure 1.1). Separate one dimensional

linear interpolation has been done for values between two subcarriers (the result: straight

line), hence the performance is not effected much for various one dimensional interpolation

algorithms.

Since we do linear interpolation the channel is assumed to be changing linearly with in thetile, this assumption might not be true depending on the symbol time. This generates a noisefloor at the receiver (Errors are generated due to addition of AWGN noise and due to thisapproximation of fading channel as a linearly varying channel. Beyond a certain value of

SNR, the BER is nearly constant for any further increase in SNR.). For larger symbol time, aswill be seen in simulations, this noise floor is reached at lower SNR, hence results in poor

performance.


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Figure: 1.5. The Block Diagram

Subcarrierallocation

Modulation scheme

(QPSK, 16QAM)

Interleaving

Convolutional

coding

Randomization

Data bits Cyclic prefixS/

P

IFFT

P/

S

fading

AWGN

fading

fading

delay

delay

ChannelEstimation

Interpolation SymbolDemappingS/

P

FFT

P/

S

Remove cyclic

prefix

Subcarrierdemapping

Timing, Frequency andpower offset detection


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1.4 Problem descript ion

For a mobile fading channel with the specifications given in the standard [1], the maximum

Doppler frequency would be around 700Hz and the corresponding Coherence time of cT =600 S . Due to scalability, the useful symbol time period is not a constant value and in some

cases (FFT size = 2048; Bandwidth of total channel = 1.75 MHz) is 1024 S. This being far

beyond the coherence time and results in fast fading (channel changes with in symbol

duration), which is difficult to track leading to poor performance.

An urban environment can suffer from a RMS delay spread ( ) of 10-25 S[2]. This would

relate to a Coherence bandwidth of 8 KHz (50% frequency correlation 1/ (5* )). The

frequency spacing between subcarriers is in some cases (FFT size = 2048; BW of total

channel = 28 MHz) is as large as 15.625 KHz; more than the coherence Bandwidth. And the

symbol will experience a frequency selective fading instead of flat fading.

In [3], INTEL corp. addresses this issue by keeping a fixed subcarrier spacing of ~11.1KHz

corresponding to an OFDM useful symbol time (usually ~1/ s ) of 89.6B S. With these

values the coherence time will span for around 6 ODFM symbols (worst case) in time domain

thus making it slow fading. The exact relation between coherence BW and the rms delay

spread is a function of channel impulse response and applied signal [2]. In [4] the required

pilot spacing for successful interpolation in time and frequency domain is given as

Maximum excess delayf

s

N

T=max -- (eq. 1.8)

is the pilot spacing in frequency domainfN

With = 20max Sand = 100sT S(symbol time including guard) we get = 5fN

For time interpolation Doppler freq should be less than

)1(2

1max

+=

st

DTN

F -- (eq. 1.8)

is OFDM symbol duration including guardsT

is Guard interval factor is Pilot spacing in time domaintN

With maxD = 700 Hz, sT = 100F S, we get min pilot spacing in time domain should be 5.7,

which is well satisfied for 89.6 Sof symbol time [3] but not all the values mentioned in the

standard (64 S ,128 S ,256 S,512 S,1024 S- ETSI)[1].

Even though the proposal from [3] works well theoretically, the performance is not as

expected, this along with all proposals from [1] are investigated by means of simulation.

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1.5 Simulation resul ts

The simulations have been made for various symbol times mentioned in the IEEE standard [1]

and the Intels recommendation [3] to see the effect of fading, multipath, operating frequency

range, and mobile speed, on system performance. The hierarchal structure given belowsummarizes all the simulations being done.

Scalable OFDMA

IEEE standard INTEL recommendation

Check Fading Multipath Modulation Fading Multipath Modulation

Vary Speed RMS delay QPSK Speed RMS delay QPSK

Frequency Amplitude 16QAM Frequency Amplitude 16QAM

Simulation settings:

Symbol time: 64 S, 128 S, 256 S , 512 S, 1024 S- ETSI

Guard time: 1/4, 1/8, 1/16, 1/32 of symbol time.

Frequency: 3, 5.9 GHz (2-11 GHz specified in [1])

Speed of mobile: 125 KMPH (peak)

Modulation: QPSK, 16-QAM.

FFT size: 2048

Assumptions

Power in guard time is not considered.

1.5.1 Performance in AWGN channel

The system model has been tested for QPSK and 16 QAM modulations with an AWGN

channel and the simulation results are shown in the Figure 1.5 and 1.7 respectively. It isconvincing to see that the theoretical and the simulation results overlap.

Please note that the signal power in the cyclic prefix is not considered while simulating. If

considered the performance degrades by around 0.96 dB for cyclic prefix of and around 0.5

dB for cyclic prefix of 1/8.

The theoretical curve is given by [2] as

=

o

b

e

N

EQP

2where is the energy per bit. --(eq. 1.9)bE

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Figure 1.6: SNR v/s BER for AWGN channel with QPSK modulation

Channel coding improves the performance significantly. The next simulation was done for

AWGN channel with QPSK modulation scheme with rate tail biting convolutional code

(G1 = 171; G2 = 133).

Figure 1.7: Response of tail biting convolutional code in AWGN channel

Similarly the theoretical curve (symbol error rate) for 16 QAM system is given by

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=

o

eN

EQP min

2*3 where is the energy per bit (minimum) -- (eq. 1.10)minE

Figure 1.8: Performance of 16 QAM modulation in AWGN channel

1.5.2 Performance in a Rayleigh Fading Channel

A Rayleigh fading channel has been simulated and the data is passed through it, followed by

addition of AWGN noise. A carrier frequency of 3GHz is considered with an FFT size of

2048. The guard time assumed was 1/8 times the symbol period. QPSK modulation will be

used in all further simulations (unless otherwise specified).

Figure 1.9 shows the simulation results for a fading channel at carrier frequency of 3GHz and

at a mobile speed of 125 KMPH. A single multipath channel has been considered for thissimulation plot.

Since there is no excess delay spread the only parameter affecting the graphs is the Doppler,

and as described in the problem description the system with smallest symbol time will

experience the most flat channel, as most of its symbols are well with in the coherence time.

The simulation results very well corroborate it.

The system performs best when the symbol time is 64 S, and the performance gradually

reduces as the symbol time is increased.

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Figure 1.9: Performance in fading channel at 3GHz (without multipath)

The variations in Channel envelope are dependent on the carrier frequency and the mobilespeed

)/(__

)(*)/()(_

smlightofspeed

HzfrequencysmvelocyffrequencyDoppler d = -- (eq. 1.11)

These variation are shown in figure 1.10, the Doppler is around 350, 680, 70, 135 Hz

Figure 1.10: Typical channel at different Doppler frequency (350, 680, 70, 135 Hz)

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Keeping all the parameters constant, we run the simulation for a carrier frequency of 5.9GHz.

Since the carrier is much higher, the Doppler increases, fading becomes fast and we get

degradation in the performance, which is shown in figure 1.11. Additionally, the response of

convolutional coding on the system with symbol time of 128 S and 256 Shas been shown.

It indicates that the response of channel coding is also dependent on the symbol time and maydiffer significantly.

The results at two frequencies become much clear when plotted together, in figure 1.12. We

see that the performance degradation is more significant for symbol time 256, 512 and

1024 S . The reason is that for these values the tile structure is not within the coherence time

of the channel.

Figure 1.11: Performance in fading channel at 5.9GHz (without multipath)

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Figure 1.12: Performance of OFDM at 2 carrier frequencies (dotted line for 3GHz and solid

line represents 5.9 GHz)

1.5.3 Effect of mobile speed

To see the effect of mobile speed on system performance the simulations were made by

keeping the system parameters same as before at frequency 5.9GHz but at different speeds 30,

80 and 125 KMPH. By changing the mobile speed, Doppler changes and the coherence time

of the channel is changed according to equation (eq. 1.4) and (eq. 1.5). For low speed, the

channel remain flat for much larger time (figure 1.10) hence the performance improves for

certain values of symbol time. This is shown for 128 and 256 S in figure 1.13 and 1.14

respectively.

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Figure 1.13: Performance of 128 S for different speed of mobile

Figure 1.14: Performance of 256 Sfor different speed of mobile

1.5.4 Performance in presence of Multipath

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Multipath delay spread leads to time dispersion and may result in frequency selective fading

depending on the subcarrier spacing (figure 1.16). To maintain orthogonality between

subcarriers, the subcarrier spacing is set as 1/symbol time (excluding guard time). Two

multipaths, in addition to one used earlier have been considered in the simulation. The delay

has been introduced as a random parameter (uniform distribution) is within the guard interval

and the amplitude was scaled by a random parameter (uniform distribution) between 0 to 0.9,for the second multipath and 0 to 0.7 for the third multi path, relative to the main path (figure

1.15). Simulation was also run with the amplitude scaling of 0.1 and found that the effect of

multipath becomes insignificant, as long as it is within the guard time. Figure 1.17 and 1.18

shows the simulation results for multipath Rayleigh channel at 125KMPH and two different

frequency bands 3GHz and 5.9GHz respectively. In figure 1.19 we see the effect of multipath

amplitude on the bit error rate.

Figure 1.15: Channel model

Rayleighfading

simulator

Rayleighfading

simulator

Rayleighfading

simulator

1

2

From

transmitter

a1 U (0,1)

a2 U (0,0.9)

a3 U (0,0.7)

To receiver

AWGN

One of the causes for performance degradation results from loss of orthogonality due to Inter

carrier interference (ICI), but by the use of cyclic prefix this can be avoided and as long as the

maximum excess delay is with in the cyclic prefix, Inter symbol interference (ISI) can be

avoided. In the simulations these conditions have been satisfied.

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Figure 1.16: Figure showing subcarriers and multipath separately, in practice a combination

is transmitted.

Figure 1.17: Performance at 3GHz in fading channel with multi path (guard time Ts/8)

Figure 1.18: Performance at 5.9GHz in fading channel with multi path (guard time Ts/8)

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The performance at Ts= 64 Sis the best if we have a guard time ofTs/4. Even though the

multipath delay spread is within the cyclic prefix (theoretically no ISI and ICI), but still when

compared with the case of no multipath, we find a degradation in performance of the system.

This is due to the different phase offset on different subcarriers, resulting form multipath [5].

Figure 1.19: Performance for different amplitudes of multipath. Amplitudes are attenuatedrelative to the main path

At the receiver we get an added version of pure sine waves (delayed) on each subcarrier. The

addition does not destroy the orthogonality, because we do not consider the cyclic prefix for

samples in FFT period (figure 1.16),but the addition results in different phase shifts on each

subcarrier [5]. Different phase shift on pilot and data subcarriers and due to the interpolation

scheme being used to equalize the symbol, the phase shift results in performance degradation.

This is seen in figure 1.19 for a time period of 256 S .

Figure 1.20 gives the performance of 16QAM in a Rayleigh fading channel with multi path.The Doppler is ~680 Hz, and multi path is within the guard time, limited to 8 S. It should be

noted that the plot shows bit error rate versus SNR per bit.

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Figure 1.20: 16QAM in fading channel

1.5.5 Effect of multipath delay spread

So far we see that the symbol time of 64 Shas performed the best, but it should be notedthat this value cannot be used in channels having maximum excess delayof more than 16 S ,

else ISI would result. Table 1 gives a summary of various symbol time durations (ETSI [ 1])

and their corresponding maximum guard time ( times symbol time).

Symbol time inmicro sec

Maximum guard time in microsec (1/4*symbol time)

64 16

INTEL 89.6 22.4

128 32

256 64512 128

1024 256

Table 1.1: Symbol time and maximum guard time

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Figure 1.21: Performance when multipath delay is limited with in 16 S (guard time Ts/4)

Figure 1.22: Performance when multipath delay is limited with in 100 S (guard time Ts/4)

In figure 1.21 we find that, when the delay is limited to 16 S , the system with a symbol time

of 128 Sperforms the best. Hence it should be the obvious choice in environment where the

maximum excess delay profile is around 16 S . Similarly Figure 1.22 shown simulation

results at 100 Sand here also we find system with symbol time of 128 Sperforms best.

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The value of the delay is a characteristic of the channel and varies in different environments.

Hence selection of the symbol time is to be done depending on the channel, also its worth to

note that longer the guard time, more is the power wasted.

In [2], the rms delay spread is given to have a value of maximum in urban environment of

25 S, and the maximum excess delay can be 2 4 times the rms delay spread [5]. Hence byconsidering the worst case we get a maximum excess delay of 100 S(25*4). Figure 1.22

gives the system performance at this value, we see that the performance of the system with

useful symbol duration of 128, 256 Sis better compared to other symbol durations. In case

of low Doppler and same excess delay spread, system with symbol duration of 256 S

outperforms all other systems.

From the results, with simulations done at a mobile speed of 125 KMPH, we can infer that the

system with a useful symbol time of 512, 1024 Sperform worse in all cases when compared

with useful symbol time of 64, 89.6, 128, 256 S. Hence there is no point in using these

values as the useful symbol durations. Moreover 64 Sperforms better in case of low excessdelay spread, 128 S performs better in channel with high Doppler and high excess delay

spread and 256 S is optimal in channels having low Doppler and high excess delay spread.

The solution from Intel, about the use of fixed symbol time of 89.6 Sworks well only in few

channel environments, hence is not a good solution.

1.5.6 Effect of Channel coding

In practice we implement channel coding, to get a better performance (reduced BER at low

SNR). 802.16e specifies tail biting convolutional coding scheme as mandatory, forimplementation by all the devices compliant with the standard. Optionally, it provides

specification for zero terminated convolutional codes, block turbo codes, block convolutional

codes and low density parity check codes [1] [7].

Figure 1.23: Convolutional encoder [1]

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There are three possibilities of implementing convolutional codes [20] based on boundary

conditions and initial state of encoder/decoder. One possibility is that the encoding,

transmission and decoding is continuous and goes on indefinitely. But, 802.16e users are not

transmitting data continuously. Second possibility is that the encoder operates on a block of

data, starts and ends in the same state, known to the decoder. These are the most commonly

used type of convolutional codes with the known state being all zero state. The thirdpossibility is that the encoder operates on a block of data, encoder and decoder start and end

in the same state, but the state is unknown to the decoder. Tail biting convolutional codes fall

under the third category and are generated by making the encoder start state and end state

same, which is given by the last bits to be encoded. As the encoder starts with an initial state

being its last bits, it is referred to as tail biting.

The zero terminated convolutional codes start and end in all zero state, and to end in an all

zero state we require to flush the memory of the encoder by feeding extra zeros into to

encoder (we transmit extra bits). For 802.16e this will be 6 bits (constraint length 7) per block

(figure 1.23). By using tail biting convolutional coding this can be avoided and thus saving 6bits per block of transmission. The result is that the decoder becomes complex as it does not

know the initial state. In Viterbi decoding, no matter what state we start in, as we move along

in the trellis, the path gets converged to the desired path. This fact is utilized at the decoder

and the viterbi decoder is made to run on the input data in a cyclic manner. First we feed the

decoder with the last few bits (equal to traceback length), followed by the original data and

finally first few bits (equal to traceback length). By the time the decoder starts decoding

original data it has converged to the desired path and initial state.

Figure 1.24: Convolutional coding performance

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Tail biting convolutional codes have been simulated in the thesis. The specified encoder is a

rate coder with constraint length 7, G1 = 171, G2 = 133 (figure 1.23). Additionally,

puncturing capability is provided to achieve rates of 2/3, 3/4 and 5/6 (simulation was done for

rate only). Data is transmitted in blocks of variable length (6 to 36 bytes), as specified in

[1]; block length depending on the modulation scheme (QPSK, 16QAM, 64QAM), encoding

rate and the concatenation rule being used. Block length of 36 bytes was used in thesimulations.

Simulation was done at 2 different Doppler frequencies for systems operating using 128 and

256 S symbol times. Rayleigh fading channel with multi path limited to 16 S has been

used and the system has a guard time of 32 S. As seen from figure 1.24, the system

performance with symbol time of 128 Sdoes not change much with Doppler changes when

compared with system at symbol time of 256 S. Due to complexity and time restrictions,

simulations were limited to system operating on the above two symbol times.

1.6 Discussion

In [1] it is proposed that for a FFT size of 2048, to get different bandwidths, one should use

symbols with different time durations. But, as we see, this does not result in good

performance in all cases. In [3] the use of a fixed time period is proposed and to get variable

bandwidth the size of FFT is to be varied i.e. scalable FFT. The solution seems to be good,

but the value of symbol time used might be worth discussing.

The coherence time and the coherence bandwidth of the channel, form the major parameters

influencing the value of the symbol time. We need to have a flat channel over the subcarriersbandwidth in frequency domain and 3 subcarriers in time domain. Also from the calculations

we find that the coherence time for the maximum speed and the highest carrier frequency is

600 S. Hence in time domain the OFDM symbol including guard band should have a span

of not more than 200 S(600/3). The recommendation of 89.6 Sin [3] well satisfies this

limit. This is also satisfied for a symbol time of 128 Sin some cases.

As mentioned in [2] the worst case rms delay spread ( ) in urban environment can be

25 S, which translates to a coherence bandwidth ( 50% correlation) of 8 KHz.cB

5

1=

c

B -- (eq. 1.12)

In an OFDM system to achieve orthogonality we need to have symbol time (excluding guard

time) as multiple of 1/subcarrier spacing. Hence for 89.6 Swe need to have a subcarrier

spacing of 11.16 KHz and for 128 S we require 7.81 KHz. This shows that 128 S is

theoretically a better option than 89.6 S. Also to remember is that 128 S is more effected

by fading than 89.6 Sand in practice fading is more dominant and the maximum rms delay

of high value is seen in very few environments (urban). By observing figure 1.18, 1.21 and

1.22, we can fix the symbol time as either 64 or 89.6 or 128 Sdepending on the rms delay

profile of the environment [2].

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PART 2: Ranging

2.1 Introduction

Total synchronization of an OFDM system is a very important criterion which should be

fulfilled to avoid any interference (ISI and ICI) leading to performance degradation. In

OFDMA it is required that all transmissions from various mobile subscriber station (MSS)

should arrive at the base station (BS) at the same time. Imagine a cell size of 20 Km. we, then

have a maximum round trip delay (RTD) of around 133.3 S. This means that, instead of

arriving at expected time at BS, data may arrive anytime within 0 to 133.3 S of delay. If the

symbol duration is 64 S, the amount of error in detection of data at BS is very high! The

whole network should be synchronized to one reference and in WiMAX this reference is theBS clock. All data is expected to arrive at the same time at the BS receiver and all data

addressed to the MSS is transmitted at same time. The MSSs, which are distributed all over

the cell, receive data at different instant of time and similarly transmit data at different instant

of time depending on their distance from the BS. A MSS at the cell boundary receives data

quite late and transmits data very early when compared with MSS close to the BS.

When a new MSS is seeking entry into the network, its distance, with reference to the BS, is

not known hence the RTD is not known. The MSS does not have any idea as to what time or

power should be used for transmitting the initial signal. This is the BSs job to detect this new

MSS, find the misalignment between the new MSS and the network, and then send response,to correct it.

Reducing system complexity without compromising with performance has been the major

focus of the testing various methods in this part of the thesis. It has been shown that a better

approach is to use frequency domain correlation by using IFFT, which is very simple and

efficient in implementation.

The below section starts with a brief description of this problem and how 802.16e system

handles this situation. Next the report explains the calculation of timing, frequency and power

offset. Major focus is to reduce the complexity of the system, and still maintain the systemperformance at an acceptable level. Methods to estimate the timing offset, using both time and

frequency domain correlation have been explained and later corroborated with simulation

results. It is shown that, in time domain, the complexity of implementing a full cross-

correlation is very high and can be significantly reduced if the CDMA code at the receiver is

quantized and represented using just 2 bits.

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2.2 Problem Descript ion

An OFDM system performance highly depends on synchronization between the transmitter

and the receiver. When a new SS or MSS is trying to enter a network, it is not synchronized.

Hence it tries to achieve coarse synchronization by listening to the transmissions from the BS,

and then starts transmitting to achieve fine synchronization. The MSS starts transmission by

the least possible power, each time increasing it by a level, if nothing is heard back from the

BS. The BS is to detect the new MSS and calculate the time offset, frequency offset and

power offset, then reply back to the MSS to correct its transmitting parameters before

transmitting data. The process goes on (maximum 16 number of times) until the MSS has

achieved synchronization. This process of obtaining synchronization and logging onto the

network is known as initial ranging.

The BS requires that all the signals received at the BS be time synchronized, irrespective of

the source location in the cell. During start of initial ranging (process of establishing

synchronization) the timing offset between MSS and BS can be more than the RTD, which is

quite high. The subcarriers carrying data from the new MSS might have frequency offset and

are delayed (compared with signal from other MSS) causing loss of orthogonality over the

FFT period, hence resulting in ICI. Moreover if the new MSS uses more power, its probability

of getting detected is more, but it leads to increased interference to the data on other

subcarriers. Hence the requirement is to detect the new MSS at BS with the least possible

power.

In 802.16e OFDMA system [1], [7], code division multiple access (CDMA) codes are used toimprove the system efficiency in detecting the new user. A new MSS will transmit one of the

predefined CDMA codes, which should be detected at the BS. The BS is not only to detect the

new MSS but also to calculate its timing, frequency and power offset. Power offset can be

detected just by calculating the difference between the required power and the received

power. Next section describes in detail the process of recovering the timing and frequency

offset information and the causes for performance degradation.

2.3 Timing offset calculation

2.3.1: Cross-correlation in time domain

The CDMA code (defined in frequency domain) which is used for ranging is defined such that

it has very good autocorrelation properties. This fact is harnessed in detection of the timing

offset. The received signal in time domain is stored and cross-correlation is performed

between the received signal and different predefined CDMA codes (to know which code was

transmitted) in time domain. For the same code we get a correlation peak, index of which

gives the time offset. In presence of multipath, we get many peaks corresponding to each

multipath signal.

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Data is represented by using double precision in MATLAB, this is correlated with double

precision representation of the CDMA code (in time domain) to find the correlation peak

corresponding to the time offset (eq. 2.1).

( ) ( )= +=n

kkmxkcmR

11)( tm ,......3,2,1= -- (eq. 2.1)

Where R is cross-correlation outputc is the code

x is the received data

n is the length of code

t is length of x n

The amount of processing required to achieve is quite high for the digital signal processor

(DSP) also the BS has a limited time to process this information as the MSS is expecting

ranging response from the BS. Instead, another way is to quantize the CDMA code in just twolevels (consider sign bits only) and correlate it with the received data. Of course the

performance is degraded and to overcome it, the received signal strength of the ranging MSS,

at the BS, is required to be slightly more.

2.3.2: Auto-correlation in time domain

The time offset can be very large depending on RTD, and this corresponds to a lot of samples

to operate on. Instead of running cross-correlation over all the available samples, a much

faster method is to locate the probable time of transmission of the signal and then crosscorrelate this localized data with different codes. To locate the probable time of transmission

auto-correlation (process is described below) of the received data is performed, this will result

in a peak or a plateau at coarse estimate of the location on time axis. But for the process to

give satisfactory results, it is required that the received signal strength at BS from the ranging

SS be very high, this might cause interference to other users on data subcarriers.

Figure 2.1: Initial ranging transmission symbol structure (figure 240 from [1])

The process is made possible only because of the repetitive structure in which the data istransmitted. Note that the second symbol does not have a cyclic prefix but a post fix! This fact

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is utilized here. It is performed by considering few samples (one OFDM symbol) at a time

(window a) and multiplying it with the same size consecutive data (window b) and computing

the sum over it (Figure 2.2). This represents the correlation value at that instant of time.

Similar process is repeated over the entire data, by sliding the window one sample every time.

Correlation output starts increasing form when the window a goes over the cyclic prefix

(figure 2.1). When the window a hold data representing the code and window b hold thesame, the multiplication and sum produces a high peak (peak remains for a period of 2 *

cyclic prefix) representing the time offset of the user (eq. 2.2). After this peak the correlation

output starts to fall.

( ) ( )+

=

+=1

)(Nm

mk

kNxkxmR tm ,......3,2,1= -- (eq. 2.2)

Where R is cross-correlation outputx is the received dataN is the length of code

t is length of x 2n

Figure 2.2: Process of finding peak corresponding to time offset

The above procedure can be simply represented as one time correlation over the total window

and the next step is nothing but one subtraction (of the first value) and one addition (of the

last value) to the previously obtained value. Thus the calculation is significantly reduced.

After locating the time offset plateau, cross-correlation is performed between the localized

data with different codes. One single code will have a good cross-correlation value and we gettwo correlation peaks because of repetition of the code twice as in Figure 2.1. This is the

bases for calculation of frequency offset.

2.3.3: Correlation in frequency domain

The received signal is a combination of data from synchronized users and CDMA code from

the ranging subscriber. Until now we considered correlation in time domain, where the

correlation is performed over the entire received signal. Timing offset can also be found by

performing the correlations in frequency domain. The advantage gained is less interference, as

Received data

TimeWindow a Window b

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we can separate the data subcarriers from the subcarriers used for ranging. A time offset in

time domain corresponds to phase offset in frequency domain.

-- (eq. 2.3))()(jwjwnDFT

o eXennxo

Delay

Figure 2.3 frequency domain correlations

The detection of timing offset constitutes the following steps; the ranging subcarriers are

separated from data subcarriers by taking the FFT over the received signal. The data on the

ranging subcarriers are provided with different phase shifts (0 to 2 ) and correlated every

time with the CDMA code (in frequency domain), to check for the peak. The number of steps

in between 0 to 2 provides us with the resolution in detection of timing offset. For detecting

integer sample delay, the number of steps should be equal to size of the FFT. As shown in the

figure 2.3, the FFT is considered over the received data and since there is a delay involved,

only one of the three FFTs would correspond to the circularly shifted code. It is this FFT

output, that gives us the timing offset value (second FFT in figure 2.3). Then we multiply itwith phase component to get the required phase shift corresponding to integer number of

samples in time domain, finally we correlate it with the CDMA code. Simulation results prove

that this system works very well because of reduced interference from the data subcarriers,

but the complexity is very high because of the high number of multiplications required to

achieve the phase shifts.

Instead of providing phase shifts by multiplication of phase component, a very simple method

is to take an IFFT after multiplication of code in frequency domain. IFFT is equivalent to

providing phase shift (figure 2.4), hence reduces a lot of complexity.

FFT window FFT window FFT window

CDMA code CDMA codeRx data

FFTSignal in

)( 0nnx )(/2 0 kXeNknj

=

=1

0

/2)(1

)(N

n

NknjenxN

kX

=

=1

0

/2)(1

)(N

k

NknjekXN

nx

IFFTFor 0nn=

we get a peak.)()(/2 0 kXkXe

Nknj

CDMA code )(kX

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Figure 2.4: Frequency domain correlation using IFFT.

Of all the methods investigated here, the frequency domain correlation method mentioned

above, turns out to be the simplest in terms of complexity. Moreover the performance, as seen

from the simulation results is better. Hence, the frequency domain correlation method is a

better choice for detecting the time offset of the ranging MSS.

2.4 Frequency and Power of fset calculation

The phase of the correlation output in time domain is equal to the phase drift between samples

that are sizeFFTTimeSymbol __ seconds apart. Hence frequency offset can be obtained by

dividing correlation phase by T2 [5].

-- (eq. 2.4))()()( oo wwjDFTnjw eXnxe

When the MSS achieves coarse synchronization, it decodes the UCD message which contains

information as to the maximum power that the BS can receive and the power which was

transmitted by the BS. The MSS calculates the Received signal strength and computes the

losses in the channel and calculates the maximum power that it can use for transmitting the

ranging request (CDMA code). After ac

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Signal Processing Requirements for Wimax (802.16E) Base Station

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