7/27/2019 IEEE a 2.4 GHz Radio Solution
1/1170 International IC Taipei Conference Proceedings
A 2.4 GHz radio solutionfor Bluetooth and wireless
home networking
AbstractA complete radio transceiver solution for the 2.4GHz ISM band
demonstrating a high level of integration along with reduced
component count and board size has been constructed. A single
conversion receiver and a direct conversion transmitter have
been fabricated on a single integrated circuit (IC). The archi-
tecture and functionality of the system blocks are described in
detail. In addition, the performance of the transceiver is evalu-
ated for several emerging wireless data and voice standards in-
cluding Bluetooth.
The first section of the paper concentrates on the system
requirements of the transceiver as used in the 2.4GHz ISM bandincluding Bluetooth, HomeRF, and upbanded DECT standards.
The subsequent sections will present details of the various cir-
cuit blocks in a complete 2.4GHz ISM transceiver solution and
the enabling BiCMOS transceiver IC. Conclusions will be
drawn about the impact on size, cost, and performance of mo-
bile terminals.
I. IntroductionEmerging applications in the unlicensed 2.4 GHz Industrial,
Scientific & Medical (ISM) band have generated a great amount
of interest in the wireless communications industry worldwide.
One of the main reasons for this is the availability of the spec-trum around the world. As shown in Table 1, spectrum within
the 2.4 GHz frequency range is common to Europe, Japan, and
the USA. This commonality allows manufacturers to develop
one product to address the global market with minimal changes.
Couple this with the need for connectivity in the exploding
Home PC market, handheld service access devices, and infor-
mation appliances and you have a very compelling segment of
the wireless communications market.
New standards have also been developed in accordance to
this available frequency range and FCC part 15 [1]. Standard
specifications are also attractive to product manufacturers. Not
only do the open standards provide an overview of protocol,
product capabilities and features, but they also help to guaran-
tee reliable high-quality radio links, and interoperability be-
tween products of any brands. This equates to customer accep-
tance, and market penetration.
There are some specific applications, however, where stan-
dard protocol may not be an immediate necessity. For example,
in wireless home networks and cordless phones, several com-
panies are providing proprietary solutions available in the mar-
ket today. Much recent development activity is occurring in
Bluetooth and HomeRF solutions [2],[3]. These two standards
have hundreds of member companies in their respective work-
ing groups, and have recently released version 1.0 specifica-
tions. Both Bluetooth and HomeRF are loosely based on the
Digital Enhanced Cordless Telephony (DECT) standard popu-
larized in Europe [4]. Both of these standards are specifically
designed with frequency hopping algorithms to work well inthe presence of microwave ovens, and also include paging
modes to maximize battery life.
II. StandardsBluetooth advertises Wireless Connections Made Easy. Ini-
tially it is a standard that will deploy radio-based wireless ports
replacing the IR-based wireless ports (IRDA) with 10m range.
Bluetooth offers freedom from IRDAs line-of-sight require-
ments and allows users point-to-multipoint connectivity
Bluetooth acts like a radio-based wireless pico-LAN. The
objective of the Bluetooth Group is to get the radio-based wire-
less port adopted on the motherboard, in peripherals, into PDA
products and new digital cell phones. Whereas Bluetooth aims
at connecting portable devices (10-100m), the Shared Wire-
less Access Protocol -Cordless Access or SWAP-CA, com-
monly referred to as HomeRF, is focused on the Home Net-
working (300 m).
William O. KeeseHead of the Applications Group
National Semiconductor
Vikas VinayakSenior Applications Engineer
National Semiconductor
*except Spain and France
Table 1: Frequency Allocation
Christopher LamApplications Engineer
National Semiconductor
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The mission of the HomeRF Working Group is to enable
the existence of a broad range of interoperable consumer de-
vices via RF digital communications, in and around the home.
HomeRFis a PC-centric wireless solution that provides a gate-
way between the PC, the PSTN and other HomeRF nodes. It
supports Isochronous (I), Asynchronous (A) and combined I/A
data transfer.
HomeRF operates at 100 mW transmit power at 50 Hops/s.
It also supports up to 6 wireline quality voice connections basedon 32 Kb/s ADPCM and DECT call processing. Some of the
system specifications are compared in Table 2.
Another option providing a quick time to market solution is
moving DECT to the 2.4 GHz band. Several manufacturers have
introduced products based on 2.4 GHz DECT available today.
Because DECT is a mature high performance cordless com-
munications standard, there exist many advantages in this tech-
nology [5]. The radio architecture and TDMA controller can
remain the same as for DECT.
Higher protocol layers can remain unchanged, and full
DECT throughput can be maintained. Interaction with WANs,
POTS, ISDN, and frame relay has already been specified and
proven in Europe.One thing worthy of noting is the typical DECT modulation
(1.152Mbits/s GFSK, modulation index = 0.5) is not 20dB down
at 1 MHz away as specified in [1]. Two options exist to achieve
compliance: 1) lower the bit rate, or 2) decrease the modulation
bandwidth. Lowering the bit rate to 1Mbits/s while keeping the
same BT product and modu-
lation index will decrease the
Tx spectrum enough to meet
the spectral requirements, and
have a minimal impact on the
link channel budget. By
changing the BT product andmodulation index to 0.35, the
spectrum can also be met at
full DECT data rates, but the
increased ISI will cause deg-
radation in the link channel
by several dB. Theoretically,
you could also keep the origi-
nal 1.152MHz channel spac-
ing, because the FCC states
only that the channels must
have at least 1MHz spacing.
However, since 2.4GHz
DECT operation exists as
largely proprietary solutions,
it is feasible to change to a
narrower 1.0MHz channel
spacing, and even change the
time slot structure to accom-
modate non zero blind slot
operation.
An overview of the dif-
ferent standards comparing
some of the key parameters
is shown in Table 2. Param-
eters from Bluetooth andHomeRF are derived directly
from [2] and [3] respectively.
As stated previously, 2.4GHz
DECT solutions today are
mostly proprietary solutions, therefore are subject only to FCC
part 15, rather than a rigorous type approval. Specifications
for upbanded DECT have been adapted from [4] where appli-
cable for comparison. The similarities and differences between
the standards are evident from examining Table 2. The
Bluetooth and HomeRF performance requirements are sig-
nificantly relaxed from the DECT standard. Notably, the mini-
mum sensitivity is reduced considerably, and the in-band
blocking levels are also not as stringent. HomeRF particu-larly has a relaxed interference specification, with no adja-
cent channel specification, only -10dB of C/I required at 3
channels away, and an intermodulation protection (IMp) re-
quirement of only -39.5dBm.
The receiver intermodulation performance is typically mea-
sured using 3 signals: a desired signal and 2 undesired block-
ing signals. All three signals are on 3 different RF channels.
One of the unwanted signals is a modulated signal and the
other is not modulated (CW). The frequencies of the unwanted
signals are chosen such that one of their third-order
intermodulation products (IMp) will appear as an unwanted
interferer on the same channel as the wanted signal, or in other
words as a co-channel interferer.For example in Bluetooth, the level of the wanted signal is
specified to be PREF = -64 dBm and the level for each of the
two unwanted signals is PINT = -39 dBm. The CCI needed is
11dB. Assuming the wanted signal is much greater than the
sensitivity limit of the device, no contribution from the noise
Table 2: Parameter Comparison for 2.4GHz Applications
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floor of the wanted band is added to the CCI number.
For IMp, the following is needed:
System IIP3 = 1/2( PINT - PREF +CCI) + PINT =
+ 1/2[(-39) - -64 + 11] -39 = -21dBm
Since there is no direct co-channel interferer (CCI) specifi-
cation in HomeRF [3], it is inferred by the clear channel
assesment level of -80dBm. With an input level 3 dB above thesensitivity limit, -73 (same as IMp) this gives a CCI of -7dBc.
The phase noise numbers are a combination of the Tx and
Rx requirements and are also inferred from the specifications.
In order to relate the noise as dBc/Hz, the spectrum must betranslated to dBc/Hz, which can be done as:
10log(2*BW) = 10log(1 MHz) = 60dBc/Hz
For the transmit, the VCO phase noise is assumed as the
only contributor to the noise generated in adjacent channels.
For example, with a transmit power of 100mW, the noise in the
+/- 1MHz adjacent channels due to the VCO is determined for
Bluetooth as:
- 10log(100mW/1mW) - 60dBc/Hz =
- 80dBc/Hz @ 550 kHz
In the receive mode, reciprocal mixing can occur when an
interfering signal mixes with residual noise of the VCO creat-
ing an in-band interferer (CCI). For Bluetooth, the required C/
I is 11 since it is a Co-Channel Interferer that is created. For
example, the requirements of the +/- 1MHz adjacent channel
are calculated as below (adding 60dBc/Hz for the 1MHz mea-
surement BW):
0dB - 11dB - 60 dBc/Hz =-71dBc/Hz @ 1 MHz
Note, that if no channel filtering is provided at IF, and a
second down conversion is to be used, the sum of the two VCOsphase noise is required to meet the above requirements. Some
additional margin may be desired as the actual adjacent chan-
nel covers offsets of N*1.0MHz +/- the modulation spectrum.
III. RF TransceiverA radio transceiver was designed for Frequency Modulation
(FM) based schemes such as GFSK [7]. These modulation
methods are popular for low cost, low power applications. The
high tolerance to system non-linearity allows for decreased
operating current and lower voltage headrooms. In particular,
the transmitter side of the radio benefits enormously. There is
only a single frequency modulated carrier which is insensitive
to amplifier non-linearities[8],[9]. Non-coherent demodulationmeets the Bit Error Rate (BER) performance requirements of
these protocols and
translates to simpler
transceiver architectures
reducing the cost of the
solution[10],[11].
The block level
schematic diagram of
the transceiver is shown
in Figure 1. It consists
of a BiCMOS radio
transceiver IC, a power
amplifier, VCO, voltageregulator, Transmit/Re-
ceive (T/R) switch, ce-
ramic filter and SAW
filter.
The LMX3162
BiCMOS IC [13] con-
tains the phase lock loop
(PLL), transmit and re-
ceive functions. The
1.3GHz PLL is shared between transmit and receive sections.
The transmitter portion of the LMX3162 includes a frequency
doubler and a high frequency buffer and employs direct VCOmodulation. The receiver part consists of a 2.5GHz low noise
down converting mixer, an intermediate frequency (IF) ampli-
fier, a high gain limiting amplifier, a frequency discriminator, a
received strength signal indicator (RSSI), and an analog DC
compensation loop. The receiver section has single conversion
architecture and the received signal is demodulated by a Quadra-
ture Discriminator[12]. The IC features an on chip regulators
to allow supply voltages ranging from 3.0 to 5.5 volts, and is
specified for operation from -10 to +70 degree centigrade. Two
additional voltage regulators provide a stable supply source to
external discrete stages in the TX and RX chains.
The ceramic filter is shared between transmit and receive
sections, while the SAW filter provides selectivity at IF. The
RSSI output may be used for channel quality monitoring and
regulation of transmitted power, as required by HomeRF and
Bluetooth respectively. The external regulator supplies the VCO
and prevents frequency pulling. The T/R switch enables either
the transmit section or the receive section to connect to the an-
tenna. The power amplifier is implemented using a discrete bi-
polar transistor. An external VCO module provides the local
oscillator.
A. PLL SectionThe transceiver contains a phased locked loop (PLL). The PLL
runs at one half of the ISM band (2.4 - 2.5 GHz) and employsan integrated frequency doubler to synthesize the desired fre-
quencies as shown in Figure 1. This architecture alleviates the
disturbance to the local oscillator (LO) when the power
Figure 1: Transceiver Schematic
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amplifier (PA) is switched on. The radiation is isolated by off-
setting the PA output frequency from the LO frequency.
The PLL is shared between transmit and receive sections
since HomeRF, Bluetooth and Upbanded DECT are half-du-
plex TDMA systems. The data is transmitted and received in
different time slots. The length of time is determined by the
protocol employed. The PLL must be able to hop to the desired
carrier frequency in a given amount of time before data trans-
mission and reception starts. This time is called the lock time.It is defined as the time the PLL takes to settle down to an
acceptable frequency error. Table 2 shows that the demanding
requirement for lock time comes from HomeRF. Requirements
for Bluetooth (220 ms) and Upbanded DECT (416.67 ms with 1
blind slot) are more relaxed. Most Upbanded DECT solutions
employ blind slot operation, which allows certain latency and
also provides enough time to acquire lock.
To improve performance as well as reduce the cost of imple-
mentation, the transceiver transmits and receives data in the
open loop mode [6]. The PLL is first locked at the desired car-
rier frequency and then shut down during the data transmission
or reception. In this short duration, the VCO is subsequently
modulated by the baseband signal in transmit mode or idling in
the receive mode. There are two issues of concern with open
loop mode of operation.
The first issue is the drift of the VCO as the loop capacitors
discharge or charge because of the leakage currents associated
with the charge pump of the PLL and the varactor of the VCO.
This implies the drifting of the VCO must be negligible com-
pared to the carrier frequency. In order to achieve low frequency
drift (
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frequency is doubled and is also used as the PLL input. The
frequency doubled signal is then buffered and is output from the
LMX3162 at -7.5dBm. This signal needs to be high pass fil-
tered to reduce the level of what is now a sub-harmonic at half
the frequency. The level of this unfiltered harmonic is about
11dB below the desired output. The amount of amplification
required by the next stage, off-chip amplifier depends upon the
protocol being employed. Table 2 shows the required levels.
This transmit section of this radio was tested for Bluetooth op-eration, in the low power mode. The transmit power required is
0dBm. Adding to this 1dB of loss in the T/R switch and 2dB
loss in the post amplifier ceramic bandpass filter, the power
output from the PA desired is 3dBm, computing to a gain of
10.5dB. The amplifier was designed on a Siemens BFP420, a
25GHz bipolar transistor. The amplifier runs off a voltage regu-
lator internal to the LMX3162, which has a 2.7volt output rated
at 10mA. The transistor was biased at a collector-emitter volt-
age of 1.7volt with a quiescent current of 10mA. It delivers
+3dBm when its input is connected to the output of the
LMX3162.
Some Bluetooth options, and also other protocols, require
higher transmit powers. These are in the range of +20dBm andabove. This mandates the use of high efficiency power amplifi-
ers. Some suitable parts are the ITT2302 for upbanded DECT
and the ITT2304 for Bluetooth and HomeRF from ITT
GaAsTEK [16]. These amplifiers interface to the -7.5dBm out-
put of the LMX3162 directly, and also offer an on board T/R
switch, 14dB Gain LNA and 3.6volt operation. The BFP420
based PA is not required for the ITT2304.
After power amplification, the transmit signal is filtered to
remove all the harmonics and the sub-harmonics of the carrier.
This is required by regulatory organizations such as FCC and
also by the 2.4GHz protocols. The spurii levels acceptable are
dependent on the protocol employed. The filter chosen is a twostage ceramic filter from Murata, part number
DFC22R44P084LHA. It has a 3dB bandwidth of 84MHz cen-
tered at 2442MHz, and a typical insertion loss of 2dB. For higher
performance, Murata offers a 3 stage ceramic filter in the same
footprint. The wideband output and unmodulated transmit spec-
tra are shown in Figure 3 and Figure 4 respectively[v1]. The
sub-harmonic meets the Bluetooth specification (-36dBm)
without any filtering. A low pass filter can be added for addi-
tional margin. The ceramic filter is shared by the TX and the
RX modes, and a TX/RX switch is placed between the PA and
the T/R switch.
The unmodulated signal shows the phase noise of the trans-
mit signal. This signal is also used as a Local Oscillator (LO)
in the receive mode, and has to satisfy certain requirements of
phase noise in both the TX and RX modes. These requirements
are shown in Table 1.
Figure 5 shows a PRBS15 modulated Bluetooth signal,
which requires baseband Gaussian filtering with BT=0.5 andFSK modulation with typical peak-to-peak frequency devia-
tion of 320kHz. Figure 5 [v2]also shows the superimposed spec-
trum of a similar signal generated by a HP ESG4433 signal
generator. The trace with the larger first sideband [v3]is from
the signal generator. FCC requires the radiated power at off-
sets of 500kHz or more to be 20dB less than the peak power
being transmitted, all measurements being made with in a
Figure 3: Harmonics of Tranmitter Output
Figure 4: Phase Noise of Local Oscillator
Figure 5: Bluetooth Modulated Signals
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100kHz bandwidth. The Bluetooth signal easily meets this
requirement.
The VCO gets its power supply from an external low noise
voltage regulator, the National Semiconductor LP2980. This
regulator isolates the supply to the VCO from the power supply
to the other parts of the radio and various transients. Most im-
portant of these transients occurs when the Power Amplifier
(PA) is switched on. The peak power required to be delivered
can be in excess of +20dBm. At typical operating voltages of2.7 to 3.6 volts, the supply currents of these PAs are greater
than 150mA. This can result in IR voltage drops from the inter-
nal resistance of the battery and traces on the PCB. The VCO
has a worst case pushing figure of approximately 5MHz/V. The
VCO used oscillates at half the desired transmit frequency so
the pushing figure also doubles to 10MHz/V. Most 2.4GHz pro-
tocols require frequency accuracies of the order of +/-50kHz
(See Table 2). This is the total frequency accuracy required of a
transmitter and includes effects due to the temperature coeffi-
cient and initial accuracy of the crystal employed for the PLL,
the frequency jump resulting from opening the PLL, and fre-
quency pushing of the VCO. A typical error budget distribution
would allocate +/-20kHz out of the total +/-50kHz to errorscaused by pushing. Dividing this by 10MHz/V, the allowed noise
on the power supply pin of the VCO is +/-2mV. Now consider
that the connection of the battery to the PA and the VCO, and
that the two traces have 2cm in common. Assuming 35 micron
thick copper traces which are 12mil wide, the series resistance
of the trace evaluates to about 22 milliohms, assuming resistiv-
ity of 20 nano-Ohm/m for copper. The internal resistance of a
typical Ni-MH battery used in handsets is about 0.25 Ohms.
The total series resistance becomes 0.272 Ohms[v4], and the
IR drop computes to 40.8mV. This is far in excess of what is
allowed. The voltage regulator shields the VCO from all such
switching events.
C. Receive SectionA ceramic filter selects the desired frequency band received by
the antenna. The LMX3162 processes the received signals at
an IF of 110.5MHz, and the LO is chosen to be 110.5MHz be-
low the RF Input. The ceramic filter rejects the image frequency
221MHz below the RF Input. It also removes or attenuates out
of band blockers.
While the transceiver meets the BER requirements of
Bluetooth and HomeRF, some applications require the use of
an LNA external to the LMX3162. A LNA built using a BFP420
can achieve 13dB gain and 2dB noise figure. It can run off the
second voltage regulator internal to the LMX3162. A typical
schematic for the LNA is shown in Figure 6.
Once the received signal enters the LMX3162, it is
downconverted by the low noise 2.5GHz mixer to 110.5 MHz.
The mixer has 17dB gain with 11dB Noise Figure. The OIP3 of
this mixer is 7.5dBm. The LO is derived from the frequency
doubler internally, and in the receive mode the TX output buffer
of the LMX3162 is shut down to conserve power.
After the RF signal gets downconverted to IF, it needs to be
bandlimited for channel selection. Filtering is very helpful in a
multi-signal environment. The IF filter provides the desired se-
lectivity and prevents generation of spurii in the Limiter and
the Quadrature Detector. Limiters are inherently very nonlin-ear devices and in a hostile environment stronger, unwanted
signals can capture the receiver if no IF filtering is employed
[18]. Bandlimiting is also required to provide an optimum BER,
the primary requirement for which is to limit the noise power
as much as possible. This filtering can be done before the
Quadrature Detection and after it. However, it is a known fact
that discriminators have non-linear performance with respect
to the input signal-to-noise ratio [19]. This is called the Thresh-
old Effect. The discriminator needs to see a signal-to-noise ra-
tio that is above a certain threshold, otherwise its performance
degrades very rapidly. Hence there is a requirement to filter the
IF signal early on in the chain.
In the radio the IF Filter is implemented by a high volume,
readily available DECT SAW Filter. The SAFU110.6MSA40T
[17] Murata SAW filter is centered at 110.6 MHz, and has a3dB bandwidth of 1.5 MHz. The minimum insertion loss is about
3dB. The SAW filter is matched to LMX3162 using two induc-
tors and two capacitors. As the market for 2.4GHz products
increases, SAW filters that have been designed specifically for
these protocols will be made available by various vendors and
improve the performance of these radios.
After the initial filtering, the IF signal is amplified by the IF
Amplifier and fed to the IF limiter. A resistor and two capaci-
tors provide some filtering.
The LMX3162 demodulates the IF signal by quadrature de-
modulation [12]. The phase shifting tank consists of a capaci-
tor, an inductor, and a varactor. The varactor is used to tune the
tank, along with its associated parasitics, at exactly 110.5MHz.
The voltage at the output of the quadrature discriminator is mea-
sured by the baseband controllers ADC and compared to a ref-
erence, and the generated error signal is amplified and con-
verted back to an analog signal by using a Digital to Analog
Converter (DAC). The scheme is shown in Figure 7.
Using elements with different Qs and values can change
the Q of the tank. Higher Qs mean a narrower bandwidth and
greater sensitivity. This translates to better SNR and also greater
ISI [20][21]. Conversely, lower Qs result in slightly lower SNR
and lesser ISI. The LMX3162 has a 1pF quadrature shift ca-
pacitor internal to it.
Once the signal has been demodulated, it is passed throughan active low pass filter. This further limits the noise band-
width of the system and provides larger peak to peak voltage
output. The quadrature tank and the active LPF are shown in
Figure 7.
Figure 6: PA and LNA Schematic
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The LMX3162 provides a feature that eliminates the effect
of any initial frequency offset between the Transmitter and the
Receiver. Most RF protocols have an initial Synchronization
Field during which a string of alternating 1s and 0s is trans-
mitted. During this time, the demodulated signal is well known
and its average represents the nominal center frequency of the
transmitter. This average level can be captured by a Sample-
And-Hold (S&H) circuit and used as a reference level for a
data slicing comparator which converts the demodulated signal
into logic level 0s and 1s. The LMX3162 features an onboard
S&H. An internal resistor of 3kW and external capacitor
together provide the averaging time constant. The S&H actionis controlled by the S_Field input of the LMX3162, as shown
in Figure 7.
IV. Performance MeasurementsA. BERThe transceiver achieves a BER of 1E-3 for a Bluetooth modu-
lated signal at an input power of -83dBm, with the received
signal being offset by the +/-
115kHz frequency offset al-
lowed by Bluetooth. In this
measurement the transceiver
does not employ an LNA and
uses a low cost DECT SAW
filter. This easily meets the
Bluetooth minimum sensitiv-
ity requirement of -70dBm.With the use of a high per-
forming LNA and an opti-
mally chosen SAW filter, sen-
sitivity of -96dBm can be
achieved. The eye diagram of
the received signal for 1E-3
BER is shown in Figure 9,
and Figure 10 shows the eye
diagram for an input level of
-40dBm.
B. Synthesizer
The frequency offsetsallowed are given in Table 2[VV5]. The transmitter must be
within this limit and the receiver must be able to receive with
this frequency offset. A typical jump on opening the loop and
open loop frequency drift for a 500uS period at room tempera-
ture is shown in Figure 8. The parameters affecting these per-
formance criteria have been discussed in Section III.A. The fre-
quency jump is independent of the operating temperature, but
the frequency drift doubles every 10 degree centigrade follow-
ing the same law as the leakage currents it is caused by do. The
typical drift measured indicates a combined LMX3162 charge
pump, VCO varactor and PCB surface leakage current to be
around 55pA. The maximum operating temperature specifiedby Bluetooth is 35 Degrees ambient. Adding to this 10 degrees
due to self-heating of the product, the drift specification must
be met at 45 degrees Centigrade. This is 20 degrees above the
room temperature, and quadruples the leakage current to 220pA.
The maximum continuous Transmit time for Bluetooth is 5 slots,
or 3.125ms. Bluetooth has a channel spacing of 1 Mhz, but the
IF of the radio is at 110.5MHz. This means the LO must be
synthesized with a frequency resolution of 0.5MHz. The PLL
output is doubled in frequency, and this halves the resolution
required from the PLL to 0.25MHz. For 0.25MHz frequency
resolution, the phase comparison frequency that must be used
is 0.25MHz. The lock time requirements for Bluetooth are
220uS, and budgeting for margins and programming time re-
duces this to 180uS. To achieve a lock time of less than 180uS
with a phase comparison frequency of 0.25MHz requires a
total loop filter capacitance of about 5nF. Using this value, the
frequency drift on account of leakage is calculated as:
Df = ILeakage Dt KVCO/C
The drift in 3.125ms is about 20kHz. This meets the
Bluetooth requirement of 40kHz. This is because the National
Semiconductor proprietary charge pump provides excellent
leakage characteristics (typical 700pA at 85C).
As discussed in Section III.A, the lock and drift require-ments of HomeRF are demanding. The Alps VCO employed
had a KVCO of 140MHz/V, referred to 2.4GHz. The phase
detector comparison frequency was 250kHz. A second order
Figure 7: Quadrature Demodulation
Figure 8: Open Loop Frequency Jump and Drift
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loop filter is used. The loop bandwidth is 40kHz and the phase
margin is 5.
Figure 2 shows the component values of the filter. The lock
time is measured from the falling edge of signal that shuts down
the PLL (PLL_PD) to 1225 MHz (2450 MHz at Tx Output)
within 5 kHz (10 kHz at Tx Output). The lock time is 111ms as
shown in Figure 11.
The frequency drift due to the current leakage is measured
starting from the rising edge of PLL_PD. Figure 12 demon-
strates the frequency drift of the VCO due at 85C. The very low
drift of 778Hz/ms is much better than what is required for
HomeRF.
Figure 12 shows two frequency jumps. The first is due to
the phase noise of the PLL and appears as a jump due to the
finite response time of the modulation domain analyzer. Thesecond jump is due to the common mode operating range of the
op-amp. Even though the control voltage line varies a few hun-
dred mVs, the frequency jumps tens of Kilohertz because of the
large VCO gain and the bias current coming out of the op-amp.
Placing a 10-MW resistor in parallel with the loop filter will
sink away the bias current of the opamp and prevent saturation.
The occurrence of the second jump depends on the selection of
the opamp and the CMOS switch. In the current design, the
second jump occurs only after the loop is open for a long time
(>100 ms). In case of HomeRF, the loop will not be idle or
open for such long time.
C. InterferenceAnother difference between ideal and real operating conditions
for a receiver is the presence of signals other than the wanted
ones. This is especially true for an unlicensed environment like
the 2.4GHz band. A typical transceiver must operate in a multi-
transmitter environment, with unwanted signals that are either
on the desired or on adjacent channels. These hostile signalscan capture the desired channel, cross-modulate information
on to the desired signal, or simply reduce the demodulated sig-
nal to noise ratio and lower the BER sensitivity of the radio.
Large, out of band signals can even desensitize the radio
Figure 9: Eye Diagram for BER=1E-3
Figure 10: Eye Diagram for -40dBm Input
Figure 11: PLL Lock Time
Figure 12: Open Loop Frequency Drift at 85C
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completely. The performance of the radio is shown in Table 3
and is seen to meet all the requirements.The second test in the table is Co-Channel Interference
(CCI). The radio meets a BER of 1E-3 for CCI 11dB below for
a desired input down to -80dBm. At the specified desired sig-
nal level of -60dBm, the radio even meets the BER for a CCI
that is only 6db. Bluetooth requirements have relaxed for a pe-
riod of three years. During this period, the radios may meet a
relaxed CCI specification of 14dB. For this CCI level, this ra-
dio suffers no degradation in BER sensitivity and performs at -
83dBm. Figure 13 shows the eye diagram for the Bluetooth
CCI test signal.
The third test is first of the Adjacent Channel Interference
(ACI) requirements the radio has to meet. A radio must per-
form at 1E-3 BER with an undesired, Bluetooth signal interfer-
ing at 1 MHz offset. The desired signal must be at the same
level as the interferer, both of them being at -60dBm. The three
year relaxed specification for the 1MHz test is an ACI 4dB be-
low the desired signal. The radio meets the relaxed specifica-
tion. The low cost, off-the-shelf-available SAW filter selected
for this radio has a 3dB bandwidth of 1.5 MHz and is wider
Table 3: Interference Test for Bluetooth
Figure 13: Eye Diagram for Bluetooth CCI Test
Figure 14: Eye Diagram for Bluetooth ACI 1MHz Test
Figure 15: Eye Diagram for Bluetooth ACI 2MHz Test
Figure 16: Demodulated Signal for Bluetooth ACI 2MHz Test
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than what is required for a Bluetooth signal. Figure 14 shows
the eye diagram for the Bluetooth ACI 1MHz test signal.
The radio meets the 2MHz ACI specification for levels from
-81dBm to -30dBm. Figure 15 shows the eye diagram for the
Bluetooth ACI 2MHz test signal. Figure 16 shows a single trace
of the demodulated signal, and a 2 MHz signal is clearly visible.
The fifth test measures the performance of the radio for a
Bluetooth ACI that is 3MHz offset. The radio meets the speci-
fication from -72 to -35dBm. Figure 17 shows the eye diagramfor the Bluetooth ACI 3MHz test signal. Figure 18 shows a
single trace of the demodulated signal, and a 3MHz signal is
clearly visible.
The seventh test measures the intermodulation properties
of the radio. The test requires the desired signal to at -64dBm,
a -39dBm static sine wave at 3 MHz offset and another
Bluetooth signal at -39dBm at 4MHz offset. The radio meets
the requirements. Figure 19 shows the Bluetooth
intermodulation test signal.
The radio also meets the performance requirements for out
of band interferers.
D. TransmitterThe transmitter section of the radio too must meet some speci-
fications. Being unlicensed radiators, these radios have to com-
ply with the FCC requirement that the radiated power at offsets
of 500kHz or more must be 20dB less than the peak power
being transmitted, all measurements being made in 100kHz
bandwidth. In addition, Bluetooth requires the power at 2 and
3MHz offsets to be below -20dBm and -40dBm respectively,
all measurements being made in 100kHz bandwidth.
Figure 5 shows that that the radio meets these requirements
when used as a Bluetooth transmitter. Table 4 shows the out of
band spurious emission requirements of Bluetooth, and Figure
3 shows that these are met.
V. ConclusionThis paper has covered key design, specification and test is-
sues for emerging 2.4GHz wireless communication applications.
Three 2.4GHz protocols and their impact on transceiver speci-
fications were discussed. A new single chip BiCMOS radio
transceiver was presented. A radio built around this chip was
made and its performance measured for the 2.4GHz protocols.
The single chip transceiver itself contains most of the function-
ality required for 2.4GHz radios and is an enabling component
of low cost, small size wireless applications.
This paper is published with due permission from Penton Publishing for the
IIC-Taipei 2000 conference.
Figure 17: Eye Diagram for Bluetooth ACI 3MHz Test
Figure 18: Demodulated Signal for Bluetooth ACI 3MHz Test
Figure 19: Bluetooth Intermodulation Test Signal
Table 4: Out of Band Spurious Emission for Bluetooth
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AcknowledgementsThe authors would like to thank Eric Lindgren, Doug Steen,
Tai Wong, Erik Ankney, John Lund, Finn Anderson, and Jim
Stubstad whose contributions were instrumental in developing
the single chip transceiver solution.
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Authors contact detailsVikas Vinayak
National Semiconductor
Wireless Communications Division
2900 Semiconductor Drive
M/S D3-500, Santa Clara, CA 95052-8090 USA
Phone: 1 408 721 2228
E-mail: [email protected]
William O. KeeseNational Semiconductor
Wireless Communications Division
2900 Semiconductor Drive
M/S D3-500, Santa Clara, CA 95052-8090 USA
Phone: 1 408 721 4494
E-mail: [email protected]
Christopher Lam
National Semiconductor
Wireless Communications Division
2900 Semiconductor Drive
M/S D3-500, Santa Clara, CA 95052-8090 USAPhone: 1 408 721 5724
E-mail: [email protected]
This paper is published with due permission from
Penton Publishing for the IIC-Taipei 2000 conference.