2894 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 58, NO. 9, SEPTEMBER 2010
Electromagnetic Coupling Reduction in High-ProfileMonopole Antennas Using Single-Negative Magnetic
Metamaterials for MIMO ApplicationsMohammed M. Bait-Suwailam, Graduate Student Member, IEEE, Muhammed Said Boybay, Member, IEEE, and
Omar M. Ramahi, Fellow, IEEE
Abstract—Single-negative magnetic metamaterials are used inorder to reduce mutual coupling between high-profile antennasused in multiple-input multiple-output systems. The magnetic per-meability of the developed single-negative inclusions have negativeeffective response over a specific frequency band. The inclusionsconsidered here are composed of broadside coupled split-ring res-onators. The single-negative magnetic inclusions are inserted be-tween closely-spaced high-profile monopole antenna elements. It isshown that mutual coupling between the antenna elements can bereduced significantly by incorporating such magnetic inclusions.Effective response of the constitutive parameters of the developedmagnetic inclusions are incorporated within the numerical models.Good agreement is obtained between the experimental and numer-ical results.
Index Terms—Artificial magnetic materials, metamaterial,monopole antennas, multiple-input multiple-output (MIMO),mutual coupling.
I. INTRODUCTION
D ECORRELATING multiple antenna elements whenplaced in a small platform is very important in order to
improve the performance and capacity of an antenna systemcompared to a single-input single-output (SISO) system [1].Usually antenna elements need to be spaced by in orderto have suitable isolation and low correlation between theelements. However, degradation would result upon placingthe antenna elements in close proximity, due to near-fieldeffects, diffraction from finite-ground planes, and strong in-ductive and capacitive coupling between the elements [2]–[4].Therefore the performance of antenna arrays or multiple-inputmultiple-output (MIMO) systems can be degraded when theantenna elements are in close proximity.
In many outdoor and indoor wireless communication envi-ronments, there are many obstacles and scatterers, like build-ings, mobile terminals, and offices. A problem that is encoun-tered in such environments is multipath fading. This fading phe-
Manuscript received July 17, 2009; revised February 03, 2010; acceptedMarch 20, 2010. Date of publication June 14, 2010; date of current versionSeptember 03, 2010. This work was supported by in part by the Sultan QaboosUniversity of Oman, Research in Motion Inc., and in part by the NationalScience and Engineering Research Council of Canada under the NSERC/RIMIndustrial Research Chair and Discovery Programs.
The authors are with the University of Waterloo, Department of Electrical andComputer Engineering, Waterloo, ON N2L 3G1, Canada.
Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TAP.2010.2052560
nomenon limits the performance and capacity of the antennasystem. One of the ways to combat that multipath fading is touse multiple antennas at either transmit, receive, or both ends.In MIMO systems, more than two antennas are often employed.Thus, the correlation between any two antennas within the arrayis often required and needs to be kept as low as possible forbetter performance of the MIMO antenna system.
Compactness and low-weight have become a highly desir-able feature in an antennas’ design. As such, reduction of elec-tromagnetic coupling or interference between the containingantenna elements becomes a challenging design task. Mutualcoupling is a common problem in the antenna and electromag-netic interference (EMI) communities. It significantly affectsmost types of antenna arrays. The study of the mutual couplingproblem started several decades ago, and several research effortshave been devoted to combat the mutual coupling between cou-pled antennas, not just from antenna engineers, but also fromother disciplines such as communications where multiple an-tennas are frequently encountered, like in MIMO systems.
Anderson et al. in his work [5] introduced the possibility ofconnecting a lossless network between the input ports and theantenna ports, such that no coupling between antennas is en-countered. It was shown that antenna mutual impedances shouldbe purely reactive at the resonance in order to realize a de-coupling network and hence isolate the antenna elements whenplaced close to each others. Other methods have also been sug-gested in [6], [7], where transmission lines are used as antennadecouplers. Although the capacitive decouplers work well, theyare inherently narrowband, and limited by the antenna band-width.
An alternate way of reducing the electromagnetic couplingbetween radiating and/or receiving antenna elements sharing acommon ground plane or chassis is to introduce resonant de-fects or slits in the ground plane [8]–[10]. By proper choice ofdimensions, the slits resonate and can trap some of the energybetween the radiating elements. Another strategy to mitigate thecoupling between radiating antennas is to use electromagneticband-gap (EBG) structures [11]. EBG structures were used ex-tensively in mutual coupling reduction in planar and low-profileantennas [12], [13].
Another mechanism proposed previously to decorrelate (i.e.,isolate) highly-coupled monopole antenna elements is by using180 hybrid couplers [7], [14], [15]. The method is based on themode-decomposition network, in which a multi-port network isinserted between the antennas and their driving ports.
BAIT-SUWAILAM et al.: ELECTROMAGNETIC COUPLING REDUCTION IN HIGH-PROFILE MONOPOLE ANTENNAS 2895
Ferrer et al. introduced the idea of using capacitively loadedloop (CLL) magnetic resonators in order to decorrelate twomonopole antennas [16]. Although the coupling had beenreduced in [16], the antenna elements were not well-matched.In [17], split-ring resonator (SRR) magnetic inclusions wereused to reduce the coupling between closely-spaced monopoleantennas. Higher suppression of mutual coupling was achievedin [17] in comparison to [16].
In this work, single-negative magnetic (MNG) metamaterialsare developed in order to efficiently suppress the electromag-netic coupling between closely-spaced high-profile monopoleantenna elements. One way of realizing MNG metamaterials isto use SRRs [18], [19]. Once these inclusions are excited usingproper polarization, an induced electric current develops on theinclusions. If a large number of inclusions are arranged eitherperiodically or aperiodically, the inclusions behave as magneticdipole arrays. Those dipole elements result in having negativeeffective permeability , over a certain frequency band. Thisin turn prevents the existence of real propagating modes withinthe MNG metamaterials. Thus, it is plausible that a MNG meta-materials layer can block electromagnetic energy radiated byone antenna element from being transmitted to a nearby antennaelement within an array system. This implies that the ensembleof resonators, comprising the MNG layer, act as a decouplinglayer. The mutual coupling and envelope correlation betweentwo monopole antennas, separated by an MNG layer, are inves-tigated.
This paper is organized as follows. Section II gives a con-spectus of the theory and motivation behind using single-nega-tive magnetic metamaterials as antenna decoupler. The designmethodology for the magnetic materials considered in this workis also discussed with emphasis on the numerical characteriza-tion of the magnetic metamaterials. Experimental and numer-ical setups are presented and discussed in Section III. Section IVpresents the results obtained from both the numerical and exper-imental setups. Moreover, an experimental analysis and investi-gation on the effectiveness of the developed magnetic materialsas antennas decoupler is also presented. Conclusion and sum-mary are given in Section V.
II. SINGLE-NEGATIVE MAGNETIC METAMATERIALS AS
ANTENNA DECOUPLER
When mounting closely-spaced high-profile monopole an-tenna arrays, degradation in impedance matching, field pattern,and strong mutual coupling become a performance challenge.Although the impedance can be tuned and/or optimized, the mu-tual coupling still remains as a bottleneck that deteriorates theperformance of the antenna system. Due to the nature of the cur-rent distribution on monopole antennas, a magnetic field circu-lates around the antennas, leading to radiation into free-space,and possibly to nearby elements. A discussion on mutual cou-pling effects between linear antenna elements can be found in[2].
Displacement current is one of the contributors to the strongmutual coupling in antenna arrays. Such a current can exist inmany types of matter, even in free-space. To have more phys-ical insights into that kind of current, consider having two cou-pled antennas and spatially spaced in free-space. We introduce
Fig. 1. (a) Schematic model for the coupling path with the two antenna systemand (b) its equivalent network model.
here the “coupling path” term, as shown in Fig. 1(a), in order toaccount for the mutual coupling between the antennas and thecurrent path through the displacement current. That current pathis also expected to strongly couple the antenna elements, andwithin which a magnetic field exists around the path betweenthe antennas. This coupling path is indeed critical in high-profileantennas. Therefore, mitigating it through the use of an appro-priate synthesis technique or introducing a decoupling networkis desirable in many applications.
Let us consider highly-coupled high-profile antennas in free-space. For simplicity, two antenna elements are considered inthis analysis. The model comprising the two antennas with thecoupling path, as shown in Fig. 1(a), is analyzed systematicallyin terms of scattering parameters for ease of analysis. Then, weshow that the magnetic resonators indeed work as a decouplingnetwork, in which the mutual impedance should be purely reac-tive at the resonance frequency in order to decouple the antennaelements (i.e., at resonance.) Such a condition isnecessary for the existence of a decoupling network. This nec-essary condition had been discussed and verified in [5], [6] and[20]. Fig. 1 shows the two antenna elements as a two-port linearnetwork with the following scattering parameters
(1)
where represents the coupling path between the twoelements, clearly designated in Fig. 1(a). Let us assume that thetwo coupled antennas are lossless, reciprocal, matched and havepoor isolation (i.e., ), then (1) becomes
(2)
Obviously what we need in order to efficiently decouple the an-tenna elements is to have high-isolation between them, tobe zero, which is an ideal case at resonance. It follows from (2)that the antenna mutual impedances , should be re-active, and thus all the mutual impedances should ideally be re-active. Although this might not be intuitive, it will be motivatedand validated with antenna mutual impedance results. Note thatthe condition holds quite often at a single resonant frequency,but nonetheless, it could be satisfied over a frequency band de-pending on the designed decoupling network [5].
Generally, natural materials are characterized by constitutiveparameters: the electrical permittivity , which is related to theresponse of a material to an electric field and the magnetic per-meability , which gives the response of the material to a mag-netic field. At microwave frequencies, and are both posi-tive for natural materials, which can be termed double-positive(DPS) materials.
2896 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 58, NO. 9, SEPTEMBER 2010
Fig. 2. The developed single-negative magnetic metamaterials based on the(BC-SRRs): (a) Split-ring resonator (SRR) unit cell with its dimensions, (b)waveguide structure used for the characterization of MNG metamaterials. Notethat �-field points in y-direction,�-field is in z-direction, and the propagationis x-direction.
An interesting phenomenon of strong relevance to the widerange of problems where electromagnetic interference is of con-cern, is the elimination of propagation when enhanced mag-netic structures are employed. In such a scenario, transmissionof electromagnetic energy would cease since purely evanescentwaves (i.e., no real energy flow) exist within the magnetic de-coupling layer. In this work, the decoupling layer has a complexeffective magnetic permeability, with a negative real partabove the enhanced resonance and positive real part below theresonance. In [21]–[23], several types of MNG metamaterial de-signs were discussed. In this work, without loss of generality, weconsider one of the popular types, namely the broadside coupledrectangular split-ring resonators (BC-SRRs). Fig. 2(a) shows aunit cell of the developed artificial magnetic inclusion.
A unit cell of the BC-SRR metamaterial is designed in orderto decouple the antenna system while at the same time maintainlow correlation between the antenna elements. Initial design di-mensions of the SRR inclusion unit-cell were estimated numer-ically using the characterization model that will be discussedlater. Next, the optimized dimensions were obtained such thatthe inclusion resonance takes place at the middle of the fre-quency band of interest. The SRR rings considered here haveequal sides of lengths with strip width of 1 mm.The rings with opposite cut openings are etched on the sidesof a dielectric substrate (Rogers RO4350, ,
) having a thickness of 0.762 mm. The cut gaps within themetallic rings is 1 mm. The size of the dielectric substrate, ,[see Fig. 2(a)] is 16 mm, which is much less than the operatingwavelength at a frequency of 1.24 GHz. The SRR rings are madeof copper with a thickness of 20 .
The characterization and retrieval of MNG metamaterials areconsidered next. The model adopted in our work is based on theretrieval extraction method reported in [24]. Basically, one hasto compute the scattering parameters of a unit cell SRR inclu-sion, from which the effective refractive index and impedanceZ are first computed. The effective permittivity and perme-ability are then calculated from and .
In order to numerically characterize the SRR inclusions, anair-filled waveguide with a unit cell positioned at the center ofthe waveguide is used, as shown in Fig. 2(b). Notice that the cellis positioned such that the incident magnetic field is perpendic-ular to the inclusion’s trace (or surface). The model mimics atransverse electromagnetic mode in both ports [see Fig. 2(b)],with top and bottom sides of the air-filled waveguide assigned
Fig. 3. Scattering parameters of a MNG unit cell inclusion extracted from thewaveguide structure shown in Fig. 2(b).
Fig. 4. Effective response of the MNG unit cell constitutive parameters, ex-tracted using the retrieval method in [24].
as perfect electric conducting walls, while its sides as perfectmagnetic walls.
Fig. 3 shows the scattering parameters, and , com-puted using two commercially available full-wave simulationtools (Ansoft HFSS [25], and CST Microwave Studio [26]).Clearly, the transmission coefficient, is below 20 dB atresonance. That dip in the transmission coefficient is attributedto the magnetic resonance nature of the developed inclusionswhen a magnetic field impinges normally to the inclusions axis.As such, the energy is sustained within the inclusions and resultsin no transmission at the enhanced magnetic resonance. The ex-tracted real and imaginary parts of the effective electric permit-tivity and magnetic permeability of the magnetic resonators areshown in Fig. 4.
As mentioned earlier, the antenna mutual impedance shouldbe purely reactive at resonance with its real part being nearlyzero at resonance. In order to show that condition, a numericalmodel comprising two monopole antennas with the single-nega-tive magnetic decoupling network is made (see Fig. 5). The twoantenna system with and without magnetic inclusions are com-pared.
Fig. 6(a) shows the reactive part of the computed mutualimpedance, , between the two monopoles with and withoutthe magnetic decoupler layer. Around the resonance of thedecoupling network, the reactive part is inductive, while it iscapacitive over the antenna frequency band for the air case.Fig. 6(b) shows the real part of the mutual impedance .It is noticed that the computed resistances do not turn out tobe exactly zero at resonance. Those losses are mainly dueto the losses within the MNG decoupling network. It is ofinterest to note that the necessity condition in [5] was providedparticularly for lossless decoupling networks. The decoupling
BAIT-SUWAILAM et al.: ELECTROMAGNETIC COUPLING REDUCTION IN HIGH-PROFILE MONOPOLE ANTENNAS 2897
Fig. 5. Lateral view of the two-monopole antennas with SRR inclusions. Dis-tances are given in terms of an operational wavelength � � ��� ��. Theplastic support is not shown for clarity purposes.
Fig. 6. Computed mutual impedance, � , between the two monopoleantennas with and without the MNG decoupling network: (a) reactive part and(b) resistive part.
Fig. 7. Fabricated monopole antenna system with the MNG metamaterial. Notethat one monopole antenna is visible and another antenna element is behind theMNG layer.
network presented here is based on magnetic resonators, whichare lossy and inherently narrowband [27]–[29].
III. EXPERIMENTAL AND NUMERICAL SETUPS
Fig. 5 shows the schematic of two monopole antennas sepa-rated by a distance of , where is the wavelength cor-responding to the resonant frequency of the separate monopoleantennas. The two antennas have been designed to operate ata frequency of 1.24 GHz. A finite copper ground plane of size
is used. Stacks of MNG inclusions, as designedand discussed in Section II, are aligned vertically between thetwo monopole antennas. In this work, 4 SRR inclusion pairs areetched on both sides of the dielectric substrate. The periodicityof the magnetic inclusions is an important parameter in the de-sign of such inclusions, and it needs to be much smaller than theoperating wavelength. The periodicity, or separation distancebetween each strip of the developed magnetic inclusions (seeFig. 5), corresponds to an electrical thickness of at theresonance frequency. The developed MNG inclusions have highpermeability values below the resonance frequency. Those highpermeability values enhance the inclusions resonance frequencyand sustain much of the induced magnetic field within the MNGlayer.
The model setup for the monopole antennas, shown in Fig. 5,was fabricated using two brass rods of length 57 mm and di-ameter of 1.3 mm, soldered to 50 coaxial (SMA) connec-tors. Fig. 7 shows the fabricated model with the MNG meta-materials. Fig. 8 shows the reflection coefficient of the antennasystem, , obtained using an HP8722ES VNA, and Fig. 9shows the mutual coupling, , between the two antenna el-ements. A two-antenna system with no spacer (air case) is usedas a reference for comparison purposes. Another case whenplacing a PEC screen between the antenna elements is consid-ered for completeness, although it seems very intuitive froma PEC screen to reflect back incident waves. In order to sus-tain the SRR inclusions in a vertical position, a plastic supportlayer ( , ) has been used. The inclusionswere sandwiched within the plastic holder with a uniform gapspacing between the MNG strips of 16 mm (see Fig. 7). Notethat the spacing between the SRR inclusions corresponds to thespacing used in the characterization model of the MNG unitcell in Fig. 2(b). The scattering parameters have been computedusing Ansoft HFSS and compared with measurements.
Due to computational memory constraints, the effective re-sponse of the magnetic inclusions (see Section II and Fig. 4) isused in order to mimic the response of the magnetic inclusions.The artificial magnetic structures are in general inherently dis-persive and anisotropic. The MNG materials considered in thiswork are based on the BC-SRRs [19]. Such inclusions provideenhanced permeability only in the direction normal to the inclu-sions plane ( -direction in Fig. 5), and enhanced permittivity inthe directions tangent to the plane ( - and -directions in Fig. 5).In this work, the orientation of the magnetic inclusions is as-sumed in -plane, as shown in Fig. 5. In such a case, the ef-fective permeability and permittivity of the magnetic structurescan take the following tensor forms, respectively
(3)
2898 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 58, NO. 9, SEPTEMBER 2010
(4)
where represents the averaged permittivity betweenthe host medium ( with a slab thickness of
) and air gaps between successive MNG strips( with an air-gap thickness of — ).
It is instructive to note that within the numerical models, ef-fective response of the constitutive parameters need to be care-fully oriented. In more details, the effective permeability in the
directions correspond to ( , 1, 1), while the effec-tive permittivity response corresponds toin respectively, as can be seen from (3) and (4).
IV. RESULTS
A. Scattering Parameters
By placing the MNG inclusions between the antenna ele-ments, the mutual coupling has been reduced by almost 25dB at the resonance frequency, while at the same time main-taining good impedance match for the two-antenna system.The high values below resonance of the MNG inclusions,see Fig. 4, have contributed to the mutual coupling suppres-sion, where the transmission coefficient, , starts to dropbelow 10 dB around 1.17 GHz. Moreover, the negativevalues above resonance have stronger influence on the mutualcoupling suppression. This is attributed to the existence ofevanescent fields within the negative region of the magneticinclusions (i.e., above resonance), which blocks the EM energyfrom being transmitted from one antenna element to another. Asizeable dip in (below 40 dB) was noticed at a frequencyof 1.22 GHz. We note here that the antennas without magneticinclusions (air case) were not well-matched due to the strongmutual coupling between the two monopole antennas. Goodagreement between the numerical results and experimentalones can be seen. The modeling of the effective response ofthe developed MNG inclusions, rather than modeling the realstructure shows its effectiveness from both computational timeand memory requirements. The mutual coupling suppressionlevel achieved as a result of using the BC-SRRs inclusions ishigher than the suppression level presented in [16], especiallyat the antennas’ resonance. Most interestingly, however, is thatunlike [16], the antennas in this work are well matched at theoperating frequency despite the severe limitation placed ontheir separation distance. (The antennas were separated by adistance of only , and in another case . In termsof the operating wavelength, the separation distances betweenthe antennas considered here were relatively smaller than theantennas separation distance of given in [16].)
The field generated by antennas can be considered as a super-position of evanescent and propagating waves. The interactionbetween antennas is a combination of the effects of these planewave components. In order to eliminate the coupling, the prop-agating waves and the evanescent components with low decayconstants must be suppressed. To analyze the relevant evanes-cent spectrum, we assume that coupling due to the evanescentcomponents that decay more than 10 times within a dis-
tance can be ignored. Therefore the evanescent fields that sat-isfy are ignored, where is the compo-nent of the wave vector (see Fig. 5). Using dispersion rela-tion and the assumption stated above, it can be deduced thatthe components that cause the interaction between the antennaelements are both the evanescent and propagating waves with
. As a result the minimum spatialwavelength in the plane to be suppressed is
. Since the periodicity of the developed SNGmagnetic metamaterial inclusions is smaller than (i.e.,around ), the SNG layer can be assumed as an effective ho-mogenized medium for these components.
B. Envelope Correlation
Next, the performance of antenna arrays applicable to MIMOsystems is studied. In a rich scattering multipath environment,the correlation between the antenna elements can be expressedusing far-field components [4], mutual impedances between theantenna elements [30], [31], or directly from the scattering pa-rameters measured at the ports [32]. Under the assumption ofuniform incident waves and good impedance matching, the en-velope correlation can be calculated from the scattering param-eters much more readily than from the far-field patterns. In thiswork, the use of scattering parameters is incorporated in the en-velope correlation computation and compared with that usingfar-field method [4].
The envelope correlation of a two antenna system usingthe far field method [4] can be calculated using
(5)
where represents the field radiation pattern of the an-tenna system when port is excited and all other ports are ter-minated in a matched 50 load [32], and is the Hermitianproduct. The envelope correlation based on the scattering ma-trix can be calculated using [32], [33]
(6)
In the case of a 2 2 MIMO system , the envelopecorrelation between antenna element and is givenas
(7)The envelope correlation for the antenna system with the
MNG inclusions is computed using (6) and compared with thefar field formula in (5). Results presented in Fig. 10 show com-parison with the case without spacers between the antennas.Good agreement between the envelope correlation computedusing (6) and that obtained using the far field method in (5) canbe seen for the air case. The computed envelope correlation wasabove 0.3 in average.
BAIT-SUWAILAM et al.: ELECTROMAGNETIC COUPLING REDUCTION IN HIGH-PROFILE MONOPOLE ANTENNAS 2899
Fig. 8. Magnitude of � for the monopole antenna system with and withoutthe MNG metamaterial slab.
Fig. 9. Magnitude of � for the monopole antenna system with and withoutthe MNG metamaterial slab.
Fig. 10. Envelope correlation, � , for the monopole antenna system with andwithout MNG inclusions.
The assessment of envelope correlation using (6) is more suit-able for lossless cases (i.e., no lossy decoupling networks be-tween the antennas) rather than to use the Far-field method (5).This is because the scattering parameters formulation is morecost effective and can be obtained from a simple measurementof scattering parameters instead of using the far-field radiationpattern, which is a time-consuming process. The presence of theartificial magnetic medium between the antennas introduces en-ergy dissipation which can be accounted for if a three-port net-work model is conceived for the system at hand (in such model,the third port will be terminated with a resistive network to ac-count for the losses.) Therefore, since the coupling between thetwo antennas is diminished as measured by the parameter,the 3-port S-parameter model used for calculating the correla-tion coefficient, while not fully descriptive of the energy transfer
between the two antennas, is more relevant to analyzing thesystem presented in this work.
From Fig. 10, the envelope correlation results using (6) forthe MNG case shows low correlation between the antenna ele-ments below 0.2 over the antennas frequency band as comparedto the air case. The correlation then starts to increase after 1.3GHz, which corresponds to diminishing of the mutual couplingsuppression when using the developed magnetic inclusions (seeFig. 9). We emphasize here that the increase in envelope corre-lation beyond 1.3 GHz will result in non-orthogonal radiationpatterns. Nonetheless, the 1.3 GHz is at the edge of the opera-tional band of the two-monopole antenna system. The envelopecorrelation results obtained using (5) show that the antenna ele-ments are well-decorrelated below 0.1 for frequencies up to 1.25GHz where losses of the magnetic inclusions are inherently pro-nounced above 1.25 GHz.
An important conclusion based on the assessment of the en-velope correlation between antennas is that the scattering matrixbased formula is inaccurate, especially in situations when lossydecoupling networks are placed between the coupled antennaelements, which is simply because the evaluation of correlationbased on the scattering parameters do not account for the an-tennas efficiency. Similar observations were reported in [33].
Although the difference between envelope correlation calcu-lations is attributed to the electromagnetic loss of the MNG ma-terial, Figs. 4 and 10 show that the deviation between envelopecorrelation calculations correspond to the frequencies where theloss tangent of the MNG material is low. This counter intuitiveobservation is explained as follows. The first thing to note is thatthe decay and energy dissipation relation in the case of a reg-ular double positive lossy material is different than the relationin the case of a single negative material. Therefore, a straightforward relation between the imaginary part of the and theelectromagnetic loss within the negative material that states “thehigher the imaginary part, the higher the loss” is not correct inall cases. Basically, the decay within MNG material is a resultof the high negative value of the real part of the . When the realpart of is highly negative, the field is suppressed faster and thefield would not penetrate to the MNG material. Therefore, theamount of the field within a slab with highly negative real isnot as much as the amount of the field within a MNG materialwith negative closer to zero. As a result, even if the imagi-nary part is higher for highly negative MNG materials, they donot dissipate as much energy as MNG materials which have anegative closer to zero. This behavior is also affected by thethickness of the slab and the mode, direction and evanescenceof the incident field. This phenomenon is described in Fig. 11,where the electric field intensity is plotted within the MNG ma-terial at different frequencies. Although the imaginary part islower at higher frequencies, since the field penetrates more, it ispossible to lose more energy. Note that for the presented plot, itis assumed that the incident field is a TEM plane wave.
In addition, the total power consumed within the metamate-rial layer (using effective medium parameters) was calculatedusing CST Microwave Studio and presented in Fig. 12. Thepower consumption is higher at higher frequencies although theimaginary part of the permeability is lower.
2900 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 58, NO. 9, SEPTEMBER 2010
Fig. 11. Computed electric field intensity within the MNG material at severalfrequencies.
Fig. 12. Computed Power Loss within the MNG layer using CST MicrowaveStudio.
Fig. 13. Computed gain pattern for the two monopoles with and withoutmagnetic inclusions at 1.22 GHz (a) E-plane (��-cut, � � �� ), (b)H-plane(��-cut,� � �� ).
C. Far-Field Results
Far-field radiation patterns for the monopole antennas withand without the MNG metamaterial are numerically computedusing Ansoft HFSS. In the numerical simulation, one antennawas connected to a 50 coaxial (SMA) connector while theother element is terminated with a 50 load. Fig. 13 shows thefar-field pattern for the two antennas with and without the mag-netic inclusions. The computed results [see Fig. 13(a)] show that
Fig. 14. Computed gain pattern for the two monopoles with magnetic inclu-sions at 1.2 GHz (a) E-plane (��-cut,� � �� ), (b) H-plane (��-cut,� � �� ).
the use of MNG inclusions enhances the antennas’ potential tosteer or concentrate energy into a more specified direction, thusincreasing the gain in specific directions, whereas the counter-part antenna system (i.e., antennas without SRRs) yield the typ-ical donut shape. For the resonant frequency of 1.22 GHz, thegain has increased from 4.2 dB for the air case to 5.6 dB for thecase when SRR magnetic inclusions were used between the twoantennas. We also notice a dip in at the resonance frequency.Fig. 13(b) shows the gain in the H-plane (xy-plane, ) forthe aforementioned cases. It is observed that the MNG inclu-sions help in achieving quasi-orthogonal patterns and reducingthe back radiation in comparison to the air case. This feature ofachieving orthogonal or quasi-orthogonal patterns is desirablein order to maintain low correlation between antenna elements.As there is a trade-off between mutual coupling suppression andattained envelope correlation when incorporating the magneticinclusions, the two antenna system, when used for MIMO appli-cations, can still operate at a frequency of 1.2 GHz where highsuppression level for mutual coupling and lowenvelope correlation below 0.1 were achieved. Fig. 14 shows thefar-field patterns for the two antenna elements with MNG in-clusions at frequency of 1.2 GHz. The antenna system achievequasi-orthogonal patterns when using the magnetic inclusions.Due to the nature of the magnetic resonators losses, the radia-tion efficiency of the antenna system with magnetic inclusionslayer dropped to 60% at the SRR’s resonance of 1.24 GHz. De-spite the encountered losses, the gain has been enhanced whenusing the magnetic inclusions at a frequency of 1.2 and 1.22GHz where the radiation efficiency was almost 93% and 80%respectively.
D. MNG Decoupler Effectiveness
The effectiveness of the MNG as an antenna decoupleris experimentally investigated. Parametric analysis based onvarying the number of MNG strips is considered. Results are
BAIT-SUWAILAM et al.: ELECTROMAGNETIC COUPLING REDUCTION IN HIGH-PROFILE MONOPOLE ANTENNAS 2901
Fig. 15. Parametric study for the measured mutual coupling between themonopole antenna elements when MNG strips are varied.
Fig. 16. Snapshots of the H-field in the transverse plane (i.e. ��-plane) forthe monopole antenna system with: (a) MNG metamaterials ���� � ����, (b)MNG metamaterials ���� � ������, and (c) without any spacers ���� �������. Note that a maximum H-field strength of 1 A/m is used within the snap-shot plots.
shown in Fig. 15. It is observed that by inserting 2 strips ofthe MNG metamaterials, the effectiveness of such materialsin suppressing the mutual coupling is diminished. However,when using 8 strips, a minimum of 44 dB suppression wasachieved, and the 10 strips case is as illustrated before andshown in Fig. 9. A case study was also conducted to investigatethe effects on the envelope correlation between the antennaswhen varying the number of MNG strips from 2 to 12 at theantennas resonant frequency. Although it is not shown, it wasobserved that the use of 8 MNG strips is sufficient to achievelow correlation between the antenna elements, below 0.25.
Fig. 16 illustrates how the displacement current is mitigatedusing the developed single-negative magnetic materials. Inorder to show the effectiveness of the developed MNG inclu-sions, one of the antennas is fed while the other element is
terminated with a 50 load. The HFSS full-wave solver isused in this context. Two normalized separation distances of
and 0.125 are shown for two antenna elements withthe developed MNG inclusions and compared with air case(no spacers). By placing the MNG metamaterials between thetwo antennas, much of the radiated energy is blocked and notransmission of energy is observed within such materials. Thestrength of the coupling path can be seen for the case of nospacers between the antenna elements, due to the strong mutualinteraction between the antennas. The MNG layer effectivelyreduced the mutual coupling between the antennas and miti-gated the displacement current and as such the coupling patheffect diminished.
V. CONCLUSION
Mutual coupling between closely-spaced high-profilemonopole antennas was investigated with particular focus onmultiple-input multiple-output (MIMO) systems. Guidelinesfor the design and analysis of single-negative magnetic meta-materials as an antenna decoupler were discussed in detail.The metamaterials constructed in this work have high negativeeffective magnetic response. Effectiveness of the magneticinclusions was experimentally investigated with emphasison the effect of the number of MNG strips inserted betweenclosely-spaced monopole antenna elements. The magneticinclusions show their effectiveness in terms of reducing the mu-tual coupling and their shielding effectiveness in suppressingthe displacement current. Thus the MNG layer is an efficientmagnetic shielding wall that can be advantageous in antennaapplications and EMI problems.
Numerical and experimental results show that more than 20dB reduction in mutual coupling between antenna elements wasachieved. Moreover, good impedance matching can be achievedwhen using the MNG inclusions, unlike the case without MNGinclusions. An interesting added advantage when using theMNG metamaterials is the ability to increase the directivity,which can be useful for point-to-point communications. Theenvelope correlation between high-profile monopole antennaelements was computed and assessed using scattering param-eters formulation and far-field radiation pattern method. It isshown that the correlation based on the far-field pattern is moreaccurate than the correlation based on scattering parameters.
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Mohammed M. Bait-Suwailam (S’98) receivedthe B.E. degree from Sultan Qaboos University,Muscat, Oman, in 2001 and the M.A.Sc. degreefrom Dalhousie University, Halifax, Nova Scotia, in2004, both in electrical and computer engineering.He is currently working toward the Ph.D. degree atthe University of Waterloo, Waterloo, ON, Canada.
In 2001, he was appointed as a Lecturer at SultanQaboos University, where he was involved withteaching, academic advising and research. From2002 to 2004, he was a teaching assistant at Dal-
housie University, while pursing his masters degree. His research interestsinclude electromagnetic bandgap (EBG) structures, metamaterials for EMIreduction in antenna systems, computational electromagnetics and EMI/EMCapplications.
Mr. Bait-Suwailam is the recipient of two scholarships from the Sultan Qa-boos University. Bait-Suwailam is an IEEE student member since 1998.
Muhammed Said Boybay (S’07–M’09) receivedthe B.S. degree in electrical and electronics engi-neering from Bilkent University, Turkey, in 2004 andthe Ph.D. degree in electrical and computer engi-neering from the University of Waterloo, Waterloo,ON, Canada, in 2009.
From 2004 to 2009, he was a research andteaching assistant in the Mechanical and Mecha-tronics Engineering, and Electrical and ComputerEngineering Departments of the University of Wa-terloo. Currently, he is a Postdoctoral Fellow in the
Department of Electrical and Computer Engineering, University of Waterloo.His research interests include double and single negative materials, near fieldimaging, electrically small resonators, electromagnetic bandgap structures andEMI/EMC applications.
Omar M. Ramahi (F’09) received the B.S. degree inmathematics and electrical and computer engineering(summa cum laude) from Oregon State University,Corvallis and the M.S. and Ph.D. degrees in electricaland computer engineering from the University of Illi-nois at Urbana-Champaign.
From 1990 to 1993, he held a visiting fel-lowship position at the University of Illinois atUrbana-Champaign. From 1993 to 2000, he workedat Digital Equipment Corporation (presently, HP),where he was a member of the alpha server product
development group. In 2000, he joined the faculty of the James Clark Schoolof Engineering at the University of Maryland at College Park as an AssistantProfessor and later as a tenured Associate Professor. At Maryland he was alsoa faculty member of the CALCE Electronic Products and Systems Center.Presently, he is a Professor in the Electrical and Computer EngineeringDepartment and holds the NSERC/RIM Industrial Research Associate Chair,University of Waterloo, Ontario, Canada. He holds cross appointments with theDepartment of Mechanical and Mechatronics Engineering and the Departmentof Physics and Astronomy. He has been a consultant to several companiesand was a co-founder of EMS-PLUS, LLC and Applied ElectromagneticTechnology, LLC. He has authored and coauthored over 240 journal andconference papers. He is a coauthor of the book EMI/EMC ComputationalModeling Handbook, 2nd Ed. (Springer-Verlag, 2001).
Dr. Ramahi serves as an Associate Editor for the IEEE TRANSACTIONS ON
ADVANCED PACKAGING and as the IEEE EMC Society Distinguished Lecturer.
