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A Current-Mode Control Techniquewith Instantaneous Inductor-Current Feedbackfor UPS InvertersHongy ing Wu*, Dong Lin*, Dehua Zhang*, Kaiwei Yao**, and Jinfa Zhang*

*Department of Electrical Engineering, Zhejiang University, Hangzhou310027, P. R. China**Hwadar ElectronicsCo., td., Shenzhen5 18067, P. R. ChinaAbsfract Acurrent-mode control technique with output filter

inductor-current instantaneously controlled is proposed forvoltage-source inverter of uninterruptible power suppliesUPSS) This technique shows good dynamic responsibility,high stability and current limiting in case of load short Small-signal analysis, parameter design, simulation and experimentalresults are given in this paper.

I . INTRODUCTIONFor Uninterruptible Power Supply (UPS) nverters, it isimportant to have high stability and reliability as well as fastdynamic responsibility particularly under nonlinear loadssuchas computers. The control technique used is one of themost significant factors affecting the whole performance ofthe system. Sine pulse width modulation (SPWM) techniqueis preferred to obtain a sinusoidal output-voltage for thereason of simple control scheme and easy control of theharmonic content in the output-voltage. The traditionalaverage voltage feedback SPWM control only regulates theamplitude of the output-voltage with the waveform open-loop

controlled, showing very slow dynamic response to loaddisturbance and poor waveform under nonlinear loads [l].So instantaneous voltage feedback SPWM control techniqueshave been developed and widely used in the past years [2].The waveform of the output is instantaneously regulated bythe comparison of the instantaneous voltage feedback with asinusoidal reference. The influence of component nonlinearcharacteristics and dc voltage fluctuation is restrained.Dynamic performance is greatly impved, but still not goodenough for rush current and nonlinear load because onlysingle voltage-mode control is used. Furthermore, thedifficulty of the system stability design israised.To overcome these drawbacks, current-mode controltechniques were developed and have been verified to beeffective in improving the system stability and dynamicresponsibility [l], [3]. Current-mode control is basically amultiple-loop control method in which the current negative-feedback loop is commandedby the error signal of the outervoltage regulation loop. In terms of the typical current-modecontrol techniques, hysteresis current control, predictivecurrent control and SPWM current control have been

reported. Hysteresis current control has a fast transientresponse, but the switching frequency varies widely [4]. So avariety of improved constant-frequency hysteresis currentcontrol techniques were proposed with additional circuits foradaptive hysteresis band [5]-[8]. To reduce the Complica-tion, most of them are implemented by digital techniques.Thus switching fiequency is limited by operation andDIA orA/D ime, which can not well satisfy the requirement of sizereducing and fast dynarmc response of UPS Predictivecurrent control requires a good knowledge of loadparameters, in addition to having the same calculationproblem [8], [9]. In contrast, SPWM current control, withcomparison of instantaneous current error with triangularwaveform, not only maintains constant switching frequencybut also provides fast dynarmc response for U P S application.Meanwhile, the control circuit is relatively simple. Some ofSPWM current control techniques with regulation of filtercapacitorcurrent for LC filter-VSI inverters wereproposed in[IO] [l 11. The output-voltage is differentially pre-rectifiedby the control of capacitor current. The sensitivity toparameter variations is reduced and the robustness is muchimproved. Furthermore, the scheme has very fast dynamicresponse in both linear and nonlinear load applications.In this paper a SPWM current-mode control techniquewith filter inductorcurrent instantaneously controlled isproposed. The current in the inductor includes both filtercapacitor current and load current. So the differential pre-rectification is retained and the load current can be closelycontrolled, which contribute to the fast dynamic response,high stability and current-limiting in case of load short aswell as easy load sharing for parallel operation of UPSsystems [121,

11. OPERATIONOF T H E PROPosEDCONTROLTECHNIQUEThe controldiagram of the proposed current-mode control

technique on a LC-filter inverter is shown in Fig. 1. Asingle-phase half-bridge structure is used for clear andconvenient discussion. S1 S2 are switches,Ed Ed- aredc supplies,and L C are inductor capacitor that composethe output filter, Multiple-loop control is necessary in thiscontrol method. The voltage control loop acts as the outerloop. The output-voltage feedback is compared with a sine

0-7803-5160-6/99/ 10.00 1999 IEEE. 95 1

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reference signal and he error voltage is compensatedby a PI-regulator to produce the current reference i , . The proposedinstantaneous current control loop acts as the inner loop. Theswitching current through the inductor is sensed andcompared with i,, . After compensatedby a P-regulator, theerror signal v,, is compared with a triangular waveform v , togenerateSPWM ignal for switching control of S 1 & S2.The switching control operation is represented in Fig. 2(supposing that the gain of the P-regulator equals 1). Theoperation of the current control loop is equivalent to acomparison of the inductorcurrent feedback i , and theequivalent reference i , +v , . S keeps on and S2keeps offas i , is less than i,, +v, , and i increases. As i reachesthe values higher t h n i , +v , ,SI ums off, S2 turns on, and

Ra

mS2 D2

LoadResistanceGam Of The Indudor CmatFeedbackGam Of The Output-voltage Feedback

Fig. 1. Controldiagam of theproposed controltechnique

1k +-Ts

i , decreases. It can be seen that as long as the slope of theinductorcurrent is kept less t h n that of the triangularwaveform, the SPWM switching signal c n be carried outand the high fresuency ripples of the inductorcurrent c n belimited in a window decidedby the triangular waveform.

111. STABILITYNALYSISWith small-signal averaging model, the control block canbe obtained as shown n Fig. 3(supposing resistant load forsimplicity). The definition of the parameters in Fig. 3arelisted inTABLE .

Transfer FunctionOf The PI-RegulatorGam Of The P-Regulator

1 1

kmRCS + 1

RICS +LS + R

Fig. 2. The operationofthe prcposed urrent ontrol

Gam Of The Effedive S P W M AmplifierAdmatance Of The Output Filter And Load

Fig. 3. Block diagram ofth e proposed method

TABLEDEFINITIONFTHEPARAMETERSN FIG.3

I mpedanceOf The FilterCapacitor And LoadRClS + 1952

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The -er functionscan be derived as:Open-loop transfer functionof the systemLw R kpS +1G = L - .S(RLCSz+ L +ak,k,,,RC)S+R + km] (1)

Closed-loop ransfer functionof the system

k,k,,,R(k,zS +1)- RLCZS +(L+akik,,,RC)zS2+ ( R +akik,,,+Pb,k,,,k,R)zS+ @ , k S( 2 )

-on (1) canbe re-written as1qAU IBikmkv k JG , ( S ) =-

1 ak,k, 1 ak,k,,, S[S2 +(-+- ) S + - + - ]C RC L LC RLCThen the open-loop zeros andpoles are determined:

1s - - -9 k.7z -

i1---) --r I(---Iak,k* 2 J 4RC L LCHere, Spz , , are negative reals under the conditions that

IR > and &,km > 2 , / L / C . Such conditionsak,k,C - 2 J L ccan always be satisfied under an appropriate design. Thenthe root-loci of the closed-loop system under Merent loadconditionscan be dr wn as Fig. 4. It can be seen from thefigurethat the system has stable closed-loop poles under anycondition. The asymptote of the loci is parallel to theimaginary ax is , and the intersecting point of the asymptote

1 1 ak,k,,, 12 RC L kvrwith the real a x i s lies on Q =--(-+- --) . Theasymptote will go nearer o the imaginary axis asR increases.

But si ne ~0mm0dy 2 m

and -

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That is voltage ripplesare relevant directly with the inductorcurrent

where,ripples, zero output-voltage condition is used o design L and

5 ) C. combining

(7)

The simplif~edystem is a combination of a proportionalunit and a Zndader resonant unit. The damping coefficient5; and the regulating time t , (proportional to

EdAIh c-fsw

(9)and

yields the determinationof L and C.Under current-mode control,parameter design of controlloop is focused on the inner loop. The voltage feedbackcontroller can be designed conventionally, while the currentfeedback controller should be designed carefully for thereliable comparison of the error signal with the triangularwaveform. The slope of the inductorcurrent feedbackmustbe less th nthatof the triangular waveform:

where Km is the peak-peak value of the triangularwaveform. Meanwhile, the gain k, should be selected aslargeas possible to improve the current tracing.Adesignexample is given inTABLE 11. According to thedata listed, the frequency responsesof Fquations 1) and (2)

R C Lthe load especially under light load. Fig. 6 shows themagnitude and phase responses of (5). The phase margin islarge enough for stability consideration.Iv. PARAMETER DESION are shown in Fig. 7andFig.8respectively. From Fig 7, the

1il;;ited or& to reduce the l ndE M Electro-Magnetic 50Hz o the output-voltage canwell trace the refence. TheInterference). Because the largest high frequency inductor-current ripplesare around zero output-voltageand he output- high resonant-frequency and wide frequency bandwidthensure the fast dynarmcresponse.

U w.2 UFig.6. Magnitude and phase responsesof (5)

TABLE1SPEClFICAnONS AND PARAMETERSOF THE YSTEM

PWIUlWteRj

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ala) Magnitude response

I5ms 1Oms 15ms 2OmsTime

0 Fig. 9. Simulationresults ofthe steady-state waveform of output-voltage-50 5 OA)

e m -100f i m )

-150

OA-200

m(b) Phaseresponse

Fig. 7. Bode diagram of 1) with given parameters40

ma) Magnitude response

.oA OS 5ms 10ms 15ms m m sTime

Fig. 10. Simulation results of th e steady-- waveform of indudor currenthalf load)

5ms 1Oms 1511s 2OmsTime

Fig. 11. Simulation r e s u b of the dynamic response waveform(no load to half load)

m(b) Phase response

Fig. 8. Bode diagam of(2) with givtm parameters

v. SIMULATION AND EXPERIMENTALJZSULTSprototype of lkVA current-mode controlled voltage-source U P S inverter has been built in laboratory. The

9-12 and Fig. 13-15 show the simulation and experimental- 0 s 5mS 1Oms 15m 2Omsspecifications nd pameters are as listed in TABLE I. Fig.

results under resistant load condition respectively.Time

Fig. 12. Simulation resuks of current limitmgunder loadshort

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Fig. 13. Experimentalresultsof th e outp ut-voltage and inductor-currentunder no load condition

(voltage=lOOV/div, current4Ndiv)

L

Fig.14. Experimental results ofthe outprd-vobge and inductor-currentunder half load condition

(voltage=lOOV/div, current=&Vdiv)

LFig.15. Experimentalresults of th e out put-voltage and inductor-current

under abrupt load ange fkom no loadto half load condition(voltage=lOOV/div, currmt=4A/div)

Fig. 9shows the steady-state waveform of the output-voltage. The output-voltage has a good sinusoidal waveformand very small rifles, and traces well with the reference.The total harmonic distorlion of the output-voltage iscalculated to be 1.3 .Fig. 10shows the inductorcurrent, which is composed ofa low-frequency current and high-frequency ripples. It can

be seen that the ripples are limited within a certain band andthe largest ripples happen around the zero output-voltage.The dynarmc response to the abrupt load change ftom noload to half load is shown in Fig. 11. The whole transienttime is shorter th n lms and the output-voltage has a lowdistortion.Fig. 12 shows the situation of load short. When load shorthappens, the output current is limited tightly to a given value.By appropiate designing, the output current can be limitedunder a safe value.The experimental results shown in Fig. 13-1 5are coincidewith the simulation results. Fig. 13shows the steady-statewaveforms of the output-voltage and the inductorcurrentunder no load condition. The inductorcurrent is sensed witha current transformer. The figure shows the output-voltage isnearly perfectly sinusoidal. The inductorcurrent is just thefilter capacitorcurrent. Fig. 14 shows the waveforms underhalf load condition. The inductorcurrent includes the load-current in additionto the capacitorcurrent.Fig. 15 shows the dynamic response to the abrupt loadchangmg. The inductorcurrent is under perfect control,which contributes to the good performance of the output-voltage.In the tests of the prototype, the load short experiment hasalso been accomplished. The outputanent can be limitedeffectively to a safe value for power devices. But this valueis still relatively large because it must be larger th n rushcurrent under normal operat1 11in actual application. In theexperiment this value is measured as 16A. To avoidunnecessary loss, the limit setting should be drawn to zerowhen load short is detected

VI.CONCLUSIONAn altemative instantaneous inductorcurrent controlledSPWM current-mode control technique for voltage-sourceUPS inverters has been presented, which has the bothadvantages of current-mode control and SPWM ontrol.With this control method, not only the output-voltage can

be differentially pre-rectified to improve the transientresponse, but also the output current can be directlycontrolled The stability analysis shows that the systemperforms well from no load to full load, with small over-regulating magnitude and short regulating time agatnst loadchange. It is confirmed by the simulation and experimentalresults that the system has an excellent performance with lowharmonic distortion in output-voltage, much i m p e doperation stability and reliability, as well as fast dynanucresponsi0ility.The proposed current-mode control technique can beapplied in UPS and all other inverters requiring highperformance. It is also attractive for the inherent outputcurrent limitation in case of load short and the easy paralleloperation and capacity expansion of U P S systems withinstantaneous load current sharing.

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