Architectures and RF System Design Issues for 3G transceiver systems

10/26/01 Walid Y. Ali-Ahmad 1

Architectures and RF System Design Issues for Integrated Receivers and Transmitters in

3rd Generation Wireless Handsets

Walid Y. Ali-AhmadSenior Member of Technical StaffWireless Communications Group

Maxim Integrated ProductsSunnyvale, CA


TALK OUTLINE

Introduction: Example of present level of integration in an RF chipset for CDMA cellular radioReceiver Architectures

Heterodyne ReceiverImage-Reject ReceiverDirect-Conversion ReceiverLow-IF Receiver

Transmitter ArchitecturesIF-Modulation / Up-conversion TransmitterDirect-Modulation TransmitterOffset-PLL Transmitter

Summary


Example of Present Level of Integration in an RF chipset for CDMA Cellular Radio


Heterodyne Receiver

Advantages:Down-conversion to baseband I&Q is done at an “Intermediate-Frequency” (IF) lower than RF. This results in superior I & Q matching.

Its selectivity, which measures the receiver’s capability to process a desired small channel in the presence of close-in strong interferes, is done partly at IF using a highly selective SAW filter and at baseband I&Q using low-pass baseband filters.

The use of the IF SAW filter relaxes the linearity requirements (IIP2, IIP3) of the succeeding IF and baseband stages.

DC offsets at baseband I&Q do not limit its sensitivity because they are minimized by the fact that the first LO frequency is not equal to the input RF carrier frequency.

DUPLEXER

LNARF SAW

BPF Mixer

IF SAWBPF

1st LO 2nd LO

090

IFAGC

PLL1 PLL2XREF XREF

Presenter

Presentation Notes


Heterodyne Receiver (cont’d)Design Issues:

The need to use an an off-chip passive BPF for image rejection adds cost and board space requirements.

Currently, due to technology limitations, this “image-reject” BPF and the IF SAW filter can not be integrated on-chip.

The trade-off between image rejection and channel selection is key in determining the IF frequency.

Low frequency IF SAW filters (40-150MHz) have high Q and provide high adjacent channel selectivity. However, These filters tend to be large.

High frequency IF SAW filters (150-400MHz) have a relatively smaller physical size, but provide a lower adjacent channel selectivity.

Good frequency planning is essential in order to minimize spurious responses generated in the receiver’s front-end (Fs = ±m⋅FRF ±n⋅FLO1 ±p⋅FLO2) .

The “half-IF” spurious response at (FRF+ FLO)/2 can be a serious problem in the case of low IF frequency. The front-end mixer after LNA should have a low 2nd-order distortion and a high suppression of the (2FLO×2FRF) product.

WantedSignal

RFIF

LO

fRFfLO fRF- fIF/2

Half-IF

2x2 product

WantedSignal

IF


Importance of IF Selectivity for Suppression of 3rd-order IM products

Typical Two-stages cascaded IIP3 equation, in linear format:: ;“g1” is gain of stage 1; “il” is insertion loss of IF filter

Equivalent IIP3 of IF block when including selectivity “S” (dB) ahead of IF stage:

Generalized equation for overall IIP3 of a receiver chain with M cascaded stages:

IL

S

IF Filter#1

IIP31

G1

Block #1

RF Block IF Block

IIM31

IIP32

G2

IIM32

IIP3IIM3

PI

P1 P2

Block #2

On-ChannelPassband

PCW_tone

PIIM3

Off-ChannelInterferersOn-Channel

Signal

IM3 product

(C/I)

2

1

1 331

31

iipil)g(

iipiip+=

(dBm) 2333 )

233(2332)(33 2221212 S;IIPIIPSIIPPIIPSPIIM e

oo ⋅+=⇒⋅+⋅−⋅=⋅−−⋅=

23121

12123

213

2123

12

1

1 33331

31

/M-M

M// )ss(siip

ggg)s(siip

ggsiip

giipiip ⋅⋅⋅⋅

⋅⋅⋅++

⋅⋅

⋅+

⋅+= −

L

LL


Importance of RF Selectivity for Suppression of 2rd-order IM products

Two-stages cascaded IIP2 equation, in linear format: ; “g1” is gain of stage 1; “il” is insertion loss of image-reject filter

Equivalent IIP2 of mixer’s block, including selectivity “S” (dB) ahead of mixer stage:

Generalized equation for overall IIP2 of a receiver chain with M cascaded stages:

On-ChannelPassband

IIM2

IM2 product

On-ChannelSignal

fRF fLOfLO- fIF/2

Half-IF

PI

(C/I)

IL

S

RF Filter#2

IIP21

G1

Block #1

RF Block

IIM21

IIP22

G2

IIM22

IIP2IIM2

PI

P1 P2

Block #2

1st Mixer

2

1

1 221

21

iipil)g(

iipiip+=

2121

1212

213

21212

1

1 22221

21

)ss(siipggg

)s(siipgg

siipg

iipiip M-M

M

⋅⋅⋅⋅

⋅⋅⋅++

⋅⋅

⋅+

⋅+= −

L

LL

(dBm) 222 )22(22)(22 2221212 S;IIPIIPSIIPPIIPSPIIM eoo ⋅+=⇒⋅+−⋅=−−⋅=


Image-Reject Receiver

Advantages:It facilitates the integration of the heterodyne receiver’s front-end by eliminating the use of the off-chip RF image-reject filter and accomplishing the image rejection on-chip through phasing.

It works nicely in receiver systems which do not suffer from strong out-of-band blockers and do not require interstage filter between front-end LNA and MIXER blocks.

It is suitable for receiver systems using a very low IF frequency (e.g. 10.7MHz, 45MHz), since it eliminates the need for a very high Q bandpass RF filter in order to reject the image .

Hartley Image-Reject Receiver Weaver Image-Reject Receiver

Mixer I

Mixer Q

LO1

Desir

ed

Imag

e IF

IF

RFInput

LO1

090

sin(ωLO1t)

cos(ωLO1t)

LO2

090

sin(ωLO2t)

cos(ωLO2t)BPF

LO

090

Mixer I

Mixer Q

sin(ωLOt)

cos(ωLOt)

LO

Desir

ed

Imag

e IF

IF

RFInput

R

C

C

R

LPF


Image-Reject Receiver (Cont’d)Design Issues:1– Hartley Architecture:

Image rejection is limited by amplitude and quadrature phase mismatches. Amplitude mismatches are minimum when using ωIF= 1/RC. Phase mismatches are due mainly to errors in the LO quadrature generation circuit.

Image Rejection Ratio IRR (dB) = 10∗LOG([(ΔA/A)2 + θ2 ]/4); for (ΔA/A)<<1, θ<<1rad.

For typical matching in integrated circuits, image suppression falls in the range of 30 to 40dB (0.2-0.6dB gain mismatch & 1°-5° quadrature phase mismatch). In most RF systems, 60-70dB of image rejection is required. The front-end filter (duplexer) normally makes up for the remaining required image rejection (~30dB).

For the RC-CR 90° phase-shift network, (ΔA/A) = (ΔR/R) + (ΔC/C), at ωRC≈1.

2– Weaver Architecture:

It is also sensitive to mismatches, but it is free from gain imbalances due to the RC-CR phase shift network, thereby achieving greater image rejection despite process and T° variations.

The Weaver architecture suffers from the “secondary” image problem because of the use of a second mixing operation. The LPF (or HPF) in between 1st and 2nd mixing stages is used to suppress the secondary image.

The problem of secondary image can be eliminated if we choose a Zero IF frequency at the output (FLO1± FLO2 = FRF). To its advantage also, 2nd-order distortion in the signal path can be removed by the bandpass filters following the first mixing operation.


Image-Reject Receiver (Cont’d)A 1.9GHz Wide-Band IF Double Conversion Receiver (J.C. Rudell, et al., UC Berkeley, IEEE JSSC, December 1997):


Direct-Conversion Receiver

In current cellular systems where signals are either frequency- or phase-modulated, direct downconversion must provide quadrature outputs so as to avoid loss of information, since the two sidebands of the RF spectrum contain different phase information.

Advantages:The problems of image frequency and “half-IF” spurious response are eliminated since FIF = 0.RF BPF after LNA is optional; it is only needed for additional rejection of out-of-band interferers and TX power leakage. The bulky off-chip IF SAW filter is eliminated. All channel selectivity is done at baseband with low-pass filters and baseband amplifiers.One VCO and one PLL are needed for the whole receiver.

BPF #1 LNA

LPF I

Q

Mixer I

LPF

AGC

AGC

090

BPF2

Mixer Q

Cext

PLL1 XREF


Direct-Conversion Receiver (Cont’d)

Design Issues:1– DC offsets:

Static offsets are caused by process mismatch and drift of analog circuitry that vary slowly vs. T°, aging, and current gain setting.Time-variant offsets are caused mainly by parasitic LO coupling to mixer RF port, LNA input port, and antenna port. LO “self-mixing” occurs in mixer and it produces a dc component at the mixers I & Q baseband outputs. Time-variant offsets can also be caused by a large interferer which can leak from LNA or mixer input to LO input port and self-mix with itself to produce a dc offset at mixers outputs.The time-variance is due to reflection of LO leakage against moving objects back to receiver and due to receiver movements. Maximum frequency content of time-variant DC offset due to Doppler shift = 2∗νmax/λ; where νmax: maximum moving object or car speed.

BPF #1 LNA

LPF I

Q

Mixer I

LPF

AGC

AGC

090

BPF2

1st LO

Mixer QLOLeakage

InterfererLeakage

Cext



2– DC offsets Cancellation Techniques:DC offsets can easily saturate the receiver’s final baseband output stages. Hence, DC offset removal or cancellation is required in direct-conversion receivers:

− DC blocking or High pass filtering: − it is feasible in non-burst mode systems which are receiving continuously (FDD). − In order to minimize distortion of signal, the high-pass corner should be < 0.1% of the data

rate for random binary M−ary data.− The baseband signal in the transmitter can be encoded to result in “dc-free” modulation

scheme, such as FSK with βm > 1 or wideband Direct-Sequence Spread-Spectrum signals.− DC calibration loop: In TDD systems, periodic offset cancellation can be performed during idle

times where the DC offset is stored on a capacitor and then subtracted from the received signal during actual reception.

− Adaptive DSP techniques have been used for DC offset-cancellation in TDD systems.



DC offset cancellation in Pager system using HP filtering

Eye Diagram Distortion with

HPF:a) No filtering;

b) Fc = 1% of Rb;

c) Fc = 0.1% of Rb



3– LO leakage:

LO coupling to the antenna will be radiated out and will create interference in the receive band of other users equipment using the same wireless standard.

In order to minimize this problem of LO leakage and re-radiation, it is important to use differential LO and RF inputs to the receiver IC to cancel out common mode signals. In addition, LO leakage is further reduced by fully integrating the RF VCO tank on chip.

4– Flicker Noise:

The 1/f noise of devices in the baseband section of a Zero-IF receiver can substantially corrupt the down-converted signal after the mixers I/Q outputs, especially in MOS implementations (1/f corner ~ 200kHz).

The effect of flicker noise can be reduced by the use of active mixers with bipolar transistors in switching pairs and by the use of large MOS devices for baseband filters and amplifiers.

High pass filtering at baseband, when used as part of the DC offset cancellation, can reduce the integrated 1/f noise at baseband.

Integrated total noise at baseband including 1/f noise can be expressed as following:

cornerfrequency High : corner,frequency low : corner, 1/f :/1 density, spectralpower noise Thermal :

;)()/1ln(/12

HfcfffthS

thScfHfcf

ff

ffthSnV ⋅−+⋅⋅=



Mixer Topology that minimizes 1/f noise at its baseband output

Interference due to down-converted 1/f noise and DC offset



5– I/Q mismatch:

Assuming that the received signal can be written as vin(t)= I(t)∗sin(ωCt)+Q(t)∗cos(ωCt), and the amplitude and quadrature phase imbalances in the I/Q down-converter are “ε” and “θ”, respectively, we can write the baseband I/Q outputs, after demodulation, as:

vBB,I (t)= I(t).(1+ε/2).cos(θ/2) − Q(t).(1+ε/2).sin(θ/2);

vBB,Q (t)= −I(t).(1−ε/2).sin(θ/2) + Q(t).(1−ε/2).cos(θ/2);

As we see from equations above, Gain error “ε” appears as a non-unity scale factor in the amplitude, while phase imbalance “θ” results in cross-talk between demodulated I and Q waveforms degrading the SNR (I & Q data streams are usually uncorrelated).

− In practice, ε < 1dB and θ < 5° for SNR degradation less than 1dB (for QPSK signals).

The full on-chip integration of the Zero-IF receiver and the minimization of devices mismatch reduce drastically the amplitude and quadrature phase imbalances in the I/Q down-converter.

6– Channel Filtering:

Baseband channel low pass filters in a Zero-IF receiver need to have a high dynamic range:− Receiver sensitivity can’t be compromised.− Close-in interferes should be rejected without causing in-band distortion.

External capacitors at mixers I&Q outputs can be used to provide additional selectivity to blocking and out-of-band signals (pole @ 1/RCext)



7– Even-Order Distortion:

Assume vRF(t) = A1∗cos(ω1t)+A2∗cos(ω2t), the LNA output will contain an IM term at frequency f1-f2, resulting from 2nd order non-linearity in the LNA. This IM2 product at LNA output will leak to mixer’s output because of finite feedthrough from RF input to IF output (−30…−40dBc).

Special attention to the mixer’s design is required since IM2 product can also be generated in the mixer’s RF port by two tones interferes after being amplified in LNA.

The 2nd order non-linearity in the LNA and in the mixer will also demodulate any AM component on the received signal due to fading during propagation or Nyquist filtering.

Based on input two-tone interferers level and the resultant low-frequency IM2 level at baseband output, a receiver’s 2nd order intercept point (IP2) can be derived.

Using differential LNA output and differential mixer’s input will suppress the generated common-mode 2nd-order IM products. As a result, receiver’s IIP2 can be improved.

RF

LO

IF

WantedSignal

fRFf2 f1

WantedSignal

IM2Interferer

LNA0

Feedthrough

0

f1-f2

Interferers


Low-IF Receiver

Advantages:The received signal is down converted to a low-IF frequency, which is normally one to two times the signal BW.

It has the same advantage of Zero-IF receiver in terms of the integration of channel filters.

It is less susceptible to 1/f noise.

It is less susceptible to DC offsets since the bulk of signal energy is not centered around DC.

DC offsets cancellation scheme can be simplified.

Very low frequency IM2 products can be easily blocked.

BPF #1 LNA

I

Q

Mixer I ComplexPolyphase

Filter

AGC

AGC

090

BPF2

Mixer Q

Cext

PLL1 XREF


Low-IF Receiver (cont’d)

Design Issues:The A/D converters at baseband output require to have a higher sampling rate than in the case of a Zero-IF receiver.

Image suppression is an issue; Very good amplitude and phase matching between I & Q baseband channels is required to obtain > 35dB image suppression.

Careful choice of the IF can place the image signal in the adjacent channel.

− In order to discriminate between these two signals, it is essential to process I & Q outputs as a complex pair.

− Complex Polyphase filters is essential to obtain the necessary reject in the adjacent channel

2nd -Order Distortion can still result in in-band channel intereference.


Baseband I&Q signals undergo quadrature modulation at an intermediate IF frequency (ωIF). The following IF filter (BPF1) rejects the harmonics of the IF signal. The IF modulated signal is then up-converted to (FIF ± FLO2).

The unwanted sideband imposes tough rejection requirements on BPF2, typically 50-60dB, in order to meet transmitter’s spurious emissions levels imposed by standards.

This topology does not allow full transmitter’s integration because of use of off-chip passive devices such BPF2 and BPF1.

On-chip I and Q matching is superior since modulation is done at IF and not at RF. This will lead to better EVMs and lower cross-talk between I & Q channels.

IF filtering reduces transmitted noise in RX band.

Wide power control dynamic range because control it is distributed between RF and IF sections.

IF-Modulation / Up-Conversion Transmitter

090

Mixer I

Mixer Q

I

Q

Tank

PA

LO1

cos(ωIFt)

sin(ωIFt)

TankLO2

IF RF

DUPLEXER


Direct-Modulation Transmitter

In direct-conversion transmitters, the baseband signal is directly modulated unto the RF carrier. The output carrier frequency is equal to the LO frequency at mixers inputs.

This topology is attractive for full transmitter’s integration since it does not use an intermediate IF stage with upconversion and interstage IF filter.

Its main disadvantage is the corruption through “injection pulling” of the VCO spectrum by the high level PA output. Isolation required is normally > 60dB.

The isolation can be highly improved by “offsetting” the LO frequency by using 2xLO off-chip and dividing by 2 on-chip or by adding or subtracting another oscillator.

The power control dynamic range is limited by the carrier feedthrough. A fully integrated differential transmitter architecture will minimize carrier feedthrough because of higher of common mode rejection (differential LO inputs and modulator output).

090

Mixer I

Mixer Q

I

Q

Tank

PA

2xLO

/2

carrierFeedthrough

090

Mixer I

Mixer Q

I

Q

Tank

PA

LO

carrierFeedthrough

cos(ωLOt)

sin(ωLOt)

cos(ωLOt)

sin(ωLOt)


Direct-Modulation Transmitter (cont’d)

( ) ( )[ ]

⎥⎥⎦

⎤

⎢⎢⎣

⎡

⎟⎟⎠

⎞⎜⎜⎝

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⎜⎝⎛+Φ⋅⎟

⎠⎞

⎜⎝⎛⋅+⎟

⎟⎠

⎞⎜⎜⎝

⎛⎟⎠⎞

⎜⎝⎛+Φ⋅⎟

⎠⎞

⎜⎝⎛⋅−⋅=∴

⎥⎥⎦

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⎢⎢⎣

⎡

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⎝

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⎟⎠

⎞⎜⎜⎝

⎛⋅⎟

⎠⎞

⎜⎝⎛+⋅⋅⋅⎟

⎠⎞

⎜⎝⎛⋅+⋅=∴

==

Φ+++−Φ+−Φ=∴

Φ+−Φ+Φ+++−=∴

Φ

−

22

22

22

1

22

e

)cos(21)cos(21LOG10 (dB)n Suppressio Sideband

)cos(2141sin2LOG10 (dB)n SuppressioCarrier

)sincossignal modulating tonesingle (assuming :following as calculated becan n suppressio sideband andcarrier The -2

)sin())(()()1()cos()()1)cos(tan EVP(t)

))cos()()1)cos(())sin())(()()1(( EVM(t)

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signals. basebandinput are and error; phase quadraturemodulator is

ly.respective paths, Q & I of errors amplitude are & ly;respective inputs, Q & I basebandat offsets DC are &

I

Qe

I

Q

I

Qe

I

Q

I

Qe

I

QQ

I

QeQI

I

QI

mmm

eQQIIIeQQeQ

eQQeQeQQIII

QIQI

AA

AA

AA

AA

AA

AAOA

A)(ΦOOAAO

t)(ωt); Q(t)(ω; I(t) ω

AOtQAOtIAAOtQA

AOtQAAOtQAOtIA

Q(t)I(t)

AAOO

090

Mixer I

Mixer Q

I(t)

Q(t)

Tank

PA

LO

carrierFeedthrough

cos(ωLOt)

sin(ωLOt + φe)

OI

AI

AQ

DesiredVector

MeasuredVector

ErrorVector

OQ


Direct-Modulation Transmitter (cont’d)

Modulation error or EVM in Transmitters


Baseband I&Q signals undergo quadrature modulation at an intermediate IF frequency (ωIF). Instead of upconverting the IF signal, the phase modulation in the IF signal is transferred faithfully to the TX VCO via the offset PLL topology, with condition that the loop BW of PLL is chosen properly.

This phase translation scheme to RF is only valid for constant envelop modulation signals such in GSM (GMSK).

The PLL LPF helps tremendously at suppressing the out-of-band noise generated by the modulator and, hence, meeting the stringent GSM requirements for the thermal noise in the receive band. This eliminates the need for the off-chip bulky Duplexer.

This topology is quite attractive for low-cost high performance integrated transmitters using constant-envelop modulation. However, enough isolation is required between the TX VCO and the PA in order to suppress VCO pulling by PA output noise.

Offset-PLL Transmitter

090

Mixer I

Mixer Q

I

Q

Tank

PA

LO1

cos(ωIFt)

sin(ωIFt)

Tank

LO2

LPFIF

LPFPLL Tank

TXVCO

LPFMXR

PD

OffsetMXR


SUMMARY

Need for Fully Integrated VCOs.

Need for Mixers with low level of 2nd-order Distortion (High IIP2).

DC offset cancellation schemes have to be implemented without distorting signal.

High Dynamic Range Baseband filters are key for direct-conversion receivers.

Variable Gain PAs improve power control dynamic range in Direct-Modulation Transmitters.

Low Noise Direct-Modulation Transmitters and Direct-Conversion Receivers enable low-cost radios for 3G applications.

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