Di erential feed antenna in millimeter wave Radar

Submitted byZiqiang Tong

Submitted atInstitut fur Nachricht-entechnik und Hochfre-quenzsysteme

Supervisor andFirst ExaminerUniv.-Prof. DI Dr. An-dreas Stelzer

Second ExaminerProf. Dr.-Ing. WolfgangMenzel

Co-SupervisorName of assistant

February 2020

JOHANNES KEPLERUNIVERSITY LINZAltenbergerstraße 694040 Linz, Osterreichwww.jku.atDVR 0093696

Differential feed antennain millimeter wave Radarapplications

Doctoral Thesis

to obtain the academic degree of

Doktor der technischen Wissenschaften

in the Doctoral Program

Technische Wissenschaften

To my parents,

my father Maoda Tong, and my mother Lihua Jiang

and my family,

my wife Jing and my son Michael.

i

Acknowledgements

I would like to take this opportunity to thank all the people without whom this

Doctor dissertation would not have been possible. I would like to thank my advisor,

Prof. Dr. Andreas Stelzer, for all the opportunities you have provided for me to

better understand the field of catalysis. I am also indebted to my co-supervisor

Prof. Dr. Wolfgang Menzel for his guidance and understanding in the course of this

program.

The research was made possible by the financial assistance from the Danube Inte-

grated Circuit Engineering GmbH (DICE GmbH). I appreciate the cooperation and

useful discussions of Dr. Eric Kolmhofer and Dr. Linus Maurer. I really appreciate

all their help and brilliant ideas that helped channelize my research.

I am also grateful to Mr. Ralf Rudersdorfer and Mr. Johann Katzenmayer, for their

assistance in the manufacturing process.

I thank all my committee members for their help, guidance, and knowledge you have

provided me. I would not be where I am now without your questions, suggestions,

and insights. I would like to thank my colleagues at the Institut fuer Nachrich-

tentechnik und Hochfrequenzsysteme (NTHFS) and my research group members

for their help and useful scientific discussions, especially, Dr. Reinhard Feger,

Dr. Thomas Wagner, Dr. Martin Jahn, Dr. Alexander Fischer, Dr. Abouzar

Hamidipour, Dr. Xin Wang, Dr. Christoph Wagner, etc.

Lastly, I would like to thank my family: Jing, my wife, and Michael, my son, for

believing in me and supporting me.

ii

Abstract

This thesis presents a study of differential feed antenna in millimeter radar applica-

tions. In recent years, millimeter-wave radar (30–300 GHz) has been exploited for

a variety of applications, particularly in the automotive industry (76–81 GHz). A

high-level integration of systems is desired to reduce the cost and achieve a com-

pact size for mass production of radar systems. Most of the millimeter chips have

differential topology and RF IOs, while the classic antennas are single-ended struc-

tures. Therefore, there is a growing interest to develop a differential feed antenna for

mmW radar systems. Differential antenna eliminates the need of a balun in the RF

front-end system. This reduces the system size and the system loss. Meanwhile, the

differential feed antennas also bring inherited benefits like lower cross polarization,

etc.

The main focus of the thesis is to develop various differential feed antennas for

mmW radar systems. Three groups of differential feed antennas have been studied.

The first group are differential antennas on a microstrip structure. The microstrip

structure is the most popular layer stack in mmW radar systems. In this part, the

first differential feed microstrip patch antenna is designed based on a rectangular

patch antenna. Then two different antenna arrays – H-plane array and E-plane array

– are developed for increasing the gain of the antennas. The H-plane array provides

wide bandwidth, while the E-plane array provides better radiation performance.

In the second group, the differential feed antenna is implemented as radiation ele-

ments in the transition devices which connect planar structures and air-filled waveg-

uide structures. A couple of novel transitions are designed based on various differen-

tial feed patch antennas. These transitions provide a smooth connection from planar

structures (coupled microstrip lines) to vertical structures (rectangular waveguide

structures). This facilitates the integration of a waveguide antenna – like horn

antenna – in the radar front-end systems. Advances in 3-D printed technology

development mean that this solution has more and more wide applications.

The third group is the differential feed antenna integrated in package. Antenna in

package has much higher integration levels compared with antenna built in printed

circuit boards. The eWLB package is a promising package solution for mmW ap-

plications. It brings new facilities for the antenna in package development. In this

part, differential feed antenna concepts are further extended. Three types of dif-

ferential feed antenna in package are designed in the eWLB package: folded dipole

antenna, folded dipole antenna with cavity in PCB, and dual patch antenna. To

improve the radiation performance, two dielectric lens – hemisphere lens and rod

lens – which are mounted on top of the package are also developed and verified.

As part of this study, all of the antennas/transitions are manufactured and tested.

Measured results are presented and discussed, validating design and simulations.

Meanwhile, the theories for antenna/transition measurement are also developed.

iii

The antennas are implemented in a couple of different systems. For instance, planar

structures are suitable for middle- and long-range radar applications, while waveg-

uide structures are good candidates for high-performance radar where long trans-

mission lines are needed. The antenna-in-package solution is a promising candidate

for ultrashort range (< 20 m) applications, etc.

iv

Contents

Contents v

List of Figures vii

1 Introduction 1

1.1 Radar . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

1.1.1 FMCW radar . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

1.1.2 MIMO radar . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

1.1.3 Radar in millimeter-wave applications . . . . . . . . . . . . . . . . . . . . 3

1.2 Motivation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

1.3 Simulation tool . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

1.4 Thesis structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

2 State of Art for mmW Radar Antenna 8

2.1 Waveguide antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

2.2 Lens antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9

2.3 Reflector antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

2.4 Planar antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

2.4.1 Open-ended transmission line antenna . . . . . . . . . . . . . . . . . . . . 13

2.4.2 Grid antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

2.4.3 Patch antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

2.4.4 Substrate integrated waveguide (SIW) antenna . . . . . . . . . . . . . . . 18

2.5 High-integration antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

2.5.1 Antenna on chip . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

2.5.2 Antenna in package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

2.6 Examples of antenna in realized systems . . . . . . . . . . . . . . . . . . . . . . . 21

3 Differential Microstrip Patch Antenna 24

3.1 Microstrip antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

3.1.1 Microstrip structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

3.1.2 Microstrip patch antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

3.1.3 Cavity model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26

3.2 Differential feed microstrip patch antenna . . . . . . . . . . . . . . . . . . . . . . 30

3.2.1 Prior art work of differential antenna . . . . . . . . . . . . . . . . . . . . . 30

3.2.2 Cavity model analysis for impedance of DMPA . . . . . . . . . . . . . . . 31

3.2.3 mmW-DMPA design at 79 GHz . . . . . . . . . . . . . . . . . . . . . . . . 40

3.3 Differential feed antenna array . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47

v

CONTENTS

3.3.1 H-plane DMPA array . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48

3.3.2 E-plane DMPA array . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54

3.4 Transition for DMPA measurement . . . . . . . . . . . . . . . . . . . . . . . . . . 60

3.5 Application of DMPA/array . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65

4 Coupled Microstrip Line Feed Waveguide Transition 66

4.1 Introduction of transitions from planar transmission lines to waveguide . . . . . . 66

4.2 Novel transition concept with DMPA: First prototype in E-band . . . . . . . . . 69

4.2.1 Design of DMPA transition at 79 GHz . . . . . . . . . . . . . . . . . . . . 69

4.2.2 Manufacturing and measurement of DMPA transitions at WR12 . . . . . 75

4.2.3 Material comparison in transition design . . . . . . . . . . . . . . . . . . . 82

4.3 Improvement for common-mode signal suppression: Prototype design in W-band 83

4.4 Further bandwidth improvement by extended ground: Prototype for E-band

transition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88

4.5 Summary and applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 91

5 Differential Antenna in eWLB Package 93

5.1 Introduction of antenna in package . . . . . . . . . . . . . . . . . . . . . . . . . . 93

5.2 Folded dipole AiP with eWLB package . . . . . . . . . . . . . . . . . . . . . . . . 97

5.2.1 eWLB structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 97

5.2.2 Folded dipole AiP design . . . . . . . . . . . . . . . . . . . . . . . . . . . 98

5.2.3 Manufacturing and measurement . . . . . . . . . . . . . . . . . . . . . . . 99

5.3 Folded Dipole AiP with cavity in PCB . . . . . . . . . . . . . . . . . . . . . . . . 102

5.3.1 Antenna design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102


5.4 Dual patch type AiP . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 105

5.4.1 Antenna design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 106


5.5 Lens over eWLP AiP . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111

5.5.1 Effects of package on radiation performance . . . . . . . . . . . . . . . . . 111

5.5.2 Hemisphere lens design . . . . . . . . . . . . . . . . . . . . . . . . . . . . 112

5.5.3 Rod lens antenna design . . . . . . . . . . . . . . . . . . . . . . . . . . . . 113

5.6 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 120

6 Conclusions and Future Topics 121

6.1 Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 121

6.2 Future topics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 122

Appendix A 123

.1 Dipole antenna resistance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 123

.2 Horizontal electric dipole . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 124

Acronyms 125

References 127

vi

List of Figures

1.1 Radar basic form . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

1.2 FMCW basic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

1.3 MIMO radar . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

1.4 Long-range radar from Bosch . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

1.5 Gauge radar from Siemens . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

1.6 FMCW radar front-end block diagram . . . . . . . . . . . . . . . . . . . . . . . . 5

1.7 Solver Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

1.8 Transient Solver . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

2.1 Waveguide antenna in mmW applications . . . . . . . . . . . . . . . . . . . . . . 8

2.2 Slot antenna on narrow wall of waveguide . . . . . . . . . . . . . . . . . . . . . . 9

2.3 Spherical lens antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9

2.4 Lens antenna fed by planar array . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

2.5 Artificial lens at 76 GHz . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

2.6 Lens antenna for 77 GHz: plano convex and planar lens . . . . . . . . . . . . . . 11

2.7 Cylindrical parabolic reflector antennas . . . . . . . . . . . . . . . . . . . . . . . 12

2.8 Printed folded reflector antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

2.9 Open-ended transmission line antenna . . . . . . . . . . . . . . . . . . . . . . . . 13

2.10 Differential fed grid antenna array on RO3003 . . . . . . . . . . . . . . . . . . . . 14

2.11 Grid antenna on LTCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

2.12 Series-fed patch antenna array . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

2.13 Series-fed patch array in phase-shift receiver system . . . . . . . . . . . . . . . . 16

2.14 Dual-fed phased array . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

2.15 Dual linearly polarized microstrip patch antenna array . . . . . . . . . . . . . . . 17

2.16 Circularly polarized microstrip antenna array . . . . . . . . . . . . . . . . . . . . 17

2.17 Patch antenna in Bosch automotive radars . . . . . . . . . . . . . . . . . . . . . . 18

2.18 SIW antenna on flex substrate . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

2.19 Slot-pair SIW antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

2.20 AoC at 77GHz . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

2.21 A 79GHz LTCC radar front-end . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

2.22 Wide bandwidth LTCC radar front-end . . . . . . . . . . . . . . . . . . . . . . . 21

2.23 AiP by QFN packaging . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

2.24 AiP by eWLB packaging . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

2.25 Examples of antenna in automotive radar . . . . . . . . . . . . . . . . . . . . . . 23

3.1 Cross section of microstrip line structure . . . . . . . . . . . . . . . . . . . . . . . 24

3.2 Electric and magnetic field lines at low frequencies with static approximation . . 25

vii

LIST OF FIGURES

3.3 Microstrip patch antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

3.4 Magnetic wall of microstrip patch antenna . . . . . . . . . . . . . . . . . . . . . . 27

3.5 Electric field and magnetic surface current distributions along the periphery for

various modes of a rectangular microstrip antenna . . . . . . . . . . . . . . . . . 29

3.6 Differential-fed antenna in centimeter-wave applications . . . . . . . . . . . . . . 31

3.7 Differential-fed antenna in mmW applications . . . . . . . . . . . . . . . . . . . . 31

3.8 SMPA and DMPA configuration . . . . . . . . . . . . . . . . . . . . . . . . . . . 32

3.9 Normalized input resistance of DMPA – middle feed . . . . . . . . . . . . . . . . 36

3.10 Resistance ratio between DMPA and SMPA – middle feed . . . . . . . . . . . . . 37

3.11 Antenna impedance of DMPA with middle feed . . . . . . . . . . . . . . . . . . . 37

3.12 DMPA with middle feed and edge feed . . . . . . . . . . . . . . . . . . . . . . . . 38

3.13 Real(Z) of DMPA with edge feed . . . . . . . . . . . . . . . . . . . . . . . . . . . 40

3.14 Real(ZDMPA) of DMPA middle and edge feed . . . . . . . . . . . . . . . . . . . . 40

3.15 Imag(ZDMPA) of DMPA middle and edge feed . . . . . . . . . . . . . . . . . . . 41

3.16 DMPA with middle feed MSL and edge feed MSL . . . . . . . . . . . . . . . . . . 41

3.17 Coupled MSL structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42

3.18 Simulated differential mode characteristic impedance of coupled MSL at 79 GHz 43

3.19 Single patch DMPA simulated return loss (RL) in Smith Chart . . . . . . . . . . 43

3.20 Simulated differential mode E-field distribution at 79 GHz . . . . . . . . . . . . . 44

3.21 Parameter study of single patch DMPA – Lp . . . . . . . . . . . . . . . . . . . . 44

3.22 Parameter study of single patch DMPA – y1 . . . . . . . . . . . . . . . . . . . . . 45

3.23 Relative bandwidth of DMPA and SMPA with different Wp/Lp ratio . . . . . . . 45

3.24 Simulated reflection coefficient for the optimized DMPA . . . . . . . . . . . . . . 45

3.25 Single patch DMPA prototype for S-parameter measurement . . . . . . . . . . . 47

3.26 Simulated and measured reflection coefficient of DMPA . . . . . . . . . . . . . . 47

3.27 Single patch DMPA prototype for far-field measurement . . . . . . . . . . . . . . 48

3.28 Normalized radiation pattern of a single DMPA for E-plane (y-z) . . . . . . . . . 48

3.29 Normalized radiation pattern of a single DMPA for H-plane (x-z) . . . . . . . . . 49

3.30 DMPA H-plane / E-plane extension indication . . . . . . . . . . . . . . . . . . . 49

3.31 Four-element H-DMPA array structure . . . . . . . . . . . . . . . . . . . . . . . . 50

3.32 Four-element H-DMPA array matching network design . . . . . . . . . . . . . . . 51

3.33 Four-element H-DMPA array final dimension . . . . . . . . . . . . . . . . . . . . 51

3.34 Four-element H-DMPA array E-field distribution at 79 GHz . . . . . . . . . . . . 51

3.35 Four-element H-DMPA array prototype for S-parameter measurement . . . . . . 52

3.36 Four-element H-DMPA Array S-parameter measurement and simulation results . 52

3.37 Four-element H-DMPA array prototype for far-field measurement . . . . . . . . . 53

3.38 Photo of far-field measurement setup . . . . . . . . . . . . . . . . . . . . . . . . . 53

3.39 Normalized radiation pattern of four-element H-DMPA array for E-plane . . . . 54

3.40 Normalized radiation pattern of four-element H-DMPA array for H-plane . . . . 54

3.41 Normalized radiation pattern of four-element H-DMPA array for frequency squint

in H-plane . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 55

3.42 Three-element E-DMPA array Structure . . . . . . . . . . . . . . . . . . . . . . . 55

3.43 Three-element E-DMPA array matching network design . . . . . . . . . . . . . . 56

3.44 Three-element E-DMPA array final dimension . . . . . . . . . . . . . . . . . . . . 56

3.45 Three-element E-DMPA array E-field distribution at 77 GHz . . . . . . . . . . . 57

3.46 Three-element E-DMPA array S-parameter measurement and simulation results . 57

viii

LIST OF FIGURES

3.47 Three-element E-DMPA array prototype RO3003 . . . . . . . . . . . . . . . . . . 58

3.48 Three-element E-DMPA arrray prototype for far-field measurement . . . . . . . . 58

3.49 Normalized radiation pattern of a three-element E-DMPA array for E-plane . . . 59

3.50 Normalized radiation pattern of a three-element E-DMPA array for H-plane . . . 59

3.51 Normalized radiation pattern of the three-element E-DMPA array for frequency

range 76 GHz to 79 GHz . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 60

3.52 Cross section of wideband transition type 1 for DMPA measurement . . . . . . . 61

3.53 PCB part of the transition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61

3.54 Electric field of tapered transition . . . . . . . . . . . . . . . . . . . . . . . . . . . 62

3.55 Simulation results of taper transition . . . . . . . . . . . . . . . . . . . . . . . . . 62

3.56 Cross section of wideband transition type 2 for DMPA measurement . . . . . . . 63

3.57 Photo of tapered transition prototype (shim type) . . . . . . . . . . . . . . . . . 64

3.58 Simulation and measurement results of the return loss of transition . . . . . . . . 64

3.59 DMPA array application examples . . . . . . . . . . . . . . . . . . . . . . . . . . 65

4.1 Classic single-ended transition prior art work . . . . . . . . . . . . . . . . . . . . 67

4.2 Differential port transition prior art work . . . . . . . . . . . . . . . . . . . . . . 68

4.3 E-field in rectangular waveguide – TE01 mode . . . . . . . . . . . . . . . . . . . . 69

4.4 3-D view of the transition structure . . . . . . . . . . . . . . . . . . . . . . . . . . 70

4.5 Cross section of the transition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70

4.6 Top view of PCB part for single DMPA transition . . . . . . . . . . . . . . . . . 71

4.7 Cross section (x-y plane) of E-field of microstrip patch antenna in open air and

with shielded wall . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72

4.8 Simulated return loss of transition for different d . . . . . . . . . . . . . . . . . . 72

4.9 Relative bandwidth of 10 dB return loss for the patch in waveguide . . . . . . . . 73

4.10 Simulation results of the transition with single patch DMPA . . . . . . . . . . . . 74

4.11 Top view of the transition with gap-coupled DMPA . . . . . . . . . . . . . . . . . 75

4.12 Simulation results of the transition with gap-coupled DMPA . . . . . . . . . . . . 75

4.13 Tolerance of d1 and d2 in gap-coupled DMPA transition . . . . . . . . . . . . . . 76

4.14 Photo of top mount part of transition . . . . . . . . . . . . . . . . . . . . . . . . 76

4.15 Photo of test structure of transitions with single DMPA and gap-coupled DMPA

type . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 77

4.16 Measurement results of B2B structure of the transition with single patch DMPA 77

4.17 Measurement results of B2B structure of the transition with gap-coupled DMPA 78

4.18 Measurement results of GC DMPA transition on TLE95 material vs simulation . 78

4.19 Measurement results of transition with spiral load wt/wo absorber material . . . 79

4.20 Block diagram of measurement of LRdR . . . . . . . . . . . . . . . . . . . . . . . 80

4.21 Photo of test board of GC DMPA transition in E-band for LRdR measurement

setup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82

4.22 LRdR measurement results and simulation results of the transition . . . . . . . . 83

4.23 Measurement results of GCP transition on RO3003 and TLE95 material . . . . . 84

4.24 Structure of vertical transition between rectangular waveguide and coupled MSLs

– short-ended parasitic patch . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84

4.25 Top view of the PCB of the transition with common-mode suppression . . . . . . 85

4.26 E-field distribution of transitions with common-mode suppression . . . . . . . . . 86

4.27 Simulated S-parameters of the transition with common-mode suppression . . . . 86

4.28 Photo of W-band transition with common-mode suppression . . . . . . . . . . . . 87

ix

LIST OF FIGURES

4.29 Measurement and simulation results of B2B structure of the transition with dif-

ferent lengths . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87

4.30 Repeatability test of transition B2B structures . . . . . . . . . . . . . . . . . . . 88

4.31 Structure of transition with extended ground DMPA . . . . . . . . . . . . . . . . 89

4.32 Cross-section of transitions comparison . . . . . . . . . . . . . . . . . . . . . . . . 90

4.33 Enlarged details of PCB design in extended ground DMPA transition . . . . . . 90

4.34 Simulated S-parameters of an extended-ground transition . . . . . . . . . . . . . 91

4.35 Measured S-parameters of the B2B structures of the transitions, classic and pro-

posed . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 91

4.36 Application of GCP transition in polarimetric mmW radar system . . . . . . . . 92

5.1 The geometry of the aperture-coupled microstrip patch antenna with LTCC so-

lution . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 93

5.2 LTCC antenna designs at 77/79 GHz radar applications . . . . . . . . . . . . . . 94

5.3 Antenna-in-package solution of superstrate structure . . . . . . . . . . . . . . . . 95

5.4 Parasitic stacked patch antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . 95

5.5 QFN packaging solution for AiP . . . . . . . . . . . . . . . . . . . . . . . . . . . 96

5.6 Radiation beam optimization of eWLB AiP by stack structure . . . . . . . . . . 96

5.7 Comparison of standard WLP and fan-out WLP . . . . . . . . . . . . . . . . . . 97

5.8 Schematic process flow for a fan-out wafer level package . . . . . . . . . . . . . . 98

5.9 Cross section of AiP with MMICs in eWLB package . . . . . . . . . . . . . . . . 98

5.10 Folded dipole and equivalent regular dipole . . . . . . . . . . . . . . . . . . . . . 99

5.11 Simulated S11 of folded dipole AiP in eWLB packaging . . . . . . . . . . . . . . . 100

5.12 Photo of manufactured folded dipole AiP . . . . . . . . . . . . . . . . . . . . . . 100

5.13 Radiation pattern measurement of AiP configuration . . . . . . . . . . . . . . . . 101

5.14 Measurement and simulated radiation pattern of folded dipole AiP . . . . . . . . 102

5.15 Cross section of folded dipole plus cavity in PCB . . . . . . . . . . . . . . . . . . 103

5.16 Top view of folded dipole AiP with cavity simulation model and simulated an-

tenna impedance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103

5.17 Simulation model and simulated S11 of the AiP - FD with and without cavity . . 104

5.18 Photo of the fabricated package of folded dipole with cavity on PCB and test

board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 104

5.19 Measurement and simulated radiation pattern of folded dipole AiP with cavity

at 76.5 GHz . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 105

5.20 EIRP of AiP – folded dipole with cavity in PCB . . . . . . . . . . . . . . . . . . 105

5.21 Cross section comparison of AiP with eWLB package and superstrate structure . 106

5.22 Simulation model of AiP – dual patch . . . . . . . . . . . . . . . . . . . . . . . . 107

5.23 E-field distribution of the AiP – DP for differential signal . . . . . . . . . . . . . 107

5.24 Antenna impedance of AiP – DP . . . . . . . . . . . . . . . . . . . . . . . . . . . 108

5.25 Simulated return loss of the AiP DP . . . . . . . . . . . . . . . . . . . . . . . . . 108

5.26 Bottom view of the AiP DP package . . . . . . . . . . . . . . . . . . . . . . . . . 109

5.27 Power measurement of the AiP DP . . . . . . . . . . . . . . . . . . . . . . . . . . 110

5.28 EIRP of AiP DP . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 110

5.29 Measured and simulated radiation patterns of the AiP DP at 76.5 GHz . . . . . 110

5.30 Radiation pattern of the primary antenna with different mold size . . . . . . . . 111

5.31 Displacement currents of the molds with different dimensions . . . . . . . . . . . 112

5.32 Cross section of hemisphere lens on eWLB AiP . . . . . . . . . . . . . . . . . . . 113

x

LIST OF FIGURES

5.33 Simulated radiation pattern of hemisphere lens on eWLB AiP – FD with cavity . 113

5.34 Photo of eWLB AiP test board with hemisphere lens . . . . . . . . . . . . . . . . 114

5.35 Measured and simulated radiation pattern of hemisphere lens on eWLB AiP –

FD with cavity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 114

5.36 EIRP of hemisphere lens on AiP . . . . . . . . . . . . . . . . . . . . . . . . . . . 114

5.37 Cross section of the eWLB package and the rod antenna . . . . . . . . . . . . . . 115

5.38 Simulated gain of the AiP for different heights of the rod lens . . . . . . . . . . . 116

5.39 Simulation of S11 with the enhanced model . . . . . . . . . . . . . . . . . . . . . 116

5.40 Photo of AiP-DP test PCB with rod lens . . . . . . . . . . . . . . . . . . . . . . 117

5.41 Photographs of the measurement setup in the absorber chamber and the AiP

with lens mounted . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 117

5.42 Measured EIRP of AiP with and without lens . . . . . . . . . . . . . . . . . . . . 118

5.43 Simulated and measured gain of AiP with and without lens . . . . . . . . . . . . 118

5.44 Measured and simulated radiation pattern of AiP rod lens at 78.3 GHz . . . . . . 119

5.45 Measured and simulated radiation pattern of AiP rod lens at 72 GHz . . . . . . . 119

5.46 Measured and simulated radiation pattern of AiP rod lens at 81 GHz . . . . . . . 119

Chapter 1

Introduction

This chapter first introduces the basics of radar and its application in millimeter-wave opera-

tions. Later, the motivation of the thesis and the chosen simulation tools are shown. In the

last section, the whole structure of the thesis is presented.

1.1 Radar

Radar is acronym for radio detection and ranging. It is used to locate distant objects by sending

out radio waves and analyzing the echoes that return. Radar can detect the range, the speed,

the angle, and even the shape of the object.

The first radar was invented by Christian Huelsmeyer in 1904 [1]. In this patent, he intro-

duced for the first time an apparatus for detecting a distance object using electromagnetic wave

(EM wave). Since then, many radar systems have been developed and implemented in many

military and civil applications.

Figure 1.1: Radar basic form.

Figure 1.1 shows the radar basic form. A radar system can be simplified into two parts:

(1) baseband block and (2) radio frequency (RF) front-end block (including antenna). The

baseband block defines the waveform of transmitting radio signals and analyzes the received

1

1. INTRODUCTION

signals. The RF front-end generates, transmits, and receives the radio signal. The antenna

is the interface of EM wave propagation between the circuit and free space. The EM wave

is transmitted by the transmitting antenna (Tx antenna), and the reflected wave is received

by the receiving antenna (Rx antenna). The target information is calculated by comparing

the received signal (Srx(t)) and the transmitted signal (Stx(t)) in time, frequency, and phase

domain, etc.

Different radar systems implement different topologies for target detection. For instance,

the pulse radar detects the range of the target by measuring the time delay of the echo signal.

The continuous-wave radar (CW radar) detects the velocity of the target from the doppler

shift of the received signal. The frequency-modulated continuous-wave (FMCW) radar has the

advantage of both range and speed detection. Therefore, FMCW radars are widely used in

many applications, such as the automotive radar.

1.1.1 FMCW radar

FMCW stands for frequency-modulated continuous-wave. From the Institute of Electrical and

Electronics Engineers (IEEE) Standard Radar Definitions (686-2008), frequency-modulated

continuous-wave radar is a radar transmitting a continuous carrier modulated by a periodic

function such as a sinusoid or sawtooth wave to provide range data.

This section shows the basic principle of FMCW radar [2]. Figure 1.2 shows a brief explana-

tion about FMCW radar. The transmitting signal (Stx(t)) is a frequency-modulated triangular

wave (solid black line). The reflected signal (Srx(t)) (dashed red line) from the target is par-

tially mixed with the transmitted wave. In this example, the target is indicated as approaching

the radar.

The time delay (delta T) between Stx(t) and Srx(t), as well as doppler frequency, maps to

the frequency of the beat signal (LO signal). They have the following relationship:

fBU = fR − fV (1.1a)

fBD = fR + fV (1.1b)

where fR and fV are the range frequency and velocity frequency, respectively.

The positive sign in the formula represents the frequency of the beat signal (downbeat fBD)

obtained where the transmitter frequency falls. The negative sign represents the frequency (up-

beat fBU ) of the beat signal obtained where the transmitter frequency rises. More descriptions

of the FMCW radar are shown in the [2]. If the target is moving away from the radar, the sign

selection in Eq. 1.1 will be opposite.

1.1.2 MIMO radar

From the antenna design point of view, the angle resolution is defined by the aperture size

of the antenna. Since multiple-input and multiple-output (MIMO) configuration may exceed

the antenna aperture over the physical aperture size by multiplexing Tx and Rx channels, it

greatly improves the angular resolution of the radar system, or in other words, it reduces the

antenna size for a given angular resolution in a system. In addition, in most radar systems,

digital beamforming (DBF) technologies have been widely implemented [3, 4, 5, 6]. It enables

fast scanning compared with physical beam scanning. With MIMO and DBF, the radar size

2

Figure 1.2: FMCW basic. In this example, the target is indicated as approaching from the ecoradar.

becomes more compact, and the total cost is also largely reduced. Both enable the radar in

wider applications. Figure 1.3 shows the classic configuration of the MIMO antenna array

in which two Tx antennas and three Rx antennas are included. The antenna aperture can

be derived from the convolution of the two individual aperture distributions [7]. Three Rx

antennas are placed with uniform spacing d2, while two Tx antennas are placed with spacing

d1 in between them. The typical value of d1 is λ/2, where λ is the free space wavelength of

the operating frequency to compromise angular resolution and grating lobes in antenna arrays.

The distance between Tx (d1) is usually a multiple of the distance between Rx (d2). In this

configuration, the virtual Rx antennas are realized between Tx1 and Tx2. The number of Rx

antennas is effectively doubled so that the radar performance in terms of angle detection is

improved.

In MIMO systems, coherent signals are transmitted through Tx antenna 1 and Tx antenna

2. The Tx signal between the channels are orthogonal either in time domain or in frequency

domain, etc. The reflected signals from the target reach different Rx channels with different

time and phase. DBF technologies enable high angular resolution based on those time and phase

information. The spacing between Tx and Rx channels varies widely. For instance, compact

Tx channels’ spacing may be combined with sparse Rx channels’ spacing, etc. More detailed

discussions are referred to [8, 9].

1.1.3 Radar in millimeter-wave applications

Radar, especially millimeter-wave radar (mmW radar), has a wide range of civil applications,

including automotive applications [10] [11]. It has been used in the automotive industry since the

1970s [12], [13]. After more than three decades of development, it has become a key for accident-

free driving and autonomous driving in the future. The frequency regulation of automotive

applications includes 24 GHz and 77/79 GHz for long range. More and more development are

focusing on 77/79 GHz radar systems. Figure 1.4 shows a typical radar module in automotive

applications [14]. It supports driver-assistance functions, such as autonomous cruise control

(ACC), etc.

3

1. INTRODUCTION

Figure 1.3: MIMO radar.

Figure 1.4: Long-range radar from Bosch:LRR3 c© 2011 IEEE [14].

Figure 1.5: Gauge radar from Siemens. c©2003 IEEE [15].

Another example is the tank level gauge with a 24 GHz FMCW radar system [15]. Figure

1.5 shows the basic configuration of the level gauge radar. It measures the liquid level inside

the tank. Higher operating frequency improves the range accuracy of the radar.

1.2 Motivation

This section explains the motivation for the development of differential feed antenna in a

millimeter-wave (mmW) radar system.

First and foremost, the motivation is the system integration within silicon monolithic mi-

crowave integrated circuits (MMICs). Silicon technologies strongly drive mmW radar develop-

ments [16]. It enables high integration in the radar front-end and low cost of radar systems.

A simplified radar front-end MMICs block diagram can be demonstrated as Figure 1.6. It in-

cludes a voltage-controlled oscillator (VCO) for RF signal generation, power amplifier (PA) for

Tx channel, and mixer for Rx channel.

In all of those MMICs blocks, differential topologies are widely implemented. For instance,

reference [17] demonstrates VCO design, reference [18] demonstrates differential PA design,

reference [19] shows mixer circuits, many works are reported for the whole transceiver (TRx)

[20, 21, 22, 23], etc. All those differential topologies MMICs have some common advantages

such as [24]

• wide swing range,

4

Figure 1.6: FMCW radar front-end block diagram.

• rejection of common mode (CM) noise,

• DC offset reduction, and

• zero IF (direct conversion).

Besides those, differential topologies also benefit layout design, for example, neglect the

ground routing as in single-ended topologies. The radio frequency input/outputs (RF IOs) are

preferred to be differential, while the traditional antennas for mmW radar are single-ended. It

is natural to implement differential feed antenna in mmW radar systems. This removes the

balun for compact systems and reduces the transmission loss between MMICs and antennas.

The second reason is that the differential feed antenna has superior radiation performance,

for instance, lower cross-polarization, etc.

Last but not least, the differential feed antenna introduces an additional option for mul-

tichannel integration. Thus, it supports highly integrated multichannel radar. The balun

structure from the classic system is eliminated.

1.3 Simulation tool

The simulation tool used in this work is Computer Simulation Technology Microwave Studio

(CST MWS). Its transient solvers are suitable for analyzing the wide frequency behavior of the

devices with less port numbers (see Figure 1.7).

CST MWS is a general-purpose electromagnetic simulator based on the finite integration

technique (FIT) first proposed by Weiland in 1976/1977 [25]. This numerical method provides

a universal spatial discretization scheme applicable to various electromagnetic problems ranging

from static field calculations to high-frequency applications in time or frequency domain [26].

FIT discretizes the integral form of Maxwell’s equations rather than the differential one.

Transient Solver :

The CST MWS transient solver allows the simulation of a structure’s behavior in a wide

frequency range in just a single computation run. Consequently, this is an efficient solver for

most driven problems, especially for devices with open boundaries or large dimensions.

The transient solver is based on the solution of the discretized set of Maxwell’s grid equations.

5

1. INTRODUCTION

Figure 1.7: Solver Selection (Courtesy of CST AG, Darmstadt, Germany).

To better understand the explanations, let us look at how the transient solver calculates S-

parameters. The transient solver operates with time pulses, which can be easily transformed

into the frequency domain via a fast Fourier transformation (FFT). The S-parameters can then

be derived from the resulting frequency domain spectra:

Figure 1.8: Transient Solver. [26]

For instance, a division of the reflected signal by the input signal in the frequency domain

yields the reflection factor S11. Within just one simulation run in time domain, the full broad-

band information for the frequency band of interest can be extracted without the risk of missing

any sharp resonance peaks. It is very efficient for wide bandwidth design. CST MWS is a pop-

ular simulation tool in mmW antenna design and is used in this project for electromagnetic

(EM) simulations. More details are referred to [26].

1.4 Thesis structure

This dissertation has six chapters in total. Chapter 1 is the introduction (this chapter). Chapter

2 gives a brief view of the state of art of mmW antenna development.

Chapters 3 to 5 explain the different feed antennas in planar form, waveguide integrated

form and antenna in package form, respectively. Chapter 3 discusses the differential feed an-

tenna in microstrip structures. It first analyzes a single patch feed by differential signal. Then

the antenna design extends to E-plane arrays and H-plane arrays. Chapter 4 discusses the dif-

ferential feed antenna integrated with air-fill waveguide structure. A couple of wide bandwidth

6

designs are discussed. Chapter 5 discusses the differential feed antenna in package. A novel

fan-out package technology is implemented in the designs.

Lastly, Chapter 6, presents the conclusion and future development of differential feed an-

tenna.

7

Chapter 2

State of Art for mmW Radar

Antenna

This chapter provides an overview of the state of art for antenna in mmW radar applications.

Automotive radar is the most popular application of mmW radar, and the examples in the

chapter are mainly for these applications [7, 27]. At the end of the chapter, some antennas

from realized systems are shown as examples.

2.1 Waveguide antenna

Waveguide antenna is a traditional antenna that has also been developed in many radar appli-

cations. It offers advantages because of its mechanical stability and high gain property. Horn

antenna is the most popular waveguide antenna, but it is bulky in size. In mmW radar systems,

because of its low-profile property, slot waveguide antennas attract more interest. Prof. Ando

presented a couple of high-gain waveguide antenna designs in [28]. Figure 2.1 presents a couple

of different solutions, including cophase feed, alternating phase feed, radial line slot antenna,

post-wall waveguide, etc. They are all good candidates for high-gain antenna designs. Prof. K.

Sakakibara proposed slot antenna on the narrow wall of a waveguide with an alternative feed

mechanism for the grating lobe suppression in [29]. It is a new way for MIMO antenna array

configuration since each array of slot has narrow width (see Figure 2.2). In addition, unlike

Figure 2.1: Waveguide antenna in mmW applications. Copyright c© 2010 IEICE [28]

8

classic metal waveguide antenna, the manufacturing of antenna was done by metal injection

molding for cost reduction. It provides another possibility for reducing the cost of waveguide

antenna.

Waveguide antennas offer great advantages because of their low-loss transmission line, which

brings higher efficiency for the antenna. But because of cost and manufacturing difficulties,

these antennas have limited applications in the mass production of radar systems.

Figure 2.2: Slot antenna on narrow wall of waveguide. Copyright c© 2000 IEICE [29]

2.2 Lens antenna

Lens antenna is another type of antenna with a long history. Spherical lens is a classic design of

lens antenna. Spherical lens antenna is based on the refraction of electromagnetic waves at the

(a) Schematic of the spherical lens antenna system[30].

(b) Wide scanning array with 33-beam [31].

Figure 2.3: Spherical lens antenna: (a) schematic of single antenna and (b) photo of scanningarray. c© 2002 IEEE [31]

9

2. STATE OF ART FOR MMW RADAR ANTENNA

(a) Photograph of radar system circuit boards. (b) Near-field behavior of the lens used in the systemat 77 GHz.

Figure 2.4: Lens antenna fed by planar array. c© 2014 IEEE [33]

lens’ surfaces (in the case of isotropic homogenous lenses) or within the lens’ dielectric material

in the case of nonuniform refractive index lenses [32]. Schoenlinner presented a dielectric spher-

ical lens antenna for 77 GHz automotive radar applications in [30, 31], first for a single antenna,

and later, he extended it to a wide scanning array. The feeding antenna is a finline tapered-slot

antenna. Figure 2.3 shows a schematic layout and a photo of the manufactured samples. The

scanning capability was realized by multiple feeding antenna for different directions. It requires

many Tx/Rx channels to cover a large field of view (FoV).

Dielectric lens may combine with multiple Tx/Rx antennas [35]. Such configuration gen-

erates multiple beams within detection range. The target angle information can be calculated

by comparing the phase and the magnitude information between the beams. It is a promising

solution for long-range radar applications, which requires approximately 20 degrees field of view

(FoV). In [33], Lutz extended the concept by adding elevation angle scanning. The Tx and Rx

(a) Sketch of the internal view of double-focusedlens.

(b) Photo of a foam-made double-focused lens withhorn primary antenna source at 76 GHz.

Figure 2.5: Artificial lens at 76 GHz. c© 2003 IEEE [34]

10

antennas are placed in different vertical positions (see Figure 2.4(a)).

Gallee presented another type of lens antenna—an artificial lens—which is composed of a

group of parallel-plate waveguides in [34]. In this example, the artificial lenses consist of stacked

parallel-plate waveguides of various lengths. The shape and length of the plate and the distance

between the plates are the degrees of freedom for lens design. Unlike the dielectric lens, the

equivalent refractive index of the artificial lens is smaller than 1. Figure 2.5 shows a foam-made

double-focused lens with horn primary antenna source at 76 GHz.

Some other lens antennas have been presented by Prof. Chen’s group in 2015 (see Figure

2.6). In [36], a plano convex lens antenna was designed for 77 GHz, while in [37], the plano

convex lens was replaced by a planar lens. These works presented different lens antennas as

well as a combination with substrate integrated waveguide (SIW) antenna.

In general, lens antennas are bulkier compared with planar antennas. They also have a

limitation of mounting position in automobiles.

(a) Photo of plano convex lens for 77 GHz [36]. (b) Photo of planar lens for 77 GHz [37].

Figure 2.6: Lens antenna for 77 GHz by Prof. Chen’s group: (a) plano convex lens c© 2015IEEE [36] and (b) planar lens c© 2015 IEEE [37].

2.3 Reflector antenna

A reflector antenna uses either planar shape or other forms of metal to reflect electromagnetic

waves. It has been used since the discovery of electromagnetic wave propagation in 1888 by

Hertz.

The most popular type reflector antenna is the parabolic reflector antenna. The research

group at Karlsruhe Institute of Technology (KIT) has developed a couple of different reflector

antennas for automotive radar sensors. Park presented an offset solution for cylindrical reflector

antenna fed by waveguide Luneburg lens in 2003 [38], while Beer presented a more compact

solution with high-integrated Yagi-Uda antenna as source antenna [39]. Figure 2.7 shows the

principle and photo of both solutions. Parabolic antenna has a sophisticated theory for design

but requires high accuracy for manufacturing in mmW applications. In addition, it requires

high maintenance for assembling in cars.

In 1999, Prof. W. Menzel introduced another type of reflector antenna–printed folded

reflector antenna for 77 GHz automotive radar [40]. This type of antenna uses two printed

11


(a) Side view of the cylindrical reflector antenna fedby waveguide Luneburg lens [38].

(b) Cylindrical parabolic reflector with Yagi-Uda an-tenna [39].

Figure 2.7: Cylindrical parabolic reflector antennas from the research group at KIT. c© 2003,2003 IEEE [38, 39]

substrates to build a polarized grid on top and a twist reflector on the bottom. With adjusting

the twisting and focusing requirement, the overall plane waves are focused and passed to the

top grid plane. The antenna has good radiation performance. It supports very narrow half-

power beam width (HPBW = 2.7 degrees) and low side lobe level (SLL = 24 dB). Folded

reflector antennas have been successfully implemented in series productions. Figure 2.8 shows

the layout of the reflector. The scanning capability can be realized by mechanical scanning

method—tilting the reflector plane. In 2001, MA-COM demonstrated a convex plane design

for folded reflector antenna [41]. It increases the detection angle of the radar but introduces

extra manufacturing challenges.

Figure 2.8: Printed folded reflector antenna by Prof. Menzel. c© 1999 IEEE [40].

Like lens antennas, reflector antennas are good candidates for high gain antenna solution,

but they are bulky and have high demand of manufacturing.

12

2.4 Planar antenna

Planar antennas are the most popularly implemented antenna when realizing mmW radar

systems because of its low profile and low cost. There are many types of planar antennas –

such as wire antenna, grid antenna, patch antenna, etc. – which have been developed for mmW

radar applications in the last two decades. In the following part of this subsection, the author

gives a number of examples for different types of planar antennas.

2.4.1 Open-ended transmission line antenna

Open-ended transmission line can be implemented as a radiating element for antenna designs.

It is easy to realize different polarizations by tilting the line angle. The application of this

antenna in automotive radar was first reported by Toyota in 2000, to the best knowledge of

the author. Iizuka from Toyota reported his work on a 45-degree polarized wire antenna (open-

ended transmission line) in [42] and [43].

Figure 2.9 shows the configuration of the proposed antenna by Iizuka. The antenna is a

series fed by microstrip line. The radiating elements – open-ended half wavelength (λr/2)

microstrip lines – are placed alternatively on either side of the feeding line. The separation

of the wire element is λr/2. The connection is direct coupling. Unlike the classic comb wire

antenna, the proposed wire antenna has 45-degree polarization for reducing the interference

between the incoming cars.

(a) 45-degree polarized wire antenna configuration. (b) Photo of a 45-degree polarized wire antenna ar-ray.

Figure 2.9: Wire antenna (open-ended transmission line) proposed by Toyota. Copyright c©2002 Toyota CRDL [43]

2.4.2 Grid antenna

Grid antenna is another type of planar antenna. It has a periodic rectangular loop structure.

Each rectangular loop integrates the connecting line (long side of the loop) and radiating line

(short side of the loop); thus, the total antenna is formed in a very compact way. Grid antennas

date back to 1964 and 1981. Recently, grid antennas were extensively investigated by Prof.

Zhang [44] in mmW applications. The research group from Ulm University presented a couple

13


of designs in grid antenna at 79 GHz for automotive radar applications. Frei presented a grid

antenna based on soft substrate—RO3003 in [45]. The antenna has a novel feeding structure

with differential input signals. Figure 2.10 shows the antenna configuration and photo of the

manufactured sample.

(a) Grid antenna array configuration. (b) Photo of grid antenna on RO3003 with waveg-uide feeding.

Figure 2.10: Differential fed grid antenna array on RO3003 (h = 256 µm). c© 2011 IEEE [45].

Bauer demonstrated another grid antenna design based on a multilayer structure – low-

temperature co-fired ceramic (LTCC) substrate. A system solution of radar front-end is also

shown in [46]. The antenna was designed as a microstrip structure by back fed of coaxial

structure. Two antennas make a subarray pair to support a stable radiation pattern in broadside

direction over a wide frequency bandwidth. The feeding network implements a laminated

waveguide (LWG) structure, which gives low transmission loss compared with a microstrip line.

Figure 2.11 shows the photo of the RF front-end as well as the antenna configuration.

(a) Photo of RF front-end built on LTCC material. (b) Top view (top) and cross section (bottom) ofgrid antenna array on LTCC.

Figure 2.11: 79 GHz radar front-end with grid antenna on LTCC: (a) photo of front-end and(b) antenna structure. c© 2013 IEEE [46].

14

2.4.3 Patch antenna

Patch antennas are the first structure to be introduced for the microstrip antenna and are still

the most popular structure in radar applications. A lot of research has been done on patch

antenna for automotive radar applications in the last two decades.

Series-fed patch array is the most common configuration of microstrip antenna array. Schoebel

summarized a couple of designs for series-fed patch array in [47]. Those antennas include uni-

form array, amplitude-tapered array, amplitude-tapered inclined array, phase and amplitude

optimized and high-gain array, etc. (see Figure 2.12). In some other designs, the Wilkinson

dividers in high-gain antenna were replaced by T-junction dividers [12, 48].

Figure 2.12: Series-fed patch antenna array examples: (a) patch column with uniform seriesfeed, (b) amplitude-tapered patch column, (c) amplitude-tapered inclined patch column, (d)phase and amplitude optimized column and (e) high-gain antenna array – 8-column/12-patcharray using Wilkinson dividers with mounted resistors. c© 2012 Schoebel J, Ituero Herrero P.Published in [47] under CC BY 3.0 license.

Since MIMO configuration is becoming more and more popular in radar applications, the

research group at Toyota developed a 16-patch series-fed array and applied it in a phase-shift

receive system [49, 50]. It implements tapered patch width for side lobe level optimization. The

separation between the arrays are 0.6λ0, where λ0 is the wavelength of operating frequency in

free space. The photo of the manufactured antenna and layer stack is shown in Figure 2.13.

To realize elevation scanning capability, a novel concept for adjusting antenna beam has

been proposed by Topak [51]. In this work, the classic series-fed patch array was fed from

both ends of the array (see Figure 2.14). The radiation beam can be controlled by tuning

the amplitude and phase of the two feeding signals. It introduces a new solution for elevation

information detection for automotive radar applications.

Many other types of patch antennas have been tried for automotive radar application re-

cently. Shin developed an inclined antenna array by combining gap-coupled patch and direct-

coupled patch [52] for wide bandwidth. Dewantari proposed a novel design for side lobe sup-

pressing by a complementary split ring resonator (CSRR) structure [53]. Hamberger from the

Technical University of Munich (TUM) presented new designs for different polarizations of patch

antenna in the 77 GHz band [54, 55]. Reference [54] shows a dual-polarized patch antenna array

which is a potential candidate for polarimetric radar. The radiating element in this design is a

15


(a) Phase array Rx antenna - 16 series-fed arrays. (b) Layer stack-up of the Rx antenna board.

Figure 2.13: Series-fed patch array in phase-shift receiver system. c© 2013 IEEE [49].

(a) Block diagram of the dual-fed phased array pro-totype.

(b) Photograph of RF board employing a linear ar-ray antenna and MMIC phase shifters on the mea-surement platform.

Figure 2.14: Dual-fed phased array block diagram (a) and photo of RF board (b). c© 2013IEEE [51].

square patch which is fed by two MSLs from the adjacent edges of the patch (see Figure 2.15).

Each feeding line is coming from a separate channel of MMICs. In [55], Hamberger extended

the design to circular polarization antenna. The two feeding lines were replaced by a single

feeding line connected with a power divider at a 90-degree phase offset (see Figure 2.16). The

challenge to these proposals is using very narrow MSL (width<0.1 mm) within the design. For

this, a laser etching system was implemented in manufacturing. The mass production methods

are still under investigation.

Xu from Southeast University (SEU) introduced a new Tx antenna design for combining

long-range and middle-range applications in one single antenna [56, 57]. The antenna beam

patterns were optimized as a shoulder shape – high gain in the broadside direction and middle

gain for the off-broadside direction. The power distribution parts were based on substrate inte-

16

(a) A single element of the array column includingthe feed network.

(b) Antenna array model designed on a RO3003 sub-strate.

Figure 2.15: Dual linearly polarized microstrip patch antenna array: (a) single element modeland (b) antenna array model. c© 2016 IEEE [54]

(a) Simulation model in CST microwave studio. Thesplitter has an additional quarter wave section (90-degree phase shifting) in up branch of the feedinglines.

(b) Photo of the array with measurement setup.

Figure 2.16: Circularly polarized antenna array: (a) simulation model and (b) photo of thearray. c© 2017 IEEE [55]

grated waveguide (SIW) structures, which have less radiation loss than microstrip structures.

The proposed antennas were verified by different RF substrate materials such as Taconic TLY-5

[57] and Rogers RO3003 [56], respectively. It shows a novel configuration of the antenna beam

pattern while the system implementation is under development.

There are many patch antenna/series-fed arrays which have been implemented for automo-

tive radar mass production (see Figure 2.17). For instance, in Bosch long-range radar (LRR3),

four rectangular patches are implemented as source for the lens antenna. There are two side

patches for bandwidth enhancement. In middle-range radar (MRR), two Tx and four Rx an-

tennas are all series-fed patch array. One of the Tx antenna uses a short pitch between the

patch (< λr/2) for shifting the maximum radiation beam from the broadside.

17


(a) Bosch long-range radar (LRR3). (b) Bosch middle-range radar (MRR).

Figure 2.17: Patch antenna in Bosch automotive radars.

2.4.4 Substrate integrated waveguide (SIW) antenna

Another type of planar antenna is substrate integrated waveguide (SIW) antenna. The SIW

structure was first promoted by Deslandes in 2001 [58]. It integrates waveguide structure on a

soft substrate material. The broad walls of waveguide are formed by top and bottom metals,

while the narrow walls are formed by metallized vias array or grooves. Because of its low-

profile and low-cost (compared with air-filled waveguide) property, many SIW antennas have

been studied for automotive radar applications.

The classic SIW antenna uses slot as a radiating element. The slots are in parallel with

the longitudinal axis of the waveguide, in other words, the propagation direction of the waves.

The radiating slots are placed on the broad wall of the SIW—top side. Cheng proposed such

a design of SIW antenna based on a flexible substrate – Kapton HN polyimide foil [59]. It

shows a potential solution for mounting radar module on a convex surface (see Figure 2.18).

(a) Top view (up) and side view (down) of the single-column SIW antenna.

(b) Photograph of the folded SIW-based 4 by 4 slotarray antenna.

Figure 2.18: SIW antenna on flex substrate: (a) top view and side view of the antenna config-uration and (b) photo of the manufactured folded SIW antenna. c© 2009 IEEE [59].

18

(a) Top view (up) and cross section (down) of slot-pair SIW antenna.

(b) The photo of the fabricated 4-column high-gainantenna array.

Figure 2.19: Slot-pair SIW antenna at 77 GHz: (a) sketch of single antenna array and (b) photoof the four-column array. c© 2014 IEEE [60].

Prof. Wang presented another design of SIW antenna based on a RO5880 substrate [60]. The

radiating elements are slot-pair, which are perpendicular to the longitudinal axis. The whole

antenna is composed of four columns of arrays, and each array has 22 slot-pair elements (see

Figure 2.19). Massen designed a 3×15 subarray of SIW antenna at 79 GHz and implemented it

into a MIMO radar design [61]. In the design, the slot widths are optimized for side lobe level.

The SIW antenna is a good candidate for mmW radar antenna and has been implemented

in realized radar systems.

2.5 High-integration antenna

There is another trend of antenna development for high-integration antenna. A lot of research

work has been done in mmW radar applications, such as antenna on chip (AoC), antenna in

package (AiP), etc. This section gives a short introduction for such type of antennas.

2.5.1 Antenna on chip

To the best of the author’s knowledge, the first publication of antenna on chip (AoC) at 77

GHz applications has been reported by Babakhani in 2006 [63]. It has been addressed that the

AoC has very low efficiency compared with classic antennas on PCB since the substrate layer

is very thin. In [63], Babakhani proposed a solution to radiate from the bottom of the silicon

and combine it with the lens.

Hasch reported a novel solution of AoC for 77 GHz [62]. In his works, a parasitic resonator

element is added on top of the patch antenna on silicon to increase antenna efficiency (see Figure

2.20). The parasitic resonator element is formed by a quartz glass whose length is equal to half

the wavelength size. Hasch further investigated the system performance of AoC in [64]. In [64],

AoC was integrated in LRR module by replacing the original source antenna—patch antenna.

The system performances were measured and compared with LRR module. In general, AoC

has high integration level but low radiation efficiency and high cost.

19


(a) Photo of AoC with parasitic resonator elementon complete transceiver.

(b) Photo of AoC in LRR3 module housing.

Figure 2.20: AoC proposal by Bosch: (a) photo of AoC and (b) AoC in LRR3 module. c© 2010IEEE [62].

2.5.2 Antenna in package

Antenna in package (AiP) is another solution for high-integration system. It balances the cost

and efficiency between antenna on chip and antenna on PCBs. It is getting more and more

attention in the mmW radar development.

There are also some recent works on multilayer structure antenna.

Vasanelli presented a multilayer structure, aperture-coupled antenna design based on Rogers

material [65]. Mosalanejad showed another proposal for multilayer structure [66].

Besides the soft substrate, because of its low-loss and high dielectric constant property,

low-temperature co-fired ceramic (LTCC) is getting more and more attention in automotive

radar research studies. X. Wang presented a sophisticated solution for the antenna design in

(a) Photo of single antenna array. (b) Photo of LTCC RF front-end.

Figure 2.21: A 79 GHz LTCC radar front-end with 45-degree polarization antenna: (a) photoof single array and (b) photo of front-end. c© 2015 IEEE [67].

20

(a) LTCC antenna 3-D view and layer stack. (b) Photo of device: back view and front view.

Figure 2.22: Wide bandwidth LTCC radar front-end: (a) antenna structure and (b) photo offront-end. c© 2017 IEEE [68]

LTCC material [67] (see Figure 2.21). Sickinger presented a very compact solution for the RF

front-end of a 79 GHz radar system [68]. The total front-end size is only 2 cm by 3 cm. The

antenna has a very wide bandwidth (>5 GHz) (see Figure 2.22).

The multilayer material facilitates for complex feeding network in antenna designs, but it is

not a full packaged solution for high-integrated antenna since the on-chip-off-chip connections

are from either the bonding wire or the additional packaging.

The improvement in packaging technology brings new chance for the AiP solution. One of

the earliest solutions is AiP based on quad-flat no-lead (QFN) package. The first QFN-based

AiP was proposed by Gaucher in 2004 [69]. Prof. Zwick gave an insightful summary for QFN

AiP development [70]. Figure 2.23(a) shows a classic QFN package outlook, and Figure 2.23(b)

shows an off-chip AiP configuration by QFN package.

Since another package technology—embedded wafer-level ball grid array (eWLB) —is avail-

able for mmW applications [71, 72], a variety of AiP technologies in mmW applications have

been developed [73]. Figure 2.24 shows some examples of AiP developed by eWLB for 77 GHz

radar applications. Compared with a QFN package, eWLB eliminates the bonding wire in the

packaging connections.

2.6 Examples of antenna in realized systems

This section gives a couple of examples of mmW antenna in industry products and gives a

short comparison of them. Figure 2.25 shows antennas in four different automotive mmW radar

products: middle-range radar from Bosch, long-range radar from Continental and short-range

radars from Autoliv and Delphi, respectively.

The antenna in MRR of Bosch is a series-fed patch array structure. There are two Tx

antennas. One is a high-gain antenna with maximum beam at broadside direction. The other Tx

antenna is designed for tilted beam from the broadside direction. It is for elevation information

evaluation. The distances between Rx antennas have nonuniform space. It is for better angle

resolution.

The antenna in Continental LRR is a series-fed wire (open-ended transmission line) antenna.

21


(a) The initial concept drawing for an antenna ontop of chip within a QFN package [69].

(b) The concept of an off-chip antenna and wire-bond interconnect within a QFN package [74].

Figure 2.23: QFN AiP proposal of antenna on top of chip (a) and an off-chip interconnectiondrawing (b). c© 2017 IEEE [70].

(a) AiP array at 60 GHz [75]. (b) AiP MIMO array at 77 GHz [76].

Figure 2.24: eWLB-based AiP solution of antenna array at 60 GHz (a) and MIMO array at 77GHz (b). c© 2018 IEEE [73].

There are some dummy antennas placed beside the Tx and Rx antennas. The purpose of the

dummy antenna is to reduce the surface wave propagation inside the substrate.

In Autoliv SRR radar, there are three Tx antennas and four Rx antennas. The virtual

antenna array concept is implemented for large equivalent aperture. All of those antennas

are same series-fed patch array. It largely reduces the design procedure of the antenna. The

separation between Tx channels has two different distances. The larger spacing is for better

antenna resolution. The smaller spacing supports phase correction among different Tx channels.

Delphi SRR has a complex structure of RF front-end. It separates MMICs and antennas

on different sides of PCBs. The antenna structure is SIW antenna. This configuration reduces

the parasitic radiation from the feeding structure, etc. Meanwhile, it also increases the system

complexity and cost.

22

(a) Middle-range radar from Bosch [77]. (b) Long-range radar from Continental [78].

(c) Short-range radar from Autoliv [79]. (d) Short-range radar from Delphi [80].

Figure 2.25: Examples of antenna in automotive radar from: (a) Bosch, (b) Continental, (c)Autoliv and (d) Delphi.

All of these antennas are selected planar structures because of their low profile and low

cost in manufacturing. Two out of four antennas are patch antennas. The other two are wire

antenna and SIW antenna, respectively. The RF substrate in most of the automotive radars is

soft substrate like Rogers or Taconic laminate.

23

Chapter 3

Differential Microstrip Patch

Antenna

3.1 Microstrip antenna

3.1.1 Microstrip structure

Microstrip line structure was first proposed by Grieg in 1952 [81]. Since then, it has become

perhaps the most popularly used transmission line for radio frequency (RF) and microwave

integrated circuits (ICs). This popularity and widespread use are because of its planar nature,

ease of fabrication using photolithographic processes, easy integration with solid-state devices,

easy combination with heat sink, good mechanical support and vast design information [82],

[83].

The geometry of a microstrip structure is shown below in Fig 3.1. A patterned conductor is

printed on thin, fully grounded dielectric substrate of thickness h and relative permittivity εr.

The wave traveling on microstrip is of the quasi-TEM mode. Figure 3.2 [84] shows a sketch of

the field diagrams for the static approximation. The parallel-plate capacitor field dominates Ez,

while at the conductor edges, the fringing fields dominate Ez and Ex. In the higher operating

frequency, both E-field and H-field have small longitudinal components Ey and Hy.

Figure 3.1: Cross section of microstrip line structure.

3.1.2 Microstrip patch antenna

Microstrip structures were popular in circuit designs and, later on, were also used in antenna

design. Microstrip patch antennas are planar antennas which are constructed under microstrip

24

Figure 3.2: Electric and magnetic field lines at low frequencies with static approximation.

structures.

The first microstrip antenna concept was proposed by Deschamps in 1953 [85]. Microstrip

antennas inherit the merit of microstrip structure and easy integration with planar structure

circuits. However, it took 20 years before the first practical antennas were developed by Howell

[86] and Munson [87]. A microstrip antenna in its simplest configuration consists of a radiating

patch on one side of the dielectric substrate and a ground plane on the other side [88]. Ideally,

the dielectric constant, εr, of the substrate should be low (2 < εr < 4) to enhance the fringe

fields that account for the radiation.

Figure 3.3 illustrates a basic configuration of microstrip patch antenna. It consists of a

very thin metallic strip (patch) placed a small fraction of a wavelength above a ground plane

[89]. The radiating patch is a rectangular patch, fed by microstrip line. The thickness of the

substrate is usually much less than the wavelength in the dielectric (h λr).

Figure 3.3: Microstrip patch antenna.

Microstrip antennas are referred to as patch antennas. The radiating patch may be square,

rectangular, thin strip (dipole), circular, elliptical, triangular, or any other configuration. The

radiating elements and the feed lines are usually photoetched on the dielectric substrate. The

radiation mechanism of the patch can be considered as magnetic current (M) along the pe-

riphery of the patch. The ground plane acts as mirror and will double the equivalent magnetic

current of the patch.

Feeding methods of microstrip patch antenna can be categorized as line feed, coaxial feed,

25

3. DIFFERENTIAL MICROSTRIP PATCH ANTENNA

aperture-coupled feed and proximity-coupled feed [89].

Microstrip patch antennas have several advantages compared with other microwave antennas

[88]. Some of the principal advantages of microstrip antennas are as follows:

• Lightweight, low-volume and thin profile;

• Low fabrication cost, suitable for mass production;

• Linear and circular polarizations are possible;

• Dual-frequency and dual-polarization antennas can be easily made;

• Fully ground structure;

• Easily integrated with microwave integrated circuits;

• Feed lines and matching networks can be fabricated simultaneously with the antenna

structure.

Meanwhile, some disadvantages of microstrip antennas are as follows:

• Narrow bandwidth and associated tolerance problems;

• Somewhat-lower gain;

• Large ohmic loss in the feed structure of arrays;

• Most microstrip antennas radiate into half-space;

• Complex feed structures required for high-performance arrays;

• Polarization purity is difficult to achieve;

• Poor end-fire radiator, except tapered slot antennas;

• Extraneous radiation from feeds and junctions;

• Lower power handling capability (-100 W);

• Excitation of surface waves;

• Microstrip antennas fabricated on a substrate with a high dielectric constant are strongly

preferred for easy integration with MMICs RF front-end circuitry. However, use of high

dielectric constant substrate leads to poor efficiency and narrow bandwidth.

In particular mmW applications, for instance, a 77 GHz adaptive cruise control (ACC)

radar, some of the disadvantages are minimized:

• Radiation into half-space is suitable for automotive radar applications;

• Few percent relative bandwidth is sufficient for the 76–81 GHz radar applications;

• Broadside radiation is desired instead of end-fire radiation in the realized systems;

• Power handling capability is limited with tens milliwatt.

Some of the other disadvantages may be improved by system designs like as follows:

• Use differential feed mechanism to improve polarization purity;

• Improve antenna aperture by implementation of MIMO antenna configurations;

• Improve RF front-end integration level with differential interface MMICs.

Therefore, microstrip patch antenna (MPA) is one of the most popular antennas in millimeter-

wave radar applications.

3.1.3 Cavity model

There are many methods to analyze microstrip patch antenna, such as transmission line model,

multiport network model and cavity model. The cavity model method gives physical insight and

accurate results for microstrip patch antenna. It was first introduced by Lo and Richards in 1979

[90, 91]. This section gives a brief discussion of the cavity model and shows the fundamental

results of the analysis.

There are three assumptions for the cavity model based on observation of the MPA on thin

substrates (h λr). The following derivation is according to [92].

26

The fields in the interior region do not vary with z (that is, ∂/∂z ≡ 0) because the

substrate is very thin (h λr);

The electric field is z directed only, and the magnetic field only has the transverse compo-

nents Hx and Hy in the region bounded by the patch metallization and the ground plane.

This observation provides for the electric walls at the top and bottom;

The electric current in the patch normal to the edge of patch metallization is zero, which

implies that the tangential component of ~H along the patch periphery is negligible, and

a magnetic wall can be placed there. Mathematically, ∂Ez/∂n = 0.

The field distribution in the patch can be divided into two regions: the interior fields and

the exterior fields. The interior fields are useful in determining the input impedance of the

antenna and the currents responsible for radiation. The exterior fields are the fields outside the

cavity region that determine the radiation characteristics of the patch antenna.

With fringing effects, the magnetic wall is placed at a distance ∆ away from the edges of

the patch (see Figure 3.4).

Figure 3.4: Magnetic wall of microstrip patch antenna.

Consider the region of the antenna between the patch metallization and the ground plane.

Because the dielectric substrate is thin, the field distribution in this region can be described by

TM to z modes with ∂/∂z ≡ 0. As a result, there are only three components of the fields Ez,

Hx and Hy. The interior electric field ~Ei must satisfy the inhomogeneous wave equation.

∇×∇ ~Ei − k2 ~Ei ≡ −jωµ0~J (3.1)

or

∂2Ez

∂X2+∂2Ez

∂y2+ k2Ez = jωµ0J (3.2)

where k2 = ω2µ0ε0εr, ~J is the excitation electric current density caused by either due to

the coaxial feed or the microstrip feed, z is a unit vector normal to the plane of the patch, and

∇ is the transverse del operator with respect to the z axis.

27


In addition to satisfying the wave equation, the fields must also satisfy the following bound-

ary conditions:

n× ~Ei = n× ~Ee on the top and bottom conductors (3.3)

and

n× ~Ei = n× ~Ee

n× ~Hi = n× ~He

on the walls. (3.4)

Here, n is the unit outward normal to the walls, ~Ei and ~Hi are the fields in the interior

region, and ~Ee and ~He are the fields in the exterior region.

Under the magnetic wall assumption, Equation 3.4 reduces to

n× ~H = 0 on the magnetic walls (3.5)

It is now easy to determine the interior fields.

The electric field in the patch cavity can be written as

Ez = jωµ0I0

∞∑m=0

∞∑n=0

φmn(x1, y1)φmn(x0, y0)j20(mπd

2a)

k2 − k2mn

(3.6)

~H =1

jωµ0z ×∇Ez (3.7)

where ω is the angular frequency and µ0 is the permeability of vacuum.

φmn(x, y) =√ε0mε0n/aebe cos(mπx/ae) cos(nπy/be) (3.8a)

k2mn = (mπ/ae)

2 + (nπ/be)2 (3.8b)

k2 = k20εr(1− jtanδ) (3.8c)

k0 = ω/c = 2πf/c (3.8d)

j0(x) = sin(x)/x (3.8e)

The magnetic current (M) around the periphery of the patch may be calculated as

M = −2n× Ez (3.9)

Figure 3.5 shows the first several modes of MPA, TM01, TM10, TM20, TM21, etc. When

the feeding point is located at (a/2,0), the fundamental is TM01. When the feeding point is

located at (0, b/2), the fundamental is TM10.

Since the electric current on patch is negligible, the radiation performance can be calculated

from M . The uniform M distribution edges represent radiation edge since they are main

28

contributors to the radiation of the patch. The nonuniform M distribution edges are non-

radiation edges. Here, non-radiation edges mean those edges that have very little radiation on

the principle plane. The higher-order modes introduce feed reactance and cross-polarization

radiation of the patch [89].

(a) TM01 mode (b) TM10 mode

(c) TM20 mode (d) TM21 mode

Figure 3.5: Electric field and magnetic surface current distributions along the periphery forvarious modes of a rectangular microstrip antenna.

The input impedance can be calculated as

Zin =VinI0

(3.10)

where Vin is the RF voltage at the feed point. It is computed from Equation 3.6 as

Vin = −Ez(x0, y0)h (3.11)

= −jωµ0hI0

∞∑m=0

∞∑n=0

φ2mn(x0, y0)j2

0(mπd

2a)

k2mn − k2

e

(3.12)

29


Therefore, the input impedance becomes

Zin = −jωµ0h

∞∑m=0

∞∑n=0

φ2mn(x0, y0)j2

0(mπd

2a)

k2mn − k2

e

(3.13)

k2e = εr(1− jδe)k2

0 (3.14)

Equation 3.13 will yield the input impedance as reactive because all the quantities under

the summation sign are real if the substrate is lossless. The effect of radiation and other losses

on the input impedance has been included in the model in an ingenious manner [91, 92]. The

substrate loss tangent is increased artificially for the power loss from the antenna. The new

loss tangent denoted δe was determined as

δe = tanδ +∆

h+

Pr

ωWT(3.15)

Pr is the power radiated from the patch and can be calculated by integrating the radiation

field over the hemisphere above the patch.

3.2 Differential feed microstrip patch antenna

3.2.1 Prior art work of differential antenna

Traditional differential feeding antennas are Vivaldi antenna, dipole or dipole-like antenna, etc.

For those antennas, there are two separate radiating elements which are fed by each signal line,

respectively.

There is another group of differential antennas where only a single radiating element is

used. For instance, in a patch antenna, the differential signals are fed to both edges of the

patch. The focus of this chapter will be a single radiating patch with differential feeding

signal. The development of differential antenna, or more general, the differential feed patch

antenna, can be traced back to 1980s [93] [94], from the author’s knowledge. In reference [94],

the authors promoted the low cross-polarization radiation properties of the differential feed

patch antenna. Since then, many designs of differential feed patch antennas have been reported

[95, 96, 97, 98, 99, 100, 101, 102, 103, 104]. Those antennas are developed with different

feed structures. For example, [96] proposed a microstrip feeding structure, and the differential

signals are fed to both edges of the patch by microstrip lines. Prof. Zhang proposed a couple of

designs of DMPA with coaxial feed structure [98, 105]. Similar to coaxial feed structure, folded

plate feeding structure has been reported by Chin in [99, 106].

Among the reports, differential feed patch antennas show advantages not only in radiation

performance [95] but also in IC integration [96, 97, 100, 101, 104]. Meanwhile, different feed-

ing mechanisms are developed, such as coaxial feeding [95, 98], microstrip feeding [96, 103],

proximity feeding [100] and aperture-coupled feeding [97, 101].

In centimeter-wave (cmW) applications, for instance, at frequency below 10 GHz, there are

many different feeding mechanisms available. In mmW applications, those feeding structures

are not easy for manufacturing. Therefore, only planar structure, like microstrip line feeding,

or aperture-coupled feeding are suitable for mmW applications.

30

(a) Microstrip line feed design at 2.5 GHz c© 1999IEEE [96].

(b) Coaxial line feed design at 2.5 GHz c© 2007 IEEE[105].

Figure 3.6: Differential-fed antenna in centimeter-wave applications.

Akkermans reported a differential antenna with aperture-coupled structure for 60 GHz ap-

plications [101] (see Figure 3.7(a)). Hamouda presented a microstrip line feeding DMPA [107]

also in the 60 GHz band (see Figure 3.7(b). Ou proposed a DMPA for antenna on chip with

a proximity-coupled feeding structure [108] for 77 GHz applications. Bisognin proposed a dif-

ferential feed patch array by feeding the series-fed patch from different ends of the antenna

[102].

(a) Aperture coupled feed design at 60 GHz c© 2009IEEE [101].

(b) Microstrip line feed design at 60 GHz c© 2013IEEE [107].

Figure 3.7: Differential-fed antenna in mmW applications.

3.2.2 Cavity model analysis for impedance of DMPA

Figure 3.8 shows a DMPA example, as well as SMPA. Usually, the feed position of SMPA is

along the middle line of the patch (x1 = a/2). Naturally, the first DMPA example is also

considered as feed position along the middle line x = a/2. Here, we call it middle-fed DMPA.

31


The y1 and y2 are symmetrical along the y = b/2. Therefore, the two feed positions of DMPA

are (a/2, y1) and (a/2, y2), respectively.

Since the antenna has two feeding points, each feeding point is injected with RF signals

which have a 180-degree phase difference. The natural choice from single-ended microstrip

patch antenna (SMPA) is to put two feeding points symmetrically according to patch. Feeding

distance is defined as the distance from the edge of the patch to the feeding position. The dual

feed points are symmetrical around the line y = b/2.

(a) SMPA configuration. (b) DMPA configuration.

Figure 3.8: Single-ended microstrip patch antenna (SMPA) (a) and differential feed microstrippatch antenna (DMPA) (b).

The rest of this section will present the cavity model analysis of DMPA. Two different feed

mechanisms are discussed, and one practical DMPA in mmW applications is shown in detail.

From Section 3.3, the application of DMPAs are extended to arrays, and both E-plane array and

H-plane array are shown. In Section 3.4, a wide bandwidth transition, which is implemented

in far-field measurement of DMPA, is shown. A short summary with different DMPA/array in

applications is given at the end.

A. Middle-fed DMPA

The cavity model in the previous section is also suitable for DMPA since the assumptions

are still valid. The antenna impedance of DMPA can be derived from multi-feed analysis [91]

[98].

The antenna impedance of SMPA (ZSMPA) has a cosine-squared relationship with different

feeding positions (yi). This can be derived simply from the cavity model.

First, let us recapitulate Equation (5) in [91] for antenna impedance with single-ended feed.

Zs = jωµ0t

∞∑m,n=0

φ2mn(x1, y1)j2

0(mπde2ae

)

k2mn − k2

e

(3.16)

where ω is the angular frequency, µ0 is the permeability of vacuum, and de is the “effective

width” of a uniform strip of z directed source current of 1 A. Since the patch exhibits fringing

effects, the physical dimensions of the patch (a,b) are replaced with the effective dimensions

32

(ae,be).

φmn(x, y) =√ε0mε0n/aebe cos(mπx/ae) cos(nπy/be) (3.17a)

k2mn = (mπ/ae)

2 + (nπ/be)2 (3.17b)

k2e = εr(1− jδe)k2

0 (3.17c)

k0 =ω

c=

2πf

c(3.17d)

j0(x) =sin(x)

x(3.17e)

where c is the velocity of light in free space, f is frequency, δe is the effective loss tangent

of dielectric, εr is the relative permittivity, and ε0m = 1 for m = 0 and 2 for m 6= 0.

The subscript s of Zs denotes single-ended feed. The subscript (m,n) indicates the mode

indices in the x and y axis. Typically, x1 is selected as ae/2, and y1 represents feed distance

measured from the edge of the patch to the feed point.

Zs is a sum of series and can be rewritten as

Zs =

∞∑m,n=0

Zs,mn (3.18)

where

Zs,mn = jωµ0tφ2mn(x1, y1)j2

0(mπde2ae

)

k2mn − k2

e

(3.19)

Zs,mn represents the antenna impedance under TMmn mode and is called mode impedance

here. Comparing (3.17a)-(3.17e), for fixed mode number (m,n), Re[j/(k2mn − k2

e)] reaches

maximum value when k2mn = εrk

20. Therefore, the peak value of Re[Zs] occurs at its resonant

frequency (fmn) and drops quickly away from the resonant frequency. In other words, the

value of Re[Zs] around the fundamental resonant frequency (f01) is dominated by the value of

Re[Zs,01].

The resonant resistance Rs is defined as the value of Re[Zs] at f01, and it can be calculated

as

Rs(y = y1) = Re[Zs]f01 ≈ Re[Zs,01]f01 ∝ cos2(πy1/be) (3.20)

Comparing (3.17a) with (3.20), we obtain the following relationship for Rs(y = y1) over the

feed distance (y1):

Rs(y = y1) = Rs(y = y0) cos2(πy1/be) (3.21)

where Rs(y = y0) represents the antenna impedance with feed point at the edge of patch

[109]. It proves that the impedance of patch antenna with single-ended feed exhibits cosine-

squared behavior over the feed distance. Does the antenna impedance of DMPA have a similar

behavior? Next, we address antenna impedance of DMPA. It may be calculated using the

Z-parameters:

33


Zd =VdI

= 2(Z11 − Z12) = 2(Z22 − Z21) (3.22)

where

Z11 = jωµ0t

∞∑m,n=0

φ2mn(x1, y1)j2

0(mπde2ae

)

k2mn − k2

e

(3.23a)

Z12 = jωµ0t

∞∑m,n=0

φmn(x1, y1)φmn(x2, y2)j20(mπde2ae

)

k2mn − k2

e

(3.23b)

The subscript d of Zd denotes differential feed. Z11 and Z12 are called self-impedance and

mutual impedance, respectively. Comparing (3.23a) and (3.23b) with (3.16), we can find that

Z11 is the same as Zs, while Z12 is different. Now we compare Z12 with Z11.

Similar to Zs, we introduce mode impedance Zd,mn for Zd:

Zd =

∞∑m,n=0

Zd,mn = 2

∞∑m,n=0

(Z11,mn − Z12,mn) (3.24)

where

Z11,mn = jωµ0tφ2mn(x1, y1)j2

0(mπde2ae

)

k2mn − k2

e

(3.25a)

Z12,mn = jωµ0tφmn(x1, y1)φmn(x2, y2)j2

0(mπde2ae

)

k2mn − k2

e

(3.25b)

Obviously, the self-mode impedance Z11,mn is equal to Zs,mn in (3.19). It has a relationship

with the mutual mode impedance Z12,mn as follows:

Z11,mn

Z12,mn=

φmn(x1, y1)

φmn(x2, y2)=

cos(mπx1/ae) cos(nπy1/be)

cos(mπx2/ae) cos(nπy2/be)(3.26)

For the typical feed positions, x1 = x2 = ae/2, and y1 = be − y2, we can obtain

Z11,mn

Z12,mn=

cos(nπy1/be)

cos(nπ(be − y1)/be)=

1, n is even number

−1, n is odd number(3.27)

Insert (3.27) into (3.24):

Zd =

∞∑m,n=0

Zd,mn = 4

∞∑m=i,n=2i+1,0

Z11,mn = 4

∞∑m=i,n=2i+1,0

Zs,mn (3.28)

Up to this point, we have built the relationship between Zd and Zs using the mode impedance

Z11,mn(Zs,mn). Equation (3.28) implies that Zd,mn is zero if mode index n is an even number,

while Zd,mn is four times of Zs,mn if n is an odd number. The fundamental mode of the

antenna is TM01, and the next two higher-order modes are TM20 and TM21. Here, we assume

34

that 1.5be > ae > be. Therefore, Zd,20 and Zd,21 are zero, while Zd,01 is four times of Zs,01.

Similar to Rs, we define the resonant resistance Rd as Re[Zd] at f01. The assumption of

(3.20) is still valid. Therefore, we calculate the Rd as follows:

Rd(y = y1) ≈ Rd,01 = 4Rs,01 ∝ cos2(πy1/be) (3.29)

Equation (3.29) implies some interesting results:

First, the resonant resistance of the antenna with differential feed exhibits cosine-squared be-

havior over the feed distance, which is same as for antenna with single-ended feed:

Rd(y = y1) = Rd(y = y0) cos2(πy1/be) (3.30)

where Rd(y = 0) represents the antenna impedance with differential feed points at the edges

of patch.

Secondly, the impedance match position for the antenna with differential feed can be calcu-

lated from Rs:

Rd(y = y1) ≈ 4Rs(y = y1) = 4Rs(y = 0) cos2(πy1/be) (3.31)

Equation (3.31) shows that the resonant impedance of the differential feed antenna is four

times that of the single-ended feed antenna for the same feed distance (y1).

Usually, the reference impedance of the differential feed antenna is selected as 100 Ohm,

which is double that of the single-ended feed antenna. Thus, the impedance match feed distance

of the differential feed antenna (yd,100Ω) and that of the single-ended feed antenna (ys,50Ω) are

related by comparing (3.21) and (3.31):

cos(πyd,100Ω/be) = cos(πys,50Ω/be)/√

2 (3.32)

If we normalize the feed distance y by y = y/be, Equation (3.32) may be rewritten as

cos(πyd,100Ω) = cos(πys,50Ω)/√

2 (3.33)

Obviously, the normalized feed distance of the DMPA yd,100Ω is larger than that of the

SMPA ys,50Ω. That means the feed point of the former antenna is close to the patch center

compared with that of the latter one.

Full-wave simulation of the DMPA example – DMPA with middle feeding points

Here is an example of DMPA antenna at 79 GHz.

The dimension of the patch can be calculated according to [89]. The width of a single patch

is given by Equation 3.34

W =c

2fr

√2

εr + 1(3.34)

35


with c as the free-space velocity of light, εr( = 3) as the relative permittivity, and fr as the

resonance frequency. The length L of a single patch is calculated to Equation 3.35

L =1

2λr − 2∆l (3.35)

∆l

h= 0.412

(εeff + 0.3)(Wh + 0.264)

(εeff − 0.258)(Wh + 0.8)

(3.36)

with εeff as the effective permittivity and λr as the relative wavelength. They can be

calculated according to [89]. In Equation 3.36, h ( = 0.127 mm) is the height of the substrate.

The patch parameters are calculated as follows: L1 = 0.98 mm and W1 = 1.3 mm.

A model of DMPA, as in Figure 3.8(b), was built in CST MWS. Two ports are set as

50 Ohm lumped port. The differential antenna impedance are calculated from mixed-mode

methods [110].

The feed distance (y1) varies from 0 mm to 0.4 mm, with 0.1 mm step. The patch impedances

are simulated by CST MWS. Figure 3.9 shows the normalized input resistance of DMPA versus

feed distance. Taking the fringing effects (∆) into account, the b in Figure 3.8 is replaced by

be. It shows cos2 behavior of the resistance, which is same as SMPA.

Figure 3.9: Normalized input resistance of DMPA – middle feed.

Another comparison is with SMPA. For a fair comparison, the same size of patch (0.98 mm,

1.3 mm) is fed with single-ended signal. The ratio of the resonant resistances of DMPA and

SMPA is shown in Figure 3.10. It fulfils the prediction in Equation 3.31.

Figure 3.11 shows the antenna impedance of DMPA with middle feed, 3.11(a) real part

and 3.11(b) imaginary part. The dashed lines are the self impedance Z11,s (blue lines) and

the mutual impedance Z21,s (green lines). The solid lines are the differential mode impedance

(black lines). It shows clearly the mode cancellation for differential feed antenna.

The impedance matching feed distance for DMPA and SMPA is 0.2 mm and 0.3 mm,

respectively.

Till now, we have derived three important conclusions between DMPA and SMPA:

1) The resonant resistance of the fundamental mode of DMPA is four times that of SMPA;

36

Figure 3.10: Resistance ratio between DMPA and SMPA – middle feed.

(a) Real(Z)

(b) Imag(Z)

Figure 3.11: Antenna impedance of DMPA with middle feed: (a) real part and (b) imaginarypart.

2) The resonant resistance of DMPA exhibits cosine-squared behavior over the feed distance

similar to SMPA;

3) The impedance matching feeding point (100 Ohm) of DMPA is roughly 1.4 times the

impedance matching feeding point (50 Ohm) of SMPA.

B. Edge-fed DMPA

The middle feed mechanism of DMPA is suitable for coaxial cable feed, where the cable can

be easily connected from the back of the patch [105]. In microstrip line feed structure, it re-

quires long feed line and bending before the patch [96] [107]. In mmW, this bending introduces

37


additional loss and radiation. Therefore, another feeding mechanism, which shifts the feeding

position from the middle of the patch (x = a/2) to the edge of the patch (x = 0), provides a

more compact solution for the RF front-end in mmW systems.

Figure 3.12 shows the comparison of both feeding methods. In both configurations, the dual

feed points are symmetrical around the line y = b/2. In Figure 3.12(a), the feed points are

arranged along the vertical center line of the patch (a/2, yi). In Figure 3.12(b), they are placed

at the edge of the patch (0, yi).

(a) middle feed (b) edge feed

Figure 3.12: DMPA with (a) middle feed and (b) edge feed.

The discussion in the previous section can be extended for the edge feed DMPA.

Here, we recapitulate Equation 3.18 for single-ended feed antenna impedance. The single-

ended feed antenna impedance (Zs) can be calculated by using the mode impedance Zs,mn:

Zs =

∞∑m,n=0

Zs,mn (3.37)

The subscript (m,n) indicates the mode indices in the x and y axis. It was shown in the

previous section that: (i) the differential antenna impedance Zd,mn can also be calculated as

the sum of impedances over all modes; (ii) for symmetrical differential feed mechanisms, only

the modes (m,n = 2i+ 1) exist [111].

Zd =

∞∑m,n=2i+1

Zd,mn (3.38)

Different feed points introduce different mode impedances. For instance, since TM11 mode

requires zero E-field along x=a/2, it cancels out in a middle feed configuration but exists for

the edge-feed antenna. Thus, the differential antenna impedance with middle feed (Zd,center)

can be written as

38

Zd,center =

∞∑m=2i,n=2i+1

Zd,mn (3.39)

and the differential antenna impedance with edge feed (Zd,edge) as

Zd,edge =

∞∑m=0,n=2i+1

Zd,mn (3.40)

Comparing Equations 3.39 and 3.40, we observe that

Zd,edge =

∞∑m=0,n=2i+1

Zd,mn

=

∞∑m=2i+1,n=2i+1

Zd,mn +

∞∑m=2i,n=2i+1

Zd,mn

=

∞∑m=2i+1,n=2i+1

Zd,mn + Zd,center (3.41)

It is shown in former section that Zd,mn is zero if the mode index n is even, while Zd,mn

is four times Z11,mn if n is odd. Z11,mn denotes the mode impedance of the Z-parameter Z11.

It is also identical to the mode impedance of the single-ended feed antenna impedance Zs,mn.

Therefore,

Im[Zd,edge

]=

∞∑m=2i+1,n=2i+1

Im[Zd,edge

]+ Im

[Zd,center

](3.42)

= 4

∞∑m=2i+1,n=2i+1

Im[Zs,mn

]+ Im

[Zd,center

](3.43)

Since Im[Zs,mn

]is always positive (m,n > 1) around frequency f01, f01 is the resonant

frequency for the TM01 mode. Thus, Im[Zd,edge

]is bigger than Im

[Zd,center

]around f01. This

implies that for the same feed distance yi,

(i) the fundamental resonant frequency of the edge-feed differential antenna is higher than that

of the middle-fed, and

(ii) the edge-feed antenna requires a larger value of the electrical separation condition ξ.

Full-wave simulation of the DMPA example – DMPA with edge feeding points

Same as before, full-wave simulation results are provided for DMPA with the edge feeding

method. The same patch (L = 0.98 mm, W = 1.3 mm) shown in the previous section is reused

for comparison.

Figure 3.13 shows the real part of antenna impedance of DMPA as well as single-ended multi-

feed antenna. The simulation results show that a couple of modes from single-ended feed are

cancelled in DMPA. For instance, the mode TM10, TM20, and TM30 are cancelled with differ-

ential feed mechanism.

39


Figure 3.14 shows the real part of the antenna impedance of DMPA with edge feed and middle

feed. It shows that (i) the edge feed MPA has more modes than middle feed, for instance, TM11,

TM31, etc., and that (ii) around the fundamental mode TM01, both DMPAs have a similar real

part of antenna impedance Real(Z).

Figure 3.15 shows the imaginary part of the antenna impedance of DMPA with edge feed and

middle feed. It shows that around the fundamental mode TM01, the Imag(Z) of edge feed is

higher than that of middle feed DMPA. The reason is that the higher-order mode TM11 raises

the Imag(Z).

Table 3.1 shows the simulated results of the patch. For the same patch, the edge feed requires

less yx for the impedance matching. That is mainly because of the influence of the higher-order

mode TM11.

The theory of edge-feed analysis has been published by the author in [112].

Table 3.1: Simulated DMPA with middle feed and edge feed

Port type Feed method W L yx

lumped middle feed 1.30 mm 0.98 mm 0.3 mmlumped edge feed 1.30 mm 0.98 mm 0.2 mm

Figure 3.13: Real(Z) of DMPA with edge feed.

Figure 3.14: Real(ZDMPA) of DMPA middle and edge feed.

3.2.3 mmW-DMPA design at 79 GHz

This subsection shows a practical design of DMPA antenna at 79 GHz, in which antenna patch is

fed by coupled microstrip lines (coupled MSLs) instead of lumped port in the previous section.

40

Figure 3.15: Imag(ZDMPA) of DMPA middle and edge feed.

Two feeding configurations, middle feed and edge feed, are further discussed with coupled

microstrip feeding lines [107, 113] (see Figure 3.16).

Figure 3.16(a) shows the middle feed configuration. The coupled MSLs connect to the

radiating edges – the top and bottom side of the patch. It uses an inset to adjust the feed

positions on the patch. The advantage of the middle feed configuration is less high-order mode

resonance. But it has length and bending feeding lines, which introduce loss and radiation in

system integration. In addition, the inset destructs the radiating edge of the patch.

Figure 3.16(b) shows the edge feed method. The coupled MSLs connect to the patch from

the non-radiating edge – the left side of the patch. The impedance matching can be realized

by adjusting the geometry of the couple microstrip lines (Wm and Sm). It has a compact and

straight feeding line structure in system integration. The radiating edges of patch are kept

completely. Furthermore, edge feeding is more feasible for use in an antenna array. Therefore,

in this work, edge feeding structures are proposed for DMPA design.

(a) middle feed (b) edge feed

Figure 3.16: DMPA with (a) middle feed MSL and (b) edge feed MSL.

The initial DMPA model as in Figure 3.16(b) is built in CST MWS. The patch sizes are

inherited from the previous section, where Lp = 0.98 mm and Wp = 1.3 mm. The feeding

points in the previous sections are replaced by coupled MSLs.

Since the feeding line is DMSL, it is necessary to give an analysis of DMSL. Figure 3.17

shows the structure of coupled MSLs. It has two modes of signals: common mode signals and

differential mode signals. The characteristic impedance of the coupled MSLs also has different

41


values under the different modes [114]. For a fixed substrate (dielectric constant εr and thickness

h), coupled MSLs have two parameters (Sm and Wm) to adjust the characteristic impedance,

while single-ended MSL has only one parameter (Wm). This greatly assists the impedance

matching of the DMPA.

Figure 3.18 shows the simulated differential mode characteristic impedance of the coupled

MSL at 79 GHz. There are multiple pairs of (Sm, Wm) having the same Zd as 100 Ohm,

for instance, (Sm, Wm) = (225 µm, 260 µm), or (325 µm, 280 µm), etc. When coupled

MSLs connect to the patch, different pairs correspond to different feeding positions of the

patch. Following the example in Table 3.1, it is straightforward to select the initial geometry of

coupled MSLs as feeding lines. Here, we consider that the feeding point of coupled MSLs has

the middle of the lines. The initial feeding point is selected as y1 = 0.2 mm, which corresponds

to Wm + Sm = L− 2 ∗ y1 = 0.58 mm.

Till now, we set up a simulation model of DMPA with coupled MSLs edge feed. The model

has been built and simulated in CST MWS. The simulation port is a rectangular waveguide

port, which is more accurate for port de-embedding in the simulation. CST MWS supports

the setup of high-order modes for the waveguide port. In this way, both common mode and

differential mode behaviors are simulated.

Figure 3.19 shows the simulated reflection coefficient of the initial model, both differential

mode and common mode. The differential mode reflection coefficient shows the resonance

around 80 GHz, which corresponds to the resonance mode TM01. For the common mode signal,

the resonant frequency is on TM10 and TM20 mode, which is below 65 GHz and above 120 GHz,

respectively. In the desired frequency range (76-81 GHz), the common mode signal behaves as

open-ended with the chosen length (Wp) of transition line. Therefore, the antenna has good

differential-to-common mode rejection ratio.

Figure 3.17: Coupled MSL structure.

Figure 3.20 shows the E-field distribution of differential mode signal at 79 GHz. It shows

uniform distribution along x-axis and nonuniform distribution along y-axis. It proves the fun-

damental mode of the patch is TM01 mode.

The parameters of the single patch DMPA are studied accordingly. Figure 3.21 shows

that varying Lp corresponds to a resonant frequency change; an increase in Lp will decrease

the resonant frequency fr. Figure 3.22 shows that the feeding distance y1 shifts the antenna

impedance from high Ohmic range to low Ohmic range.

Figure 3.23 shows the relative bandwidth of DMPA for the Wp/Lp ratio varying from 0.7 to

1.6. It shows that the maximum relative bandwidth occurs when the ratio of Wp/Lp is between

1.3 and 1.4. Further comparison between the proposed DMPA and SMPA fed by MSL are also

42

Figure 3.18: Simulated differential mode characteristic impedance of coupled MSL at 79 GHz.

Figure 3.19: Single patch DMPA simulated return loss (RL) in Smith Chart.

shown here. Compared with SMPA, DMPA shows wider relative bandwidth when Wp/Lp < 1.5.

The maximum BW% reaches 4.8% for DMPA, while it is 4.55% for SMPA. The major reason

is that the inset of the patch degrades the bandwidth of SMPA. When Wp/Lp > 1.5, the

TM11 mode starts to degrade DMPA bandwidth. DMPA and SMPA show the same BW when

Wp/Lp = 1.6.

The DMPA has been further optimized for center frequency at 79 GHz and bandwidth. The

S-parameters of the final optimized patch are shown in Figure 3.24. In the desired frequency

range, the single patch DMPA shows 3.3 GHz bandwidth for 10 dB return loss of differential

signal, while the reflection for common mode (CM) signal is very high (below 1 dB return loss).

During the optimization procedure, some design rules are concluded. For instance,

1) Lp controls the differential mode resonant frequency (TM01) of the patch;

2) yi controls the impedance match; and

3) Wp controls the relative bandwidth, as well as the common mode resonant frequency.

43


Figure 3.20: Simulated differential mode E-field distribution at 79 GHz.

Figure 3.21: Parameter study of single patch DMPA – Lp.

The designed single patch DMPA has been manufactured and measured. The RF material

is selected as RO3003 with 127 µm thickness. The first step is the S-parameter measurement.

Figure 3.25 shows the prototype DMPA for S-parameter measurement. Since it is not easy to

have direct measurement of the differential signal response, the DMPA was first measured as

a two-port device. A taper structure was built between coupled MSLs and the ground-signal-

ground (GSG) probe stub (see Figure 3.25). The vector network analyzer(VNA) and frequency

extender were used as the measurement devices. The S-parameters as well as the radiation

patterns of the prototype can then be measured.

The antenna was measured as a two-port device, and the reflection coefficient was calculated

by mixed-mode matrix [110].

44

Figure 3.22: Parameter study of single patch DMPA – y1.

Figure 3.23: Relative bandwidth of DMPA and SMPA with different Wp/Lp ratio.

Figure 3.24: Simulated reflection coefficient for the optimized DMPA with size Lp =0.98 mm,Wp = 1.34 mm,Wm = 0.26 mm,Sm = 0.20 mm.

45


M =1√2

[1 −1

1 1

](3.44)

SMM = M · SSE ·M−1 (3.45)

with SSE as the measured single-ended two-port S-parameters and SMM as the calculated

one-port differential S-parameters.

Figure 3.26 shows the measurements in comparison with the simulation results of the reflec-

tion coefficient of the two antennas. The measured bandwidth of a single patch DMPA is 4.7

GHz. It is wider than the simulation results because of the losses in the substrates. Meanwhile,

the common mode return loss shows high reflection from 60 GHz to 90 GHz.

The next step is to measure the radiation pattern of the single patch DMPA. In a mmW

measurement setup, coaxial cable is expensive and difficult to use to fulfill the measurement

setup. Active devices with waveguide interface, like harmonic mixer, are more popularly im-

plemented. Therefore, it is convenient to build a transition which can convert signal from the

rectangular waveguide to the symmetric MSL.

Figure 3.27 shows the prototype of DMPA for far-field measurement. It is composed of

a transition from a waveguide to coupled MSLs and a single patch DMPA. The transition

converts signal from rectangular waveguide to the differential mode signal on coupled MSLs.

The DMPA prototype was measured as a receive antenna. Section 3.4 gives further details of

the transitions.

The normalized E-plane and H-plane, co-polarization and cross-polarization radiation pat-

terns of the single patch DMPA are plotted in Figures 3.28 and 3.29. The measurement shows

that the half power beamwidth (HPBW) of single patch DMPA is 88 degrees in E-plane and

62 degrees in H-plane, respectively. The simulation results of the cross-polarization in H-plane

are neglected here because of its quite low level.

Additional noise is observed within the far-field measurements. Therefore, average values (20

samples per degree) were used for plotting the results. Because of the height of the transition

cap, the last several degrees of the radiation patterns in the H-plane [85 to 90 degrees] are

disturbed.

Within the far-field measurements, some ripples were observed in the radiation pattern,

stronger in the E-plane than in the H-plane. Radiation from the surface wave at the edges of

PCBs could be the most likely explanation. It is a well-known fact for a microstrip type of

antenna [115]. Therefore, an absorbing material was used on the edges of PCBs. This reduced

the surface wave effects but also distorted the radiation patterns at the angle ranges [+(-)80 to

+(-)90 degrees].

The gain of single patch DMPA was measured using a comparison with a standard horn

antenna. The calibrated gain for the single patch DMPA is 6.2 dBi at 79 GHz.

From the measurement results of the prototype, the concept of single patch DMPA was

approved. The results were first reported by the author in [113].

46

Figure 3.25: Single patch DMPA prototype for S-parameter measurement.

Figure 3.26: Simulated and measured reflection coefficient of DMPA.

3.3 Differential feed antenna array

In many mmW radar applications, single patch DMPA does not support enough gain and beam

pattern for the system. Therefore, it is natural to build DMPA arrays for different system

applications. There are two possibilities to extend single patch DMPA to DMPA arrays. The

first one is to extend in H-plane, and the other is E-plane.

Figure 3.30 shows the E-field distribution of the single patch DMPA. The E-plane is y–z

plane, and H-plane is x–z plane. The field is uniform in y-axis (E-plane) and balance mode

in x-axis (H-plane). Therefore, it is natural to use single-ended MSL as a connection line for

E-plane extension while having coupled MSLs for H-plane array extension. In this section, we

introduce first the design of a four-element H-plane DMPA and then a three-element E-plane

DMPA. For both antenna arrays, the measurements of prototypes are shown and discussed.

47


Figure 3.27: Single patch DMPA prototype for far-field measurement.

Figure 3.28: Normalized radiation pattern of a single DMPA for E-plane (y-z).

3.3.1 H-plane DMPA array

This subsection shows a design example of an H-plane DMPA (H-DMPA) array at 79 GHz. The

H-plane DMPA (H-DMPA) array is shown in Figure 3.31. The antenna array is composed of

four antenna patches which are connected by a pair of coupled lines in between. The differential

signal is fed from the first patch (left patch) and series transferred to the rest of the patches.

The last patch behaves as a matched load for the array; therefore, the H-DMPA array functions

as a travelling wave antenna. It supports wide bandwidth over resonant array.

The antenna array is designed as a symmetric structure for symmetric radiation pattern,

which means the middle two patches have the same dimension and the two side patches have

the same size, while the middle patches can be different from the side patches. The initial patch

size was inherited from the single patch DMPA design in the previous section.

Different from single patch DMPA design, in the DMPA array design, the radiation pattern

48

is an important factor to consider. It depends on the patches’ location and the excitation signal

on the patches.

The pitch between the antenna patches (center to center) is a critical parameter influencing

the radiation pattern. From the application point of view, we need maximum radiation at the

normal direction of PCB. It requires all patches to have an equal phase. Since the antenna

patches are aligned in the H-plane (x-z), the signal input and output of patches have no phase

offset. It is different from the standard single-ended patch array, where the patches give a

180-degree phase offset for the input and output signals. The distance of the antenna patches

is optimized based on the simulation. To keep each antenna patch radiating in phase at the

center frequency, a distance of 3.48 mm (0.92λ0) is selected based on simulation optimization.

It equals the sum of the width of the patch (W1) and the gap between the antenna patches

(Lc). This larger pitch creates a narrower beam, but if the pitch is too large, there are risks for

the grating lobe. In this design, we keep uniform pitch for all patches.

The patch dimension is also optimized for the wider bandwidth. All the patches are selected

to have the same width (Wi=1,2,3,4) for keeping the same connection lines between the antennas.

While the patch lengths are not all the same, L1 = L4 and L2 = L3 for the symmetric structure

Figure 3.29: Normalized radiation pattern of a single DMPA for H-plane (x-z).

Figure 3.30: DMPA H-plane / E-plane extension indication.

49


Figure 3.31: Four-element H-DMPA array structure.

configuration. L1 and L2 are selected differently for wider bandwidth.

Simulation results show that the impedance of H-DMPA array falls on the upper part of the

Smith Chart (see Figure 3.32). Compared with the single patch DMPA, the H-DMPA array

introduces more inductance in the antenna impedance. It is not possible to reach impedance

matching by only adjusting the feeding position as for the single patch DMPA. Therefore, a

matching network is designed here for the impedance matching. The matching network consists

of an inset on the first patch and quarter-wavelength coupled MSLs between the first antenna

patches and feeding lines. Figure 3.32 demonstrates the function of the matching network.

The inset works as a section of transmission line. It converts the antenna impedance from the

inductive part in Smith Chart (curve 1, red) to the high resistance part of real axis (curve

2, green). Then the quarter-wavelength high-impedance coupled MSLs transform the antenna

impedance to 100 Ohm differential impedance (curve 3, blue).

The final optimized four-element H-DMPA array has the following dimensions: d0 = 2.08

mm, L2 = L3 = 1.1 mm, L1 = L4 = 0.96 mm and W1 = W2 = W3 = W4 = 1.4 mm,

Li=0.20 mm, Wq= 0.16 mm, Lq=0.60 mm and Sq=0.20 mm. The coupled MSLs are same as

before Wm/Sm/Wm=0.26 mm / 0.20 mm / 0.26 mm. The total length of the array is 12.44

mm (including the matching network), and the width of the array is 1.1 mm = 0.29λ0. The

impedance bandwidth derived from the simulation results is 4.1 GHz, and the antenna gain is

12.0 dBi.

The E-field distribution of the four-element H-DMPA array at 79 GHz is shown in Figure

3.34. It shows all patches are radiating in-phase at the desired frequency.

Prototypes of the designed H-DMPA array were built for both S-parameter and far-field

measurements. The measurement procedure is same as for single patch DMPA. Figure 3.35

shows the photo of prototype for S-parameter measurements.

Figure 3.36 shows the measurements in comparison with the simulated reflection coefficients.

The measured 10 dB return loss of the differential mode signal has 4.6 GHz bandwidth, which

is wider than the simulation results. This is because of the losses in the substrates.

The radiation patterns of H-DMPA are also measured. The prototype for the radiation

patterns measurement is shown in Figure 3.37. The same waveguide transition structure, which

is used for single patch DMPA measurement, is implemented here.

The average values (20 samples per degree) were used for plotting results to suppress the

noise during the measurement. Within the far-field measurements, some ripples were observed

50

Figure 3.32: Four-element H-DMPA array matching network design.

Figure 3.33: Four-element H-DMPA array final dimension.

Figure 3.34: Four-element H-DMPA array E-field distribution at 79 GHz.

in the radiation pattern, stronger in the E-plane than in the H-plane, stronger for the single

patch antenna than for the antenna array. Radiation from the surface wave at the edges of

51


Figure 3.35: Four-element H-DMPA array prototype for S-parameter measurement.

Figure 3.36: Four-element H-DMPA Array S-parameter measurement and simulation results.

PCBs could be the most likely explanation. Absorbing material was used on the edges of PCBs

in the later measurement setup. This reduced some surface wave effects but distorted radiation

patterns at the angle ranges [+(-)80 degree to +(-)90 degree].

The normalized E-plane and H-plane, co-polarization and cross-polarization radiation pat-

terns of both antennas are plotted in Figures 3.39 and 3.40. The simulation results of the

cross-polarization in H-plane are neglected here because of its quite low level. The measure-

ments show that the HPBW of four-element H-DMPA is 100 degrees in E-plane and 14 degrees

in H-plane. The measurement results match the simulation prediction very well.

The gain of the antenna/array at 79 GHz is also measured. Similar to single patch DMPA,

the measured gain is compared with calibrated horn antenna. The calibrated gain for four-

element H-DMPA is 12.8 dBi. It is very close to the simulation results.

The side lobe level (SLL) of the H-DMPA is about 7 dB. It is due to the large pitch between

the elements.

52

Figure 3.37: Four-element H-DMPA array prototype for far-field measurement. c© 2008 IEEE[113].

Figure 3.38: Photo of far-field measurement setup. c© 2008 IEEE [113]

The radiation pattern of H-DMPA at different frequencies are measured for squint beam

analysis. The results are shown in Figure 3.41. The maximum radiation angle of the array

spreads within 10 degrees from 76 GHz to 81 GHz range. It is a genetic character for the

series-fed array.

In this section, we demonstrated the concept of H-DMPA array with an example of four-

element H-DMPA array at 79 GHz. The advantages of the H-DMPA are wide bandwidth (4.6

GHz for 10 dB return loss), very compact geometry for array configuration (the width of the

array is only 0.29 λ0), low cross-polarization (18 dB below co-polarization). The disadvantages

of H-DMPA include the high SLL and the frequency squint. The work has been first published

by the author in [113].

53


Figure 3.39: Normalized radiation pattern of four-element H-DMPA array for E-plane (y-z) at79 GHz. c© 2008 IEEE [113]

Figure 3.40: Normalized radiation pattern of four-element H-DMPA array for H-plane (x-z) at79 GHz. c© 2008 IEEE [113]

3.3.2 E-plane DMPA array

This subsection presents another possibility of building DMPA array – E-plane DMPA (E-

DMPA) array. In contrast to H-DMPA array, E-DMPA array is configured as center-fed, which

means that the differential signal fed to the center patch and the side patches is placed in +y

and -y directions with symmetrical dimension. Such configuration brings a straight result that

an E-DMPA array has an odd number of patch elements.

The E-DMPA array is comparable with a conventional single-ended series-fed array. It is

a resonance antenna array. The main difference is that E-DMPA is fed from the center of the

array, while conventional single-ended series-fed array is fed from the end of the array. The

center-fed configuration brings benefits, such as the power distribution over the patches for SLL

reduction and stable radiation pattern over the frequencies.

54

Figure 3.41: Normalized radiation pattern of four-element H-DMPA array for frequency squintin H-plane.

The top view of such a three-element E-DMPA array is shown in Figure 3.42. The E-

DMPA array has three radiating elements—one main patch and two side patches. The main

patch is similar to the single patch DMPA and is connected to the coupled MSLs. Two side

patches were added in parallel with the radiating edge of the center patch. The side patches

have smaller patch width to reduce the side-lobe level of radiation patterns. The symmetrical

alignment of the patches ensures that the maximum radiation gain is in the normal direction

of the PCB plane at all frequencies. Since the antenna patches are aligned in the E-plane, the

patch introduces approximately 180-degree phase difference for the connection line. Therefore,

different from the H-DMPA, the E-DMPA array needs a connection line with a phase of 180

degrees. The connection lines are MSLs with characteristic impedance larger than 50 Ohm

(Zc > 50 Ohm).

Figure 3.42: Three-element E-DMPA array Structure.

Compared with the single patch DMPA, the three-element E-DMPA array introduces more

inductance in the antenna impedance. As a result, the antenna impedance is in the upper

part of the Smith Chart (see Figure 3.43). For impedance matching, a matching network is

55


Figure 3.43: Three-element E-DMPA array matching network design.

Figure 3.44: Three-element E-DMPA array final dimension.

designed. The function of the matching network is similar as for H-DMPA array. It is shown

in Figure 3.43. The impedance loci of the array lie in the inductive part of the Smith Chart

(curve 1 in Figure 3.43, blue). Here, an inset is cut in the main patch to act as one section of

the transmission line, which reduces the total size of the matching network. Adding the inset

shifted the impedance loci to the right of real axis in the Smith Chart (curve 2, red). The

quarter-wavelength coupled MSLs, whose characteristic impedance is higher than 100 Ohm,

transform the antenna impedance to 100 Ohm differential impedance (curve 3, black).

After the optimization by the EM simulator, the final dimension of three-element E-DMPA

array is shown in Figure 3.44. The E-field distribution of three-element E-DMPA array at 77

GHz is shown in Figure 3.45. It shows that the patches are radiating in phase. The simulated

S-parameters are shown in Figure 3.46. The bandwidth of the E-DMPA is relatively smaller

than H-DMPA since it is a resonant antenna array.

Prototypes of the designed E-DMPA array are fabricated and measured for both S-parameter

56

Figure 3.45: Three-element E-DMPA array E-field distribution at 77 GHz.

Figure 3.46: Three-element E-DMPA array S-parameter measurement and simulation results.c© 2011 IEEE [116].

and far-field characteristics measurements. For S-parameter measurement, the antenna array

was measured as a single-ended two-port device (see Figure 3.47). Then the differential S-

parameters were calculated using the mixed-mode matrix [110], with S as the measured single-

ended two-port S-parameters and Sdm as the calculated one-port differential S-parameters (see

Equation 3.44).

The simulated and measured S-parameters are plotted in Figure 3.46. The measurement

shows that the 10 dB return loss of differential mode has a bandwidth from 76.2 GHz to 78.2

GHz. The common mode return loss at the same frequency is less than 3 dB.

For the radiation performance measurements, the same transition from differential mi-

crostrip line to waveguide is used (see Figure 3.48). The radiation patterns at 77 GHz for

both E-plane and H-plane are plotted in Figures 3.49 and 3.50.

The 3 dB beam widths of co-polarization were 28 degrees and 68 degrees in the E-plane and

57


Figure 3.47: Three-element E-DMPA array prototype RO3003.

the H-plane, respectively. The side lobe level of E-DMPA is 15 dB, which is a good improvement

from H-DMPA. The cross-polarization was at least 25 dB lower than the co-polarization in the

E-plane and had its minimal point in the normal direction of the patch. In the H-plane, the

cross-polarization was around 20 dB lower than the co-polarization. The simulation results for

cross-polarization in the H-plane are too weak to plot. The measurement results are in good

agreement with the simulation results.

Figure 3.48: Three-element E-DMPA arrray prototype for far-field measurement.

The radiation patterns of the array in the E-plane were also measured at various frequencies

(see Figure 3.51). The results show that the radiation patterns of the antenna array are stable

from 76 GHz to 79 GHz. The maximum gain is always in the normal direction of the PCB

plane within the bandwidth.

The calibrated gain of the array is 11.4 dBi after taking into consideration the loss from the

feeding line and transition. It is close to that of the simulation (11.5 dBi).

58

Figure 3.49: Normalized radiation pattern of a three-element E-DMPA array for E-plane (y-z)at 77 GHz. c© 2011 IEEE [116].

Figure 3.50: Normalized radiation pattern of a three-element E-DMPA array for H-plane (x-z)at 77 GHz. c© 2011 IEEE [116].

The three-element E-DMPA array is shown in this section. The measurement results of

the prototype antenna verified the concept. Compared with the H-DMPA array, the E-DMPA

has better radiation performance, for example, better SLL, no frequency squit of the radiation

beam, etc. But it has less impedance bandwidth since it is a resonant antenna structure. The

work from the author was published in [116].

Till now, three different DMPA/arrays have been demonstrated. Table 3.2 gives a sum-

mary of the performance of all three antennas. From the table, single patch DMPA has wide

beamwidth for E-plane and H-plane. The gain is about 6 dBi. It is suitable for ultrashort

range, especially when the operating frequency is high (above 100 GHz) and the propagation

loss of transmission line is significant.

H-plane array has wide bandwidth and higher gain than single patch. But its SLL is only 7

dB. It is suitable for short-range applications, in which SLL is acceptable. Another advantage,

not shown in the table, is that H-plane array is very suitable for MIMO application because of

59


Figure 3.51: Normalized radiation pattern of the three-element E-DMPA array for frequencyrange 76 GHz to 79 GHz. c© 2011 IEEE [116].

its compact width (0.29 λ0) as shown later on in Chapter 3.5.

E-plane array has the good radiation performance but narrow bandwidth (2 GHz). It is

suitable for long-range applications where range resolution is less critical.

Table 3.2: Summary of DMPA/arrays performance

Antenna type BW Gain HPBW(E/H) Squint SLL Co./Cro.

Single Patch 4.7 GHz 6.2 dBi 88 Deg./62 Deg. No - 15 dB

4-element H-DMPA 4.6 GHz 12.8 dBi 100 Deg./14 Deg. Yes 7 dB 18 dB

3-element E-DMPA 2.0 GHz 11.4 dBi 28 Deg./68 Deg. No 15 dB 20 dB

3.4 Transition for DMPA measurement

During the far-field measurement of DMPA/array, we have implemented a wide bandwidth

transition from waveguide to coupled MSLs in the antenna far-field measurement. This section

gives a detailed discussion for the transition design.

Many transitions have been presented in the last two decades, for example, [117], [118],

[119], [120], etc. In those prior art works, the transitions are to connect rectangular waveguide

to the single-ended MSLs. A balun structure is required when we connect to DMPA. Different

from prior art works, this work will design a transition from the rectangular waveguide directly

to the coupled MSLs interface. It supports seamless connection with DMPA/array for the far-

field measurement. It is helpful for removing the balun impact in the measurement, such as the

bandwidth and radiation of the balun, etc. The transition was designed in E-band, and center

frequency is selected at 75 GHz.

The cross section of the transition (looking from the microstrip port) is shown in Figure

3.52. The transition is composed of a metallized cap and a bottom mount with a dielectric

structure in between. The cap mount is a short-end WR12 waveguide with an open channel for

the microstrip port. The inner height of the cap mount equals λg/4 (λg is the guide wavelength

in waveguide). The bottom mount is a sustainer which has a WR12 waveguide interface.

The dielectric substrate between the mounts plays an important role in the transition. Figure

60

3.53(a) shows the top layer of the substrate. Inside the waveguide area (dashed line in the

figure), there is a pair of triangular patches. One side of the triangular patch connects to the

waveguide conductor, while the other side of the patch connects to the microstrip port formed

by a pair of coupled microstrip lines. The taper structure behaviors as impedance convertor

between different transmission lines. The ground plane of the substrate within the waveguide

area is removed.

A C-shape guarding ring structure is built for preventing the surface wave in the substrate

in +/-y and -x directions. The guarding ring was built by vias in the substrate for the first

prototype and later by a thin metal piece for the second prototype.

The shadowed parts in the Figure 3.53(a) are metallized areas on the substrate. The top

and bottom mounts are made of brass. The dielectric substrate in the structure is RO3003 with

0.127 mm thickness.

Figure 3.52: Cross section of wideband transition type 1 for DMPA measurement. c© 2008IEEE [121].

(a) Top view (b) Bottom view

Figure 3.53: PCB part of the transition : top view(a) and bottom view (b).

The E-field distribution of the transition derived from EM simulation is shown in Figure

61


3.54. It can be observed that the dominant TE10 mode waveguide signal is guided by the

triangular patches to the coupled microstrip lines. Therefore, it supports a large bandwidth

and suppresses the common mode of the coupled microstrip lines. A section of ungrounded

coupled microstrip lines (inside the waveguide area) works as quarter-wavelength transformer

for impedance matching.

Figure 3.54: Electric field of tapered transition. c© 2008 IEEE [121].

Simulation results are shown in Figure 3.55. There is around 23 GHz bandwidth of 15 dB

return loss for the differential mode in both the microstrip port and the waveguide port. The

insertion loss of the waveguide port is less than 0.5 dB within the bandwidth. The return loss

of the common mode in the microstrip port is below 0.5 dB. It gives quite good common mode

suppression.

Figure 3.55: Simulation results of taper transition. c© 2008 IEEE [121].

The prototypes of the transition were manufactured in the university workshop. The vias

were realized by conductive glue fillings in the first prototype. Because of the difficulties with

62

Figure 3.56: Cross section of wideband transition type 2 for DMPA measurement. c© 2008IEEE [121].

via realization on the substrates in the university lab, another structure type-2 transition is

also tested within this work. The cross section of this transition is shown in Figure 3.56. In

this structure, the C-shape slot around the tapered structure is removed from the substrate

and replaced by a piece of metal sheet. This metal sheet forms the guarded ring instead of the

via’s guarded ring in type 1. The remaining part of the substrate, with the tapered structure,

is 0.4 mm wider than the waveguide size to provide better contact with the top mount. Similar

S-parameter results are reached by optimizing the triangular patches.

Fabrication and Measurements:

Both types of the transitions are fabricated and measured. A pair of spiral structures is con-

nected to the microstrip port of the transition. In combination with the absorbing material

on top of the spiral structures, the spiral structures behave as a matched load for the coupled

MSLs port. The return loss of the spiral load is measured below 15 dB for the whole E band.

Figure 3.57 shows a photograph of the transition with the spiral loads. The return loss of

the waveguide port is measured using a vector network analyzer. The measurement and the

simulation results are plotted in Figure 3.58(a). The measurement results show good agreement

with the simulation results.

There is a frequency extension module used within the measurement to generate signals

from 67 GHz to 90 GHz. Noise in the measurement results above 67 GHz is due to the used

equipment.

There is a ripple in the return loss within the bandwidth of the measured results. The

period of the ripple is approximately 4.5 GHz, which corresponds to 0.22 ns propagation delay

in the time domain. The distance between the spiral load and the DMSLs port of the transition

is 20 mm. Therefore, the ripple is due to the imperfect match of the spiral load.

The fabrication tolerances and the misalignment during the assembly influence the perfor-

mance of the transition. These inaccuracies introduce some mismatch, resulting in a difference

between the simulation and the measurement results. But the large bandwidth of the transition

63


supports its wide usage in millimeter-wave applications.

The simulation and measurement results of the type-2 transitions (metal sheet guarded ring)

are shown in Figure 3.58(b). The measurement results agree very well with the simulation. The

compact size and the simple fabrication enable the transition to be employed in a number of

millimeter-wave applications. The work of this transition was published in [121].

In this section, a wide band transition from waveguide to coupled MSLs in E-band is demon-

strated. The transition provides a simple connection between differential active circuits and

rectangular waveguide structures. The transition was fabricated and measured. The measure-

ment results show a bandwidth of 19 GHz, which is 25% relative bandwidth. The low costs for

fabrication and small dimensions enable the transition to be used in many applications.

Figure 3.57: Photo of tapered transition prototype (shim type).

(a) Simulation and measurement results of thereturn loss of transition type 1 – vias type.

(b) Simulation and measurement results of thereturn loss of transition thin metal type 2 –shim type.

Figure 3.58: Simulation and measurement results of the return loss of transition thin metaltype 1 – vias type (a) and type 2 – shim type (b).

64

3.5 Application of DMPA/array

Here are some examples of the applications of DMPA/array [122, 123, 124, 125]. Reference

[122] demonstrated a 122 GHz radar system, in which single patch DMPAs are implemented in

both Tx and Rx antennas. The loss of transmission line is significant in such high frequency.

It is difficult to realize a high-gain antenna. A compact single patch DMPA brings benefits for

the system.

For the classical MPA array, it is difficult to have 0.5 λ0 spacing. The solutions are either to

increase the spacing of the antenna array [126] or to reduce the antenna width, which decreases

the antenna bandwidth [127]. Since H-DMPA has compact width (0.29 λ0), it is very suitable

for MIMO radar configuration. References [123, 124, 125] show a couple of different MIMO

radar systems for 79 GHz and 94 GHz.

(a) Single patch DMPA application in a 122 GHzradar sensor c© 2011 IEEE [122].

(b) H-DMPA application example A – in a 79 GHzradar sensor c© 2009 IEEE [123].

(c) H-DMPA application example B – in a 79 GHzradar sensor c© 2011 IEEE [124].

(d) H-DMPA application example C – in a 94 GHzradar sensor c© 2012 IEEE [125].

Figure 3.59: DMPA array application examples.

65

Chapter 4

Coupled Microstrip Line Feed

Waveguide Transition

Waveguide antennas have many advantages, such as high gain, high efficiency, etc. Many

waveguide antennas, like horn antenna and slot waveguide antenna, have been developed in

mmW radar applications [29, 28, 128]. On the other hand, MMICs are mounted on PCBs with

microstrip line structures. Therefore, in a mmW radar system with waveguide antenna, there

is a need for a transition from planar transmission line to waveguide. This chapter extends the

concept of DMPA to transition design in mmW frequency.

4.1 Introduction of transitions from planar transmission

lines to waveguide

In the last two decades, various types of vertical transitions from microstrip line (MSL) to

waveguide (WG) have been proposed in the literature. The classic transition is from single-

ended transmission line to waveguide, like [129, 130, 131, 132, 117]. Yao proposed a ridge

waveguide type of transition in [129]. With a stepped ridge waveguide structure, the signal

propagates from the microstrip line to the air-filled waveguide. It requires a complex waveguide

structure and is expensive to fabricate. Grabherr reported a multilayer structure of vertical

transition in [117] (see Figure 4.1(a)). It implemented a slot-coupled structure with a multilayer

of substrate inside the waveguide. This type of transition needs multilayer structures, which

make fabrication costly. The so-called Quasi-Yagi antenna transition [130, 131] is of the in-line

type (see Figure 4.1(b)). It supports a wide bandwidth, but it is difficult to fabricate because

the antenna must be clamped inside the waveguide structure.

Thiel proposed a transition structure which integrates an EM coupling element – patch –

and a reflector. It brings very broad bandwidth performance [133] (see Figure 4.1(c)). The

EM coupling element can be replaced by a probe structure [118] or fed offset from the middle

of the waveguide [132]. In all those designs, the common part is a short-ended waveguide

reflector which gives rise to additional fabrication costs. Villegas proposed another type of

wide bandwidth transition design [134] (see Figure 4.1(d)). It implements iris waveguide as a

coupling element between planar transmission lines and waveguide.

All of those transitions are from single-ended port to waveguide port. Recently, since MMICs

with differential RF IOs are becoming more and more popular in mmW applications, there are

66

(a) Transition with multilayer substrate c© 1994IEEE [117]

(b) Transition with Quasi-Yagi antenna c© 1999IEEE [130]

(c) Transition with patch as EM coupling elementc© 1998 IEEE [133]

(d) Transition with iris waveguide as EM couplingelement c© 1999 IEEE [134]

Figure 4.1: Classic single-ended transition prior art work.

new designs of transition from coupled transmission lines to waveguide.

Henawy proposed an in-line transition with antipodal finlines [135] (see Figure 4.2(a)). Giese

further developed compact-size transitions with elliptic shape finline structures and reported in

[136, 137] (see Figure 4.2(b)). The in-line structure transitions have wide bandwidth, but it is

difficult for assembling.

The author proposed a vertical transition from the coupled MSLs to waveguide [121] (see also

in Chapter 3.4). It implements an impedance-tapered structure and a short-ended waveguide

reflector. This type of transition has very wide bandwidth, but needs complex mechanical

mount. Yuasa presented a novel vertical transition design with full microstrip structure for

the transition from slot line to waveguide [138] (see Figure 4.2(c)). It requires a very complex

mechanical part for the impedance matching.

A couple of new transitions have been reported for LTCC structures in 77 GHz applications

[139, 140] (see Figure 4.2(d)). All of those transitions are planar multilayer structure which

convert signals from coupled microstrip lines to dielectric filled waveguide. It is difficult to

67

4. COUPLED MICROSTRIP LINE FEED WAVEGUIDE TRANSITION

realize in soft substrate material for multilayer structures.

(a) In-line transition with antipodal finlines c© 2012IEEE[135]

(b) In-line compact-size transition with ellipticshape finline c© 2015 IEEE [137]

(c) Vertical transition with short-ended slot line c©2014 IEEE [138]

(d) Planar transition in LTCC structure c© 2010IEEE [139]

Figure 4.2: Differential port transition prior art work.

Microstrip patch antennas have been widely used in transition design [141, 142]. Figure

4.3 shows the E-field of fundamental mode (TE01) in rectangular waveguide [143]. It shows a

good match with the fundamental mode (TM01) of patch antenna (see Figure 3.20). Since the

dielectric constant of the RF substrate is around 3, the size of the patch is smaller than the air-

filled waveguide geometry. For instance, the patch for 77 GHz is about 1 mm by 1.2 mm, which

is smaller than the corresponding air-filled waveguide WR12 – 1.55 mm by 3.1 mm. Therefore,

it is feasible to embed a microstrip patch antenna inside WR12 as a coupling element.

From the work in Chapter 3 of DMPA, it is a natural extension from the previous planar

DMPA design to a microstrip line feed waveguide antenna by introducing a transition from

coupled MSL to waveguide. The implementation of DMPA into waveguide transition leverages

a large group of waveguide antennas with high gain and high efficiency, such as horn antenna,

slot waveguide antenna, etc. In this work, the concept was first verified in 77 GHz with WR12

waveguide. Later on, various types for different optimization purposes in different bandwidths

68

are also discussed and verified by prototypes. The following sections of this chapter give detailed

explanation of these transition designs and application examples.

Figure 4.3: E-field in rectangular waveguide - TE01 mode.

4.2 Novel transition concept with DMPA: First prototype

in E-band

Figure 4.4 shows a 3-D view of the proposed transition. It is composed of a mechanical hous-

ing part – top mount – and a microstrip structure PCB part. The top mount is a modified

rectangular waveguide section with a channel cut on the narrow side of the wall. The channel

is for the coupled microstrip line connection. The key component of the transition is DMPA

which is integrated inside the top mount part. The DMPA acts as a radiating element for

signal coupling between the coupled microstrip lines and the waveguide transmission line. The

transition has a compact size and a simple fabrication.

There are two advantages of using differential microstrip patch antennas instead of single-

ended microstrip patch antennas in transitions. The first is that in DMPAs, the patch is fed from

the non-radiating edge. Hence, there is no need for cutting inset into radiating edges of patch

and cutting channel into the long side walls of the waveguide. Therefore, the coupling between

the patch and the waveguide is more efficient. The other advantage is that the bandwidth

behavior of a DMPA is better than that of a single-ended MPA, especially when the width of

the patch is small. This brings the possibility to insert parasitic patches into the waveguide

and increase the bandwidth of the transition.

In the simulation model, there are two waveguide ports set up – one for the coupled MSLs

and the other for the waveguide interface. The port for coupled MSLs is set as two modes in

the simulation analysis. It helps for the analysis of the common-mode behavior and differential-

mode behavior simultaneously.

In this section, two transitions are designed with a center frequency of 79 GHz for a WR12

waveguide (E-band). The first one uses a single patch DMPA, and the second one is a gap-

coupled patch DMPA. Both designs were verified by measurement of prototypes. In the follow-

ing sections, various designs for different optimization purposes are given.

4.2.1 Design of DMPA transition at 79 GHz

The transition has two parts: top mount (made of brass) and PCB part (made of RF substrate

material). Here, the design for the top mount is shown first. The top mount is a modified

waveguide line. In this work, waveguide size is selected as WR12 (a = 3.1 mm, b = 1.55 mm),

69


Figure 4.4: 3-D view of the transition structure.

which has an operating frequency range from 60 GHz to 90 GHz. The cross section of the

transition is shown in Figure 4.5.

Figure 4.5: Cross section of the transition (x-z plane). c© 2010 IEEE [144].

The width of the channel (Wc) is larger than the narrow wall (b) of the waveguide, while

the height of the channel (Hc) needs some trade-off. If Hc is too high, there are risks of wave

leakage from the top mount; if Hc is too low, the top mount will interfere the wave propagating

along the coupled MSLs. In this design, Hc is selected as 1 mm, which is about eight times

of the substrate thickness. It is an optimized value for the trade-off leakage suppression and

high-order mode suppression [145]. It is worth to mention that the top mount part is linked to

waveguide band. It can be reused in same waveguide band applications.

The key component of the transition is the DMPA part. The following part of the section

gives two examples of the design. The top mount parts are the same and can be shared for

both designs. The differences are the DMPA part designs. The first design is a single patch

DMPA inside the transition, and the second is a gap-coupled (GC) DMPA inside the transition.

The RF substrate is selected as Taconic TLE-95 (εr=2.95, h=0.127 mm, tanδ=0.004). Here,

designed details for both DMPA designs are given respectively.

70

A. Single patch DMPA transition

The simplest way to build such a transition is to place a DMPA which has been verified

in the previous chapter into a waveguide. Therefore, the first design is a single patch DMPA

transition. The top view of the PCB is shown in Figure 4.6. The waveguide area is marked

as rectangular shape with the size of a by b, where a = 3.1 mm and b = 1.55 mm. The patch

antenna is placed in the center of the waveguide area. Since the patch size is smaller than the

waveguide size (a by b), it is feasible to put a patch inside the waveguide. The radiating edges of

the patch are in parallel with the broad walls of the waveguide. The feeding lines are from the

narrow wall of the waveguide. In such configuration, the coupling between the patch and the

waveguide are maximum maintained. A C-shaped vias array surrounds the patch antenna area.

The vias array is outside of the waveguide area. It is for suppressing surface wave propagation

inside the substrate. The vias have a diameter of D=250 µm and pitch of d=400 µm.

Figure 4.6: Top view of PCB part for single DMPA transition, a=3.1 mm, b= 1.55 mm, D=250µm, d=400 µm, Wc=7 mm. c© 2010 IEEE [144].

The DMPA is the key component of the transition. The main difference between the dif-

ferentially fed antenna and the single-ended antenna is that the former one is fed through the

non-radiating edge of the patch. Therefore, no inset at the radiation edge of the patch is

necessary. Integrating a differentially fed antenna into a transition also removes the need for

cutting into the long side walls of the waveguide. Hence, good coupling effects between the

antenna patches and the waveguide walls are retained. Furthermore, the waveguide walls work

as electric shield walls for the patch. Therefore, more fields are in free space compared with

the classic microstrip patch antenna in open air. This reduces the Q factor of the patch. In the

remainder of this section, the design procedures are shown in detail. Later on, the results are

presented and discussed.

After building the first model, let us have a look at the difference to DMPA in open air. The

waveguide can be simplified as shielded wall around the patch, together with vias array. Figure

4.7 shows the model of patch in open air and waveguide, and d is the distance from the patch

edge to the waveguide wall. Figure 4.8 gives simulation results for three different cases. The

middle curve (blue) shows the patch in open-air situation, and d = infi here means the shielded

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(a) E-field of microstrip patch antenna in openair.

(b) E-field of microstrip patch antenna withshielded wall.

Figure 4.7: Cross section (x-y plane) of E-field of microstrip patch antenna in open air (a) andwith shielded wall (b). Top view of PCB part for single DMPA transition. c© 2010 IEEE [144].

wall has infinite distance to the patch. After adding the shielded wall, there are two kinds of

effects. If the distance is relatively large, for instance, twice the thickness of the substrate (h),

the resonant frequency shifts to the higher part (red curve). If the distance is relatively small,

for instance, half of h, the resonant frequency will shift to the lower part (orange curve). In

both cases, the relative bandwidth of the patch is bigger than the patch without the shielded

wall.

Those effects can be explained by further study of the E-field distribution in the transition

cross section. Figure 4.7 shows the cross-section E-field distribution of patch with and without

the shielded wall. In an open-air situation (see 4.7(a)), the E-field is concentrated in the

substrate. By adding shielded wall, the E-field is distracted by the wall. In other words, the

E-field beneath the patch is getting less (see 4.7(b)). This fact is responsible for high shifting

the resonant frequency of the patch.

Figure 4.8: Simulated return loss of transition for different d, where d is the distance from thepatch edge to the waveguide wall.

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There is another effect while putting patch inside the waveguide, and that is the increasing

capacitance between the edge of the patch and the shielded wall. It will become the dominant

effect when the shielded walls are close to the patch edges, for instance, less than the thickness of

the substrate (h). Therefore, bandwidth is increased by inserting the patch inside the waveguide.

Combining both effects, in most of the cases, because of manufacturing limitations, the spacing

between the patch edges and the waveguide walls are larger than h.

In the DMPA design procedure, the feeding position of the patch determines its impedance.

This also holds true for the transition design. A wider spacing in the DMSL corresponds to a

higher impedance of the microstrip port.

Figure 4.9 shows the simulation results for different patch width-to-length ratio (W/L). The

maximum bandwidth occurs when the W/L ratio is 1.1. It is different from DMPA in open-

air design and keeps the DMPA part compact in transition design. Figure 4.9 also adds the

bandwidth of DMPA in open air for comparison. The simulation results show that the relative

bandwidth for 10 dB return loss of single patch inside the waveguide reaches 6.6% while in open

air, it is about 4%.

Figure 4.9: Relative bandwidth of 10 dB return loss for the patch in waveguide.

Such a transition model was built in CST MWS and optimized for 78 GHz. Figure 4.10

shows the simulation results with optimized dimension. The coupled MSLs have a dimension

of Wm = 260 µm and Sm = 240 µm. The bandwidth for 15 dB return loss of the coupled MSLs

port in differential-mode reaches 2.66 GHz, while 10 dB return loss is 4.5 dB. The common-

mode signal is strongly suppressed for the whole frequency range because both the patch and

the waveguide do not support the common-mode wave propagation. The common-mode return

loss of the coupled MSLs port is less than 1 dB in whole E-band (60–90 GHz). The simulated

insertion loss is about 0.3 dB. Theoretically, the common-mode return loss should be 0 since the

structures are fully symmetric. But because of asymmetric structure in the realized examples,

there are certain level common-mode signals observed in the measurement.

B. Gap-coupled patch DMPA transition for wider bandwidth

The second design is intended to increase the bandwidth of the transition. There is a

common method in antenna design for bandwidth improvement – adding a parasitic patch. It

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Figure 4.10: Simulation results of the transition with single patch DMPA, w = 1.10 mm,l = 1.02 mm, Wm = 0.26 mm, Sm = 0.24 mm. c© 2010 IEEE [144].

can be easily implemented in the transition design, especially with DMPA structure.

The top mount is identical as in the first design. The modification is only the DMPA part

on PCB. The top view of the gap-coupled patch DMPA transition is shown in Figure 4.11. In

single patch DMPA transition design, the patch has a compact size (1.02 mm by 1.10 mm) –

almost a square shape. It brings benefit for adding additional patch in the waveguide area.

The DMPA has two patches: the main patch and the gap-coupled (or parasitic) patch. The

main patch is similar to the single patch DMPA. The parasitic patch is placed alongside the

non-radiation edge of the main patch. The main patch is connected with coupled MSLs, and

the parasitic patch is gap-coupled with the main patch.

With different dimensions of the two patches, two different resonant frequencies could be

reached. The patch length of parasitic patch l2 has a smaller size than the main patch l1. Since

the length of patch determines the resonant frequency, the main patch corresponds to a lower

resonant frequency while the parasitic patch corresponds to a high resonant frequency.

The differential signals are injected in the main patch through the coupled MSLs. Sub-

sequently, through coupling effects, the electromagnetic power on the main patch excites the

parasitic patch. Therefore, the bandwidth of the transition is almost doubled here. It is dif-

ferent from the classic single-end MPA transition where the patch is center placed, and there

is less space for a second patch within the waveguide area. After adding the second patch, the

impedance matching cannot be reached without a matching network. The matching network is

composed of two sections of transmission lines. The first one, from A-A’-plane to B-B’-plane, is

shorter than a quarter of the wavelength. It converts the patch impedance into a value greater

than 100 Ohm on the real axis of the Smith Chart. The second one, from B-B’-plane to C-

C’-plane, has a high characteristic impedance. It works as a quarter-wavelength transformer

and matches the 100 Ohm differential-mode impedance of coupled MSLs. The simulated S-

parameters of the optimized GC DMPA transition are shown in Figure 4.12. It shows 7 GHz

bandwidth for 15 dB return loss which is more than double of single DMPA transition. The

common mode return loss is slightly high at 78 GHz.

Assembling tolerances are studied by simulation for parameter d1 and d2. d1 and d2 are the

74

misalignment of the top mount of the PCB in x-axis and y-axis, respectively. The tolerance

range is selected as 100 µm for both parameters. Simulated results are shown in Figure 4.13.

The results show that the transition has robust performance for assembling tolerance.

Figure 4.11: Top view of the transition with gap coupled DMPA. c© 2010 IEEE [144].

Figure 4.12: Simulation results of the transition with gap coupled DMPA, w1=0.46 mm,w2=0.40 mm, w3=1.00 mm, w4=1.00 mm, l1=1.00 mm, l2=0.94 mm, g1=0.10 mm, g2=0.35mm, s0=0.76 mm, s1=0.24 mm, s2=0.56 mm, s3=0.24 mm. c© 2010 IEEE [144].

4.2.2 Manufacturing and measurement of DMPA transitions at WR12

Both transitions were manufactured and measured. The transition can be treated as a three-

port device, including coupled MSLs port (differential mode and common mode) and waveguide

port. There is no direct measurement setup for supporting such devices. In this work, the mea-

surement were performed in three different measurement setups.

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Figure 4.13: Tolerance of d1 and d2 in gap-coupled DMPA transition.

Measurement step A: Two port measurement for back-to-back test boards

Since the transitions have an asymmetric port structure – waveguide port and coupled

microstrip line port – the most common way is the back-to-back (B2B) test structure. Both

designs were first verified in back-to-back test structures. Figure 4.14 shows a photo of the

manufactured top mount which is made of brass. The top mount was designed with WR12

compatible pin and screws. Figure 4.15 shows the PCB part of the back-to-back (B2B) test

structure of both transitions.

Figure 4.14: Photo of top mount part of transition.

The measurements were performed by two-port vector network analyzer (VNA). Figures

4.16 and 4.17 show the measurement results for both designs. There are some frequency shifts

because of the manufacturing tolerance. The measurement of the second manufacturing of gap-

coupled DMPA transitions B2B test board, as well as simulation results show good agreement

(see Figure 4.18). From the B2B measurement, the functionality of the transitions has been

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(a) B2B structure of the transition with singleDMPA.

(b) B2B structure of the transition with gap-coupled DMPA.

Figure 4.15: Photo of test structure of transitions with single DMPA (a) and gap-coupledDMPA type (b). c© 2010 IEEE [144].

Figure 4.16: Measurement results of B2B structure of the transition with single patch DMPA.c© 2010 IEEE [144].

verified. The insertion loss as well as the return loss can be estimated. The ripples of return

loss reduce the measurement accuracy. This is improved in measurement step B.

Measurement step B: One port measurements of test boards with spiral struc-

ture

In step B, the transition is measured as one port device from waveguide port. The other

port of transition – the coupled MSLs port – is terminated with spiral structure. In this step,

gap-coupled transition with spiral structure at the coupled MSLs port has been manufactured

and tested. The spiral structure, together with the absorber material on top, behaves as a good

matching load. Figure 4.19 shows the measured return loss (waveguide port) of such test struc-

ture with and without the absorber material. The return loss of transition without the absorber

material has ripples in the measured bandwidth. The ripples are coming from the reflection

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of coupled MSLs port. In the measurement of the transition with the absorber material, the

ripples are gone. Therefore, the absorber material shows improvement for the matching load. A

couple of differential absorber materials are tested. Among them, the best materials are QR13

and CRAM369. In general, the transition shows 10 dB return loss bandwidth from 74 GHz

to 82 GHz. It matches the simulated results of the transitions. It is slightly larger than the

simulation results because of loss in the substrate.

Measurement step C: LRdR for coupled MSL port measurement

Till now, the concept of the transition has been verified by insertion loss and return loss of

the waveguide port. More difficult is the return loss of the coupled MSLs port under different

modes. In this part, the author developed a LRdR (load, reflect, delayed reflect) method for

verifying the return loss of the coupled MSLs port.

The transition can be treated as a three-port device, include asymmetric MSL port and

Figure 4.17: Measurement results of B2B structure of the transition with gap-coupled DMPA.c© 2010 IEEE [144].

Figure 4.18: Measurement results of GC DMPA transition on TLE95 material vs simulation.

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Figure 4.19: Measurement results of transition with spiral load wt/wo absorber material.

waveguide port. It is possible to simplify the three-port device to a two-port device in the

measurement. The solution is using three different terminators on waveguide port in three

measurements. Hietala introduced a method for determining two-port S-parameters from only

one-port measurement results in 1999 [146]. The same model can be extended to three-port

device. The measurement setup of the transition can be described as a loaded two-port network.

Figure 4.20 shows a block diagram of a loaded two-port network and its equivalent S-

parameter model. The transition is depicted as a device under test (DUT), which has two types

of port. Port 1 is coupled MSL port, and port 2 is waveguide port. The matrix [S] represents the

S-parameters of the transition. Standard connectors are connected to port 2. Γs,i denotes the

reflection coefficient of the i-th standard connector. Γi is the reflection coefficient measured at

port 1 with the i-th standard connector at port 2. It must be noted that there are two different

modes of a coupled microstrip line (differential mode and common mode); therefore, Γi is

different for each mode. The coupling between models are neglected here because of symmetric

structures. Γi must be selected for the correct mode when extracting the S-parameter matrix

of the transition.

Considering Figure 4.20, it is easy to deduce the relationship among Γi, Γs,i and [S] as

follows

Γi = S11 +S12 · S21 · Γs,i

1− S22 · Γs,i(4.1)

Taking the reciprocal property of the transition into account (S12 = S21), there are three

unknowns in Equation 4.1: S11, S12, and S22. Therefore, using the reflection coefficients Γs,i

of the three standards, [S] can be calculated. Closed-form formulas of calculation are derived

on the basis of matrix calculation.

Multiplying both sides of Equation 4.1 with (1− S22 · Γs,i) results in

S11 − S11 · S22 · Γs,i + S212 · Γs,i + S22 · Γs,i · Γi = Γi (4.2)

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(a) Block diagram of measurement of LRdR. c©2011 IEEE [147].

(b) Signal flow diagram.

Figure 4.20: Block diagram of measurement and signal flow diagram.

To obtain a linear equation, W is introduced here:

W = −S11 · S22 + S212 (4.3)

Inserting Equation 4.3 into Equation 4.2, a linear equation is reached for the three unknowns:

S11, W , and S22:

S11 +W · Γs,i + S22 · Γs,i · Γi = ·Γs,i · Γi (4.4)

With three different standards – matched load, reflector, and delayed-reflector – we obtain

solutions as follows: 1 Γs,ML Γs,ML · ΓML

1 Γs,R Γs,R · ΓR

1 Γs,d−R Γs,d−R · Γd−R

S11

W

S22

=

ΓML

ΓR

Γd−R

(4.5)

S11, W and S22 can be calculated as

S11

W

S22

=

1 Γs,ML Γs,ML · ΓML

1 Γs,R Γs,R · ΓR

1 Γs,d−R Γs,d−R · Γd−R

−1 ΓML

ΓR

Γd−R

(4.6)

After obtaining the array, [S11, W and S22], S12 can be calculated according to Equation

4.3.

S12 = ±√W + S11 · S22 (4.7)

The sign of S12 can be determined from prior information about the phase.

It is important to note that none of the above derivations depend on the port types of the

DUT. The crucial point is that the Γs,i must be measured with the same port impedance of

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port 2 of DUT. This method is suitable for measuring a transition with different types of port.

However, it is not limited to transitions; it can be used to extract the S-parameters of other

hybrid 2-port device.

The test board, as well as measurement setup, is shown in Figure 4.21. The design of gap-

coupled transition is implemented as DUT. The test board has a waveguide port interface and

a coupled microstrip lines interface. Because the coupled microstrip lines cannot be measured

directly by G-S-G probe tips, we convert the coupled microstrip lines to two single-ended

microstrip line ports. Those two single-ended MSL ports can be measured by G-S-G probe tips

of a VNA.

Since open is not a good choice for waveguide port, in the project, the three standard

terminators are matched load, short, and delayed short. This leads to the name LRdR (load,

reflector and delayed reflector).

The measurement procedure consists of two steps. The first step is to measure the reflection

coefficients Γs,i of the three standards. This is a one-port measurement of the waveguide port.

The second step is to measure Γi of coupled MSLs port with different standard connectors at

the waveguide port. This is a two single-ended MSL ports measurement. Subsequently, the Γi

of two different modes can be calculated from these measurements.

Calculation steps:

The coupled microstrip line has two different modes: differential mode and common mode.

Therefore, the first step of the calculation is to convert the single-ended measurement results

into a mixed-mode matrix according to the following equations in [110].

M =1√2

[1 −1

1 1

](4.8)

SMM = M · SSE ·M−1 (4.9)

where SSE denotes the two-port single-ended measurement result. SMM is the mixed-mode

matrix, in which S11,MM stands for the reflection coefficient in differential mode, and S22,MM

stands for the one in common mode. Using this conversion, we obtain Γi of the two different

modes.

The second step is to calculate the S-parameters of the transition using Γi and Γs,i. The

model and formulas 4.5, 4.6, 4.7 are used here to calculate the S-parameters either in differential

mode or in common mode.

The calculated results are plotted in Figure 4.22. To provide a better comparison, the results

include simulation results for a single transition and measurement results from a B2B structure

of the transition.

Figure 4.22(a) shows the return loss of the waveguide port from the simulation, the presented

method, and the B2B structure. The bandwidth of the return loss from the B2B structure

measurement is difficult to determine, since there were several ripples that distorted the results.

But from the presented method, the ripples are much more moderate. It is easy to determine

that the 15 dB return loss bandwidth is 7.4 GHz, which matches the simulation results (6.6

GHz). The center frequency shift is caused by the fabrication tolerance.

Figure 4.22(b) shows the return loss of the coupled MSLs port, the differential-mode and

common-mode return loss of the coupled MSLs port are plotted for comparison, including both

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Figure 4.21: Photo of test board of GC DMPA transition in E-band for LRdR measurementsetup. c© 2011 IEEE [147].

simulated and calculated results. It must be noted that these results cannot be obtained from

any two-port measurement of B2B structures of the transition. The calculated results show that

15 dB return loss bandwidth of the differential mode is much wider than that of the common

mode. This agrees with the simulation results. The narrow bandwidth of the common mode

is caused by the parasitic patch within the transition. There are still some moderate ripples in

the return loss because of the feeding lines.

The improvement that our method provides is even clearer in the calculated insertion loss

(differential mode of coupled MSLs to waveguide) (see Figure 4.22(c)). The strong ripples in

the B2B measurement are removed completely when using LRdR method. The higher loss in

the measurement is coming from the fabrication tolerances and surface roughness.

Finally, in Figure 4.22(d), the insertion loss of both differential mode and common mode is

plotted together for comparison. Theoretically, the common-mode insertion loss will be very

high, but because of fabrication and measurement limitation, it is about 6 dB lower than the

differential-mode insertion loss. Taking the high reflection of the common-mode signal into

consideration, the common-mode propagations are highly suppressed by the transition. This

fulfills the design target.

4.2.3 Material comparison in transition design

The same layout of B2B structures of gap-coupled transition is manufactured with two different

RF substrates for verification – TLE95 and RO3003. Figure 4.23 shows the measurement results

of test structures with different RF substrates – RO3003 and TLE95. The measurement results

show both test structures have 10 dB return loss from 74 GHz to 80 GHz. It proves the

transition can be implemented with both RF substrates. The insertion loss of the sample of

RO3003 is less than the sample of TLE95. This can be explained by the different loss tangent

of the material.

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(a) Return loss of the waveguide port. (b) Return loss of the coupled MSLs port.

(c) Differential-mode insertion loss. (d) Common-mode and differential-mode in-sertion loss.

Figure 4.22: LRdR measurement results and simulation results of the transition. (a) Returnloss of the waveguide port from the simulation, the presented method, and the B2B structure.(b) Return loss of coupled MSL port from the simulation and the presented method. (c)Differential-mode insertion loss from the simulation, presented method, and B2B structure. (d)Comparison: insertion loss of differential mode and common mode from the presented method.c© 2011 IEEE [147].

4.3 Improvement for common-mode signal suppression:

Prototype design in W-band

In the previous section, the gap-coupled transition has a narrow band resonance at 79 GHz

for the common-mode signal. Further study shows that the common-mode resonance is coming

from the parasitic patch. When the concept extends to W-band, the common-mode signal rejec-

tion is also worse. The return loss of common-mode signals may reach 3 dB. Such common-mode

signals within the transition either leak to the substrate or radiate from the waveguide port.

To suppress the common-mode signal, a new design with center frequency at 94 GHz is shown

in this section. In addition, the matching network design is more robust for manufacturing.

The structure of such a transition is shown in Figure 4.24. It has a similar structure as

before, while the waveguide is selected WR10 for W-band applications. The main change is

the DMPA part. In this work, one side of the parasitic patch is connected to the ground to

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Figure 4.23: Measurement results of GCP transition on RO3003 and TLE95 material.

facilitate the matching network design and suppress the common-mode signals. The substrate

material of the PCB is Taconic TLE-95 (εr=2.95, h=0.127 mm, tanδ=0.004). The conductivity

of the coupled MSLs is 5.8e7 S/m. The transition housing has a waveguide interface for WR10

and a channel for coupled MSLs.

The transition housing is a modified rectangular waveguide section (see Figure 4.24). A

channel which allows signal propagation along the coupled MSLs is cut into the short side wall

of the waveguide. The dimension of the channel (Hc and Wc) should not be so small as to

interfere with the wave propagation on the PCB. However, if Hc is too high, it causes waves to

leak from the transition. In this work, Hc was chosen to be 1.0 mm, which is about eight times

the thickness of the substrate h. Wc was chosen to be 5.0 mm.

Figure 4.24: Structure of vertical transition between rectangular waveguide and coupled MSLs– short-ended parasitic patch. c© 2012 IEEE [148].

The top view of the PCB is shown in Figure 4.25. The antenna includes the radiation part

(the main patch and the parasitic patch) and the matching network. The main patch of the

antenna is fed by coupled MSLs. The parasitic patch is side placed to the main patch. The

electromagnetic power is coupled through the non-radiating edge of both patches. Different

84

from the previous design, the other side of the parasitic patch – the right side of the patch in

the figure – is connected to the ground by vias. This modification brings two advantages in the

design: (1) it suppresses the common-mode signal in the center frequency, and (2) it facilitates

the matching network design.

Figure 4.25: Top view of the PCB of the transition – W-band transition with common modesuppression. c© 2012 IEEE [148].

There are two kinds of propagation modes in the coupled MSLs: the differential mode and

the common mode. Figure 4.26 shows the simulated E-field distribution of the antenna under

both mode of input signals. The radiating edges of the patches, which have uniform E-field dis-

tribution, are parallel to the long side walls of the waveguide (x-axis) for the differential-mode

feeding signals, whereas they are parallel to the short side walls (y-axis) for the common-

mode feeding signals. Consequently, the fundamental modes of the patch are TM01 mode for

differential-mode signals and TM10 mode for common-mode signals. TM01 mode also matches

with the fundamental mode of the waveguide. The match of signal modes allows the transmis-

sion of energy.

In the case of differential signals, the resonant frequency of the TM01 mode on each patch

is determined by the patch length (l3, l4). It is slightly less than λr/2 because of fringing

effects. Here, λr is the wavelength in the dielectric layer. Dual resonant frequencies are realized

by setting l3 and l4 to different values. The patch widths (w3, w4) are the main factors in

determining the bandwidth of the transition. Therefore, for the main patch, w3 is slightly

larger than l3. For the parasitic patch, because of the short-end edge, w4 is much longer than

l4 to support the desired TM01 mode.

In the case of common-mode signals, the resonant frequency of the TM10 mode on the patch

is determined by the patch lengths (w3, w4), and because of the feeding lines of the main patch

and the short-ended edge of the parasitic patch, the TM10 modes are suppressed in the desired

frequency range. But w4 should not be longer than 1.5λr; otherwise, higher order modes arise.

The patch impedance (in the A-A’-plane) has an inductive part. Therefore, two sections

of coupled MSLs are built for the impedance match. The matching network is similar to

the previous design. The first one, from A-A’-plane to B-B’-plane, is shorter than a quarter

of the wavelength. It converts the patch impedance into a value greater than 100 Ohm on

85


the real axis of the Smith Chart. The second one, from B-B’-plane to C-C’-plane, has a

high characteristic impedance. It works as a quarter-wavelength transformer and matches the

100 Ohm differential-mode impedance of coupled MSLs. If the inductive part of the patch

impedance (at the A-A’-plane) is too high, the matching network is difficult to realize because

of fabrication limitations. The short-ended parasitic patch reduces the inductive value of the

patch impedance, which facilitates realization of the matching network.

(a) E-field of the differential-mode signal at 93 GHz

(b) E-field of the differential-mode signal at 100 GHz

(c) E-field of the common-mode signal at 96 GHz

Figure 4.26: E-field distribution of transitions at differential mode (a-b) and at common mode(c). c© 2012 IEEE [148].

After optimization with CST MWS, a prototype of design is achieved. The S-parameter

results of the transition with the optimized dimensions are shown in Figure 4.27. The simulation

Figure 4.27: Simulated S-parameters of the transition (l1=0.53 mm, w1=0.13 mm, S1=0.50mm, l2=w2=0.25 mm, S2=0.26 mm, l3=0.81 mm, w3=0.92 mm, l4=0.78 mm, w4=1.05 mm,g=0.11 mm, d=0.4 mm, D=0.25 mm, a=2.54 mm, b=1.27 mm, h=0.127 mm, Hc=1.0 mm,Wc=5.0, εr=2.95). c© 2012 IEEE [148].

86

Figure 4.28: Photo of W-band transition back-to-back structure with shim. c© 2012 IEEE [148].

model is also shown at the left corner. Both ports are set as rectangular waveguide port in

CST MWS. From the simulation results, the 10 dB return loss bandwidth of differential mode

signals is about 15.9 GHz (BW% = 16.6%). The return loss of the waveguide port exhibits

a similar behavior. The insertion loss of the differential mode signals to waveguide is 0.4 dB

at the center frequency (96 GHz). The common-mode return loss of the coupled MSLs port is

only 0.5 dB within the bandwidth and less than 1 dB for the whole W-band. The insertion loss

of the common mode to waveguide (or to differential mode) is too low to plot.

The back-to-back (B2B) structures of the designed transition with different connecting line

lengths were fabricated and subsequently measured. For ease of fabrication, the transition

housing was simplified into a 1 mm thick transition slice – shim (see Figure 4.28). Connecting

it to a standard WR10 waveguide flange results in the transition housing depicted in Figure

4.24. It is a cheaper solution compared with a full-metal housing as in Figure 4.14.

Figure 4.29: Measurement and simulation results of B2B structure of the transition with dif-ferent lengths of the connecting lines, 21 mm and 31 mm, respectively. c© 2012 IEEE [148].

87


The measurement and the simulation results of the B2B structure are plotted in Figure 4.29.

The bandwidth is 14.5 GHz for 10 dB return loss and 10.8 GHz for 15 dB return loss. It matches

the simulation results. The frequency shift is caused by fabrication tolerances. The insertion

losses were 3.7 dB and 5.0 dB at 96 GHz for short (21 mm) and long (31 mm) connecting lines,

respectively. Therefore, the insertion loss for a single transition is 0.5 dB. The loss tangent of

the material (tanδ=0.004) quoted by Taconic is only up to 20 GHz. This introduces a difference

between the measured and the simulated insertion loss. If loss tangent (tanδ) is taken as the

dominant factor, neglecting the surface roughness of conductors, a value of tanδ=0.008 fits the

measurement well.

Repeatability Test

The repeatability of the test boards were tested in two different ways. The measurement results

are shown in Figure 4.30. In the first test, the top mount part was dismounted and reassembled

five times (see Figure 4.30(a)). It verifies the mounting repeatability on the same board. In

the second test, three PCB samples with the same design were measured to verify the PCB

manufacturing stability (see Figure 4.30(b)). In both tests, the measurement results show very

good repeatability. It proves the transition design is a robust solution. It also shows the simple

solution of shim supports good repeatability.

(a) Repeated measurements of the same board ofB2B structures of the transitions.

(b) Measurements of the different boards of B2Bstructures of the transitions.

Figure 4.30: Repeatability test of reassembling one same board (a) and different PCB samples(b).

4.4 Further bandwidth improvement by extended ground:

Prototype for E-band transition

This section introduces a simple method for further extending the bandwidth of a transition

from rectangular waveguide to coupled microstrip lines (differential mode). Reducing the gaps

between microstrip patch antenna and ground on top of the substrate reduces the Q-factor of

the antenna and thus increases the bandwidth of the transition. A prototype of such a transition

with center frequency of 79 GHz was designed, and back-to-back structures were fabricated.

The measurement of prototype verifies the design target. The measurement results show that

the bandwidth of 10 dB return loss is improved from 7.2 GHz to 9.2 GHz. The new structure

does not introduce additional cost for fabrication.

The transition structure (see Figure 4.31) is composed of the transition housing (brass) and

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the PCB. The transition housing is same as the gap-coupled transition in Section 4.2. It is a

section of WG and has an opening channel for coupled MSLs. The RF material is using RO3003

substrate (εr=3, tanδ=0.0017) with a thickness (h) of 0.127 mm.

Figure 4.32 shows a cross section of both former design and new proposal to illustrate the

difference. The ground (GND) on top of the substrate is extended into the WG by a distance

c. This results in two improvements over the transition presented in Section 4.2. Firstly, closer

proximity to the ground decreases the effective dielectric constant of the patch (εeff ). Secondly

– and more importantly – smaller gaps between ground and antenna patch edges increase the

capacitances at the antenna edges. Both effects result in a broader bandwidth of the transition.

Figure 4.31: Structure of transition with extended ground DMPA (parameters: a=3.10 mm,b=1.55 mm, Hc=1 mm, Wc=7 mm).

The antenna embedded in the waveguide is the critical part of the transition. It is composed

of two patches and an impedance matching network. Figure 4.33 shows the antenna on PCB

in detail. The two-patch structure introduces two resonant frequencies which depend on the

lengths of the patches (l3, l4). A shorter length corresponds to a higher resonant frequency and

vice versa. Compared with the previous design, the proposed transition has smaller εeff and

thus smaller patch sizes. The parasitic patch is smaller than the main patch because higher

frequency signals can easily be coupled through the gap.

The parasitic patch configuration increases the equivalent width of the antenna patch and

introduces a higher inductance value of the antenna impedance. In other words, the impedance

loci of the antenna are located in the upper part of the Smith Chart. Therefore, a two-section

transmission line matching network is used to match 100 Ohm, differential-mode characteristic

impedance of the coupled MSL. The first section, from plane A-A’ to B-B’, with a length of l2,

converts antenna impedance loci to the real axis of the Smith Chart (>100 Ohm). The second,

from plane B-B’ to C-C’, with a length of l1, acts as a quarter-wavelength impedance trans-

former. Antenna reactance, which is defined as the imaginary part of the antenna impedance,

increases with decreasing εeff . Therefore, a higher characteristic impedance transmission line

is required in the second section of the matching network.

The simulated S-parameters of single transition (after optimization of the dimensions) are

shown in Figure 4.34. The bandwidth of differential mode return loss is 11.5 GHz for 10

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(a) (b)

Figure 4.32: Cross-section of transitions in OO’-plane transition without extended ground inwaveguide (a) and with extended ground in waveguide (b).

Figure 4.33: Enlarged details of PCB design in extended ground DMPA transition.

dB (BW% = 14.6%) and 9.2 GHz for 15 dB. The insertion loss is around 0.2 dB within the

bandwidth. Except around 88 GHz, the common-mode return loss of the coupled MSLs port

(S11,CM ) is below 1 dB within the whole E-band.

Measurement results:

Transitions prototypes were fabricated in back-to-back (B2B) structures and measured. The

transmission lines in between are 2 cm long. Figure 4.35 shows the results and includes, for

better comparison, the measured S-parameters of transition in Section 4.2. As can be seen,

both transitions have similar insertion loss at center frequency, but the new transition has a

2 GHz wider bandwidth for |S11,DF | < −10 dB. Insertion loss of B2B structure at 77 GHz is

2.5 dB. The propagation loss of the transmission line is about 1 dB/cm, therefore the single

transition has an insertion loss of less than 0.3 dB and thus matches the simulation results.

This section presents a new transition from coupled MSLs to WG. At no additional fabri-

cation cost, the bandwidth of the transition is increased by 28% for 10 dB return loss. The

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Figure 4.34: Simulated S-parameters of an extended-ground transition: s1=0.56 mm, s2=0.24mm, l1=0.58 mm, l2=0.38 mm, l3=0.94 mm, l4=0.89 mm, w1=0.10 mm, w2=0.26 mm, w3=1.00mm, w4=0.85 mm, g1=0.11 mm, g2=0.13 mm, g3=0.14 mm, g4=0.31 mm, c=0.20 mm, d=0.40mm, D=0.35 mm.

insertion loss within the bandwidth is around 0.3 dB for a single transition.

4.5 Summary and applications

Summary

This chapter shows four designs for the waveguide transitions in E-band and W-band. All

transitions are connecting waveguide port with coupled MSLs port. Table 4.1 gives a summary

for the performance comparison from the back-to-back measurement results. From the compar-

ison, single patch DMPA transition has the simplest structure. The gap-coupled (GC) DMPA

Figure 4.35: Measured S-parameters of the B2B structures of the transitions, classic and pro-posed.

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largely increases the bandwidth of the transition. The extended ground inside the waveguide

may further enlarge the bandwidth. The transition for W-band has broadest bandwidth be-

cause of its higher operating frequency.

Table 4.1: Summary of transitions performance

DMPA type in Transition 10dB RL BW% IL f0 Waveguide port

Single DMPA 4.5 GHz 6.6 % 0.3 dB 77 GHz WR12

GC DMPA 7.2 GHz 9.7% 0.3 dB 77 GHz WR12

GC DMPA + Ext. GND 9.2 GHz 12% 0.3 dB 77 GHz WR12

Short-ended GCP DMPA 14.5 GHz 15% 0.5 dB 96 GHz WR10

Application

Figure 4.36: Application of GCP transition in polarimetric mmW radar system. c© 2012 IEEE[149].

There are a couple of potential applications for the transition in mmW systems. Here,

an application example in polarimetric radar is given. Reference [149] shows an application

of waveguide transition in a polarimetric radar system. Figure 4.36 shows the front end of

the polarimetric radar. There are three TRx horn antennas in the system. Two are vertical

polarization (A2 and A3 in the figure), and the other is horizontal polarization (A1 in the

figure). The horn antennas were mounted on PCB. The chip set output ports are differential

signal. The GC DMPA transition has been implemented here for the connection to horn

antenna. Besides connection with horn antenna, the transition also supports feasibility with

slot waveguide antenna, etc.

This chapter presents transitions designs where the DMPA is key element in the transition.

With development of 3-D printed technologies, the proposed transition may stimulate more and

more interest in mmW applications.

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Chapter 5

Differential Antenna in eWLB

Package

5.1 Introduction of antenna in package

Highly integrated antenna is gaining more and more attraction for mmW antenna development.

There are two popular ways of improving the integration level of the system: first, antenna on

chip, (AoC) and second, antenna in package (AiP) [150]. In both ways, the RF front-end systems

can be made compact. Comparing these groups of solutions, antenna on chip (AoC) has a higher

integration level compared with antenna in package (AiP) but has poor efficiency [63, 62]. The

antenna on package is the best compromise for integration level and antenna performance. This

chapter demonstrates the development of differential feed antenna in package.

Many different types of AiPs have been developed in the last two decades, for instance, an-

tenna on multilayer structure using low-temperature co-fired ceramics (LTCC) or liquid crystal

polymer (LCP) substrate, AiP by quad-flat no-lead (QFN) solutions [70], and embedded wafer

level ball grid array (eWLB) package solutions [73], etc.

Figure 5.1: The geometry of the aperture-coupled microstrip patch antenna with LTCC solution:top view (left) and cross section (right). c© 2011 IEEE [151].

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5. DIFFERENTIAL ANTENNA IN EWLB PACKAGE

LTCC has been considered as an AiP solution for many years. It has a generic multilayer

structure and supports the flexible configuration of the antenna and feeding network. Many dif-

ferent types of antenna have been tried with LTCC solutions in mmW applications. Lamminen

proposed an aperture-coupled microstrip patch antenna design at 60 GHz [151]. It implements

the multilayer advantages of LTCC for wide bandwidth antenna design. Figure 5.1 presents the

top view and cross section of the antenna. It is an aperture-coupled patch antenna which im-

plements three metal layers. Prof. Zhang proposed a couple of solutions for grid array antenna

on LTCC structure [152], [153]. [152] introduced a dual feeding structure and [153] extended

it to multiple feedings. The radiation patterns of the grid array antenna show great stability

over wide frequency ranges.

X. Wang designed a couple of antennas for 77 GHz / 79 GHz radar applications by low-loss

LTCC material [140, 154, 155, 67]. In [154], a half-lambda grid array structure was developed for

a very broad beam pattern in azimuth plane. LTCC structures provide a low-loss transmission

line solution – laminated waveguide (LWG) – which benefits complex feeding network design.

In further work of X. Wang [155] and [67], he demonstrated two designs of dual patch sub-array

antenna with LWG feeding network, one for vertical polarization and the other for a 45-degree

polarization solution (see Figure 5.2).

(a) The fabricated LTCC RF front-end of the an-tenna side (left) and MMICs side (right). c© 2015[67].

(b) Dual patch subarray aerial view of the 3-D model(top) cross-sectional view (bottom). c© 2013 [155].

Figure 5.2: LTCC antenna designs at 77/79 GHz radar applications.

The research group in IBM introduced a superstrate structure in AiP designs. The concept

was first realized in LTCC material and later in soft substrate – LCP material for cost-reduction

reasons [156, 157]. Figure 5.3 shows the package stack-up of both proposals. An embedded

cavity is realized by multilayer structures. The MMICs were mounted with a flip-chip BGA

package. The specified structure sets air (εr = 1) below the radiating element and the dielectric

substrate (εr > 1) above the radiating element. Therefore, this kind of antenna has the best

combination of bandwidth and antenna gain performance.

Another type of antenna that has been realized in mmW application is parasitic stacked

patch antenna. The research groups in from IBM and IMEC proposed such designs with soft

substrate material [158, 159]. Figure 5.4 shows the two examples in W-band and E-band radar

applications. The parasitic stacked patch brings a very broad bandwidth of antenna. Soft

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(a) LTCC package stack-up for embedded cavity an-tenna.

(b) LCP package stack-up for embedded cavity an-tenna.

Figure 5.3: Antenna-in-package solution of superstrate structure by (a) LTCC material and (b)LCP substrate. c© 2010 [156]

substrate material reduces the material cost compared with LTCC, but the manufacturing for

multilayer structure is a big challenge.

Multilayer structure makes the antenna design feasible, which supports various antennas

with good RF performance. But as a packaging solution, the connections between MMICs and

antenna are either flip-chip BGAs or bonding wires. This still limits the integration level of the

whole RF front-end and increases the loss between MMICs and antenna.

A complete package solution which supports antenna integration is very attractive. In 2004,

IBM proposed a single package solution for integrated chip and antenna in one QFN package

[69]. Since then, many different antennas were developed by QFN package. Recently, Goettel

reported a new design for a lens integrated with AiP on QFN [160]. In this work, the AiP is

combined with a lens antenna for increasing gain. Figure 5.5 shows the cross section of the

(a) Cross section of the stacked patch antenna byLCP for W-band radar by IBM. c© 2014 [158].

(b) Photo of stacked patch antenna array for 79 GHzradar by IMEC. c© 2017 [159].

Figure 5.4: Parasitic stacked patch antenna examples from (a) IBM and (b) IMEC.

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Figure 5.5: QFN packaging solution for AiP: IC mounted directly on the silicon lens. c© 2017IEEE [160].

concept. [70] gives a good summary of QFN-based AiP in the last decade. Among all of the

QFN solutions, the bonding wires are still implemented in the package. They increase the

assembling cost but also improves the performance of the connection in mmW applications.

Since a new package technology – embedded wafer level ball grid array (eWLB) – is available

for mmW applications [71, 72], a variety of AiP in mmW applications have been developed [73].

Compared with QFN package, eWLB eliminates the bonding wire in the packaging connections.

A couple of new designs have been proposed recently for mmW radar applications. For instance,

[161] shows a stack director solution for improving the radiation beam. [162] presented a stacked

metallization solution on top of rhombic antenna in eWLB. The stacked metallization solution

largely improved the radiation performance of the antenna. In addition, a couple of rhombic

antenna were developed for AiP with eWLB package [163].

From the next section, this chapter is concerned with a differential feed AiP in eWLB

package.

(a) Stack director on folded dipole AiP. c© 2012[161].

(b) Stack metallization on rhombic antenna AiP. c©2013 [162].

Figure 5.6: Radiation beam optimization of eWLB AiP by (a) stack director and (b) stackmetallization.

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5.2 Folded dipole AiP with eWLB package

5.2.1 eWLB structure

The term eWLB stands for embedded wafer level ball grid array. It is a fan-out wafer-level

package technology which was first introduced in 2006 [71, 72]. Figure 5.7 shows the comparison

between standard wafer-level packages (WLP) and fan-out wafer level packages (fan-out WLP).

In standard WLP, the ball grid arrays (BGAs) are directly under the MMICs. The I/Os number

is limited by the MMICs’ size and pitch size. In eWLB, the package size is largely increased by

introducing the redistribution layer (RDL). RDL is the addition of metal and dielectric layer

onto the surface of wafer. RDL reroutes I/Os from MMICs to a large area of BGAs. Therefore,

the interconnection gap between MMICs and PCBs is largely reduced. Secondly, RDL supports

the connection from MMICs to BGAs instead of the bonding wire and laminate substrate. It

benefits the implementation in mmW applications.

Figure 5.7: Comparison of standard WLP (top) and fan-out WLP (bottom). c© 2006 IEEE[71].

Figure 5.8 shows the process flow for the eWLB manufacturing. The starting point is a front-

end processed wafer. When the wafer is singulated, the chips typically stick on an adhesive

carrier foil with a dicing frame (see Figure 5.8(a)). The singulated chips are picked and placed

onto a carrier with large spacing. Standard “Pick and Place” equipment is used to place the

dice onto a metal carrier. In the next step, the dice on the metal carrier is molded (see Figure

5.8(b)). The encapsulation process is the core process in the embedded die technology. On

the obtained “Reconfigured Wafer”, the redistribution layer is deposited (Figure 5.8(c)), and

afterwards, the balls for the second level are interconnected (Figure 5.8(d)). More details are

referred to [71, 72].

The RDL has a much higher resolution compared with transmission lines on PCBs. For

instance, the 100 Ohm coupled transmission line on PCB is about W/S/W = 240 µm/200

µm/240 µm, while in RDL, it is W/S/W = 35 µm/20 µm/35 µm. It brings a new concept

for building the antenna in this layer. The following sections of this chapter give a couple of

design examples for the differential feed AiP in eWLB packaging. The impact of MMICs and

package size on antenna performance is also analyzed. Different design proposals for ultrawide

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Figure 5.8: Schematic process flow for a fan-out wafer level package. c© 2006 IEEE [71].

bandwidth, high-gain antenna, etc., are discussed in the following sections.

5.2.2 Folded dipole AiP design

The cross section of eWLB AiP is shown in Figure 5.9, together with MMICs. The antenna was

built in the RDL layer. The ground on PCB beneath the antenna area behaves as a reflector

for the AiP. The maximum gain of radiation pattern is about the broadside direction of the

antenna. The distance between RDL and ground plane is defined by the height of BGAs, which

has a typical value of 180 µm.

In RDL, the coupled transmission line has very compact size with W/S/W = 35 µm/20

µm/35 µm. From the discussion in Chapter 3, it is required to have a minimal distance for

the electric separation of a single patch DMPA. The spacing between the line – 20 µm – is

Figure 5.9: Cross section of AiP with MMICs in eWLB package. The antenna is built in RDLlayer. c© 2012 IEEE [164].

98

much smaller than the required separation. It is necessary to develop other types of antenna

for eWLB AiP.

The first AiP in eWLB packaging designed in this work is a folded dipole antenna. The

reason for dipole-like antenna is that dipole antenna is suitable for tight coupled transmission

line. It matches the coupled transmission line in RDL. The classic dipole is a half wavelength

open-end transmission line. The typical antenna impedance is 73 Ohm without ground plane

below the antenna. In eWLB package, the ground-to-antenna distance is 180 µm, which is

about 0.05 λ0 at 79 GHz. Taking this effect into consideration, the dipole antenna impedance

is around 20 to 30 Ohm [165]. It is far from the 100 Ohm port impedance of MMICs. It

requires a bulky matching network for impedance matching. Therefore, a folded dipole antenna

is proposed for a compact solution. The principle of the folded dipole is shown in Figure 5.10.

Since the impedance of the folded dipole antenna is four times that of a dipole antenna (see

Equation 5.1 [166]), it is feasible to build AiP without any matching network.

Zfolded−dipole = 4× Zdipole (5.1)

The initial size of the folded dipole is calculated from half wavelength in the mold material.

Figure 5.11 shows the top view of the antenna as well as the simulation results. The final

optimized length of the folded dipole is 1.24 mm. The simulation results show that the folded

dipole antenna AiP has 6 GHz bandwidth for 10 dB return loss. It covers the full bandwidth

of automotive radar applications (76 GHz to 81 GHz).

5.2.3 Manufacturing and measurement

The designed AiP – folded dipole (FD) was manufactured and tested. Figure 5.12 shows the

manufactured AiP as well as the test board.

The folded dipole AiP was manufactured together with MMICs. Figure 5.12(a) shows

the bottom view of the manufactured folded dipole type AiP. Antenna is integrated with an

×18 frequency multiplier in a 6 mm by 6 mm eWLB packaging [167]. The MMIC includes a

Figure 5.10: Folded dipole and equivalent regular dipole.

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frequency multiplier [167], an amplifier and a balun. It generates the RF signal (76.5 GHz) by

multiplying the frequency of the input signal of LOin (4.25 GHz) by a factor of 18. The MMIC

has a differential topology and RF output interface. Figure 5.12(b) shows the test board with

mounted AiP. LOin and LOout signals are aligned on the right and left side of the board with

SMA connectors. The power supply signals is aligned on the bottom side of the boards.

The performance of AiP was measured with an active setup, which means antenna was

measured together with MMIC. Figure 5.13 shows the measurement configuration and photo

of the setup in the antenna chamber. The antenna under test is set as Tx antenna. The input

LO signal of the DUT was generated by an Agilent E8257D signal source. The RF signal was

transmitted by the AiP and received by an E-band standard horn antenna, which was placed

at a distance d = 1.8 m in front of the transmitter. The received signal was measured by

a spectrum analyzer (Rohde & Schwarz FSQ40) combined with a harmonic mixer (Rohde &

Schwarz FSZ90). The equivalent isotropic radiated power (EIRP) of the antenna is calculated

Figure 5.11: Simulated S11 of folded dipole AiP in eWLB packaging, LFD = 1.24 mm, linewidth = 0.035 mm, line spacing = 0.02 mm. c© 2012 IEEE [164].

(a) Photo of manufactured folded dipole AiP (bot-tom view).

(b) Photo of test board for folded dipole AiP (topview)

Figure 5.12: Photo of folded dipole AiP (bottom view) (a) and test board (top view) (b). c©2012 IEEE [164].

100

from the measurement as

PEIRP (dB) = Pr −Gr + LFS (5.2)

where Pr is the received power in the spectrum analyzer and Gr is the gain of the receive

antenna. In this measurement setup, Gr is 20 dBi. LFS is the propagation loss which can be

calculated from free space path loss equation:

LFS(dB) = 20log10(4πd

λ) (5.3)

The gain of AiP (Gt) can be further calculated as follows:

Gt = PEIRP − Pout (5.4)

where Pout is the measured output power of the MMICs.

The calculated gain of folded dipole AiP is about 7 dBi over a wide frequency range.

(a) AiP measurement configuration (b) Photo of measurement setup of folded dipole AiPin chamber.

Figure 5.13: Radiation pattern measurement of AiP configuration (a) and photo of chamber(b). c© 2012 IEEE [164].

The measured radiation patterns at 76.5 GHz are shown in Figure 5.14, together with the

simulated results for comparison. From the measurement of the radiation pattern, a notch for

beam pattern in H-plane has been found. This is mainly because of the package size and the

MMICs’ back metallization. More detailed discussions are shown in Section 5.5.1.

The first design of folded dipole AiP proves the concept for the eWLB AiP. From the next

section, different designs for bandwidth and radiation pattern improvement will be shown step

by step.

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(a) (b)

Figure 5.14: Measurement and simulated radiation pattern of folded dipole AiP at 76.5 GHz:(a) E-plane and (b) H-plane. c© 2012 IEEE [164].

5.3 Folded Dipole AiP with cavity in PCB

5.3.1 Antenna design

In this section, a new design for ultrawide bandwidth (BW% > 20%) is proposed. From the

dipole antenna theory, the antenna bandwidth is related to the distance from the antenna to the

ground layer (h). Larger values of h correspond to wider bandwidths. In the previous design,

the distance h is defined by package technology, which is typically about 200 µm. It is not easy

to modify the height of BGAs for increasing the distance, but it is possible to add a cavity in

the PCB beneath the AiP. In this way, the value of h can be increased. Figure 5.15 shows a

cross section for the proposed idea. The size of cavity is chosen as 1.55 mm by 3.1 mm, the

same size as the waveguide dimension WR12.

The radiating element is a folded dipole antenna which is inherited from the previous design.

The simulation model with different height of cavity was built in CST MWS. Figure 5.16(a)

shows a top view of the model. The antenna is aligned to the center of the cavity. The cavity

on PCB has a size of Lcavity by Wcavity. A ring-shaped BGA is placed around the cavity.

Figure 5.16(b) shows the antenna impedance in Smith Chart with a different height of h.

EWLB package is a superstrate structure. To have a comparison with the classic folded dipole

antenna, the models are built for both classic folded dipole antenna and superstrate structure.

Two effects are observed from the simulation results:

(1) The impedance loci of folded dipole antenna shrink to a smaller region when h in-

creases. It leads to a wider bandwidth but requires a matching network, for instance, a quarter-

wavelength transmission line with high characteristic impedance.

(2) Superstrate structure folded dipole antenna shows a very similar trend as classic folded

dipole antenna.

The final value of h is chosen to be 0.8 mm for a very broad bandwidth and optimal gain.

It is also more robust for manufacturing. The simulation results show that the bandwidth of

10 dB return loss increased from 6 GHz without cavity to 29 GHz after adding the cavity. It

brings large tolerance for manufacturing. A quarter-wavelength transmission line with high

characteristic impedance is used for the impedance matching. The length of the matching

network is 0.635 mm. A comparison of return loss for the folded dipole antenna with and

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without cavity is shown in Figure 5.17. The final size of folded dipole is 1.16 mm in length. It

is smaller than the folded dipole without cavity. Therefore, the gain of the antenna is slightly

degraded after adding the cavity.


A prototype of the antenna, as shown in Figure 5.18(a), was fabricated for testing. The whole

package comprises two parts: the MMIC at the lower part and the AiP at the upper part. The

MMIC is the same as in the previous design [167].

The antenna is placed in the fan-out area of the package and is surrounded by solder balls.

They constitute the shield walls for the antenna and the sustainer of the package on the PCB.

The antenna is composed of a radiating element – folded dipole and impedance matching

network. The differential feed line has the dimensions W/S/W = 35 µm/20 µm/35 µm with

100 Ohm differential characteristic impedance.

Figure 5.15: Cross section of folded dipole plus cavity in PCB. c© 2012 IEEE [168].

(a) Top view of folded dipole AiP with cavity struc-ture.

(b) Smith Chart of superstrate structure with differ-ent height.

Figure 5.16: Top view of folded dipole AiP with cavity simulation model (a) and simulatedantenna impedance (b).

103


The cavity on PCB was drilled by a milling machine and covered by conductive glue after-

wards. Figure 5.18(b) shows a photo of the cavity before adding the conductive glue. After

building the cavity, the AiP package was manually populated on the test board.

Measurement results

The measurement setup is similar as in Figure 5.13(a). The AiP test board is measured as

Tx antenna. Measured radiation patterns are shown in Figure 5.19. There are stronger ripples

for both E-/H-plane radiation pattern because of lower gain of the AiP FD with cavity. The

measurements match the simulation results very well, except for the notch in H-plane. Like in

the first design, the similar notch in H-plane around 30 degrees is observed. That is because

of the package size and MMICs effects. The HPBWs for E-/H-plane are 56 degrees and 120

degrees, respectively, neglecting the notch effects. The EIRP of AiP is calculated as Equation

5.2. Figure 5.20 shows the EIRP of AiP-FD with cavity over frequency range 76-81 GHz. The

Figure 5.17: Simulation model and simulated S11 of the AiP - FD with and without cavity.Folded dipole line width = 0.035 mm, Ld = 1.16 mm, LMN = 0.635 mm, and gap of matchnetwork = 0.06 mm. c© 2012 IEEE [168].

(a) Photo of the fabricated package of folded dipolewith cavity on PCB, bottom view.

(b) Photo of test board for folded dipole AiP withcavity in PCB.

Figure 5.18: Photo of the fabricated package of folded dipole with cavity on PCB (a) and testboard (b). c© 2012 IEEE [168].

104

average EIRP is 8.5 dBm within the frequency range. From Equation 5.4, the gain of AiP

folded dipole with cavity is calculated from EIRP. The gain of AiP-FD with cavity is 5 dBi,

which is smaller than FD without cavity, 7 dBi.

(a) (b)

Figure 5.19: Measurement and simulated radiation pattern of folded dipole AiP with cavity at76.5 GHz: (a) E-plane and (b) H-plane. c© 2012 IEEE [168].

5.4 Dual patch type AiP

In the section 5.3, the bandwidth of folded dipole type AiP has been greatly enlarged by adding

cavity. Meanwhile, it also increases PCB manufacturing cost by requiring a precise cavity. In

this section, a new type of AiP has been investigated. It supports very broad bandwidth, gain,

and moderated manufacturing complexity.

First take a close look at the eWLB structure. Figure 5.21 shows the cross sections of

eWLB package and superstrate structure. From the comparison, the eWLB is very comparable

with superstrate structure, which brings broad bandwidth and high gain for patch antenna

[156, 157, 169].

Figure 5.20: EIRP of AiP – folded dipole with cavity in PCB. c© 2012 IEEE [168].

105


(a) Cross section of AiP with eWLB package. (b) Cross section of superstrate structure.

Figure 5.21: Cross section comparison of AiP with (a) eWLB package and (b) superstratestructure. c© 2012 IEEE [170].

5.4.1 Antenna design

The proposed AiP has a dual patch configuration, sometimes also called a fat dipole antenna

[171]. Compared with the dipole antenna, the two arms of dipole are extended to two patches.

The differential signals are fed from the corner of the patches. Similar to a patch antenna, the

length of the patch dominates the resonant frequency, while the width of the patch helps he

bandwidth increment as well as impedance matching.

Figure 5.22 shows the simulation model of AiP-dual patch (DP) in CST MWS. The antenna

is placed in the middle of mold package. The height of mold (hm) and the height of BGA (hb)

are 0.5 mm and 0.18 mm, respectively, in the model. The distance from the antenna edge to the

mold package edge is Ldis. The total antenna length is LDP , which is equal to two times the

patch length (Lp) plus the gap (Sm). The gap between the patch is 20 µm, the same dimension

as for the coupled feeding lines. The patch width is WDP . The initial value of LDP and WDP

is selected as 1.0 and 0.25 of λr, where λr is the wavelength in the dielectric substrate. The

reason for narrow WDP is to suppress TM01 mode, etc.

Figure 5.23 shows the E-field distribution of the dual patch under differential injection

signal. The differential signal is composed of the positive signal at P+ and the negative signal

at P-. Therefore, its E-field distribution can be analyzed by superimposing the E-fields induced

by each signal. For instance, one of the patches (left one) is fed by a positive signal. With

proper selection of the patch dimension (LDP and WDP), the patch resonates in TM10 mode

at the desired frequency. Meanwhile, the other patch (right one) acts as a parasitic patch (the

other port is loaded), and it also resonates in TM10 mode. That means, the E-field on the

adjacent edges of the two patches has a similar amplitude but a 180-degree phase difference. It

is the same for the right patch fed by the negative signal. Therefore, the dual patches fed by

differential feed signals contribute the same TM10 mode on both patches. Simulation results

confirm the analysis.

E-field distribution of the patches also proves that the resonant frequency of the antenna

depends mainly on the patch length (LDP), while the antenna impedance can be adjusted by

the patch width (WDP). Shorter LDP corresponds to higher resonant frequency, and wider WDP

corresponds to lower resonant impedance. Figure 5.24 shows the antenna resonant frequency

(fr) and resonance impedance (Re(Z)) with different values of LDP and WDP. Here, fr is

defined as the frequency when Im(Z) is zero. Figure 5.24(a) plots the imaginary part of the

106

Figure 5.22: Simulation model of AiP – dual patch (DP), mold size of 3×3 mm2. c© 2013 IEEE[172].

Figure 5.23: E-field distribution of the AiP – DP for differential signal. c© 2012 IEEE [170].

antenna impedance Im(Z) for various patch lengths (LDP). The value of fr increases when

LDP decreases. Figure 5.24(b) shows the real part of the antenna impedance Re(Z) for various

patch widths WDP. A wider patch width corresponds to a lower Re(Z). The coupled feeding

lines have the dimensions width/space/width = 35/20/35 µm, with a differential characteristic

impedance of 100 Ohm.

Adjusting WDP, the antenna impedance was optimized to 100 Ohm. Wp is shorter than

Lp to suppress other resonant modes, such as TM10, etc. In other words, cross-polarization

radiation is suppressed. After optimizing the antenna parameters, an AiP antenna was designed

at 76.5 GHz with the dimensions: LDP/WDP= 0.78λr/0.24λr, where λr is the wavelength in

the dielectric substrate. The simulated S-parameter results show a 17 GHz bandwidth for 10

dB return loss (see Figure 5.25).

Since the AiP – dual patch size is bigger than AiP – folded dipole with cavity, the average

107


(a) (b)

Figure 5.24: Imaginary part of the AiP – DP impedance Im(Z) for different values of LDP,L1,2,3 = 1.64/1.72/1.80 mm (a) and real part of the antenna impedance Re(Z) for differentvalues of WDP, W1,2,3,4 = 0.3/0.4/0.5/0.6 mm (b). c© 2013 IEEE [172].

gain is improved from 5 dBi to 7 dBi.


A prototype of the AiP as shown in Figure 5.26 was fabricated for testing. The whole package

has a size of 6 × 6mm2 and comprises the MMIC at the lower part and the AiP at the upper

part. The MMIC include a frequency multiplier, an amplifier, and a balun. It generates the

RF signals (76.5 GHz) by multiplying the frequency of the input LOin signal (4.25 GHz) by 18

[167].

The antenna is placed in the fan-out area of the package and surrounded by solder balls.

They constitute the shield walls for the antenna and also the sustainer of the package on the

PCB.

The whole package was measured as a device under test (DUT) in an absorber chamber

Figure 5.25: Simulated return loss of the AiP DP, LDP=1.72 mm, WDP=0.5 mm. c© 2012 IEEE[170].

108

room. The setup for measuring the radiation pattern and the gain of the package is similar

like in 5.13(a). The DUT was configured as a transmitter. The input LO signal of the DUT

was generated by an Agilent E8257D signal source. The RF signal was transmitted by the AiP

and received by an E-band standard horn antenna, which was placed at a distance d=1.59 m

in front of the transmitter. The received signal was measured by a spectrum analyzer (Rohde

& Schwarz FSQ40) combined with a harmonic mixer (Rohde & Schwarz FSZ90).

Measurement results

First, the gain of the antennas in package is measured over the frequencies. The effective

isotropic radiated power (EIRP) of the package was calculated using the Friis Transmission

Equation. The results (see Figure 5.27, 5.28) indicate that the package supports 11 dBm EIRP

over a frequency range of 75 GHz to 80 GHz.

Pout, the output power of the MMIC, was measured on the wafer. From the difference

between Pout and PEIRP , the gain of the antenna can be estimated. The results show that the

AiP has about 7 dBi gain from 76 GHz to 81 GHz. This covers the whole band for automotive

short-range radar applications.

Second, the radiation pattern of the AiP was measured at 76.5 GHz. Figure 5.29 shows

the measured and the simulated co/cross-polarization radiation patterns in both the E-plane

and the H-plane and the simulation results. The simulated cross-polarization of H-plane is too

low to plot. Here, the simulation also includes effects such as those of the MMIC and of the

interconnection lines.

Measurements and simulations are in good agreement. The ripples in the E-plane are caused

by the diffraction effects of the PCB. The notch in H-plane around 30 degrees is caused by the

effects of the MMIC.

The decrease in gain is caused by the interconnection between MMIC and antenna and the

underestimation of the loss tangent of mold.

This section presents an AiP solution using eWLB technology for 79 GHz radar applica-

tions. The AiP has a superstrate structure with a dual-patch configuration. Since it is fed by

differential signals, it enables seamless integration with a differential MMIC for mmW applica-

tions. The AiP has a wide bandwidth and stable gain. The package material is a plastic mold,

which is cost-efficient, can be fabricated reliably, and is thus suitable for mass production. This

Figure 5.26: Bottom view of the AiP DP package. c© 2012 IEEE [170].

109


Figure 5.27: Power measurement of the AiP DP. c© 2012 IEEE [170].

Figure 5.28: EIRP of AiP DP. c© 2012 IEEE [170].

(a) E-plane (b) H-plane

Figure 5.29: Measured and simulated radiation patterns of the AiP DP at 76.5 GHz: (a) E-planeand (b) H-plane. c© 2012 IEEE [170].

110

solution also has potential in wide-area mmW applications.

5.5 Lens over eWLP AiP

5.5.1 Effects of package on radiation performance

From the previous AiP measurements, there is always a notch in the H-plane radiation pattern.

This largely deforms the AiP performance. The reason of such an effect is further studied by

analysing the package size. Since it is observed in all types of AiP, like folded dipole, dual

patch, and rhombic ring, etc., it is necessary to further analyze the package effects.

The package sizes in the previous simulation models were selected as 3×3 mm2 to save

simulation time. In the realized samples, the package sizes are 6×6 mm2. In addition, the AiP

is not placed in the center of the package when integrated with MMICs.

In this step, the effects of package size are analyzed. Figure 5.30 shows the radiation pattern

of the same AiP – DP with different package sizes – 3×3 mm2 vs 6×6 mm2. The simulation

shows that when the package has a small size, the AiP has a maximum gain at broadside angle.

When the package size increases, the radiation patterns in the broadside direction become a

notch. Further simulations show that when the antenna is not placed in the center of the

package, the notch in the radiation pattern may also shift from the broadside direction to a

certain angle. This is the main reason for the asymmetric pattern of the AiP.

(a) (b)

Figure 5.30: Radiation pattern of the primary antenna with a mold size of (a) 3×3 mm2 and(b) 6×6 mm2. c© 2013 IEEE [172].

This effect can be further explained by studying the displacement current of the mold. Figure

5.31 illustrates the displacement current of the mold of the same models as in Figure 5.30. The

displacement current of the 3×3 mm2 mold shows one dominant displacement current, while

in the 6×6 mm2 mold, there are three dominant displacement currents. The direction of the

two side currents opposes that of the center current. The radiation pattern is determined by

the sum of these three current sources. This explains the notch in the radiation pattern in the

H-plane. For the 3×3 mm2 mold, the distance from the edge of the patch to the edge of the

mold (Ldis) is smaller than λr, which is the wavelength in the mold. λr at 78 GHz is 2.15 mm.

For the 6×6 mm2 mold, Ldis is larger than λr, and the first-order mode of the side current

occurs. Further simulations show that when Ldis further increases, higher-order side currents

occur. The odd-number modes of the side currents are out of phase with the center current,

111


while the even-number modes of the side currents are in phase with the center current. To

achieve maximum gain in the broadside direction of the antenna, Ldis should be smaller than

λr to generate only center current or 2λr < Ldis < 3λr for even-number modes of the side

currents.

On top of the package size effects, the MMICs’ back metallization will further strengthen

such asymmetric behaviors.

To compensate the asymmetric radiation pattern, lens antennas are designed in this work.

There are many designs for compensating the asymmetric effect of AiP after integration. This

section gives solutions for lens antenna design. The lens antenna is designed as a dielectric lens

on top of previous AiP. Two lens have been designed in this work. The first is a hemisphere lens,

and the other is a rod lens. The following section gives details of lens design and verification of

the results.

(a) mold size of 3×3 mm2 (b) mold size of 6×6 mm2

Figure 5.31: Displacement currents of the molds with dimensions (a) 3×3 mm2 and (b) 6×6mm2. c© 2013 IEEE [172].

5.5.2 Hemisphere lens design

The dielectric material of the lens chosen was PEEK polymer material with εr,PEEK = 3.2 and

tanδ=0.003. The lens antenna is mounted directly on the eWLB package. A cavity with a size

of 6×6×0.5mm3 at the bottom of the lens antenna accommodates the eWLB package. The

primary antenna is placed in the geometric center of the lens antenna.

The first lens antenna is a hemisphere lens. Figure 5.32 shows the cross section of a hemi-

sphere lens on eWLB AiP. The principle of this design is to align the AiP from the edge of the

package back to the center of the lens. In addition, the lens will focus the antenna beam to

increase the gain. The size of the lens is selected according to the following rule:

2 × λr < Ldis = Rhemi − Wp / 2 < 3 × λr

Figure 5.33 shows the simulated radiation pattern of the AiP-FD with cavity with and

without hemisphere lens at 76.5 GHz. The total antenna gain is increased from 6.5 dBi to 11.5

dBi. It is worth to mention that there is no obvious influence for the S-parameters of the AiP

with different package size.

Measurement results

The manufactured lens as well as the AiP test board are shown in Figure 5.34. Two different

AiP – folded dipole with cavity and dual patch-are implemented here as primary antenna.

112

Figure 5.32: Cross section of hemisphere lens on eWLB AiP. c© 2012 IEEE [168].

Figure 5.33: Simulated radiation pattern of hemisphere lens on eWLB AiP – FD with cavity.c© 2012 IEEE [168].

Figure 5.35 show the measured and simulated radiation pattern of hemisphere lens on eWLB

AiP – FD with cavity. The notch in H-plane is greatly moderated.

After adding the lens on top of eWLB AiP, the total antenna gain mainly depends on the

lens performance. The measurement results also show that the EIRP of hemisphere lens +

AiP DP is similar like hemisphere lens + AiP FD with cavity, see Figure 5.36. The difference

between the two primary antennas are moderated.

5.5.3 Rod lens antenna design

The second lens antenna is a rod lens antenna. Figure 5.37 shows the cross section of the AiP

with rod lens. Given the success of hemisphere lens, further improvement can be achieved when

the hemisphere lens extends to a rod antenna. The value of Rhemi is same as hemisphere lens

– 5 mm.

113


Figure 5.34: Photo of eWLB AiP test board with hemisphere lens. c© 2012 IEEE [168]

(a) E-plane (b) H-plane

Figure 5.35: Measured and simulated radiation pattern of hemisphere lens on eWLB AiP – FDwith cavity: (a) E-plane and (b) H-plane. c© 2012 IEEE [168].

(a) (b)

Figure 5.36: EIRP of hemisphere lens on AiP: (a) folded dipole with cavity and (b) dual patch.c© 2012 IEEE [168].

114

The other parameter is the height of the rod lens, which influences the antenna gain. The

simulated gain of the AiP in the broadside direction for different values of hcly is shown in

Figure 5.38. In general, for a fixed mode behavior, a greater hcly has optimal gain at lower

frequencies, while a smaller hcly has optimal gain at higher frequencies (see Figure 5.38(a)).

Figure 5.38(b) presents the average gain and gain variation from 71 GHz to 82 GHz. Here, the

measured frequency points are from 71 GHz to 82 GHz, with 1 GHz step. The average gain

and gain variation are defined as follows:

Average Gain =1

12

82GHz∑fi=71GHz

Gfi (5.5)

Gain V ariation = max(Gfi)−min(Gfi) (5.6)

where Gfi is the gain at fi GHz frequency.

hcly is chosen as 5.5 mm since this results in optimal gain between 78 GHz and 80 GHz.

Furthermore, this yields a good trade-off between the average gain and the gain variation in

the frequency range from 71 GHz to 82 GHz.

Enhanced model simulation The reflection coefficient S11 of the enhanced model simulation

exhibits a similar behavior for a simplified model, as shown in Figure 5.39. In the enhanced

model simulation, the rod lens and the MMICs are also included.

The reflection coefficient S11 of the enhanced model simulation exhibits a similar behavior

for a simplified model, as shown in Fig. 5.39. In the enhanced model simulation, the rod lens

and the MMICs are also included.

Measurement results

The rod lens were fabricated and measured with AiP test board. Here, the AiP-DP test board

is used as primary antenna. Figure 5.40 shows the photo of the test board as well as the rod

lens.

Figure 5.37: Cross section of the eWLB package and the rod antenna. c© 2013 IEEE [172].

115


(a) Simulated gain of the AiP for different heightshcly of the rod lens at single frequency.

(b) The average (avg.) gain and gain variation (var.)between 71 GHz and 82 GHz.

Figure 5.38: Simulated gain of the AiP for different heights hcly of the rod lens at singlefrequency (a) and the average (avg.) gain and gain variation (var.) between 71 GHz and 82GHz (b). c© 2013 IEEE [172].

Figure 5.39: Simulation of S11 with the enhanced model. The AiP and the rod lens have thedimensions Lp=1.72 mm, Wp=0.52 mm, Rhemi=5 mm, Lm=6 mm, hcly=5.5 mm, hm=0.5 mm,hb=0.2 mm, and the MMIC is 2×2×0.45 mm3. c© 2013 IEEE [172].

The measurement configuration is similar as in Figure 5.13(a). The AiP was measured as

a device under test (DUT). The LOin signal was generated by an Agilent signal source (Ag.

E8257D). The RF signal was transmitted by the AiP and received by an E-band standard 20

dBi gain horn antenna. The distance between DUT and horn antenna (d) was 1.6 m. The

received signal was measured by a signal analyzer (R&S FSQ40) combined with a harmonic

mixer (R&S FSZ90). Figure 5.41 shows photographs of the measurement setup in the absorber

chamber and of the AiP with lens mounted.

First, the gain of the antenna in package was measured over the frequency. The test board

was measured both with and without rod lens. The effective isotropic radiated power (EIRP)

of the package was calculated using the Friis Transmission Equation. The measured EIRP of

the AiP with lens is over 16 dBm from 71.4 GHz to 81.7 GHz (see Figure 5.42). The maximum

EIRP is 18.5 dBm at 79.2 GHz. The power output of the chip is also plotted for comparison.

116

Figure 5.40: Photo of AiP-DP test PCB with rod lens. c© 2013 IEEE [172].

Figure 5.41: Photographs of the measurement setup in the absorber chamber and the AiP withlens mounted. c© 2013 IEEE [172].

The gain of the antenna with and without lens is shown in Figure 5.43. The measured

gain of the AiP with lens is greater than 12 dBi from 71 GHz to 82 GHz and thus, except

around 79 GHz, slightly lower than in the simulation. The difference between measurement

and simulation may be because of the assembly tolerances and underestimation of the loss

tangent of the material. The lens greatly improves the gain of the AiP. On average, it increases

the antenna gain from 5.9 dBi to 13.7 dBi. This implies twice the dynamic range in radar

applications. The gain variation is 2 dB.

The antenna radiation patterns are measured also at different frequencies. The measured

radiation patterns at 78.3 GHz are shown in Figure 5.44.

The radiation patterns exhibit symmetric behavior in both the E-plane and the H-plane.

The side lobe levels in the E-plane and the H-plane are -16 dB and -13 dB, respectively. The

cross-polarization (X-pol) is 18 dB lower than the co-polarization (Co-pol) in the E-plane and

117


Figure 5.42: Measured EIRP of AiP with (wt) and without (wo) lens and power output of thechip. c© 2013 IEEE [172].

Figure 5.43: Simulated and measured gain of AiP with (wt) and without (wo) lens. c© 2013IEEE [172].

20 dB lower in the H-plane. The simulation X-pol in the H-plane is too small to plot. The

measurements and the simulation are in good agreement.

Without the lens, a big notch around 30 appears in the H-plane. It is caused by the mold

size and conductivity of MMIC. Adding the lens eliminates this notch. The 3 dB beamwidths

of the E-plane (Θ3dB,E) and the H-plane (Θ3dB,H) are 24.5 and 17, respectively. Table 5.1

shows a comparison of the rod lens AiP with hemisphere lens AiP [168]. The rod lens AiP

achieves a much narrower beamwidth. Some other types of AiP solution are also listed in Table

5.1. The beamwidth of rod lens antenna AiP is comparable to that of various types of array

antennas.

Figures 5.45 and 5.46 plot the measured radiation patterns at 72 GHz and 81 GHz, respec-

tively. They exhibit behaviors similar to that at 78.3 GHz. Thus, we verified that the antenna

118

(a) Measured and simulated E-plane radiation pat-tern at 78.3 GHz.

(b) Measured and simulated H-plane radiation pat-tern at 78.3 GHz.

Figure 5.44: Measured and simulated radiation pattern of AiP rod lens at 78.3 GHz: (a) E-planeand (b) H-plane. c© 2013 IEEE [172].

(a) Measured and simulated E-plane radiation pat-tern at 72 GHz.

(b) Measured and simulated H-plane radiation pat-tern at 72 GHz.

Figure 5.45: Measured and simulated radiation pattern of AiP rod lens at 72 GHz: (a) E-planeand (b) H-plane. c© 2013 IEEE [172].

(a) Measured and simulated E-plane radiation pat-tern at 81 GHz.

(b) Measured and simulated H-plane radiation pat-tern at 81 GHz.

Figure 5.46: Measured and simulated radiation pattern of AiP rod lens at 81 GHz: (a) E-planeand (b) H-plane. c© 2013 IEEE [172].

119


Table 5.1: Half power beam width comparison

Θ3dB,E Θ3dB,H freq. antenna type

this work 24.5 17.0 78.3 GHz patch+rod lens[168] 38 53 76.5 GHz dipole+hemisphere lens[173] 20 20 60.0 GHz 4×4 patch array[153] 20 20 60.5 GHz 4×15 grid array

radiation pattern is stable over a wide frequency range.

This section presented lens antenna for AiP performance enhancement. Two lens – hemi-

sphere and rod lens –are designed and verified. For the rod lens antenna, measurement results

show that the proposed antenna has more than 12 dBi gain in the frequency range from 71 GHz

to 82 GHz. The measured 3 dB beamwidth at 78.3 GHz is 24.5 degrees for the E-plane and 17

degrees for the H-plane. The dimensions of the whole antenna are 10 × 10 × 10.5 mm3.

5.6 Summary

This chapter demonstrates different designs for antenna in package (AiP). A couple of antennas,

such as folded dipole (FD), folded dipole with cavity, and dual patch (DP), are realized in eWLB

packaging technology. The test antennas are integrated with MMICs in the package. Because

of the package size effects and MMICs, the radiation pattern of AiP in H-plane has a notch

around 30 degrees. To enhance the radiation performance, two dielectric lens are designed to

accommodate the AiP – hemisphere lens and rod lens. The optimal combination of performance

and manufacturing complexity is rod lens plus the AiP – dual patch. It supports more than 12

dBi gain over a wide frequency range (71 GHz to 82 GHz).

120

Chapter 6

Conclusions and Future Topics

6.1 Conclusions

The aim of this work is to design, fabricate, and characterize differential feed antenna for mmW

radar applications. It first gives a background introduction of FMCW and MIMO principles in

radar system and the simulation tool used in the study – CST MWS. In Chapter 2, a broad

introduction of prior art works of mmW antenna is given. Many antennas are discussed in

groups like planar structure antennas, waveguide antennas, reflector antennas, high integration

antennas, etc.

Chapters 3 to 5 forms the core of the study, in which three different concepts of differential

antenna have been presented. The first presented concept is to build a differential antenna based

on a microstrip structure. The microstrip structure is the most popular layer stack in mmW

radar systems. In this part, a rectangular patch is selected as the radiating element. During

the study, the classic cavity model for microstrip patch antenna is extended for differential feed

model. It gives a numerical calculation for the impedance matching feed of differential feed

rectangular patch antenna. After the success of single patch differential antenna, two different

antenna arrays – H-plane array and E-plane array – are further developed. Antenna array

largely extends the antenna performance and increases the gain of the antenna. Among those

two types of array, the H-plane array provides wide bandwidth, while E-plane array provides

better radiation performance – lower side lobe and stable pattern over frequencies. A wide

bandwidth transition has been designed for the radiation pattern measurement needed.

The second presented concept is to implement differential antenna as radiation elements in a

transition design. It is also based on a microstrip structure layer stack. Four types of transitions

are designed in E/W-band. The main differences are the various differential feed patch antennas.

The first transition implements a single patch DMPA inside a WR12 waveguide. The second

transition uses a gap-coupled DMPA to increase the bandwidth. The third transition is for W-

band. It uses a short-ended gap-coupled patch to suppress the common signal in W-band. The

fourth transition is improved from the second transition. The ground plane inside the waveguide

is extended to increase the coupling effects between the patches and the ground. Therefore,

the bandwidth of the transition is further increased. The mechanical housing of the transition

is a modified waveguide with a channel cut from the narrow wall side. Later on, the housing

is simplified to a 1 mm thick metal shim, which is much easier for manufacturing. Those

transitions bring smooth connection from the planar structure (coupled microstrip lines) to

vertical structures (rectangular waveguide structures). A special measurement method – LRdR

method – has been developed for the transition characterization. Such types of transitions bring

121

6. CONCLUSIONS AND FUTURE TOPICS

a wide facility for waveguide antenna – like horn antenna – integration in the radar front-end

systems. Nowadays, 3-D printed technology attracts more and more attention in mmW antenna

designs [174, 175]. It enables highly accurate models and fast prototype. Those transitions show

high potentials for many types of high-performance radar applications.

The last presented concept is differential feed antenna in eWLB package. Antenna in package

has much higher integration levels compared with antenna built in printed circuit boards. The

eWLB package is a promising package solution for mmW applications. It allows novel designs for

the antenna-in-package development. Since the coupled line in package has very narrow spacing

(20µm), single patch is not suitable for the radiating element. Dipole (wire antenna) and dual

patch structure have been proven to be good candidates for antenna-in-package design. Three

types of differential feed antenna in package are designed in eWLB package. To improve the

radiation performance, two dielectric lens – hemisphere lens and rod lens – which are mounted

on top of the package are also developed and verified.

As a final conclusion, differential feed antennas in either microstrip structure or transition

design and antenna-in-package design present good performance and high level of integration.

They are a good candidate for mmW front-end system and contribute to improvements in the

overall system performance.

6.2 Future topics

The presented concepts show very promising results both theoretically and in practical measure-

ment examples. Nevertheless, based on the results from this work, there is still some potential

for possible future research topics, especially now that autonomous driving is becoming more

and more interesting for car development.

• Since the 5 GHz bandwidth (76 – 81 GHz) is becoming the worldwide accepted regula-

tion, a stable radiation pattern within wide bandwidth is required for the future antenna

development. For this purpose, a multilayer of RF substrate as well as back feeding are

possible solutions. A multilayer structure benefits antenna bandwidth, and the back feed-

ing supports a stable radiation pattern. In this solution, a sophisticated and low-cost

manufacturing is the key for mass production success.

• Another trend for the mmW radar is high IO integration in the package. Since more and

more mmW radar applications are developed and digital beamforming is popularly used

in all mmW radar systems, there is a high demand for a high number of RF IO channels

as well as antenna in package from the market [176]. Such high-integration radar can be

used for gesture detection, door open alarm, etc. The high number of Tx and Rx channels

with antenna integrated in the package enables a very compact size of the radar systems

with good angle resolution performance. Further improvement of the package technology

has great potential for the future antenna/system development.

122

Appendix A

.1 Dipole antenna resistance

The radiation resistance of a infinitesimal dipole (l ≤ λ/50) is:

Rr = η(2π

3)(l

λ)2 (.1.1)

= 80π2(l

λ)2 (.1.2)

where η is 120π.

The radiation resistance of a small dipole (λ/50 < l ≤ λ/10) is:

Rr = 20π2(l

λ)2 (.1.3)

The real and imaginary parts of the antenna impedance for a finite length dipole are:

Rr =η

2πC + ln(kl)− Ci(kl) +

1

2sin(kl)[Si(2kl)− 2Si(kl)]

+1

2cos(kl)[C + ln(

kl

2) + Ci(2kl)− 2Ci(kl)] (.1.4)

Xr =η

4π2Si(kl) + cos(kl)[2Si(kl)− Si(2kl)]

− sin(kl)[2Ci(kl)− Ci(2kl)− Ci(2ka2

l)] (.1.5)

where Si(x) and Ci(x) are the sine and cosine integrals. C is Euler constant 0.5772.

When l = λ/2, Rr is 73 Ohm and Xr is 42 Ohm.

123

. APPENDIX A

.2 Horizontal electric dipole

Rr = ηπ( lλ

)2[2

3−

sin(2kh)

2kh−

cos(2kh)

(2kh)2+

sin(2kh)

(2kh)3

](.2.1)

124

Acronyms

ACC : autonomous cruise control

AiP : antenna in package

AoC : antenna on chip

B2B : back-to-back

BGA : ball grid array

BW : bandwidth

CFL : Courant-Friedrichs-Levy

CM : common mode

cmW : centimeter wave

CSRR : complementary split ring resonator

CST MWS : computer simulation technology microwave studio

CW : continuous-wave

DBF : digital beam forming

DF : differential

DMPA : differential feed microstrip patch antenna

DMSLs : differential microstrip lines

DP : dual patch

DUT : device under test

EM wave : electromagnetic wave

eWLB : embedded wafer level ball grid array

EIRP : equivalent isotropically radiated power

FD : folded dipole

FFT : fast fourier transformation

FIT : finite integration technique

FMCW : frequency-modulated continuous-wave

FoV : field of view

GC : gap coupled

GSG : Ground-Signal-Ground

HPBW : half power beam width

IC : integrated circuit

IEEE : Institute of Electrical and Electronics Engineers

IF : intermediate frequency

IO : input-output

LCP : liquid-crystal polymers

LRdR: load, reect, delayed reect

LO : local oscillator

125

. ACRONYMS

LTCC : low temperature co-fired ceramics

LWG : laminated waveguide

MGEs : Maxwell’s Grid Equation

mmW : millimeter wave Radar

MIMO : multiple-input and multiple-output

MMIC : monolithic microwave integrated circuits

MN : matching network

MPA : microstrip patch antenna

MRR : middle range Radar

MSL : microstrip line

PA : power amplifier

PCB : printed circuit board

QFN : quad flat no-leads

Radar : RAdio Detection And Ranging

RF : radio frequency

RDL : redistribution layer

Rx : receiving/receiver

SMPA : single-ended microstrip patch antenna

SIW : substrate integrated waveguide

SLL : side lobe level

TEM : transverse electromagnetic

Tx : transmitting/transmitter

TRx : tranceiver

VCO : voltage-controlled oscillator

VNA : Vector Network Analyzer

WG : waveguide

WLP : wafer level packaging

126

References

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[2] M. Skolnik, Radar Handbook. McGraw Hill, 1970. 2

[3] E. Fishler, A. Haimovich, R. Blum, D. Chizhik, L. Cimini, and R. Valenzuela, “MIMO

Radar: An Idea Whose Time Has Come,” in Proceedings of the IEEE Radar Conference,

(Philadelphia, PA, USA), pp. 71–78, Apr. 26-29 2004. 2

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Di erential feed antenna in millimeter wave Radar

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