Magnetic switching techniques for high power pulse generation Keet, A.L. DOI: 10.6100/IR366609 Published: 01/01/1992 Document Version Publisher’s PDF, also known as Version of Record (includes final page, issue and volume numbers) Please check the document version of this publication: • A submitted manuscript is the author's version of the article upon submission and before peer-review. There can be important differences between the submitted version and the official published version of record. People interested in the research are advised to contact the author for the final version of the publication, or visit the DOI to the publisher's website. • The final author version and the galley proof are versions of the publication after peer review. • The final published version features the final layout of the paper including the volume, issue and page numbers. Link to publication General rights Copyright and moral rights for the publications made accessible in the public portal are retained by the authors and/or other copyright owners and it is a condition of accessing publications that users recognise and abide by the legal requirements associated with these rights. • Users may download and print one copy of any publication from the public portal for the purpose of private study or research. • You may not further distribute the material or use it for any profit-making activity or commercial gain • You may freely distribute the URL identifying the publication in the public portal ? Take down policy If you believe that this document breaches copyright please contact us providing details, and we will remove access to the work immediately and investigate your claim. Download date: 29. Jul. 2018
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Magnetic switching techniques for high power pulsegenerationKeet, A.L.
DOI:10.6100/IR366609
Published: 01/01/1992
Document VersionPublisher’s PDF, also known as Version of Record (includes final page, issue and volume numbers)
Please check the document version of this publication:
• A submitted manuscript is the author's version of the article upon submission and before peer-review. There can be important differencesbetween the submitted version and the official published version of record. People interested in the research are advised to contact theauthor for the final version of the publication, or visit the DOI to the publisher's website.• The final author version and the galley proof are versions of the publication after peer review.• The final published version features the final layout of the paper including the volume, issue and page numbers.
Link to publication
General rightsCopyright and moral rights for the publications made accessible in the public portal are retained by the authors and/or other copyright ownersand it is a condition of accessing publications that users recognise and abide by the legal requirements associated with these rights.
• Users may download and print one copy of any publication from the public portal for the purpose of private study or research. • You may not further distribute the material or use it for any profit-making activity or commercial gain • You may freely distribute the URL identifying the publication in the public portal ?
Take down policyIf you believe that this document breaches copyright please contact us providing details, and we will remove access to the work immediatelyand investigate your claim.
It is clear that for high-speed applications very thin copper strips (or high frequency litze)
and thin magnetic strips must be used. Although copper losses are not as high as the iron
losses it is still important to use the correct wire and copper cross-sections. Expression
11.18 gives the penetration depth in a half space; for round wires "Kaden" [11] gives the
calculation for the resistance (approximated by equations II.19 and II.20).
[ ro 1 3 ö J
RDC 2 ö + 4 + 32-;: 0
r for: ~>1.5
ö
(II.l9)
(11.20)
Measurements were carried out on a wire with a total copper cross-section of 2.5 mm2
(diameter 2 mm). The conductor was constructed of 48 wires of 0.26 mm <P pure copper.
The insulation round the conductor is silicon rubber and the total diameter was 9 mm </1.
To measure the high frequency resistance, an experiment was performed to measure the
temperature rise of a wire during the flow of high-frequency current. Later a DC-current
was used to heat the wire to the same temperature. In this way the HF-resistance could be
calculated.
The experiment was performed for two different frequencies, sec table II.5. This table also
presents the calculated penetration depth and the high-frequency resistance calculated by
the expressions II.l9 and 20.
measured calculated
T~ 1eff T 1oc ~F ö RHF RHF --- --- ---RDC RDC Roe
[J.LS} [A} (oC} [A} [mm} solid litze
1.2 26 40 38 2.1 0.10 5.26 1.06
I 0.8 27 71 48 3.2 0.083 6.14 1. 09
Table ll.5 Measured and calculated high-frequency resistance.
41
The experimentally observed HF-resistance is better described by the litze wire than by the
solid wire. In real litze wire the separate wires change position cyclically and are insulated
from each other. In our wire both conditions are partially met; the separate wires are
twisted and are "insulated" by the smaller contact area and the copper-oxide formed.
11.3.2 Losses in different types of magnetic materials.
The last years a lot of work has been done on losses in ferromagnetic tapes [4, 5]. The
traditional saturation wave theory divides losses into hysteresis- and eddy--current losses.
The hysteresis losses are proportional to the frequency, they can be found from the area
enclosed by the quasi-DC loop. The eddy- current losses are proportional to the frequency
squared. For very thin magnetic ribbons more sophisticated models have been developed.
These models take into account the "sandwich" and "bar" domain theory. For more details
on this theory we refer to the literature [4, 5].
For strip wound cores a voltage is induced between the strips by the changing magnetic
field. For high dB/dt these voltages are high enough to cause a voltage breakdown in the
insulating material. Therefore, a larger eddy-current will flow, and losses will be much
higher than expected. The voltage between the laminations is:
V ~h(d +d.)dB 2 'je 'dt
(IT.21)
where dfe is the thickness of the magnetic strip, di is the thickness of the insulation foil
and h is the width of the magnetic strip. For a magnetic strip of 25 #'m thick, 25 mm wide
and an insulation foil of 12 #'m and 1 T/#'s the voltage is 0.463 Volt. For higher
magnetization speeds the voltage will rise and more interlaminar insulation is needed. For
low frequencies it may be possible to rely on the roughness and the oxide layer of the
material. To reduce losses and to be sure that the losses will not increase in time an extra
coating or foil is recommended.
A coating can hold only a few volts (1-2 V) and if there is one voltage breakdown the
voltage across the insulation will double elsewhere. If a voltage breakdown is a permanent
one, the losses increases. The advantage of a coating is the better utilization of space
which can be achieved. Although the insulation foil can hold high voltages, the
42
disadvantage is that the stacking factor may drop to less than 50%. Another disadvantage
of the foil insulation is that the annealing process requires high temperatures and the
insulation foil can not withstand these.
The influence of resistive elements in a magnetic pulse compression network is discussed
in chapter III.4. The resistive elements there, represent the copper and the iron losses.
During the tests done on the larger high-repetition rate C02 laser power supply, the losses
of the different magnetic switches were measured. Although these values do not give a
complete picture of the losses, they give an impression of their importance. The losses
were obtained from measurements of the temperature rise of the core and additional
calorimetrie measurements. In table 11.6 the test results are shown.
magnet ie unsat. 2*8 dB/dt E/pulse material time s
[J.LS] [T] [T/J.LS] [J/m3]
Amorph 7505-Z 5.0 3.6 0.7 1000
Perrite T22 0.7 0.7 1.0 220
Perrite 3C8 1.2 0.7 0.6 580
Perrite 3C8 0.5 0.7 1.4 1370
Table Il. 6 Measured lossesper pulse of three types of magnetic material and different
magnetizing speed.
Losses in magnetic material can also be estimated by the enclosed area of the B-H loop; in
section II.2.2 these loops are presented for amorphous magnetic materials. For the 7505-Z
material and for the magnetizing time of 5p.s the area is about 300 [A/m] * 3.9 [T] = 1170 [J!m\ so the used method seems to be all right.
From the table it is clear that the losses of the old type of ferrite 3C8 are still smaller than
the losses of the amorphous magnetic material 7505-Z. The new type ferrite T22 has the
lowest losses.
43
11.4 Transformers for pulse applications.
Pulse transfarmers are essential for high-power pulse applications. Consirlering that the
required output voltage can be over 30 kV and the average forward blocking voltage of the
thyristor is about 2000 Volts, a gap has to be bridged. Sametimes thyristors are stacked to
achieve a high blocking voltage, but this is very expensive and not suitable for short pulse
generation. So mostly pulse transfarmers are used.
For the medium- and high-power pulse applications, two types of transfarmers can be
distinguished:
A The fast pulse transfarmers with a pulse duration larger than 10 p.s,
B The very fast pulse transfarmers for pulse durations shorter than 10 p.s. These are
applied in series and parallel magnetic pulse compression networks.
In this section only transformers with a magnetic core are discussed. The choice of the
magnetic material depends on the magnetization time. The required cross-section of the
core is calculated with equation ll.3; it depends on the application whether the core may
or may not saturate at the end of the pulse. In figure II.19 a simple equivalent circuit of a
transfarmer is given.
-------Rs : ideol :
11
~----~----~----------~~ Tl ~---0 : 1 : N : •------·
Fig. 11.19 The equivalent circuit of a transformer.
The magnetizing inductance Lm of the transfarmer is equal to the unsaturated inductance
of the magnetic switch. The magnetizing current can be seen as a constant current Imc and
the current through the inductor Lm.
The most difficult part in transfarmer design is the leakage inductance. The leakage
inductances Lop and Lus as presenled in figure 11.19 form the totalleakage inductance. It
is only possible to calculate the total leakage inductance, whlch we may refer to the
primary or the secondary side.
44
For transformers with the ooils forming a cylinder, it is possible to retrieve an expression
for the totalleakage inductance [7].
La 2 lg at+a2
+ dair) IJ. N -(--0 t b 3
(1
(11.22)
and
d. +a +a b b (l _ a1r 1 2)
a nb (11.23)
Wherein:
b Height of the coil [m]
lg Average length of the coil [m]
al thickness of the primary coil [m]
~ thickness of the seoondary coil [m]
dair thickness of the space between the coils [m]
Nt Number of turns of the coil, to which side the
inductance is referred to.
This equation is only valid for transfarmers with a magnetic core, and has proven to be
adequate for different types of core materials. Therefore it is possible to calculate the
leakage inductance for the transformer mentioned under "A".
Very fast pulse transformers (B).
In most applications, the magnitude of the magnetizing current is not so important, as the
main current is often several kiloamps. The leak.age inductance, however, is the important
factor and must be kept as small as possible. Because of the different geometrical
constructions it is difficult to calculate the leakage inductance. No study was done to
retrieve an expression.
To obtain low leakage inductances, it is possible to design the coils of the transformer
such that the secondary coil is completely surrounded by the primary. Depending on the
application the transfarmer can also be used as a magnetic switch. In that case the ratio
LiLs is important. Note that this ratio can be different for the primary and secondary
coil.
45
Experimental results.
As an example we describe a transformer constructed for a high-repetition rate co2 laser;
a saturable transfarmer for parallel compression applications. Therefore, the transformer
must have a low leakage inductance and the secondary side must have a low saturated
inductance.
The transformer specifications are:
Primary voltage 4000 V
secondary voltage 36000 V
Turn-ratio
Pulse duration
Primary current
Primary RMS current
leakage inductance
sec. sat. indoetanee
1:9
5 p.S
4.5 kA
140 A
=100 nH
5 p.H
Repetition-rate 400 Hz
To reduce the amount of magnetic material amorphous magnetic material
(Vacuumschmelze 7505-Z) is chosen. The magnetic ribbons are insulated by mylar and the
core dimeosion is 139 * 80 * 25 mm3. To obtain a sufficient amount of magnetic cross
section, six cores were used.
The transfarmer is used in a circuit of parallel magnetic pulse compression networks and
the secondary winding of the transfarmer is constructed to have a low saturable inductance
(two coils of nine turns in parallel). The primary winding is made of eighteen single turn
coils; this implies that there is a primary turn (coil) between each secondary turn. The
transfarmer is immersed in oil to obtain maximum cooling and to avoid corona between
the two coils.
The leakage indoetanee is deterrnined by short circuiting the primary side and by a
measurement of the inductance seen at the secondary side (L sec). Two tests were made; (J,
A) with all eighteen primary coils short circuited and B) with only nine primary coils short
circuited (equally distributed).
The measured leakage inductances are:
A: L sec = 7.0 p.H (J, •
B: Lu,sec. = 9.1 p.H If the leakage inductances were referred to the primary si de (N t = 1 instead of 9) the
values would be 86 and 112 nH, respectively.
These measurements show that it is possible to construct a high-voltage pulse transfarmer
with a very low leakage inductance.
46
11.5 Design of magnetic switches and transfarmers for different
magnetic pulse compression circuits.
In the area of medium to high pulse power it is possible to specify three different types of
magnetic switches and two different types of transformers for magnetic pulse compression:
A Magnetic switches for series pulse compression,
B Magnetic switches to combine the series and parallel pulse compression circuits,
C Magnetic switches to proteet thyristors during the start-up,
D A non-saturating pulse transformer,
E A saturating pulse transfarmer for parallel pulse compression.
A: Magnetic switches for series magnetic pulse compression.
The theory of series magnetic pulse compression will be discussed in section III.2. The
basic network is shown in tigure 11.20.
Fig.I/.20 Series magnetic pulse compression network.
To achieve high performance of the compression circuit, the switch must meet three
properties:
1 saturated inductance of the magnetic switch (L8),
2 The magnetic switch must saturate at the right moment.
3 Precharging of the next capacitor must be prevented,
The Low inductance during the saturated state of the magnetic switch (Ls> must be
obtained to achieve a fast energy transfer and a high-compression ratio. The number of
tums of the coil is important for the saturated inductance (L8), therefore this parameter
must be chosen with great care (see section II.2.1). For magnetic switches with a long
47
saturation time 10 JI.S) more turns can be used. For short saturation times only a few
turns can be allowed but more parallel windings can be used. By minimizing the stacking
factor of the core itself and the required amount of electrical insulation the saturated
inductance can also be minimized.
The magnetic switch must saturate at the right moment. The required volt-second integral
is calculated from equation II.3. The magnetic material chosen gives the maximum flux
density. Therefore the total magnetic cross section is fixed.
Precharging of the next capacitor is prevented if we have:
- HiJih inductance Lu during the unsaturated state of the magnetic switch
(Lu ""'> 400 * L~, LQ.w: magnetizing current (Im) during the unsaturated period of the magnetic switch.
The average magnetic field path is often determined by the necessary core window.
A special type of magnetic switch is the one needed in the MIRVOC system (which will
be discussed in chapter IV), because the required saturated inductance (Ls) must be less
than ""'20 nH. This cannot be achieved with ordinary magnetic matenals and coils of
insulated wire.
For this application a special type of magnetic switch was designed. Also two magnetic
switches in parallel were used. Many types of low-frequency magnetic matenals were tried
(3C8, N27 and 1'22) but the saturated permeability was too high to obtain a low
inductance. Also the losses were too high to obtain the necessary voltage gain. The micro
wave ferrite (type K4 from SEI) gave better results, although the maximum flux density
swing is small (± 0.5 T).
As the required inductance was very small, we used copper strips in combination with two
return strips to reduce the inductance (Ls). Figure II.21 shows the construction of this
magnetic switch with bas two turns.
Before we choose the two turn magnetic switch we compared it with a single turn
magnetic switch. To meet the required low inductance two magnetic switches were placed
in parallel in both cases. To keep the volt-second integral the same the two turn magnetic
switch was given half the core area. It turned out that the inductance of the two-turn
model was g than twice the inductance of the one turn model. The reason for the small
increase of the inductance is probably caused by the relatively large inductance of the
connecting strips. Since the magnetic switches are only one part of the total circuit, the
total inductance of the circuit is formed by the connections between pulse forming network
and magnetic switch, the magnetic switch itself, the conneetion between the magnetic
switch and the laserhead and the laserhead itself.
48
(b)
Return strips (
Return strips /
Fig.JI.21 Construction of a two turn magnetic switch with capper strip turns,
a: front view, b: side view.
As the design of these magnetic switches are essentially different from the magnetic
switches discussed in section 11.2.1, it is not possible to give a calculated value of these
switches.
Although the two-turn model has a larger inductance than the single-turn model, the
advantage of the lower magnetizing current and less losses were more important.
Operating tests with the laser showed that the inductance was sufficiently low as the laser
operated very well.
It can be concluded that it is possible to design magnetic switches with extremely low
inductance (about 30 nH) and that it is not necessary to use a single turn magnetic switch.
It is often better to use the two-turn model to reduce the losses.
49
B: Magnetic switches to combine the series and parallel pulse
compression circuits.
The combination of series and parallel magnetic pulse compression is discussed in section
III.7. Figure II.22 presents a circuit which combines the parallel and the series
compression network in a proper way.
Fig./I. 22 The rombination of parailel and series magnetic putse rompression network.
The magnetic switches L:z and L3 should be specified differently.
The magnetic switch L3 can be designed as an ordinary series magnetic switch. Due to the
fact that the volt-second integral is very small the switch will have also small dimensions.
Therefore the saturated inductance is also smalt in comparison to the inductance of the
total circuit.
The magnetic switch L2 can also be designed as a series magnetic switch but the
properties are not so critica!. Especially the inductance during the saturated state of the
magnetic switch (Ls) is not so important as the inductor is situated on the secondary side
of the transformer; this means that the inductance will be reduced by the square of the
tums-ratio of the pulse transformer. Therefore it is best to choose for a small magnetic
cross-section (to reduce losses) and a large number of turns; however the inductance in the
saturated state must not become too large.
This magnetic switch L:z must be in the saturated state at the beginning of the energy
transfer from C 1 to c2. Because it is difficult to achieve this, it is best to use magnetic
material with a very rectangular B-H loop and without any air gap. It is possible to use
two U-cores, but then the contact surfaces mu~t be polisbed to reduce the air gap.
The required volt-second integral is also not critica! as long as the core does not resaturate
at the end of the energy transfer from c2 to c3.
50
C: Magnetic switches to proteet thyristors during the start-up.
The demands for the magnetic switch for the proteetion of a thyristor can only be given
after an explanation of the background. Although thyristors are not limited in life time, if
correctly used, they are limited in their blocking voltage (2000 - 4500 V) and in the
maximum permissible rate of change of the current (400- 1000 A/Jl.s).
For high-power pulse applications it is necessary to transfer energy in a short period. The
dl/dt ratings of the thyristors are still too small to be used for this application.
Unfortunately the forward blocking voltage cannot be changed, but the maximum dlldt is
somewhat flexible. Thyristors are capable to withstand very high di/dt values once they
are well in the conductive state. So the idea is to take time to create a conductive path in
the thyristor which is large enough to withstand the high dl/dt, which occurs later [6].
This conductive path can be created by bringing the thyristor in the conductive state before
the main current is allowed to flow. During this time only a small current flows which is
large enough to create the conductive path but still small enough not to destroy the
thyristor.
To achieve this, two aspects should be considered. Firstly, a strong gate pul se has to be
applied to the thyristor and secondly the main current has to be blocked for a short time.
To create a large conductive path around the gate of the thyristor, the gate pulse must be
very strong, typical values are Ig = 15 A with a dl/dt of about 30 A/f.l.S. The blocking of
the current can be achieved by using a magnetic switch in series with the thyristor (figure
II.23a). In this figure also the snubber circuit (R1, c2) is shown; this circuit is required to
reduce the voltage rate of change during the closing of the thyristor.
(a) (b)
Fig./I. 23a: The circuit toproteet the thyristor against high dl!dt.
b: The waveforms of the current through the thyristor.
51
The magnetic switch is biased into reverse saturation. When the thyristor is triggered, only
the magnetizing current wiJl flow, and the full voltage falls over the magnetic switch L1.
The total current through the thyristor is the sum of the magnetizing current and the
current caused by the shortcircuiting of the snubber network (R1, C:2). This must be about
100 A and the time delay between the trigger pulse and the starting of the main current
about 0.6 p.s (figure II.23b). If the delay time is less than 0.6 p.s the dissipation increases
rapidly and this wîll lead to the destruction of the thyristor. For delay times much longer
than 0.6 p.s the dissipatîon will not decrease any more, therefore it is not necessary to
create longer delay times.
Three properties for this magnetic switch are:
1 The volt-serond integral (Vs) must providefora delay time of about 0.6 p.s,
2 very low inductance during the saturated state of the magnetic switch (Ls), as the
magnetic switch is part of the primary circuit of the pulse generator. This can be
obtained by the use of a large number of parallel coils,
3 the magnetizing current (Im> must be moderately large to provide a large conductive
area in the thyristor. This can be realised with different types of magnetic material,
but to reduce losses it is better to use a low-loss ferrite. The remaining part of the
current must then be provided by the snubber-circuit.
The picture on the other page shows a stack of five thyristors in series and a magnetîc
switch with ferrite ring-cores (80*40*15 mm) and eight single turn coils.
52
Picture of a thyristor stack and a magnetic switch toproteet the thyristors during stan-up.
D: A non-saturating putse transformers.
Pulse transformers are applied to provide a voltage rise. Figure 11.24 presents a series
magnetic compression network with a pulse transformer.
Fig. l/.24 Application of a non-saturating transfarmer in a series compression network.
53
The three properties are:
Sufficient volt-second integral (Vs) since the transformer must not go into saturation at
the end of the pulse,
2 low leakage inductance (La) for energy transfer in a short time (see section II.4),
3 the magnetizing current Im should be reduced by:
- High magnetizing inductance (Lm),
- Low magnetizing current (Imc).
E: A saturating pulse transformers for parallel pulse compression.
These types of transformers are applied in parallel magnetic pulse compression circuits.
Part of the time this component is used as a transformer and part of the time its core is
driven into saturation (see for the working of this circuit chapter III.3).
Figure II.25 shows a two stage parallel compression network.
Fig./1.25 7\vo stage parallel compression nerwork which saturable transformers.
The properties for these transformers are the same as for the non-saturating pulse
transformers (D).
Since the primary and the secondary coils play different roles in the pulse compression
circuit it is rather important that the coils are designed with the same properties as for the
magnetic switches (A).
54
11.6 Conclusion.
In high-power pulse applications, magnetic components can play a very useful role.
Because of the complexity of these components, magnetic switches and transformers, it is
necessary to have good design rules. Although magnetic switches have a nonlinear
characteristic it tums out to be possible to design a magnetic switch with the required
specifications. For high-speed applications it is still difficult to calculate the parameters in
advance, because of the great influence of the geometry and the behaviour of the magnetic
materials.
The two most important design values are the volt-second integral (Vs) and the inductance
of the switch in the saturated state (Ls). The volt-second integral is easy to calculate
(eq.II.3) and it is constant for different saturation times. Therefore, the moment that the
inductor goes into saturation is well defined. To calculate the saturated inductance (Ls) a
theory is presented. For this, a detailed description is given of the magnetic fields in and
around the magnetic switch. To verify these calculations two typical size cores of magnetic
switches were constructed of plastic; the measured inductances for a number of tums
matched the calculations very well. The assumed value of "one", of the saturated relative
permeability was not quite correct. The measured saturated relative permeability varies
between 2.5 and 4.0.
In this chapter results are given of the use of amorphous magnetic materials and of ferrites
in pulse applications. Because of the minimum thickness of the amorphous magnetic strips
materials these materials should be applied for pulses longer than 3 p,s, otherwise the
losses will be very high. For shorter magnetizing times, ferrite is the best choice. There
are two categories; the low-frequency ferrites have a larger flux density swing than the
high-frequency types, but have lower losses. To choose the right magnetic material it is
recommended to measure the dynamic B-H loop. For two types of amorphous magnetic
materials the loops are presented. It will be clear that for different magnetizing times and
different applications, different materials are required.
Because of the high peak current the R.M.S value of the current is quite high, therefore it
is important to use sufficient copper cross-section. As the pulses are short, the penetration
depth of the current is very small. Measurements indicate that stranded wires have good
high-frequency properties.
55
References
[1] W.S Melville; The use of saturable reactors as discharge devices for pulse
generators. Proceedings. Institution of Electrical Engineers, London , England, vol.
98, pt liL pp. 185-207, 1951
[2] H. Blok; Electromagnetische velden in de energie techniek, D.U.M. 1975
[3] R. Boll, H. Hilziger; Eigenschaften und Anwendung von amorphen
Magnetwerkstoffen. etz Bd. 102 heft 21, pp. 1096-1100, 1981.
[4] C.H. Smith, L. Barberi; Dynamic magnetization of metallic glasses. Proc. 5th IEEE
Int'l Pulse Power Conf. Albuquerque, NM, pp. 664-667, 1985.
[5] C.H. Smith; Magnetic losses in metallic glasses under pulsed excitation. IEEE
Trans. on Nuclear Science, Vol. NS-30, No. 4, pp. 2918-2920, August 1983.
[6] J. Vitins, J.L. Steine; Fast switching thyristors replace thyratrons in bigb-eurrent
pulse applications. Proc. Second European Conference on Power Electronics and
Applications, Grenoble, pp. 37- 42, 1987.
[7] E. Hommes and G.C. Paap; Elektrische machines. Technische
Hogeschool Delft, 1984.
[8] E. Philippow; Taschenbuch Elektrotechnik, band 1 Allgemeine Grondlagen, 1976
[9] L. V. Bewley; Two-dimensional fields in electrical engineering. Dover publications
inc. New York, 1963 (1948).
[10] E.Durand; Magnétostatique. Masson et cie, Paris, 1968.
[11] H. Kaden; Wirbelströme und Schirmung in der Nachrichten Technik. 1959
56
CHAPTER 111
Magnetic pulse compression circuits.
111.1 lntroduction.
In the previous chapter the principles of saturable inductors and transformers have been
explained. In this chapter the existing pulse compression circuits are discussed and a new
circuit is presented.
All magnetic pulse compression circuits are based on the non-linearity of magnetic
matcrials and on resonant energy transfer. Thyratrons can be used as a primary switch in
these circuits, but they have a limited life time. Therefore solid-state switches are of
interest, although the present industrial fast switching thyristors are not capable of directly
switching high-power pulses. To generate high-power pulses with a high-repetition rate a
suitable combination of thyristors and magnetic pulse compression circuit must be used.
First the series and parallel compression circuits are discussed [1 ,2 and 3]. After that, a
combination of these two circuits is presented.
111.2
111. 2. 1
Series magnetic pulse compression circuits.
Theory of series magnetic pulse compression
circuits.
The first paper on magnetic pulse compression was published in 1951 [1,2]. Magnetic
switches or saturable inductors were used to achieve pulse compression. The spark gaps or
thyratrons were only required to switch the smaller currents flowing during the longer
57
pulses before the compression. In other applications only one magnetic switch is applied to
reduce the initia! rate of rise of the current in thyratrons or thyristors.
The series magnetic pulse compression circuit is shown in figure liL 1. In a series pulse
compression circuit, the energy of a capacitor is resonantly transferred to the next one.
Pulse compression is achieved by a reduction of the charging time of each stage. For equal
values of the capacitors and negligible losses, the output voltage will be equal to the initia!
charging voltage of the first capacitor.
~i os
Fig.III.l Series magnetic pulse compression circuit.
In figure III.l all capacitors have equal values, L1 is a linear inductor and L2 is a
saturable inductor. To obtain maximum flux density swing, the core of the magnetic
switch is initially biased into reverse saturation. This is shown in figure liL 1 by the
current arrow Ibias and/or by the dot in the magnetîc switch symbol. The magnetic switch
in the circuit is designed to saturate when the immediately preceding capacitor c2 becomes
fully charged.
In figure III.2 the waveforms of the series compression circuit of figure III.l are
presented.
Fig.Ill.2 Theoretica! wavefarms ofthe series magnetîc pulse compression circuit offigure IJl. I.
Fulllines are capacitor voltages, dotred lines are the currents through the inductors.
58
After the primary switch s1 closes, a resonant current 11 flows from capacitor c1,
initially charged to Umax' through the linear inductor L1 until the second capacitor c2 is
fully charged. The energy transfer time is:
(III.l)
in which L11 is the total inductance of the resonant circuit (mainly the inductance of L1)
and Cn is the series conneetion of C 1 and c2 given by:
Cl Cz C I =
t cl + Cz (III.2)
Irrespective of the ratio of c1 and c2, the resonant current over the period of time T 1 is
given by:
u ~ c" sin "' t max L 1 t1
and the voltage on capacitor c2 is:
cl Uc (t) = Umax - cosw 1t]
z Cl+ Cz
(III.3)
(III.4)
During the period T 1 the magnetic switch L2 blocks the current, and therefore capacitor
c3 will not be charged. A small magnetizing current flows through inductor L2. To derive
the voltage integral across inductor L2, it is sufficient to integrate the voltage of capacitor
c2, since the voltage of capacitor c3 can be neglected, so:
Therefore if the capacitors are equal the voltage integral is:
Tl
f U~dt = % Umax T1 0
(III.S)
(IIL6)
59
The magnetizing current flows through the magnetic switch L2 and the corresponding flux
increases until the saturation level is reached. The inductance then rapidly decreases to the
saturated value and the current ~ rises, charging the capacitor c3. The charging time of
the CapacitOr C3 iS equal tO the halve period time <Tt/2) given by:
(lil. 7)
in which Lt2 is the total inductance of the resonant circuit (mainly the saturated inductance
of L2) and c 12 is the series conneetion of c 2 and c 3. In this period of time (T 2) the
voltages across the capacitors c2 and c3 are given by:
C2 + C3 COSC.Ul Uc (t>Tt) = Umax[ C C ]
2 2 + 3
(III.8)
(111.9)
wherein:
(III.lO)
1t is possible to add a second compression stage; the energy transfer to the next capacitor
occurs in the same way. To achieve pulse compression, the saturated inductance of each
next stage must be decreased.
Although this circuit is often used with equal capacitor values, it is also possible to use
different values. In tigure 111.3 a one stage series magnetic pulse compression circuit is
shown with the three options of the ratio c31c2.
60
Uc2 L2
Uc3 ÇJ
lrr:~ I rJ
(a) ÇJ J:' ICJ rJ
(b) (c) (d)
Fig.Ill.3a: A series compression circuit and the simulated wavefarms for different
capacitor values:
b: Step up mode when c2
> C3.
c: Complete energy transfer when c2
= c3
.
d: Step down mode when c2
< c3
In the case of equal capacitor values (fig.TIL3c) and in the absence of resistive losses all
the energy of capacitor c2 will be transferred to capacitor c3. Capacitor c2, initially
charged to Umax' will be completely discharged at the end of the energy transfer.
Capacitor c3 which was initially not charged, then reaches the maximum voltage Umax· ' In fig.III.3c the voltage across the capacitors and the current through the inductor are
shown. Because the energy transfer is resonant, the waveforms are sinusoidal.
In the case of c2 > c3 (fig.III.3b), only a small part of the energy of capacitor ~·
initially charged to Umax' will be transferred to capacitor c3. In this case there will be an
incomplete energy transfer. However there is a voltage gain, the theoretica! maximum
voltage gain (a factor 2) will occur when ~ > > c3.
In the case of c2 < c3 (fig.III.3d) the voltage over capacitor c2, initially charged to
U max will reverse and the voltage rise on capacitor c3 will be very small. ' In most cases the efficiency of the circuit is very important. There is however a difference
between maximum energy and maximum charge transfer. To show the differences it is
61
best to derive the relation between the starting voltage on c2 and the voltage on c3 at the
end of the energy transfer for different values of the capacitors. The same can be done for
the energy and the charge transfer. From equation III.8 and 9 we find the relations for,
Voltage-ratio:
u c3 (!ûzl=lt)
U Cz ( !ûz!=O)
Charge-ratio:
QC3(!ûzt=1t)
QC2(!ûz!=O)
Energy-ratio:
EC3(û)zt=n)
EC2((,)zt=O)
=
=
c2 2-1-2C2 umu. = c2 + c3 umu. c2 + c3
(liL 11)
c3 uc3<!ûzt=1t) 2 C3 = (111.12)
C2 Umu. c2 + c3
1f2 c3 u~g{!ûzt"'lt) = (ITI.13)
Y2C2 u!ax These relations are presented in tigure III.4; it is clear that only for equal capacitor values
complete energy transfer can take place. For this ratio there is no voltage gain and the
initial charge is also completely transferred.
62
A Ratio!
1.5
0 0 2 3
Charge
Energy ~~--... ~ ........ ~ .. ~ .. Voltage
4-c, ~
Fig.lll.4 Ratio of voltage (dotted line), charge (dashed line) and
energy (solid line) transfer for different capacitor values.
An advantage of a series compression circuit is the capacitor at the end of circuit. Some
loads, such as lasers do not allow voltage spikes across the electrodes before the start of
the pulse. The ciosure of switch S 1 should not generate a voltage spike at the end of the
circuit. eiearly capacitor e 3 acts as a "snubber".
111.2.2 Design rules for series magnetic pulse compression
circuits.
In this section we discuss four important design rules for series magnetic pulse
compression circuits. The circuit is shown in figure III.5.
1: Por good switching behaviour, the unsaturated indoetanee of Ln must be much larger
than the saturated inductance.
Design rule: Lu(n) > > Ls(n).
2: To prevent precharging of the next capacitor en+2 the unsaturated impedance of Ln+ 1 must be much larger than the saturated impedance of Ln.
Design rule: Lu(n+ l) > > Ls(n)·
3: To achieve maximum energy transfer the capacitor values must be equal.
Designrule: en Cn+l = en+2.
4: To achieve correct timing the saturable inductor Ln must go into saturation just before
the immediately preceding capacitor en reaches the maximum voltage.
Design rule: Tsat < T1h.
Fig.Ill.5 Two stages of a series compression circuit.
63
adl: The frrst design rule gives the magnetic switch its name. The switching effect is
based on the changing of the permeability of the magnetic materiaL For a good
switching behaviour the difference between the two states must be large, a practical
recommendation is:
Lu(n) > 400 * Ls(n) (Ill.14)
ad2: The second rule prevents precharging of the capacitor of the next stage. During the
transfer of energy from capacitor en to en+ 1 the high impedance of Ln + 1 prevents
an energy transfer to capacitor en+2. A practical recommendation is:
Lu(n+l)
Ls(n)
= Lu(n+2) 2: 20 Ls(n+l)
(111.15)
ad3: This rule only applies if no voltage gain is required. For maximum energy transfer,
the capacitor values must be equal (see figure III.4).
ad4: In principle the saturation of the core of Ln + 1 should occur exactly at the moment
of maximum voltage on the immediately preceding capacitor en+ 1 .In practice there
are good reasons to let the core saturate before the maximum voltage is reached,
since then a direct energy transfer takes place from en to en+2' The advantages
are: a sa ving of magnetic material and smaller capacitor losses. However, if the
advance period is too large not all the energy will be transferred to the next
capacitor en+ 2. In figure Ill. 6 shown the early saturation of Ln + 1.
64
' In+1
Fig.II1.6 Theoretica/ wavefarms of a series compression circuit as in ftg.I11.5 with earlier saturation
of Ln+ J' Fulllines are capacitor voltages, the dotted line is the current through the inductor.
111.3
111.3.1
Parallel magnetic pulse compression circuits.
Theory of parallel magnetic pulse compression
circuits.
In the parallel magnetic pulse compression circuit the magnetic switches are used in
parallel instead of in series. The magnetic switches are made in the form of transformers.
In figure III. 7 a parallel magnetic pulse compression circuit is presented.
Fig.lll. 7 Parallel magnetic pulse compression circuit.
Here transformers are the switching elements. In principle they can provide a voltage raise
and/or impedance matching. The transformer can be in the saturated or in the unsaturated
state. In figure 111.8 the equivalent circuits of both situations are given, in this case the
turns-ratio is 1:1.
(a) (b)
Fig.ll/.8 Equivalent circuits ofthe transformer,
a: unsaturated,
b: saturated.
65
In tigure III.Sa the transformer in the unsaturated state is presented by a linear pulse
transformer. The sum of the two leakage inductances, Lul and L02, is a parameter later
used to calculate the pulse duration. The inductor Lm is the relatively large magnetizing
inductance.
The coupling between the primary and the secondary winding of the unsaturated
transfarmer must be as high as possible, but in the saturated state the coupling must be as
low as possible.
In tigure III.8b the equivalent of the saturated transformer is shown. The inductors Lps
and L8s represent the inductance of the primary and secondary windings of the transformer
respectively. Because of the saturation of the core the two windings have practically no
coupling.
Using these equivalent circuits for a three stage parallel magnetic pulse compression
circuit, we can sketch three separate circuits for each energy transfer (see tigure III.9).
In case of a turns-ratio of 1: 1 and maximum energy transfer there is no voltage raise. The
transformers (T 1 and T2) and the magnetic switch (L1) are initially biased into reverse
saturation (see the dot in the transfarmer symbol), and the tirst capacitor cl is initially
charged to U max.. The magnetizing inductors Lm of the transformers arealso drawn in the
circuit diagram although their values are relatively high.
The energy transfer starts at the dosure of the primary switch s1. The current flows
through the switch s1 and the primary winding of the transformer. This current drives the
core of the tirst transformer T 1 out of saturation which means that the energy can be
transferred, and that the current can charge the second capacitor S through the primary
winding of transfarmer T2. While this transformer is biased in the same direction as I1,
the core will stay in saturation.
In tigure III.9b the equivalent circuit for this part of the process is shown.
The pulse duration can be calculated from the total capacitance, Ctl' and inductance, Ltl,
The total capacitance is given by the series conneetion of the capacitors c1 and Ci
cl c2 cl = t cl + c2
(III.l7)
The total inductance Ltl is formed by the leakage inductances Lul,l and Lu2,1 of the first
transformer T 1 and the saturated inductance of the primary winding of the second
transformer Lp82:
(III.l8)
For maximum energy transfer, the capacitors must have equal values. To achieve a short
pulse duration, it is necessary to design a pulse transformer with a very low leakage
inductance and also a low saturated inductance for both the primary and the secondary
windings.
The core of the first transformer is designed to resaturate in the opposite direction at the
end of the energy transfer from c1 to c2. In this way the primary and the secondary
windings of T 1 are largely decoupled. Now the fully charged capacitor S starts to
discharge. The reverse current in the secondary winding of the first transformer keeps the
core in saturation. This reverse current also drives the core of the second transformer out
of saturation and this restores the coupling between the primary and secondary winding of
transformer T 2.
A second energy transfer is now started, with a current 12 flowing as shown in tigure
ITI.9.c. To prevent a current flow through the load ZL a shunt inductor L1 is present. This
shunt inductance must initially be saturated to prevent a voltage rise over the load. The
total inductance Lt2 of the transfer circuit is:
A: the saturated inductance of the secondary winding of the first transformer T 1,
B: the leakage inductances of the second transformer T2,
C: the saturated inductance of the shunt inductor L1.
(III.l9)
The total capacitance in the circuit is the series conneetion of the capacitors c2 and c3.
Also in this circuit the capacitor values are equal to obtain maximum energy transfer. The
pulse duration depends on the total inductance and the capacitor values.
At the end of this energy transfer the core of the second pulse transformer T2 saturates
68
and the third and last energy transfer starts (see fig.III.9d). Since capacitor c3 is fully
charged, the current flow in the secondary of transfarmer T2 reverses and keeps the core
in saturation. The reverse current drives the core of the shunt inductor L1 out of saturation
and causes it to have a high impedance. Therefore the current must flow through the load.
The total inductance in the discharge circuit is only the saturated inductance of the
secondary winding of the second transfarmer T2. Here the decoupling of the transformer
is also very important for maximum energy transfer from c3 to the load.
To obtain high-voltage pulses, the circuits are designed to give voltage raise. This does not
change the basic operation of the circuit as discussed above, but it affects the values of the
inductors and capacitors.
If for example the tums-ratio of the first transfarmer T 1 of figure IIL9 is "1 :N" the first
energy transfer circuit (fig.IIL9b) changes into figure III.lO
1.: PS.Z
Fig.l/1.10 Energy transfer circuit as fig.lll.9b but with a transfarmer with
a turns-ratio of "I:N".
Although it is possible to relate the component values to either the primary or to the
secondary si de of the transfarmer, we chose in figure III.l 0 for the primary side.
The transformed values are:
and
c' 2
(III.20)
(IIL21)
These transformed values must be used in the formula for the pulse duration (liL 16). Due
to the factor N2 the inductance at the secondary side of the transfarmer has less influence
on the pulse duration.
69
The secondary winding of the transformer illso plays a role in the next energy transfer, so
it is important to reduce its saturated inductance. In fact, to achieve pulse compression the
tums-ratio should be limited.
111.3.2 Design rules for parallel magnatie pulse
compression circuits.
There are two important design rules for parallel magnetic pulse compression circuits,
such as shown in figure lll.ll.
These rules are:
1: To achieve voltage raise and maximum pulse compression, the turns-ratio must be kept
modest, so that the saturated inductance of the secondary winding of the transformer
Lss remains small.
2: To achieve complete energy transfer, the capacitor values must change with N2 where
Nis the tums-ratio of each transformer.
Fig.III.ll Parallel magnetic pulse compression circuit.
adl: This rule has two contradictory aspects:
70
A: for maximum voltage raise the turns-ratio must be high,
B: for maximum pulse compression the saturated inductance of the secondary
winding of the transformer Lss must be as low as possible (see chapter Il.2.1).
It is therefore hard to give a specitic design rule. It depends on the application what
aspect is more important. A good practical compromise is to design the transformer
with a turns-ratio less then 15.
Since both the primary and the secondary windings of the transformer are part of a
compression circuit, we must design both windings as a magnetic switch to obtain
the lowest possible inductance.
ad2: This rule is only valid if the voltage raise is provided by the turns-ratio of the
transformer and not by resonant voltage gain. The capacitor value must be adapted
to the turns-ratio of the transformer.
111.4 The influence of losses on the performance of
magnetic pulse compression.
In the previous sections it was assumed that dissipative losses are negligible, which allows
a 100 % energy transfer. However, three important mechanisms reduce this optimum
energy transfer: in the first place resistive losses such as copper losses, secondly iron
losses and thirdly the effect of the magnetizing current.
In figure III.l2a a series resistor is added to the basic series compression circuit to
simulate copper losses. Obviously not all the energy is transferred to the capacitor C:3· A
part of the energy is dissipated in the resistor and a part is not transferred at all.
R1 l2
+C, ui
o Tc2 Ic3o * (a) (b)
Fig.lll.l2a: Series magnetic puLre compression circuit with resistance.
b: Simulated waviforms of the basic series magnetic pulse
compression circuit with resistance.
71
Computer simulated waveforms are shown in tigure 111.12b: The initially charged
capacitor c 2 is only partially discharged at the end of the energy transfer, and capacitor
c 3 is not fully charged. The sum of the charges of course remains constant (see equation
Ill.22), but there is lost energy in the resistor. Por equal capacitors (C2 = c 3 = C) the
charges are:
(III.22)
Since the energy of a capacitor is quadratic with the voltage, a small voltage loss gives
already a double energy toss. At the end of the energy transfer, the energy is partly stored
in the capacitors, the rest bas been dissipated in the resistor.
The total dissipated energy ~) in the resistor can be calculated Prom eq. III.22 and the
condition c2 c3 = c.
(111.23)
Por most pulse power applications the series resistance is kept as small as possible and at
least smaller than the resistance value for the critically damped case. In these cases the
energy transfer time can be formally calculated from the oscillation frequency (see
eq.III.7); note that in fact the moment of maximum voltage shifts to later times at larger
damping. In the limiting case of critical damping there is no well defmed energy transfer
time and half of the energy is dissipated in the resistance; tigure III.13 shows the
differences between the capacitor charge and energy at the end of the energy transfer.
72
Q (toQ) t 1 A A
B
0.5 0
c
0 0
(a)
Fig. Jll.13 Charges and energy in the capacitors of fig.lll.l2 at the end of rhe
energy transfer.
a: Charges A: total charge (fullline), B: charge in capacitor c3
(dashed line),
C: charge in capacitor c2
(dot-dashed Une).
b: Energy A: total energy (fullline), B: energy in capacitor c3
(dashed line),
C: energy in capacitor c2
(dot-dashed line}, D: dissipated energy in R1
(vertical line in the shaded area).
The energy loss in the magnetic material and the effect of the magnetizing current will be
discussed now in combination. The magnetizing current of the magnetic switch causes
energy loss and an incomplete energy transfer since it causes precharging of the next
capacitor. In tigure III.l4 these effects are demonstrated. Note that tigure III.14b is only
given up to the moment when 12 reverses; after that moment c3 will be discharged again.
The current 12 goes to zero as a result of the hysteresis in the B-H loop.
73
Im s, - Uc2 L2
l0 ryyy\
~30 UI~ + !2 T2 Tc30
I I
I
I",
* closureS 1 (a) (b)
Fig.Ill.14a: The effect of magnetizing current on a series magnetic pulse
compression circuit.
Uc3
b: Simulated waveforms of the series magnetic pulse compression circuit.
The Juli lines are capacitor voltages, the dotted line indicates the
current through the inductor.
The capacitor c2 is initially charged to Urnroe When the switch s1 is closed, a
magnetizing current flows through the magnetic switch. Although the magnetizing current
can be found correctly from the shape of the B-H loop, the magnetizing current (dotted
line) in tigure III.14b is approximated by a constant current during the unsaturated period
of the core. This current causes a voltage drop on capacitor c2 and a voltage rise on
capacitor c3. When the flux in the magnetic switch L2 reaches the saturation level some energy has
already been transferred. Since the second capacitor c3 is already charged only the
difference in voltage over the capacitors will be transferred. Therefore still some energy is
left in capacitor c2 ("trapped energy"), which will not be transferred at all and is
therefore lost for the main pulse.
In the parallel magnetic pulse compression circuits, the magnetizing current has a different
effect as shown in tigure 111.15.
74
Fig.Il/.15 Parallel magnetic pulse compression circuit.
In tigure III.15 the magnetizing current Im flows through a magnetizing inductor Lm. The
current r1 which discharges capacitor c1 is split into two parts: the magnetizing current
Im and the charging current r2. So, not all of the energy is transferred to capacitor c2;
some energy is lost for the main pulse. Also here this energy is called "trapped energy".
To achieve high efficiency the magnetizing current should be reduced. This can be
achieved with a larger number of tums on both windings of the transformer. However,
this causes an increase of inductance of the saturated secondary winding of the
transformer, which deercases the compression rate.
From this section, it follows that maximum energy transfer can only be achieved when the
resistive losses and the magnetizing current are minimized.
111.5 Experimental results.
In this section experimental results on series and parallel magnetic pulse compression
circuits are presented.
The results of experiments with magnetic switches have already been discussed in chapter
II.
111.5.1 The bias circuit.
Usually in magnetic pulse compression circuits the cores are biased into reverse saturation
by a DC-current. There are two reasons to do so: first to make better use of the magnetic
75
materials, this implies also a higher compression rate, secondly to have a guaranteed volt
secoud area and to improve the reproducibility. The aim of the DC-circuit is to apply a
voltage to the bias windings which is large enough to reset the core in time for the next
pulse.
Mostly a simple !ow-voltage power supply is used, which introduces a smalt problem.
During the unsaturated state of the magnetic switch the two windings form a transformer.
The tums-ratio of this transformer depends on the number of tums of the magnetic switch
winding and the bias winding. Although the number of bias tums is smaller, there is still a
considerable voltage induced in the bias circuit. Since the impedance of the power supply
is very low, the induced voltage generates a large current in the bias circuit. To solve this
problem a large inductor is connected in series with the bias winding. The induced voltage
appears across the large inductor and creates only a small current change. The inductor is
usually an air-core coil with an inductance of several mH.
111.5.2 Experimental results on series compression.
Two series pulse compression circuits were built for operation with a TEA co2 laser
system: a paper on the first one was publisbed in 1988 [4], and is included at the end of
this section. A second paper describes a combination of series and parallel compression
circuits followed by a simple series compression circuit. This paper was publisbed in 1989
[5] and is included in section TII.8.
During the experiments it became clear how important timing is for the efficient operation
of magnetic putse compression. For a one stage pulse compression circuit a change in the
volt-second integral results in a varlation of the saturation moment. So, by changing the
voltage a good timing can be obtained. This also means that for varying input voltages, the
output putse will move back.ward or forward in time.
In magnetic putse compression circuits with more than one stage, this effect introduces not
only changes in timing but also a mismatch between the stages.
A small advance (early) saturation is allowed in series compression circuits, so it is better
to design the magnetic switch for a flux about equal to 90 % of the volt-second integral.
The next part of this section is the paper presented at the Seventh International Symposium
on Gas Flow and Chemical Lasers, 22 - 26 August 1988 in Vienna, Austria.
76
High voltage solid-state pulser for high repetition-rate gas lasers.
A.L. Keet, M. Groenenboom
HOLEC, Innovation and Technology, Hengelo, the Netherlands.
F.A. van Goor, W.J. Witteman
University of Twente, Enschede, the Netherlands.
ABSTRACT
An all solid-state pulser. for the excitation of a high TEA co2
laser is described. The pulser uses a fast switching thyristor, transfarmer and a two-stage magnetic pulse compression netwerk. The pulser ln~eriacea with the laser has been succesfully operated at 200 Hz with a pulse energy of 7 J, peak voltage of 20 kV, pulse duration 0.6 ps, and an overall efficiency of 70%.
INTRODUCTION
High repetition-rate co 2 lasers find application in material processing, range finding eguipment and optical pump~ng. Practical application of these lasers reguires a reliable power supply. For low repetition-rate gas lasers (<100Hz) a thyratron switch represents the best solution. However, with higher repetition-rate gas lasers the limitation of the t9yratron becomes more and more important, particulary its relatively short life-time (10 shots). Although the thyristor is not limited in this way, there is still a considerable gap between the ratings of thyristors and the reguirements of a pulsed gas laser. Typical laser ratings are, peak > 15 kV, peak current up to 10 kA and a current pulse duration less of then 0.5 ps. industrial fast switching thyristors have a maximum forward blocking voltage of about 1300 V and a maximum rate of change of current 400 A/ps. This gap in capabilities can be closed by using a pulse transfarmer and a magnetic pulse compression network [1,2].
THE SOLID-STATE PULSES
The pu1se power supply, illustrated in figure 1, consists of an ordinary d.c. power supply, a thyristor THl for voltage doubling, a main TH2, a step up transfarmer T2 and a two stage pulse compression netwerk (C3, L3, , L4, C5).
ax.coax
Cs
Figure 1. Schematic circuit diagram of the all solid-state pulser.
The ordinary power supply charges the large storage capacitor (Cl) up to 500 V whereupon the thyristor THl is triggered. The second starage capacitor (C2 ~ 25 uF) is then charged up to about 900 V, in approximately 0.5 ms. The RC netwerk over the thyristor is a snubber circuit. At the moment that the second starage capacitor C2 is fully charged and the first THl is fully "OFF", the second thyristor TH2 is By using a fast transfarmer with a turns ratio of 1:25 the first high capacitor C3 charged up to 21.5 kV in 13 The high voltage are ceramic types of 40 nF each (20 of 2 nF parallel). Because in the main thyristor TH2 (2.5 kA), and the short duration, the current would exceed the thyristors maximum
440 /SPIE Vol. 1031 GCL-Seventh International Symposium on Gas Flow and Chemica! Lasers (1988)
77
The saturable reactor LS slows the current rise until the thyristor TH2 is sufficently in conduction. In this state when more of the chip area is in conduction, the thyristor is capable of withstanding much higher rates of rise of anode current. Once the reactor L5 saturates (decreasing inductance) the current rapidly increases to its maximum value. (Figure 2.). Inthetwo-stage compression netwerk, the 13 ps pulseis first compressed to 2 ps and in the secend compression stage furter compressed to 0.6 ps, The high compression ratio (6.5 and 3.5 respectively) is achieved by using amorphous metallic magnetic material. (Vacuumschmelze 6030 Z.)
A
Figure 2. t US
Current waveferm of the main thyristor, showing the initially slowed current rise.
V
Figure 3. t US
Voltage across C3 and es illustrating the magnetic pulse compression.
The saturable reactors·L3 and L4 are in toroidal ferm. The first reactor L3 is made up of 10 cores (id = 4 cm, od = 8 cm, h = 2.5 cm) and has 18 turns. The secend reactor also has 10 cores but only 2 turns. When the first high voltage capacitor C3 is charged, the impedanèe of the saturable reactor L3 is large, thus the secend high voltage capacitor C4 will remain virtually dischanged. At the moment the first capacitor C3 reaches its maximum voltage, the fluxe B {L3) reaches its saturation level, thus reducing the series impedance of the reactor. The stared energy is then transferred to the secend high voltage capacitor C4 (Figure 3). The werking of the secend stage is the same as that described for the first, though the value of CS (6 nF) is much lower than c• (40 nF), leading toa much more rapid voltage rise. This fast voltageriseis necessary for the corona pre-icnasatien (this matter will be discussed in the presentation "Corona pre-ionizer for high repetition-rate TEA gas lasers" by F.A. van Goor). By adding a small resistor in series with the peaking capacitor CS it is possible to reduce oscillation between the capacitors C4 and CS. The value of the resistor determines the current waveferm of the laser (Figure 4.).
I R = 10 A
t US
Figure 4. Current waveferm of the laser for different value series resistors.
SPIE Vol 1031 GCL-Seventh International Symposium on Gus Flow and Chemica/ Lasers (1988) 1 441
78
CONCLUSIONS
The use of a two-stage magnetic compression netwerk for application with a corona pre-ionised TEA co 2 laser has been considered. It has been shown that commercially available thyristors can he used in conjuction with a magnetic compression network to generate the necessary pulse for the gas laser without di/dT breakdown. In this way the life-time limitations imposed by the thyratron can be overcome. The system has been succesfully operated at a repetition-rate of 200 Hz dalivering a measured energy of 7J per pulse toa laser, with an efficiency of 70%.
REFERENCE
1. T. Shimade, N Obacra and A. Noguchi Japanse journalof applied Physics. Vol. 24 No. 11 November 1985 pp L 855-t857.
2. H.J. Baker, P.A. Ellsmore and E.C. Sille. Tech Digest Cleo 1986. San Francisco (1986) P.l92.
442 I SPIE Vol. 1031 GCL-Seventh International Symposium on Gas flow and Chemica/ Lasers (1988)
79
111.5.3 Experimental results on parallel compression.
Also parallel pulse compression circuits were built to study the behaviour of these circuits.
The experiments began with only a single saturable transformer. The circuit is shown in
figure m.l6.
Umox S1
~ (o)
uj 30
(kV] 20
10
0
-10
-20
(b)
---- ~~
10
50 nF
I2 ----~~--- 0
-I
-2
-J
Fig.Ill.16a: Ihe circuit with a single saturable transformer.
b: Measured wavefarm of the circuit. 1he fullline is the capacitor voltage,
the dotted line is the current through the transformer.
The circuit is built with a spark gap SI' a coreless inductor L1, capacitors c1 (5 p.F) and
c2 (50 nF) and a pulse transfarmer T 1. The pulse transfarmer is made of strip-wound
cores of amarphous magnetic material (5 ring-cares {od=80 id=40 h=25 mm) of
Vacuumschmelze 7505-Z). The turns-ratio is 1: 10; the primary winding exists of six
80
parallel single turn windings and the secondary winding has two parallel windings of ten
turns. For maximum use of the magnetic material the transformer is biased into reverse
saturation.
We tuned the circuit to obtain saturation of the transformer at the moment the voltage of
the capacitor c2 reaches its maximum value, this was achieved both by installing an extra
inductor L1 and by changing the voltage. Figure III.16b shows measured waveforms of
the circuit. The voltage over capacitor c2 rises in the same way as in a series compression
circuit. At the end of the energy transfer (f:z - t1 = 5 p.s) the transformer saturates.
Because of the saturation the coupling between the primary and the secondary winding is
lost. The current in the secondary winding of the transformer starts to flow in the reverse
direction and keeps the core in saturation. Because there is only the saturated inductance
of the secondary winding, the capacitor~ rapidly reverses its polarity. At the end of this
period (~ - tz = 1.5 p.s) the current reverses again and the transformer comes out of
saturation, and the energy is transferred back to the primary capacitor c1, but now with
reversed polarity of the voltage. From this circuit we learned that compression with a
transformer is possible, and that a compression ratio of 3.3 can be achieved.
By installing an extra magnetic switch L2 a fast rising voltage is achieved; this circuit and
its measured waveforms are shown in tigure IJL 17. The two horizontal bars at the bottorn
of the tigure show the state of the cores of the transformer T 1 and of the inductor L2.
The extra magnetic switch ~ is made of ferrite (UlOO, 3C6 of Philips) and has four
parallel windings of four turns. This inductor ~ is also biased into saturation (see the
dot). The voltages of tigure III.17b show the difficulty of bringing the core completely in
saturation; at early times (t1 ~ tz) there is still a voltage rise over the inductor L2 until its
core completely saturates. During the energy transfer (f:z ~ t3), a current flows through
the inductor ~ and this generates a small negative voltage across inductor ~.
After the energy transfer the current reverses and transformer T 1 saturates and the core of
inductor L2 goes out of saturation, and therefore the voltage u2 drops quickly. If a laser
was connected to this power system it should ignite at this moment and the energy would
be supplied. As there is no load present the circuit will show his oscillatory behaviour.
During the next period (t3 -+ t4) only the magnetizing current of inductor~ can flow. At
the end of this period (t4) the magnetic switch L2 resaturates and a high current 12 can
flow. Since the transformer is still in saturation there is a rapid voltage reversal on
capacitor c2. The transfer time (t5 - t4) can be determined by equation III.24:
(IIl.24)
81
where Lt is the sum of the saturated inductance of the secondary winding of the
transformer T 1 and the saturated inductance of the magnetic switch L:2·
82
Umax S 1
~ (a)
~ UNSATURATED STATE
9
tsl -t U.•l
tf'"'~~
(b) D == SATURATED STATE
Fig.Ill.J7a: The circuit of a one stage parallel magnetic pulse compression circuit.
b: Measured waveforms of the circuit, the voltages over the secondary of the
transfarmer and inductor Lz The dotred line is the secondary current lz
3l I (KAJ
-I
-2
-3
In the next section the saturable inductor L2 of figure III.17 was replaced by a transfarmer
to start a second compression stage with a voltage raise, see figure III.18. The three
horizontal bars at the bottorn of the figure show the state of the cores of the two
transfarmers T 1, T 2 and of the inductor L3.
(a) -
u~~ Ut u .. [kV)
20
10
~ UNSATURATED STATE
(b) D SATURATED STATE
Fig.lll.18a: Circuit of a two stage parallel compression circuit.
b: Measured wavefarms ofthe circuit, the voltages
over the transfarmers and inductor Lz
83
The first transfarmer T 1 is the same transformer as described in figure lll.16. The second
transfarmer is the same as the saturable inductor Lz of figure liL 17, but with an
additional winding identical to the first one; which gives a tums-ratio of 1:1, another ratio
is also possible. Both windings consist of two parallel windings of four tums. The
saturable inductor L3 is constructed with ferrite (7 rings of T22 od = 80, id =40 and h = 15
mm of Thomson LCC). The winding is formed by two parallel coils of four tums. The
transformers and the saturable inductor each have a bias winding of three tums.
Both transformers T 1 and T 2 and the inductor ~ are initially biased into reverse
saturation (see the dots).
By closing the primary switch S 1 the first transformer T 1 is going out of saturation and
the second transformer will remain in saturation. At first the waveforms of this circuit are
the same as those of the one stage parallel compression circuit (shown in figure III.17b).
At the moment (~ that the current ~ reverses, the first transfarmer T 1 must go into
saturation. The reverse current 12 drives the second transformer out of saturation and
keeps the frrst one in saturation. The magnetic switch L3 is biased in the same direction as
the charging current of capacitor c3, so it also remains in saturation. Also in this circuit,
at the end of the second energy transfer (f:l ..... t4) transformer T2 goes into saturation and
the current 13 reverses, so the voltage U 4 rapidly rises. This voltage rise can be supplied
toa laser.
111.6 Concluding remarks on the series and parallel
magnatie pulse compression circuit.
The series magnetic pulse compression circuit is easy to onderstand, although the non
linearity of the saturable inductors makes it a difficult circuit for simulations. The parallel
magnetic pulse compression circuit is relatively difficult to onderstand and the design of
the transformer is critica!. In the following we compare it to the series compression
circuit.
The advantages of series compression circuit compared to parallel compression circuit are:
- The series compression circuit bas a capacitor at the end of the circuit, therefore
voltage spikes across the load are suppressed.
84
In a parallel compression circuit the saturated transformer always keeps a small
coupling between primary and secondary windings and there is an indoetanee at the end
of the circuit, so voltage changes in the beginning of the circuit are directly coupled to
the end. This willlead to an early voltage rise over the laser.
- In a series compression circuit a capacitor is placed parallel to the load so it is possible
to achieve a voltage doubling (if load impedance is high): this can be useful for igniting
a laser. In a parallel compression circuit an indoetanee is placed in parallel with the
load, so there is no possibility for voltage doubling.
- The series compression circuit is easier to onderstand than the parallel compression
circuit.
The advantages of the parallel compression circuit compared to the series compression
circuit are:
In a parallel compression circuit it is possible to achieve voltage raise, so low voltage
switches can be used in high-voltage pulsers. In a series compression circuit however,
the primary switch must hold the maximum voltage.
- In a parallel compression circuit an inductor parallel to the laser is needed, so the main
pulse can have a short rise time. Note that the voltage rise depends on the sum of the
parasitic capacitance of this inductor and the load.
111.7 The combination of parallel and series pulse
compression circuits.
111.7.1 Introduetion
In the previous section the advantages and disadvantages of the series and the parallel
magnetic pulse compression circuits have been discussed. Por high-repetition rate
application it is advantageous to use a thyristor as primary switch. A thyristor, however,
has a limited blocking voltage. Por laser applications the voltage across the laser should
not rise before the full energy can be supplied. It is therefore obvious that a parallel
compression circuit is not suitable.
85
To reduce the voltage over the primary switch a non-saturable pulse transformer can be
used. This is applied in the circuit described in section 111.5.2. Although this seems to be a
good solution there are still some improvements possible. In tigure III.19 such a putse
transfarmer for voltage raise is added to a series compression circuit.
Fig. ll/.19 Series compression circuit with a pulse transformer.
In this circuit the transformer is notallowed to go into saturation. The cross-section of the
transforrner core must therefore be designed to stay out of saturation during the charging
and discharging of the capacitor c2. The magnetic switch L1 must block the current
during the charging time (t1 .... ~) of the capacitor c2. Clearly the volt-second integral of
the transfarmer must be larger than that of the magnetic switch L1. In this circuit both the
cores of the transfarmer and of the magnetic switch are magnetized during the charging
time of the capacitor c2. In a parallel compression circuit (fig. HL 7) only the first
transfarmer core is magnetized during the energy transfer and the second transfarmer core
stays in saturation. So during the first energy transfer only the first transformer has losses.
Therefore a higher efficiency can be achieved if the first stage is a parallel cornpression
circuit.
For laser applications high-voltage pulsers are required. Clearly it is better to start with a
parallel compression circuit (for voltage raise and lower losses) and to end with a series
cornpression circuit (capacitor at the end of the circuit). In this way the advantages of both
circuits are cornbined.
86
lil. 7.2 Realisation of the combination of parallel and
series compression circuits.
To create the combination of series and parallel compression circuits it is necessary to use
a saturable pulse transformer and end the circuit with a capacitor at the output. The first
experiments were made with the circuit shown in figure 111.20. In this circuit a shunt
inductor is connected in parallel to the last capacitor.
Fig. Ill.20 A circuit in which paralleland series pulse compression is combined.
The pulse transformer T 1 is used as a step up transformer and is biased into reverse
saturation. The saturable inductor Lz is biased into saturation. The :first capacitor C1 is
initially charged to Umax· The ideal waveforms are presented in :figure III.2la. After the
switch s1 doses the current 11 flows through the capacitor c1, the primary switch s1 and
the primary winding of the transformer T 1. The secondary current Is flows through the
secondary winding of the transformer T 1, the capacitor c2 and the saturated inductor Lz· The capacitor c3 will not be charged as long as the current Is can flow through the
saturated inductor L2. The necessary forward saturation is ensured by a bias current in the
same direction as Is. At the end of the energy transfer the current Is reverses and the core
of the saturable inductor L2 automatically comes out of saturation and blocks the current.
The core of the transformer T 1 must be designed to go into saturation when the current Is
reverses and this current will then keep the core in saturation. Because the inductor Lz now blocks the reverse current the current must flow through capacitor c3. At the end of
the second energy transfer capacitor c3 is fully charged.
Because the components are not ideal, (leakage inductance, magnetizing current and
incomplete squareness of the magnetic material) the operation of the circuit is somewhat
modi:fied.
87
In the circuit shown in figure III.20 the saturable inductor Lz must be in saturation at the
beginning of the energy transfer. The core of Lz is driven into saturation with a DC
current, but it is very difficult to reach full saturation. The effect of incomplete saturation
results in some charging of the capacitor c3, until the magnetic switch Lz reaches
complete saturation. From this moment on, the capacitor c2 should be charged as
planned, but because of the precharging of the capacitor c3 there is also a fast ringing
current between the capacitor c3 and the inductor Lz· In figure III.21a the ideal waveforms are presented and in figure III.2lb the measured
ones. The used component values are Umax = 2.5 kV, c1 = 5 fi.F, c2 = C3 = 50 nF,
Turns-ratio 1: 10.
88
u I JO
[kV] 20
10
0
-10
-20
-JO
(a) 0 2 J 4
UIJO [kV] 20
10
0
-10
-20
-JO
(b) 0 3 4
u, u2
5 ---:;.... t [,us]
3 t I
21 [kA]
-1
-2
3 t I
21 [kA]
0
-I
-z
-3
a: /deal circuit.
b: Measured in an actual circuit.
Fig.IIL21 /deal and measured waveforms ofthe combination of paralleland series
compression circuits. 1he fulllines are voltages, current in L2
(. .. . ),
current in secondary ofthe transfarmer (---) and current in c3
(-.-.-).
To prevent this precharging of the capacitor c3 it is necessary to initially saturate the
magnetic switch L2 completely. As stated it is very difficult to achieve this with a DC-bias
current; therefore it is preferabie to block the current flow towards capacitor c3 during
the charging period of capacitor c2. Of course the current should not be blocked during
the charging period of capacitor c3. This can be achieved by an additional magnetic
switch between the inductor L2 and the capacitor c3 (see figure III.22).
Umox S 1
Y2500V
t 1
Fig. IJl. 22 A circuit with an improved combination of parallel and series magnetic
pulse compression circuits.
Measured voltages for this improved circuit are shown in figure III.23.
Figure III.23 shows the waveforms in the case of an optimum timing between the magnetic
switches and the transformer. It is however rather difficult to tune the circuit to this
optimum.
89
Especially the timing between the saturation of the transformer and the end of the first
energy transfer period is very critical. This can be illustrated by a discussion of the three
possibilities:
A: The transformer saturates too early.
B: The transformer saturates just in time.
C: The transformer saturates too late.
ad A The energy transfer is not yet completed, soa part of the energy is still in the first
capacitor C 1. When the transformer saturates the remaioder of the energy will cause
a ringing current in the primary winding of the transformer, and this energy is lost.
ad B The timing is correct and all energy of c1 is transferred.
ad C All the energy was already transferred to the second capacitor C:z. but now it is
flowing backwards until the transformer saturates. So also in this case not all the
energy is transferred to the second capacitor, resulting in the superposition of a fast
ringing current.
Because the above described circuit depends critically on the saturation time of the
transformer, its also depends on voltage and temperature changes (see chapter 11.3). A
change in temperature influences the saturation level of the core.
The installation of the extra magnetic switch L1 can solve this problem. This magnetic
switch is mostly already installed to rednee the initia! rate of change of the current through
the primary switching thyristor (see chapter 11.5). The circuit is shown in figure 111.24.
Umax s1
~ c1J.é1
Fig. 111.24 Combination of parallel and series magnet ie pulse compression with the
additional magnetic switch Lr
In this circuit all the magnetic components are biased into saturation, and capacitor c1 is
charged to Umax. The energy transfer starts with the closing of the primary switch s1.
90
The current 11 is limited by the magnetic switch L1 until the thyristor creates a larger
conducting area around the gate (see chapter II.5). Then the energy transfer can take
place.
The waveforms are presented in figure III.25, the horizontal bars at the bottorn of figure
III.25a show the state of each magnetic component. Figure III.25b shows the position in
the B-H loop (at times t1 to tè
By closing the switch s1 (t0) the magnetic switch L1 blocks the current and when this
switch goes in saturation the first energy transfer can take place. During the time period
from t1 to ~ the magnetic switch L2 is driven into saturation and L3 is in the unsaturated
state to prevent precharging of capacitor c3. At the end of the energy transfer (~), the
current reverses but as long as the transfarmer T 1 is in the unsaturated state, any primary
reverse current is blocked by the magnetic switch L1, because it goes out of saturation. So
the second energy transfer can only take place when the core of the transformer goes into
saturation (t4). As the current reverses the core of inductor L3 is probably in the
unsaturated state, so it will block the current for a very short time (approximately 20 ns)
until it resaturates (t5). Then the second energy transfer to c3 can start.
The magnetic switch L3 should be very small, because it increases the inductance in the
circuit of the second energy transfer. While the required volt-second integral of inductor
L3 is very small, L3 can indeed have a small value.
During the first energy transfer (~ - ~) the core of inductor ~ does not see any flux
chance. But during the second energy transfer, the core must not go into reverse
saturation. Therefore the volt-second integral must be large enough to hold off the voltage
during the charging (t5 t6) and the discharging times of capacitor c3.
The transfarmer T 1 is designed to stay out of saturation until the energy transfer is
completed. At the end of the energy transfer the current tries to reverse, but the magnetic
switch L1 goes out of saturation and blocks the reverse current, so no energy is
transferred back to the capacitor c1. The transformer must be designed to saturate before
(at t4) the magnetic switch L1 resaturates (at ls)·
91
92
inductor L 1 inductor L2 inductor L3
(b)
Fig.JII.25 Measured wavefarms ofthe combination of paralleland series magnetic pulse
compression circuits with the additional magnetic switch L1
(Fig.lll.24).
a: Measured wavefarms
b: Schematic B-H loops of the magnetic switches.
111.7.3 Design rules tor the combination of parallel and
series magnatie pulse compression.
In this section the most important design rules are explained; as a reference the circuit of
the previous section is used (figure III.24). There are five areas where specifications are
necessary:
1: To achieve maximum energy transfer.
2: To construct a saturable transformer.
3: To achieve high energy efficiency by designing the shunt inductor ~ with less
magnetic materiaL
4: To construct a blocking inductor L 3.
5: To construct a magnetic switch L 1.
adl: To achieve maximum energy transfer the capacitor values of c1 and c2 must be in
the proportion of 1 : 11N2. The capacitor values of c2 and c3 must be equal.
ad2: The transformer is a difficult component since it is both a transformer and a
magnetic switch. During the transformer action the coupling must be close to one
which means that the leakage inductances should be exceptionally low. During the
saturated state it must have a low inductance of the secondary winding. Therefore it
is necessary to limit the number of tums and also the tums-ratio. (see for more
details chapter 11.4).
ad3: The magnetic switch L2 is used as shunt inductor. This means that the inductance of
the magnetic switch is important during the first energy transfer (C1 to c2). Since
the pulse duration of this transfer depends on the total inductance of the circuit, it is
important to remember that the inductance on the secondary side of the transformer
as seen on the primary side is reduced by N2. The design is therefore not very
critical and the magnetic switch can be designed with more tums and a smaller core
area. So, there are less losses in this magnetic switch and it makes the switch
cheaper. Because of the required presaturation of the switch the applied magnetic
material must have a square B-H loop.
ad4: The blocking inductor L 3 can be very small since the required flux swing is small.
The saturated inductance must be low, like in all magnetic switches.
93
ad5: The magnetic switch L1 has two functions: First it should reduce the initial current
to give the thyristor (S1) some time to create a larger conductive area, and secondly
it should give some freedom in the timing of the moment of maximum voltage in the
capacitor c2 and the saturation of the transformer. This magnetic switch is placed on
the primary side of the transformer, so its inductance must be kept as low as
possible, and in actdition it is necessary to have a relatively high current during the
start-up of the thyristor. Usually a single or a two turn winding is used.
111.8 Experimental results.
In the previous section some experimental results have already been shown. This section
consists of the paper which was presented at the Third European Conference on Power
Electronics and Applications, 9- 11 October 1989 in Aachen F.R.G.
94
Aacnen 1989
HIGH VOLTAGE SOLID-STATE PULSER FOR HIGH REPETITION-RATE GAS LASERS Groe:ne::bc;orn. HOL~C, lnPovotion anc Techno!ogy, P.O.Box 23, 7558 AA Henge o,
Nether!on<1s
excitation of a hîgh repetition-to convertional switched
upon a series conneetien of 5 switching
series magnetic compressie~ of 8 kW, energy of 20 J anè a repetitior: voltage is kV and the pulse àx.::ation is successfully operated on a dummy-load with at maximum has been vP~<d,~n~ rnaximur:: al::..owed by the
transformer. The o!: parallel
70%
Magnetic comp=ession circ~it, High-current pulse application, Power }aser systerr:s,
INTRODUCTION
is not limited
requirement:s laser ratings are: a than 20 kV, a peak current of up a pclse duration of less than 500 ns.
industrial fast svlitching a maximlli~ forward blocking
about 2000 V and rate-cf 400
aboat a laser power of
!lecessar·v to place about
switches in to saturate when the
becomes pul se
inductance of
EPE Aachen_ 1989
this number of it is ex::remely low induc~ance required
value less ca!1 be
1127
Fig.:. Series magnetic pulse compressio:1 circuit
capa:::::itors have the same value to maxirnize er:ergy t.ransfer, sa there is no voltage gain.
u 1
Fig.
c.:.rcuit uses shows the t.ransformer states.
95
pr~~ ~ Ec,
•
la) I bi
Fig.3 ~~u~v~~~uL circuits of the
Fig.4
(a) unsaturated, (b) saturated
One parallel magnetic comoression circuit
The transfarmer is initially biased into reverse saturation. The energy transfer begins by closing the primary switch. Wavefarms are given in fig.5. A current flows from Cl, initially charged to Vmax• through the primary winding of transferroer T1. This drives the of transferroer Tl out of saturation. from Cl will be transfered to The core of transfarmer to resaturate in the opposite at the end of the energy transfer to c2. In this way the primary and windings are largely reverse current in the secondary T1 will keep this core the capacitor c2 rapedly By adding a secend transfarmer secondary circuit it is possible to a two stage compression circuit. kind of compressor circuit is capable of gain, making it possible to use a SCR primary switch.
Fig.5 wavefarms of a one stage parallel campression circuit.
The disadvantages of this high pass characteristic allows transfer of spikes and the very transformers. By using a pulse transfarmer it to reduce the withstand voltage primary switch. In the case of compression the magnetic core transfarmer must be larger than the first magnetic switch. In the amount of magnetic parallel magnetic compressor be preferred, in spite of its disa.dvan.tages In the system presented here,
96 1128
circuits have been combined, one stage parallel magnetic c~nPJre,;sj.on this stage it is possible to voltage gain to generate the voltage. The power supply of a two-stage series magnetic compressor. A good interaction between the two networks is achieved with two carefully designed saturable inductors. To achieve maximurn pulse compression the saturable inductors must be kept as as possible. Therefore, a magnetic rna.éer~al with a high saturation level is
several kinds of ma,.".,,~L.~"' which can be
strip-wound amorphous cores and The strip-wound cores have the
of a high saturation level the other hand the thickness of is limited to 25 ~· Thus, for
durations the core losses will rapidly.
ComDared with the strip-wound cores, lowrerr~ées have less losses but also
thyristors. delay time pulse and the typical delay time a current of flow, which
THE POWER SUPPLY
The aim of the project was to design a power supply with a high reliability and a high efficiency. The supply is designed for a Transversely ~tmospheric C02 laser. rt must the following specificaticns:
Energy per pulse Ignition voltage Steady state voltage Voltage rise time Peak current Average power Peak power Pulse duration Repetition rate
The power supply can be parts; the 380-, 2000-and and the pulse transfarmer compression stage, pression stages and
20 40 25 100 4 8 100 0.5 400
Joule kV kV ns kA kW MW )J.S
Hz
six
com!fig.6).
EPE Aachen. 1989
h, L2 LJ Th,
02 C2
LOOOV
Fig.6 schematic of the power supply .
The 2000 V storage capacitor Cl is charged by a coventional d.e. power supply. By using a re sonant Charging circuit the pulse capacitor C2 is charged to about 4000 V. For this, a small thyristor-stack Th! is used, consisting of 4 thyristors in series. The inductor L2 is used to achieve the required re sonant frequency. Ta obtain a short pulse, it is necessary to construct a low inductance primary circuit. Therefore the circuit is constructed as a stripline . For this application five reverse conducting thyristors (ABB) are used in series. The operating conditions of the thyristor-stack Th2 are forward voltage 3650 v, pulse duration 5 ~s and peak current 4 kA. With such a large current rate-of-change (2500 A/ ~s) it is necessary ta incorporate a saturable inductor in the circuit. In this case a delay time of 600 ns is used as conditioning time for the chip. To minimize the tatal circuit inductance, the pulse transformer is also constructed with a low stray-inductance . To achieve this lew reactance, the transformer has a single-turn, coaxially wound, primary. The vo lume of the transfomer is minimized by using a toroidal strip-wound core made of high-flux amorphous soft magnetic alloy (VAC). The pulse transformer has a turns ratio of 1:9. Thus the inductance of the secondary circuit i s almost solely determined by the inductance of the primary ciruit. The saturable inductor L4 may then be constructed with a relatively large number of turns and a small amount ef magnetic material. The picture (fig.7) gives an impression of the first compession stage including the pulse transformer. During the charging of the first high voltage capacitor C3, the inductor L4 is saturated . The functjon of the inductor L5 is to prevent a current flow towards the capacitor C4.
Fig.7 A picture of the first compression stage and the pulse transformer
EPE Aaehon . 1989
1 lp
11 29
U, U, U3 U, Us
C3 LS' LS L,
L, C, Cs RL
~:' 32 kV
The pulse transformer is designed to go inte saturation at the end of the Charging of capacitor C3. The reverse current at the secondary side of the transformer drives the inductor L4 out of saturation and the inductor LS into saturation. The capacitor C4 is charged in 1.2 ~s. The measured voltage and current waveforms are given in fig. 8.
Fig.8 Measured waveforms of the apparatus.
The last two stages are series compressien circuits, coaxially built to achieve maximum compression. To minimise the inductance, the last stages are built built directly onte the the laser. During the charging of the capacitor C4 the saturable inductor L6 will prevent the Charging of capacitor C5 until the capacitor C4 has reached the maximum voltage . The capacitor Cs is charged in 500 ns. Ourlng the Charging of capacitor C5, the inductor L7 prevents a current fl ow t owards the load .
97
1n the case of a lew 2m,peaance magnetising current In the case of a laser, the current into a small The inductor L7 designed saturation when capacitor charged. If capacitor voltage oversboot is very useful for fig. 9 the voltage dummy-load are illustated.
Fig.9 Voltage and current waveforms of the power supply with a dummy-load.
RESULTS
The power supply, designed for a glcw discharge pumped C02-laser, can be operated at pulse repetition rates up to 400 Hz. This laser has a very high impedance befere the discharge is established. During the glow discharge the laser has a low impedance. The power supply was tested with a low resistance dummy-laad. As the impedance of a gas laser is ncn-linear, the dummy-lead gives a slightly different behaviour. It is, however, suited for energy dissipation measurements. The overall electrical efficiency of the power supply at the maximum repetition rate (400Hz.) is about 70%. The power supply has successfully been operated on a corona preionised TEA co2 gas laser at the maximurn
rate allowed by the laser of about A risetime of 100 ns for the output h~.s been measured. The U-I voltage wavefarms are presented in fig. 10.
Fig.lO Voltage co2 gas
current wavefarms of the
98 1130
CONCLUSIONS
combined series- parallel circuit for a pulsed
TEA, C02 It has been shown
thyristors can conjunction with
compress~o.n networks to generate ·~~u~•~u output pulse for gas lasers. In
life-time limitations imposed énvraéron at high repetition rates can
overcome.
J.t. Steiner, A.
the Applied
OPPOrtunitv and
RE'FERENCES
Fast switching thyristors in high current
EPE Aachen. 1989
The last two series compression stages of the high voltage solid-state pulser is built
coaxially to achieve maximum pulse compression. The picture below show the two stages
(C4' C5' L6 and L7) which were built directly onto the laser system to minimise the
inductance.
A picture of the two last series compression stages
99
111.9 Conclusions.
In this chapter the theory of the magnetic switch pulse compression is given. The
advantages and disadvantages of various type of circuits are discussed. The design rules
are given to point out the difficulties of various compression circuits. Because each pulse
power application has its own characteristic requirements the specific design rules depend
on the application.
Also it is shown that it is possible to combine advantages of the basic compression
circuits. In this manner high efficiency can be achieved and power supplies can be built
with commercially available components.
Raferences
[l] W.S. Melville. The use of saturable reactors as discharge devices for pulse
generators. Proceeciin~s. Institution of Electrical En~ineers. London, England, vol.
98, pt III, pp.185-207, 1951.
[2] G.T. Coate, L.R. Swain, jr. High power semiconductor magnetic pulse generators.
Research mono~ra,ph no 39 The M.I.T. Press, Cambridge, Massachusetts, 1966.
[3] W.C. Nunnally. Stripline magnetic modulators for lasers and accelerators. Proc. 3rd
IEEE int. putse power conference, Albuquerque pp. 210, 1981.
[4] A.L. Keet, M. Groenenboom, F.A. van Goor, W.I. Witteman. High voltage solid
sate pulser for high repetition-rate gas lasers. Seventh International Symposium on
Gas Flow and Chemical Lasers, 22 - 26 August, Vienna Austria. pp. 440-442, 1988.
[5] A.L.Keet, M. Groenenboom. High voltage solid-state pulser for high repetition-rate
gas lasers. Third European Conference on Power Electtonics and Applications, 9 -
11 October, Aachen F.R.G. 1989.
100
An improved excitation circuit for
an excimer laser.
IV .1 Introduetion
CHAPTER IV
In the previous chapter the magnetic pulse compression circuits were applied to C~
lasers. In this chapter circuits are described for excimer (XeCl) lasers. In this chapter the
inductance of magnetic switches is extremely important. The applied magnetic switch is
already presenteel in section 11.5.
As the load impedance of the excimer discharge differs from that of a C02 laser (see
section 1.2.1), a different circuit has to be developed.
To achieve maximum energy transfer from the storage capacitors to the discharge,
impedance matching must be obtained. In contrast with the co2 discharge, the impedance
of the excimer discharge is not fixed. However, the voltage over the discharge is
approximately constant (the so-called steady-state voltage U88
). To obtain high-energy
pulses, at this fixed voltage and at limited discharge times, the current must be enlarged.
Therefore the inductance between the storage capacitor and the discharge must be kept as
low as possible; also a Pulse Forming Network (PFN) is used as storage capacitor.
This PFN must be charged to 2*Uss to obtain a current flow to the discharge at a voltage
level of Uss· However the ignition voltage of an XeCl excimer laser is between 5 to 6
times U ss· This causes a problem for maximum energy transfer. In the first experiments
with these lasers, the PFN was charged up to about 3 to 4 times Uss' and peaking
capacitors were used for voltage doubling. However, the voltage of the PFN was not
matched to the discharge.
1t is difficult to combine the demands of high ignition voltage and suitable impedance
matching during the steady-state in one circuit. Therefore, a combination of two circuits
has to be used. This so-called prepulse-mainpulse technique was first reported by Long et
al. [l J
101
The basic circuit for this technique is shown in tigure IV .1.
Moinpulse Prepulse
Fig.IV.l Basic prepulse-mainpulse circuit.
The prepulse capacitor c2 must be typically charged to 6*U88
and the main capacitor to
2*U88
• After proper preionization of the laser gas (by a separate X-ray source), switch s2 is closed and the voltage over the discharge electrodes rises until the laser gas breaks
down ( < 6*Uss>· After the laser gas breaks down, the main switch s1 must be closed, so
energy can flow from the capacitor c1 into the discharge. Usually capacitor c1 is
designed as a PFN.
Successful operation of the prepulse-mainpulse technique bas first been demonstrated by
Long et al. [1]. In this experiment a rail gap has been used to separate the two circuits (S1 in fig.IV.l), and an electrical to optical efficiency of 4.2% was reported for the laser and
its associated circuits.
However, for high-repetition ra te applications, a magnetic switch should be used instead of
a spark-gap. The magnetic switch, which replaces the switch s1, prevents the prepulse
energy to flow into the mainline. So, after breakdown of the laser gas, the magnetic
switch saturates and energy flows from the main circuit into the laser discharge. Circuits
using such a magnetic switch have been described by Taylor et al. [2] and Fischer et al.
[3]. In the next section we describe an improved excitation circuit using a magnetic
switch.
It is essentially a prepulse-mainpulse circuit and it combines the advantages of the Fischer
et al. [3] circuits but does not have their disadvantages . The excitation circuit is based on
resonant charging of the peaking capacitor. Therefore we called this circuit a
In the next section deals with this circuit, it is a reprint of the paper:
IV .2 High-efficiency operanon of a gas discharge XeCI laser using a
magnetically induced resonant voltage overshoot circuit.
102
High-efficiency oparation of a gas discharge XeCIIaser using a magnetically induced resonant voltage overshoot circuit
J. w. Gerritsen, A. L. Keet,"1 G. J. Ernst, and W. J. Witteman Ullil!ersity of Twente, Department of App/ied Physics, P. 0. Box 217. 7500 AE Enschede, The Netherlands
(Received 21 July 1989: accepted forpublication 18 December 1989)
An electrical to optica\ efficiency of 5% has been demonstraled in an x-ray preionized XeCI discharge laser using an improved prepulse-mainpulse circuit, based on resonant charging of the peaking capacitor. In contrast to known prepulse-mainpulse circuits, the new scheme bas no time delay between prepulse and mainpulse nor a voltage reversal on the electrodes. We present three possible configurations of the scheme and one of them bas been realized for demonstrating its high-efficiency capabilities.
High-efficiency operation of a gas discharge XeCllaser is very desirabie if the laser has to be operaled at a highaverage output power. The main problem is the large dilference between the breakdown and steady-state voltage and the related impcdanee matching of the discharge. A breakthrough was the development of the so-called prepulsemainpulse technique in which the prepulse takes care of the avalanche ionization process dUI·ing the breakdown phase, foliowed by the mainpulse with impcdanee matching during the steady state. In this way efficiencies of up to ~ 4% have been reported. In this technology, provided sufficient preionization, it is required that no energy can flow from the prepulse to the sustainer circuit during the prepulse phase and that the mainpulse follows immediately after the prepulse. The switches and the related inductances are then the key technology problems.
In the first experiment descrihing the prepulse-mainpulse technique, a rail gap has been used to separate these two circuits. 1 It is well known, however, that rail gaps are not suited to long life, high-repetition rate lasers, mainly due to electrode erosion and high-energy dissipation, A proper solution to separate the preputse and mainpulse circuits was the introduetion of a saturable magnetic inductor. This has been reported by Taylor and Leopold' and by Fisher et al.' High-efficiency operation could be achieved by slowly ( typical time approximately 1··10 11s) charging the main line to approximately twice the steady-state voltage (V"). After that, the voltage needed to obtain breakdown is superimposed. The magnetic inductor prevents the prepulse energy flowing to the main line. After breakdown of the laser gas, the magnetic inductor saturates and energy can flow from the main circuit into the laser discharge.
Fisher3 dislinguishes two distinct operational modes depending on the voltage polarities of the two circuits [Fig. I (a)]. The two modes both ha vetheir advantagesand disadvantages: When C1 and C1 have equal polarities, the current in the magnetic inductor reverses after breakdown of the
·'' Holoc, I nnovalion and Technology, P 0. Box 23. 7550AA Hengelo, The Nethcrlands.
laser gas [Fig. 1 (b), single arrow], which causes a delay between the breakdown of the laser gas and the rise in lhe current from the ma in line. When c, and C2 have opposite polarities, the current in the saturable inductor does not invert [Fig. 1 (b), double arrow], but tbe voltageovertbc laser electrades reverses after breakdown ofthe laser gas. Besides this, now a larger magnetic core area is required. Both the mentioned delay between prepulse and mainpulse and the voltage reversal may degrade the discharge quality.
In this communication we report on an improved excitation circuit, which has no time delay between prepulse and main pulse nor a voltage re versa! on the clectrodes, We have called this circuit a magnetically induced resonant voltage overshooi circuit (MIRVOC). The breakdown ofthe laser gas is oblained by the resonant charging of a peaking capacitor. This charging is initialed after saturation of a magnetic inductor. The required high breakdown voltage ( typically ~6V") can beachieved by anegative prechargeofthe peaking capacitor. Three possible designs for this principle are presenled in Figs. 2(a)-2(c).
In Fig. 2(a), the pulse forming network (PFN) and
[; -=zr"~-
{bi
FIG. I. (a) Prepulse· mainpulse circuits proposed by Fisher (Ref. 3), with two operatîonal modes depoending on the polarity of the prepulse voltage. (b) B-H characteristic with indlcated current directions for the two modes of Fischer's circuits, The single arrow corresponds to equal poJarities and the double arrow to opposite polarities ofthe preputse and mainpulse voltages.
3517 .1. Appl. Phys. 67 (7), 1 April1990 0021-8979/90/073517-03$03.00 @ 1990 American lnstitute of Physics 3517
103
kl
FIG. 2. Diagramsof the MIRVOC circuits; (a) using a saturable trans· former T; (h) replacing Tby the saturable inductor L4 , and (c) an autotrigger design where no additional circuit is needed for negative charging of CJ.
peaking capacitor ( C,) are charged to ~ 2 V" in approximately I f.tS from charging point A. C2 is charged to 4 V.,. By switching S, the voltage on C3 decreases to - 2 V" when C2
and C3 have equal valnes and core losses are negligible. Th en the transfarmer T must saturate and C3 is resonantly charged to 6 V". The time for resonant charging of C3 is determined by the inductance ofthe saturated transfarmer T and the capacitance of C3 • After proper preionization, the laser discharge breaks down shortly before the peaking capacitor is totally charged. The voltage over the laser electrodes then drops to V" maintaining the same polarity. The saturable transformer remains in saturation because the charging current of the peaking capacitor and the discharge current have the samedirection [Fig. I (b ), double arrow ].
In principle, circuit 2 (b l is a simplilkation of scheme 2(a); the transformer T bas been replaced by a saturable inductor L 4 • The principle of operation is tbe same. Although circuit 2(b) is easier to realize, it bas the disactvanlage tbat Sis not grounded on one side, wbich is preferabie when a thyratron is used for thi• switch.
In circuit 2(c), the additional circuit for precharging the peaking capacitor can be omitted: Again, the PFN and the peaking capacitor are charged to 2 V": When a charging voltage of2V,. is reached, the inductor L5 saturates and the circuit elements L 5 and C3 forrn a fast ringing circuit. In this way C3 is charged to reversed polarity (i.e., 2 V" ) . Dur-
3518 J. Appl. Phys., Vol. 67, No. 7, 1 April1990
104
Time tns I
FIG. 3. Typicalvoltage (solidline),current (da.shedline),andopticaloutput pulse ( dot-dasbed line) charaeteristics of circuit 2(b). The laser gas consisled of2.0-moor HCI, 20-mbar Xe, and 4-bar Ne.
ing this charging reversal, there wiJl be a voltage over the saturable inductor L 4 • This inductor is designed in sneb a way that it is saturated when the voltage reversal on C3 is completed. Next C3 can be resonantly charged again in the same way as in the previous circuits. Note that the voltage on C, eaunol swing back from 2V., to 2V" through L, because the current necessary for this wiJl drive L 5 out of saturation. Because this scheme only needs one conventional switch to pulse charge the PFN, it has the potential of very easy and reliable operation.
The MIRVOC scheme 2(b) has been successfully operaled on a high-pressure x-ray preionized XeCI laser. The laser has been described in detail before. 4 The laser chamber essentially consistsof a reetaugul ar stainless-steel vessel, designed for gas pressores of up to 15 bar. Tbe discharge length and electrode separation are 60 and !. 5 cm, respectively. The x-ray preionizer is basedon a cold-eatbode e beam with a Ta foil for the anode and is operaled in the transmission mode. The water tilled PFN used in our previous experiments bas been replaced by a 300-nF PFN consisting of ceramic capacitors. The characteristic impcdanee is ~ 0.3 .fl and the double transit time is - 160 ns. The peaking capacitor consistsof 14 capacitors of 460 pF in parallel to obtain a minimum inductance between these capacitors and the laser head. c2 bas a capacitance of7 .2 nF. All capacitors are of the Murata DHS Z5V series. Sis a spark gap. The sa turnbie inductor L 4
consists of two parallel race tracks. They are made from microwave ferrite blocks (SEI type K4 ). To minimize the core area this inductor is externally biased. During saturation, the inductance between the PFN and the laser head was - 30 nH as determined from the time necessary to charge the peaking capacitor. In Fig. 3 we present typical characteristics of the voltage across the laser electrodes, the current measured in the ground strip of the PFN ( using a Rogowski coil ), and the optical output pulse as a function oftime. The optica! cavity consisted of a 5-m radius maximum reflector
· and a flat 70% rellectivity output mirror separated by 1.3 m. Maximum efficiency has been obtained in a mixture
containing 2.0-mbar HCl, 20-mbar Xe, and 4-bar Ne. Using a calibrated Gentee ED 500 energy meter, an output energy of0.28 J bas been obtained with the PFN charged to 5.5 kV (4.51) and C2 to 17.4kV ( 1.1 J). In this way au efficiency of 5% bas been achieved basedon the total energy storedon the capacitors. From burn patterns we observed a beam cross
Gerritsen el al. 3518
section of ~ !. 3 X 1.0 cm2• The optica! pulsc length was 190
ns (FWHM). Longer pulses are expected by increasing the PFN length. We also expect that further improvements in efficiency can be achieved by optimization of the system components, e.g., usc of capacitors and ferrites ha ving lower losses and decreasing the inductance between the PFN and the laser head. We believe that the MIRVOC scheme is also very suitable for the generation of ultralong XeCl pulses. 5
This work was supported by the Dutch Foundation for Fundamental Research on Matter (FOM).
'W. H. Long, M. J. Plummer, and E.A. Stappaerts, Appl. Phys. Lett 43, 735 (1983).
'R. S. Taylor and K. E. Leopold, AppL Phys. Lett 46, 335 ( 1985 ). 'C. H. Fisher, M. J Kushner, T. E. De Hart, J. P. McDaniel, R. A. Pctr, and J. J. Ewing, AppL Phys. Lett 43, 1574 ( 1986).
4 J. W. Oerritsen and G. J Ernst, Appl. Phys. B 46, 141 ( !988). 'R. S. Taylor and K. E. Leopold, J. AppL Phys. 65, 22 ( 1989).
105
IV .3 Additional results and discussion.
The circuit diagram fig.2b of the paper was built and tested. The saturable inductor IA
consists of two magnetic switches in parallel, each magnetic switch has four parallel
windings (see section 11.5, fig.11.21). To achieve a low inductance conneetion between the
PFN and the laser head after saturation of the ferrite, we constructed the PFN as a
stripline with two current return strips {see figure IV.2).
The magnetic switches are made from microwave ferrite blocks (SEI type K4). Since only
blocks of ferrite were available, we improvised a core in which the air gaps were reduced
by taping the blocks together. The copper strips were insulated by sheets of mylar. In
order to use the full flux swing of the core, we reset the saturable inductor by a bias
current from an extemal circuit.
Prepul se Moinpulse
laser
Fig.IV.2 Experimentalset-up ofthe MIRVOC-system including the auxiliary components.
The magnetic switch was constructed with two tums. This has the advantage over the
one-tum contiguration that less ferrite is needed, so the core losses and the magnetizing
current will be halved. Naturally, the disadvantage is the higher self-inductance. Since the
cross-section of the ferrite is halved and not all the fluxes of the two tums are coupled, the
inductance will increase by less than a factor of two (see section 11.2). Therefore it is still
advantageous to use a two turn configuration.
106
As the saturable inductor is only a part of the total self-inductance between the PFN and
the laser head it tumed out that the total self-inductance differed by only a few percent
when the saturable inductor is operated with one or two tums . We calculated the
inductance between the PFN and the laser head from the measured charging time of the
peaking capacitor c3; the inductance tumed out to be approximately 30 nH.
Measurements have been performed on the circuit described in fig.2b (of the paper) and
figure IV.2. The voltage over the laser electrodes was measured with a resistive divider,
the current through the saturable inductor L4 and in the prepulse circuit were measured by
means of self-integrating Rogowski-coils; the current through the peaking capacitor c3 was measured with a resistive shunt.
The first experiments were performed without charging the PFN, to study the charging of
the peaking capacitors and the characteristics of the magnetic switch. Figure IV.3 shows
the measured waveforms.
2.4
9 1.6
> ""' 4.5 0.8 <
.>::
QJ
"" ..... "' 0 0
0 c: QJ .... ....
> ;::> u
-4.5
(a) Time [ 50 ns/div l -
Fig.IV.3 Measured wavefonns of the MIRVOC circuit without charging the PFN. Voltage
across the laser electrades (drawn line), current through the saturable
inductor L4
(.-.-.-.). currellt through the peaking capacitor c3
(-----),
and current through the prepulse circuit( ..... ) as aJunetion of time.
The condusion can be made that the current through the magnetic switch L4 is completely
blocked during the charging of the peaking capacitors. When the inductor L4 saturates, the
voltage swings from ""'-5.6 kV to +5.3 kV. Note that in the final circuit the current
diagnostics had to be removed to reduce the inductance of the connections and to improve
the performance.
107
In the next experiment the PFN is charged to ""' 2*U88
in approximately 1 p.s. To isolate
the charged PFN from its power supply, we introduced another saturable inductor (L5 in
fig.IV.2) so that the PFN remains at a voltage of 2*U88
for some time ( -400 ns). This
inductor had to be added because of the inevitable jitter (typical value 100 ns) of the spark
gaps s1 and s2 and the spark gaps in the generator of the X-ray source. Figure IV.4
shows the time dependenee of the voltage waveforms of the discharge and the X-ray
generator. From this figure, the experimental problems, related to the jitter of the spark
gaps will be clear.
We found experimentally that the delay time between the prepulse and the X-ray putse is a
critical parameter in the operating performance. If the delay time is too long,
photo-triggering of the laser by the X-ray generator may occur, while at very short delay
times, the preionization density is too low when the voltage is applied to the laser.
