IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 5, MAY 2011 1377

An Integrated-Controlled AC/DC Interface forMicroscale Wind Power Generation Systems

Yu-Lin Juan, Member, IEEE

Abstract—In this paper, a novel ac/dc interface with an inte-grated controller is proposed for the microscale wind power gener-ation system (WPGS). The proposed interface is mainly composedof a dynamic maximum power point tracking (MPPT) control anda half-controlled single-stage rectifier with an integrated control.The half-controlled single-stage rectifier is proposed to replace thewidely used two-stage converter for improving the current totalharmonics distortion (THD) as well as the efficiency. The analyticexpressions of the duty ratios for controlling the rectifier are alsoderived. Moreover, the dynamic response of the wind turbine andthe extracted wind power are enhanced by the integrated dynamicMPPT control. Finally, from the experimental results, it can beseen that the current THD is reduced to around 5%, and the totalefficiency is increased by about 12%–15% depending on the windspeed variations.

Index Terms—Maximum power point tracking (MPPT), single-stage rectifier, wind power generation system (WPGS).

I. INTRODUCTION

AMONG the various attractive renewable energy sources,the wind power generation system (WPGS) is the fastest

growing renewable power system in the recent years. Comparedwith the large-scale WPGS, the small-scale ones may be moresuitable for the urban environment because of space limitationand safety considerations. In a recent small wind turbine globalmarket study made by American Wind Energy Association in2008 [1], the small wind turbine market in U.S. grew 14% anddeveloped additional capacity 9.7 MW in 2007. The microscaleWPGS is usually defined as a subset of the small-scale oneswith capacity less than 1 kW. A microscale WPGS normallyconsists of a fixed-pitch micro wind turbine, a permanent mag-net synchronous generator (PMSG), an ac/dc interface, a batterymodule, and a dc load [2]–[4]. One of the two main tasks of theac/dc interface is to convert the three-phase ac power into dcpower, and the other one is to control the wind turbine withproper operation modes, such as maximum power point track-ing (MPPT). Normally, a two-stage converter consists of a fulldiode bridge rectifier, and a dc converter is adopted for a mi-croscale WPGS for cost reduction and easy implementation.However, the current total harmonics distortion (THD) result-ing from the full diode bridge rectifier is quite significant and

Manuscript received March 17, 2010; revised August 09, 2010; acceptedSeptember 13, 2010. Date of current version June 22, 2011. Recommended forpublication by Associate Editor J. M. Guerrero.

The Author is with the Department of Electrical Engineering, NationalChanghua University of Education, Changhua, Taiwan 50856, China (e-mail:yljuan0815@gmail.com).

Digital Object Identifier 10.1109/TPEL.2010.2081378

cannot be neglected especially for heavy load [5]. Also, the ef-ficiency will be reduced because of the conduction losses of thediodes. Among the various MPPT control strategies of the windturbine [4], [6]–[11], the well-known sensorless optimal torquecontrol algorithm is widely adopted. However, the wind turbineis able to extract maximum wind power only when the windturbine is under steady-state operation point because the well-known optimal torque command is derived from the steady-stateoperation points. Moreover, due to the inherent mechanic inertiaand the time varying wind speed, the actual speed of the PMSGcannot exactly and immediately track the optimal command.Therefore, the mistracking will make the wind turbine unableto fully extract maximum wind power when the wind speed israpidly changing.

In view of the aforementioned drawbacks, first, a half-controlled rectifier is adopted to replace the two-stage convertercomposed of a diode bridge rectifier and a dc converter. In ad-dition, an integrated control is also proposed to improve the ef-ficiency and the current THD. Third, to overcome the drawbackof the existing MPPT control, a dynamic MPPT control with ad-justable virtual inertia is adopted without using any mechanicalsensors [12]. It turns out that the ac/dc power-conversion effi-ciency can be increased greatly with the proposed single-stagerectifier. Also, the efficiency of the wind turbine can be increasedabout 2%–5%, depending on the wind speed variations. As a re-sult, the efficiency of the total system can be improved by about12%–15%.

II. AC/DC INTERFACE OF THE MICROSCALE WPGS

Fig. 1 shows the configuration of the proposed microscaleWPGS. It can be seen that the system is composed of a verticalaxial wind turbine with fixed pitch angle, a PMSG, a single-stagerectifier, a battery module, and a dc load. In the microscale ap-plications, the dc load may be an inverter for supplying the acpower. Fig. 2 shows the block diagram of the controller forthe proposed ac/dc interface. The controller is basically com-posed of two major blocks. One is the dynamic MPPT controllerfor maximizing the extracted wind power, and the other is theproposed integrated control of the single-stage ac/dc rectifierfor improving efficiency and current harmonics distortion. Byadopting the dynamic MPPT controller, the dynamic response ofthe wind turbine is improved as well as the amount of extractedwind energy during wind speed variations.

The aerodynamic power extracted by the wind turbine can berepresented as follows:

Pw =12ρACpv

3w (1)

0885-8993/$26.00 © 2011 IEEE

1378 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 5, MAY 2011

Fig. 1. Configuration of the proposed microscale WPGS.

Fig. 2. Block diagram of the proposed dynamic MPPT controller for themicroscale WPGS.

where Cp is the power coefficient of the wind turbine, vw is thewind speed, ρ is the air density, and A is the effective area sweptby the wind turbine. In a microscale system, because the pitchangle of the turbine is usually fixed, the power coefficient Cp

can basically be expressed as a function of the tip speed ratio(TSR) δ of the wind turbine

Cp(δ) =N∑

i=0

niδi (2)

δ =rωm

vw(3)

where ni are real constants, N is the order of the approximatepolynomial, r is the radius of the blade, and ωm is the rotorangular speed. It is well known that if the TSR is controlled tothe optimal value δopt , then the power coefficient will achievethe maximum value Cp,max . Therefore, the wind turbine willbe able to capture the maximum aerodynamic power from theair stream. The maximum extracted power can be expressed asfollows:

Pw,max =ρr3ACp,max

2δ3opt

ω3m = kω3

m . (4)

Also, the optimal torque at the maximum power point be-comes

Tw,opt =Pw,max

ωm= kω2

m . (5)

Then, the mechanical dynamic equation of the direct drivewind turbine is given as follows:

Tw − Tem = Jdωm

dt+ Bωm (6)

where Tw is the input mechanical torque from the extractedaerodynamic power, Tem is the electromagnetic torque of thegenerator, J is the total inertia, and B is the effective frictioncoefficient. The electromagnetic torque of the generator can beexpressed as follows [13]:

Tem =Pem

ωm=

3EI

2ωm=

3Pλf

4I (7)

where Pem is the electromagnetic power of the generator, E =Pωm λf /2 is the amplitude of the back emf, λf is the fluxlinkage, P is the pole number, and I is the amplitude of thegenerator output currents.

III. SENSORLESS DYNAMIC MPPT CONTROLLER

In this paper, a sensorless dynamic MPPT controller withadjustable virtual inertia is adopted to further improve the dy-namic response of the mechanical system and the dynamic ef-ficiency of the wind turbine. The block diagram of the sensor-less dynamic MPPT controller with adjustable virtual inertia isshown in Fig. 2. A two-phase-type phase-locked loop (PLL)algorithm [14], [15] is adopted as the phase frequency estimator(PFE) in Fig. 2 to estimate the rotor speed and the phase angle ofthe generator without using any mechanical sensors. The blockdiagram of the PFE is shown in Fig. 3(a).

JUAN: AN INTEGRATED-CONTROLLED AC/DC INTERFACE FOR MICROSCALE WIND POWER GENERATION SYSTEMS 1379

Fig. 3. Block diagrams of (a) PFE, and (b) ITE.

Once the PLL is locked, the quadrature signals cos θ and sin θgenerated by the quadrature signal generator (QSG) will be inphase with the input signals vα and vβ , respectively. Moreover,the phase difference Δθ will remain constant because the an-gular frequency of the output quadrature signals from QSG isdecided by Δθ. Therefore, the electrical angular frequency ωe ,and the rotor angular frequency ωm can then be obtained by

ωe =Δθ

Ts(8)

ωm =Δθ

Ts

2P

(9)

where Ts is the digital sampling period.To obtain the information about the input torque from the

wind power, an input torque estimator (ITE) is integrated inthe MPPT controller, as shown in Fig. 2. The block diagram ofthe ITE is shown in Fig. 3(b). If the input torque Tw in (6) istaken as an unknown variable, the estimation of Tw can thenbe expressed according to the algorithm of the reduced orderestimator expressed in [16] as follows:

˙Tw =

h

J(J ˙ωm + Bωm + Tem − Tw ) (10)

where h is the loop gain of the input torque estimation, and Tw

is the estimated input torque. The resulting dynamic equationof the estimation error can then be expressed by

˙Tw = −h

JTw (11)

where Tw = Tw − Tw denotes the estimation error of the inputtorque. In practice, an additional variable x = Tw − hωm is

Fig. 4. Torque commands of the DOT and SOT controls.

integrated in the estimator, as shown in Fig. 3(b), to avoid usinga differentiator.

Next, in the dynamic MPPT controller, a generator currentcommand (GCC) with the following is used to calculate theoptimal generator current reference:

I∗ =4

3Pλf[g(Tw − kω2

m ) + kω2m ] (12)

where g denotes the adjusting coefficient of the virtual inertia,which is a negative adjustable parameter. Basically, there are twoterms in the right side of (12). The first term is the compensationcomponent, which actually is used to adjust the virtual inertiaof the wind turbine for improving the dynamic response. Thesecond term, in fact, is the traditional optimal torque reference. Itcan be seen that the first term will become zero when the windturbine is operated at maximum power point, and meanwhilethe output of GCC is equal to the traditional optimal torquereference. Finally, because the dynamic response speeds of theestimators and the current controller are usually faster enoughthan that of the mechanical system, ωm can then be reasonablyassumed equal to ωm . Hence, one can get the correspondinggenerator output torque as follows:

Tem ∼= 3Pλf

4I∗ = g(Tw − kω2

m ) + kω2m . (13)

The mechanical dynamic equation of the wind turbine can bederived from (6) and (13) as follows:

Tw − kω2m = J ′ dωm

dt+ B′ωm (14)

where J ′ = J/(1 − g) and B′ = B/(1 − g) are the virtual-inertia and the virtual friction coefficient, respectively. It is seenthat the effective inertia of the mechanical system can now beadjusted to improve the mechanical dynamic response. Fig. 4shows the torque command loci of the proposed dynamic opti-mal torque control (DOT) and the steady optimal torque (SOT)control with respect to the rotor speed when the wind speed ischanging from vw1 to vw2 and vice versa.

1380 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 5, MAY 2011

Fig. 5. Defined intervals in one line cycle.

IV. PROPOSED INTEGRATED CONTROLLER OF THE AC/DCRECTIFIER

In a three-phase three-wires system with continue conductionmode (CCM) operation, all three phase currents are simultane-ously controllable if at least two phase currents are controllableat the same time. However, for the half-controlled rectifier, therewill be only one phase current controllable in some specific in-tervals of one line cycle, such as intervals B, D, and E shown inFig. 5 [17]. It turns out that the rest two phase currents will be outof control and significantly distorted. To overcome such a disad-vantage, the well-known discontinue conduction mode (DCM)power-factor-correction (PFC) technique [18] may be adoptedto control the rectifier. In [12], a quasi-synchronous rectification(QSR) technique is proposed to further reduce the conductionlosses caused by the body diodes of the active switches. How-ever, the turned-off switching losses of the switches are stillsignificant due to the large peak current in the DCM operation.

As a compromise between efficiency and controller complex-ity, an integrated control strategy composed of CCM operationin intervals A, C, and E, and DCM operation in the rest intervalsis proposed in this paper. According to the observed phase an-gle of the source voltage, the mode flag (MF) determined by thecontrol mode detector (CMD) is used to decide the operationmode, as shown in Fig. 2. Basically, for CCM control, as shownin Fig. 6(a), the phase currents ia and ib are first sensed andtransformed to synchronous reference frame, i.e., id and iq , forscalar control algorithm with the d–q axes decoupling. Then, thecorresponding modulation index of each phase may be obtained.However, because there are only two phase currents controllablein intervals A, C, and E, the duty ratios of the correspondingactive switches should be specifically modified for achievingthree-phase equivalent-balanced modulation indices.

Interval A is taken as an example to describe the modificationof the modulation indices in CCM operation. In this interval, thetwo positive phase voltages ea and ec are expressed as follows:

ea = E sin θe (15)

ec = E sin(

θe +2π

3

). (16)

Fig. 6. Block diagrams of (a) CCM controller and (b) QSR-DCM controller.

The corresponding two phase currents may be well controlledto the reference commands, which are given by

ia ∼= i∗a = I sin θe (17)

ic ∼= i∗c = I sin(

θe +2π

3

). (18)

The state-averaged dynamic equations are also expressed asfollows:

ea = Ltdiadt

+ Rsia + vAn (19)

ec = Ltdicdt

+ Rsic + vC n (20)

vAn = m1vdc =2vA0 − vC 0

3(21)

vC n = m3vdc =−vA0 + 2vC 0

3(22)

vA0 = davdc = (1 − da)vdc (23)

vC 0 = dcvdc = (1 − dc)vdc (24)

where Lt = Ls + L is the effective total inductance, m1 andm3 are the equivalent modulation indices of phases a and c, andda and dc are the duty ratios of the respective active switchesQ1 and Q3 . From (21)–(24), it is seen that the duty ratiosda anddc need to be modified from the equivalent modulation indicesm1 and m3 for properly controlling the switches Q1 and Q3 .The modification and analytic expressions of the correspondingduty ratios are given as follows:

da = 1 − (2m1 + m3) (25)

dc = 1 − (m1 + 2m3) (26)

m1 = M sin(θe − φ) (27)

m3 = M sin(

θe − φ +2π

3

)(28)

JUAN: AN INTEGRATED-CONTROLLED AC/DC INTERFACE FOR MICROSCALE WIND POWER GENERATION SYSTEMS 1381

TABLE IDUTY RATIOS IN THE CCM CONTROL INTERVALS.

where M =√

(E − RsI)2 + (ωeLtI)2/vdc and φ = tan−1

((ωeLtI)/(E − RsI)). In this interval, switch Q2 is alwaysturned on to reduce the conduction loss. According to the sym-metry of the three-phase balanced system, the closed form of thecorresponding duty ratios of all CCM intervals are summarizedin Table I.

In the other three intervals, the controller will be switchedto the DCM control mode with quasi-synchronous rectification,as shown in Fig. 6(b). All the phase currents are controlled inDCM, and the peak value of each phase current is proportionalto corresponding phase voltage automatically. After filtering outthe switching-frequency ripple components, each phase currentwill be nearly sinusoidal and approximately in phase with thecorresponding phase voltage.

The synchronous rectification technique is widely applied inlow-voltage high-current applications for reducing the diodeconduction losses [19]. Based on the idea of the synchronousrectification technique, the QSR technique is used to extend theduty ratios of the corresponding switches to reduce the conduc-tion losses of the body diodes. According to the relationshipbetween each phase current, each DCM control interval can befurther divided into two subintervals, as shown in Fig. 5. Thesubinterval B1 is taken as an example to illustrate the operationprinciple and derive the corresponding duty ratios of the DCMcontrol with the quasi-synchronous rectification technique. Thecurrent waveforms of the three boost inductors in subintervalB1 are shown in Fig. 7, and there are four operation modes forthe DCM control. In the first operation mode, t0 < t ≤ t1 allthe three active switches are turned on simultaneously to in-crease the energy stored in the three boost inductors (L). Thepeak current of the three boost inductors can then be obtainedas follow:

Ip =dTs

LV (29)

where Ip = [I1p I2p I3p ]T and V = [van vbn vcn ]T , andd = (t1 − t0)/Ts is the duty ratio of this operation mode, whichis determined by the current controller shown in Fig. 6(b). Inthe next operation mode t1 < t ≤ t2 , Q1 is turned off for trans-

Fig. 7. Current waveforms of the boost inductors in interval B1.

ferring the stored energy in the inductors to the dc side, and theother two switches are still turned on for reducing the conductionlosses. While inductor current i3 decreases to zero, active switchQ3 will be turned off. The time ratio of the second operationmode can then be determined as follows:

(t2 − t1)Ts

=−3dvc

vdc + 3vc. (30)

Therefore, the duty ratio of active switches Q1 and Q3 canbe derived as follows:

da = d (31)

dc = da +(t2 − t1)

Ts=

dvdc

vdc + 3vc. (32)

1382 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 5, MAY 2011

TABLE IIDUTY RATIOS IN THE DCM CONTROL MODE WITH QSR.

In the third mode t2 < t ≤ t3 , Q2 is still on and Q3 remainsoff, the time period of this mode can be expressed as follows:

(t3 − t2) =−2dTsvdc(vc − vb)

(vdc + 3vc)(van − vbn − vdc). (33)

In the last mode t3 < t ≤ t4 , all the inductor currents are zeroand all switches are off. Then, the duty ratio of switch Q2 canbe obtained as follows:

db = dc +(t3 − t2)

Ts=

dvdc

vdc − vab. (34)

Finally, the corresponding duty ratios for controlling thethree switches in the DCM control intervals with the quasi-synchronous rectification are given in Table II.

V. EXPERIMENTAL RESULTS

To evaluate the validity and performance of the proposedac/dc interface for a microscale WPGS, a prototype systemis implemented and some experimental results are given inthis section. The microscale vertical axis wind turbine DS-200 is manufactured by HiEnergy Technology Co. Ltd. TheMOSFETs IXFH150N15P, and Schottky diodes DSSK60–02A, are adopted as the active switches and diodes of the proposedrectifier. To precisely measure the experimental data, a precisionpower analyzer WT3000, manufactured by Yokogawa ElectricCorporation, is used. The parameters of the prototype systemare given as follows.

Radius of the wind turbine r = 0.5mRated Power Prated = 200wRated wind speed vw,rated = 12.5m/sMaximum power coefficient Cp,max = 0.282Optimal TSR δopt = 3.53Generator side inductor Ls = 300μH

Fig. 8. Practically measured curves of the generator output torque with theDOT- and the SOT-MPPT controls.

Stator resistance Rs = 0.2 ΩBoost inductor L = 100μHFilter capacitor Cf = 14.7μFSwitching frequency in CCM fsc = 60 kHzSwitching frequency in DCM fsd = 10 kHzOutput DC voltage vdc = 100VFig. 8 shows the generator output torque with respect to the

rotor speed under wind speed variation with the sensorless dy-namic MPPT control. It is seen from Fig. 8 that when the windis changing from low speed to high speed, the torque commandof the DOT controller will be made smaller instantly than thatof the SOT controller to increase the torque difference for fasteracceleration. Similarly, the torque command will be made largerinstantly if the wind velocity is suddenly slowed down. Fig. 9

JUAN: AN INTEGRATED-CONTROLLED AC/DC INTERFACE FOR MICROSCALE WIND POWER GENERATION SYSTEMS 1383

Fig. 9. Experimental results under pulse changing wind speed. (a) Generatoroutput torque and (b) power coefficient.

Fig. 10. Experimental results under pulse changing wind speed. (a) Generatoroutput power and (b) energy.

shows the experimental results of generator output torque andpower coefficient in the time domain, the solid line curves rep-resent the results of the proposed system. It is seen that as thewind speed steps up to a higher speed, the generator outputtorque with the dynamic controller will be instantly controlledto be smaller than that with the SOT controller. On the contrary,as the wind speed steps down, the torque command will then becontrolled immediately to a larger value for quicker decelerationto the optimal operating point. Fig. 9(b) shows the dynamic re-sponse of the power coefficient that is approximately fitted witha polynomial. It can be seen that the recovery speed to the maxi-mum power coefficient with the dynamic MPPT control is muchfaster than that with SOT control. Also from (1), it is obviousthat the wind power extracted by the blades will be proportionalto the power coefficient under the same wind speed. Therefore,more wind power can be captured as the power coefficient can bekept closer to the maximum value. The generator output powerand energy under the pulse changing wind speed are shown inFig. 10. In fact, for this case, approximately 4.5% more windenergy can be obtained. Next, consider the comparison of theperformance of the proposed single-stage rectifier with that ofa two-stage converter. Due to the nonlinearity of the full diodebridge rectifier, the phase current will contain serious currentharmonics distortion, as shown in Fig. 11(a). It can be seen thatthe harmonics distortion of the phase current of the proposed

Fig. 11. Measured current waveforms of (a) conventional WPGS and(b) proposed WPGS (1 A/div, 5 ms/div). C: CCM, D: DCM.

Fig. 12. Experimental data of the two-stage converter and the single-stagerectifier with proposed control. (a) current THD and (b) conversion efficiency.

WPGS is greatly reduced, as can be observed from Fig. 11(b).Fig. 12 shows the efficiency and the current THD of the pro-posed rectifier and the conventional two-stage converter. Fromthe experimental results, one can see that the proposed single-stage rectifier with the proposed integrated control is able toreduce the current THD to about 5% and increase the efficiencyabout 11%. In summary, compared with the two-stage systemwith SOT control, the proposed WPGS can further increase the

1384 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 5, MAY 2011

system efficiency from about 12% –15%, depending on the windspeed variations.

VI. CONCLUSION

In this paper, a single-stage rectifier with an integrated controlis proposed for a microscale WPGS to replace the traditionaltwo-stage converter. There is no shooting through problem inthe proposed half-controlled rectifier. Compared with the two-stage converter, both the current THD and efficiency of therectifier with proposed integrated control strategy are greatlyimproved. Moreover, to overcome the drawback of the existingMPPT control, a novel sensorless dynamic MPPT control isalso adopted to further increase the extracted wind power underrapidly changing wind speed. In the dynamic MPPT control,a virtual inertia control is integrated with the traditional SOTMPPT control to improve the mechanical dynamic response.Finally, to evaluate the performance of the proposed system,some experimental results are also given. From the experimentalresults, it can been seen that the current THD is reduced to about5%, and the system efficiency can be improved by about 12%–15% compared with the two-stage system with SOT control.

REFERENCES

[1] AWEA (2008, Jun.), “AWEA Small Wind Turbine Global Market Study2008,” in the American Wind Energy Association Small Wind Systems[Online]. Available: http://www.awea.org/smallwind/

[2] E. Koutroulis and K. Kalaitzakis, “Design of a maximum power track-ing system for wind-energy-conversion applications,” IEEE Trans. Ind.Electron., vol. 53, no. 2, pp. 486–494, Apr. 2006.

[3] M. Malinowski, S. Stynski, W. Kolomyjski, and M. P. Kazmierkowski,“Control of three-level PWM converter applied to variable-speed-typeturbines,” IEEE Trans. Ind. Electron., vol. 56, no. 1, pp. 69–77, Jan.2009.

[4] Z. Chen, J. M. Guerrero, and F. Blaabjerg, “A review of the state of theart of power electronics for wind turbines,” IEEE Trans. Power Electron.,vol. 24, no. 8, pp. 1859–1875, Aug. 2009.

[5] P. Tenca, A. A. Rockhill, T. A. Lipo, and P. Tricoli, “Current source topol-ogy for wind turbines with decreased mains current harmonics, furtherreducible via functional minimization,” IEEE Trans. Power Electron.,vol. 23, no. 3, pp. 1143–1155, May 2008.

[6] A. M. Knight and G. E. Peters, “Simple wind energy controller for anexpanded operation range,” IEEE Trans. Energy Convers., vol. 20, no. 2,pp. 459–466, Jun. 2005.

[7] S. Morimoto, H. Nakayama, M. Sanada, and Y. Takeda, “Sensorless outputmaximization control for variable-speed wind generation system usingIPMSG,” IEEE Trans. Ind. Appl., vol. 41, no. 1, pp. 60–67, Jan./Feb.2005.

[8] F. Valenciaga and P. F. Puleston, “Supervisor control for a stand-alonehybrid generation system using wind and photovoltaic energy,” IEEETrans. Energy Convers., vol. 20, no. 2, pp. 398–405, Jun. 2005.

[9] K. Tan and S. Islam, “Optimum control strategies in energy conversion ofPMSG wind turbine system without mechanical sensors,” IEEE Trans.Energy Convers., vol. 19, no. 2, pp. 392–399, Jun. 2004.

[10] W. Qiao, W. Zhou, J. M. Aller, and R. G. Harley, “Wind speed estimationbased sensorless output maximization control for a wind turbine drivinga DFIG,” IEEE Trans. Power Electron., vol. 23, no. 3, pp. 1156–1169,May 2008.

[11] B. Shen, B. Mwinviwiwa, Y. Zhang, and B. T. Ooi, “Sensorless maximumpower point tracking of wind by DFIG using rotor position phase lockloop (PLL),” IEEE Trans. Power Electron., vol. 24, no. 4, pp. 942–951,Apr. 2009.

[12] C. T. Pan and Y. L. Juan, “A novel sensorless MPPT controller for ahigh efficiency micro-scale wind power generation system with adjustablevirtual inertia,” IEEE Trans. Energy Convers., vol. 25, no. 1, pp. 207–216,Mar. 2010.

[13] P. Vas, Vector Control of AC Machines. New York: Oxford UniversityPress, 1990.

[14] T. Emura, L. Wang, M. Yamanaka, and H. Nakamura, “A high-precisionpositioning servo controller based on phase/frequency detecting techniqueof two-phase-type PLL,” IEEE Trans. Ind. Electron., vol. 47, no. 6,pp. 1298–1306, Dec. 2000.

[15] T. Emura and L. Wang, “A high-resolution interpolator for incremental en-coders based on the quadrature PLL method,” IEEE Trans. Ind. Electron.,vol. 47, no. 1, pp. 84–90, Feb. 2000.

[16] G. F. Franklin, J. D. Powell, and A. Emami-Naeini, Feedback Control ofDynamic Systems. New Jersey: Prentice Hall, 2002.

[17] J. Kikuchi, M. D. Manjrekar, and T. A. Lipo, “Performance improvementof half controlled three phase PWM boost rectifier,” in Proc. 30th Annu.IEEE Power Electron. Spec. Conf., 1999, pp. 319–324.

[18] A. R. Prasad, P. D. Ziogas, and S. Manias, “An active power factor cor-rection technique for three-phase diode rectifiers,” IEEE Trans. Power.Electron., vol. 6, no. 1, pp. 83–92, Jan. 1991.

[19] H. Mao, O. A. Rahman, and I. Batarseh, “Zero-voltage-switching DC–DC converters with synchronous rectifiers,” IEEE Trans. Power Electron.,vol. 23, no. 1, pp. 369–378, Jan. 2008.

Yu-Lin Juan (M’08) was born in Kaohsiung, Tai-wan, China, in 1979. He received the B.S. degree inelectrical engineering from the National Chen KungUniversity, Taiwan, in 2001, and the M.S. and Ph.D.degrees majoring in electrical engineering from theNational Tsing Hua University, Taiwan, in 2003 and2010, respectively.

He is currently an Assistant Professor in theDepartment of Electrical Engineering, NationalChanghua University of Education, Taiwan. His cur-rent research interests include power electronics, re-

newable energy systems, and ac motor drives.