Philips - High Voltage Bipolar Transistor

7/30/2019 Philips - High Voltage Bipolar Transistor

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Introduction Power Semiconductor ApplicationsPhilips Semiconductors

High Voltage Bipolar Transistor

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1.3.1 Introduction To High Voltage Bipolar Transistors

This section introduces the high voltage bipolar transistorand discusses its construction and technology. Specifictransistor properties will be analysed in more detail insubsequent sections and in Chapter 2, section 2.1.2.

Basic Characteristics

High voltage transistors are almost exclusively used aselectronic switches. Therefore, the characteristics of thesedevices are given for the on state, the off state and thetransition between the two i.e. turn-on and turn-off.

Therelative importance of the VCES andVCEO ratings usually

depends on the application. In a half bridge converter, forinstance, the rated VCEO is the dominant factor, whilst in aforward converter VCES is important. Which rating is mostapplicable mayalso dependon whether a slow rise networkor snubber is applied (see section 1.3.3).

The saturation properties in the on state and the switchingtimes are given at a specific collector current called thecollector saturation current, ICsat. It is this current which isnormally considered to be the practical working current ofthe device. If this device is used at higher currents the totaldissipation may be too high, while at low currents thestorage time is long. At ICsat thebest compromise is presentfor the total spread of products. The value of the base

current used to specify the saturation and switchingproperties of the device is called IBsat which is also animportant design parameter. As the device requirementscandiffer per application a universal IBsat cannot be quoted.

Device Construction

A drawing of a high voltage transistor, in this case a fullyisolated SOT186 F-pack, is shown in Fig. 1 with the plasticencapsulation stripped away. This figure shows the threeleads, two of which are connected with wires to thetransistor chip. The third lead makes contact with themounting base on which the crystal is soldered, enabling

good thermal contact with a heatsink. It is the transistorpackage whichbasically determines the thermal propertiesof the device. The electrical properties are mainlydetermined by the design of the chip inside.

A cross-section of a transistor chip is given in Fig. 2. Herethe transistor structure can be recognised with the emitterand the base contacts at the top surface and the collectorconnected to the mounting base. The thickest part in thedrawing is the collector n- region across which the highvoltage will be supported in the off state. This layer is of

Fig. 1 Cut-away View of a High Voltage Transistor

primeimportance in thedetermination of thecharacteristicsof the device. Below the n- region is an extra n+ layer,needed for a good electrical contact to the heatsink.

Fig. 2 Cross-section of a High Voltage Transistor

Above the collector is the base p layer, and the emitter n+layer with their respective metallic contacts on top. It isimportant to realise that the characteristics of the deviceare determined by the active area, this is the areaunderneath the emitter where the collector current flowsand the high voltage can be developed. The active area oftwo devices with the same chip size may not be the same.

nickel-platedcopper lead

frame

passivatedchip

aluminium

wires

tinned copperleads

ultrasonicwire bonds

Base Collector Emitter

base emitter

250V

600V

850V

1150V

n-

n+

n+p

n+

n-

special glass

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Fig. 3 Maximum Voltages vs. n- Collector Thickness

N+

P

N-

N+

N+

P

N-

N+

N+

P

N-

N+

TIP49

450 V

BUT11

850 V

BU2508A

1500 V

In addition to the basic collector-base-emitter structuremanufacturers have to add electrical contacts, and special

measures are needed at the edges of the crystal to sustainthe design voltage. This introduces another very importantfeature, the high voltage passivation. The function of thepassivation,(theexampleshownhere isreferred toasglasspassivation), is to ensure that the breakdown voltage of thedevice is determined by the collector-base structure andnot by the construction at the edges. If no specialpassivation was used the breakdown voltage might be aslowas 50%of themaximum value. Manufacturers optimisethe high voltage passivation and much work has also beendone to ensure that its properties do not change in time.

Process Technology

There are several ways to make the above structure. Thestarting material can be an n- wafer where first an n+diffusion is made in the back, followed by the base (p) andemitter (n+) diffusions. This is thewell known triple diffusedprocess.

Another way is to start with an n+ wafer onto which an n-layer is deposited using epitaxial growth techniques. Afurther two diffusions (base and emitter) forms the basictransistor structure. This is called a double diffusedepitaxial process.

Another little used technology is to grow, epitaxially, thebase p-type layer onto an n-/n+ wafer and then diffuse ann+ emitter. This is referred to as a single diffused epi-basetransistor.

The question often asked is which is the best technologyfor high voltage bipolar transistors ? The basic differencein the technologies is the concentration profile at the n-/n+

junction. For epitaxial wafers the concentration gradient ismuch more steeper from n- to n+ than it is for back diffusedwafers. There are more applications where a smootherconcentration gradient gives the better performance.Manufacturers utilising epitaxial techniques tend to usebuffer layers between the n- and n+ to give smoother

concentration gradients. Another disadvantage of epitaxialprocessing is cost: back diffused wafers are much cheaper

than equivalent high voltage epitaxial wafers.

The process technology used to create the edgepassivation isalsodiverse. Theexpression"planar" isusedto indicate the passivation technique which is mostcommonly used in semiconductors. This involves thediffusion of additional n-type rings around the active areaof the device which give an even electric field distributionat the edge. However, for high voltage bipolar transistorsplanar passivation is relatively new and the long termreliability has yet to be completely optimised. For highvoltage bipolar transistors the most common passivationsystems employ a deep trough etched, or cut, into the

device with a special glass coating. Like the planarpassivation, the glass passivation ensures an evendistribution of the electric field around the active area.

Maximum Voltage and Characteristics

Fig. 4 Switching Times and hFE vs. VCEO

200 400 800

15

25

5010

5

2.5

30 60 120hFEsat hFE0Width of n- layer (um)

0.8

0.4

0.2 1.5

3

6

tf ts

Vceo (V)

(us)

hFE

ts, tf

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High voltage and low voltage transistors differ primarily inthethickness andresistivityof then- layer. As thethicknessand resistivity of this layer is increased, the breakdownvoltage goes up. The difference over the range of Philipshigh voltage transistors of different voltages is illustrated in

Fig. 3. The TIP49 has a VCBO = 450 V, the BUT11 has aVCES = 850 V, while the BU2508A can be used up tovoltages of 1500 V.

The penalty for increasing the n- layer is a decrease inhighcurrent hFE and an in switching times. The graph in Fig. 4points this out by giving both switching times and hFE as afunctionof the breakdownvoltage. The values givenshouldbe used as a guide to illustrate the effect. The effect canbe compensated for by having a bigger chip.

Applications of High Voltage Transistors

High voltage transistors are mainly used as the power

switch in energy conversion systems. What is common to

allthesesystems, is thata current flowsthroughaninductor,thus storing energy in its core. When the current isinterrupted by turning off thepower switch, theenergy mustbe transferred one way or another. Very often the energyis converted into an electrical output e.g. in switched mode

power supplies and battery chargers.

Two special applications are electronic fluorescent lampballasts and horizontal deflection of the electron beam inTVsandmonitors. In theballast,an ac voltage isgeneratedto deliver energy to a fluorescent lamp. In the TV andmonitor a sawtooth current in the deflection coil sweepsthebeam across the screen from left to right and back again ina much shorter blanking, or flyback, period

Other ways to transfer the energy are ac and dc motorcontrol where the output is delivered as movement, orinduction heating where the output is delivered in the form

of heat.

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1.3.2 Effects of Base Drive on Switching Times

Introduction

The switching processes that take place within a highvoltage transistor are quite different from those in a smallsignal transistor. This section describes, figuratively, whathappens within high voltage transistors under various basedrive conditions. After an analysis of the charges that arepresent in a high voltage transistor, the switch-off processis described. Then comparisons are made of switching forvarious forward and reverse base drive conditions. Afundamental knowledge of basic semiconductor physics isassumed.

Charge distribution within a transistorAn off-state transistor has no excess charge, but to enabletransistor conduction in the on-state excess charge buildup within the device takes place. There are three distinctcharge distributions to consider that control the currentthrough the device, see Fig. 1. These charge distributionsare influenced by the level of collector-emitter bias, VCE,and collector current, IC, as shown in Fig. 2.

Forward biasing the base-emitter (BE) junction causes adepletion layer to form across the junction. As the biasexceeds thepotentialenergybarrier (work function) for that

junction, current will flow. Electrons will flow out of theemitter into the base and out of the base contact. For highvoltage transistors the level of BE bias is much in excessof the forward bias for a small signal transistor. The biasgenerates free electron-hole pairs in the base-emitterleading toa concentration ofelectrons in thebase in excessof the residualholeconcentration. Thisproducesanexcesscharge in the base, Qb, concentrated underneath theemitter.

Fig. 1. On-state Charge Flow

Not only is there an excess charge in the base near the

emitter junction but the injectionandbase width ensure thatthis excess charge is also present at the collector junction.Applying a load in series with the collector and a dc supplybetween load and emitter will trigger some sort of collectorcurrent, IC. The level ofIC is dependent on the base current,IB, the load and supply voltage. For a certain IB, low voltagesupply and high impedance load there will bea small IC. Asthe supply voltage rises and/or the load impedance falls soIC will rise. As IC rises so the collector-emitter voltage, VCE,falls. The IC is composed mainly of the excess emitterelectrons that reach the base-collector junction (BC). Thiselectron concentration will continue into the collectorinducing an excess charge in the collector, Qc.

Theconcentration of electrons decreasesonly slightly fromthe emitter-base junction to some way into the collector. Ineffect, thebasewidthextendsinto thecollector. DecreasingVCE below VBE causes the BC junction to become forwardbiased throughout. This creates a path for electrons fromthe collector to be driven back into the base and out of thebase contact. This electron flow is in direct opposition tothe established IC. With no change in base drive, theultimate effect is a reduction in IC. This is the classicalsaturation region of transistor operation. As VCE falls sothe BC forward bias increases leading to an excess ofelectrons at the depletion layer edge in the collectorbeneath the base contact. This concentration of electrons

leads to an excess charge, Qd.

The charge flows and excess charges Qb, Qc and Qd areshown in Fig. 1. An example of the excess chargedistributions for fixed IC and IB are shown in Fig. 2.

Fig. 2. On-state Charge Distribution (example)

The switching process of a transistor

Removing thebiasvoltage,VBE, will cause theelectron-holepairs to recombine and the excess charge regions todisappear. Allowing this to happen just by removing VBE

QQd

Qc

Qb

Vce (V)

Ic = 5 A

Ib = 1 A

B E B

Qd Qc

Qb P

N+

N-

N+

C

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takes a long time so usually turn-off is assisted in someway. It is common practice to apply a negative bias(typically 5V) to the base, via a resistor and/or inductor,inducing a negative current that draws the charge out ofthe transistor. In the sequence that follows, four phases of

turn-off can be distinguished (see Fig. 3).

1. First the applied negative bias tries to force a negativebias across the BC junction. The BC electron flow nowstops and the charge Qd dissipates as the biasnow causesthe base holes out through the base contact and thecollector electrons back into the bulk collector. When theBC was forward biased this current had the effect ofreducing thetotal collectorcurrent, so now thenegative VBEcan cause the total collector current to increase (this alsodependsontheload). Although thebasehas been switchedoff the load current is maintained by the stored chargeeffects; this is called the transistor storage time, ts.

During this stage the applied negative bias appears as apositive VBE at the device terminals as the internal chargedistributions create an effective battery voltage. Depletingthe charge, of course, lowers this effective battery voltage.

2. The next phase produces a reduction in both Qb, Qcand, consequently, IC. The BC junction isno longerforwardbiased and Qd hasdissipated to provide the negative basecurrent. The inductance in series in the base path requiresa continuation in thebasecurrent. Theinjectionof electronsinto the base opposes the established electron flow from

emitter to collector via the base. At first the opposingelectron flows cancel at the edge of the emitter nearest thebase contacts. This reduces bothQb andQc in this region.QbandQc becomeconcentrated in thecentreof theemitterarea. The decrease in IC is called the fall time, tf.

3. Now there is an extra resistance to the negative basecurrent as the electrons flow through the base under theemitter area. This increase in resistance limits the increasein amplitude of the negative base current. As Qb and Qcreducefurther so the resistance increasesandthe negativebase current reaches its maximum value.

As Qb and Qc tend to zero the series inductance ensuresthat negative base current must be continued by othermeans. The actualmechanismis by avalanchebreakdownof the base-emitter junction. This now induces a negativeVBE which is larger than the bias resulting in a reverse inpolarity of the voltage across the inductance. This in turntriggers a positive rate of change in base current. Thenegative base current now quickly rises to zero while thebase-emitter junction is in avalanche breakdown.Avalanchebreakdown ceaseswhen thebase current tendsto zero and the VBE becomes equal to the bias voltage.

Fig. 3. Phases during turn-off

B E B

Qd Qc

Qb P

N+

N-

N+

C

B E B

Qc

Qb P

N+

N-

N+

C

B E B

P

N+

N-

N+

C

0Qb

Qc 0

B E B

P

N+

N-

N+

C

Qr

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4. If a very small series base inductor is used with the 5Vreverse bias then the base current will have a very fast rateof change. This will speed up the phases 1 to 3 and,therefore, the switching times of the transistor. However,there is a point when reducing the inductor further

introduces another phase to the turn-off process. Highreverse base currents will draw the charges out closest tothe base contact and leave a residualcharge trapped deepin the collector regions furthest away from the base. Thischarge, Qr, must be removed before the transistor returnsfully to the off-state. This is detected as a tail to IC at theend of turn-off with a corresponding tail to the base currentas it tends to zero.

The switching waveforms for a BUT11 in a forwardconverter are given in Fig. 4 where the four phases caneasilybe recognised. (Because of thesmall base coil usedboth phases in the fall time appear clearly!).

1 - Removal of Qd until t 0.7 s ts2 - Qc and Qb decrease until t 1.7 s ts

3 - Removal of Qb and Qc until t 1.75 s tf

4 - Removal of Qr until t 1.85 s tf

Note the course of VBE: first the decrease in voltage due tothe base resistance during current contraction and second(because a base coil has been used) the value of VBE isclamped by the emitter-base breakdown voltage of thetransistor. It should be remembered that becausebreakdown takes place near the surface and not in theactive region no harm comes to the transistor.

Fig. 4. BUT11 waveforms at turn-off

The influence of forward drive on storedcharge

Fig. 5 shows how, for a transistor in the on-state, at a fixedvalue of IC and IB the three charges Qb, Qc and Qd dependupon VCE. The base charge, Qb, is independent of VCE, itprimarily depends upon VBE. For normal base drive

conditions, a satisfactory value for VCEsat is obtained,indicated by N in Fig. 5, and moderate values for Qc andQd result.

Fig. 5. Charges as a function of VCE

With the transistor operating in the active region, forVCE 1V, there will be a charge Qc but no charge Qd. Thisis indicated by D in Fig. 5. At the other extreme, with thetransistoroperatingin thesaturationregion Qcwillbehigherand Qd will be higher than Qc. This is indicated by O inFig. 5. In thisconditionthereare moreexcess electron-holepairs to recombine at switch off.

Increasing IB causes Qb to increase. Also, for a given IC,Qc and Qd will be higher as VCE reduces. Therefore, for agivenIC, thestoredcharge inthetransistor canbe controlledby the level of IB. If the IB is too low theVCE will be high withlow Qc and zero Qd, as D in Fig. 5. This condition is calledunderdrive. If the IB istoohightheVCE will be low with highQc and Qd, as O in Fig. 5. This condition is calledoverdrive. The overdrive condition (high forward drive)gives high stored charge and the underdrive condition (lowforward drive) gives low stored charge.

Deep-hole storage

As the high free electron concentration extends into thebase and collector regions ther must be an equivalent holeconcentration. Fig. 6 shows results obtained from acomputer model which illustrates charge storage as afunction of VCE. Here the hole density, p(x), is given as afunction of depth inside the active area; the doping profileis also indicated. It can be seen that overdrive, O, causesholes to be stored deep in the collector at the collector -substrate junction known as "deep-hole storage", this is themain reason for the increase in residual charge, Qr.

Q Qd

Qc

Qb

0.2 0.5 1.0 Vce (V)

O N D

Ic

Vce

IbVbe

1 A/div

1 A/div

200 V/div

5 V/div

0.5 us/div

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During overdrive not only Qd becomes very big but alsoholes are stored far away from the junction: this thus leadsnot only to a longer storage time, but also to a large Qrresulting in tails in the turn-off current.

Fig. 6. Deep hole storage in the collector region

Desaturation networks

A desaturationnetwork,as shown in Fig. 7, limits thestoredcharge in the transistor and, hence, aids switching. Theseries base diode, D1 means that the applied drive voltagenowhastobeVBE plusthe VF of D1. Theanti-parallel diode,D2 is necessary for the negative IB at turn-off. As VCEreduces below VBE + VF so the external BC diode, D3,becomes forward biased. D3 now conducts any furtherincrease in drive current away from the base and into thecollector. Transistor saturation is avoided.

With a desaturation network thecharge Qd equalszero andthe charge Qc is minimised. When examining thedistribution of the charge in the collector region (see Fig. 6)it can be seen that deep hole storage does not appear.Desaturation networks are a common technique forreducing switching times.

It should be realised that there is a drawback attached tooperatingoutof saturation: increaseddissipationduring theon-state. Base drive design often requires a trade-offbetween switching and on-state losses.

Fig. 7. Desaturation network(Baker clamp)

Breakdown voltage vs. switching times

For a higher breakdown voltage transistor the n- layer (seeFig. 1) will be thicker and of higher resistivity (ie a lowerdonor atom concentration). This means that when

comparing identical devices the values for Qd and Qc willbe higher, for a given IC, in the device with the higherbreakdown voltage.

In general:

- the higher BVCEO the larger Qd and Qc will be;

- during overdrive Qd is very high and there is a chargelocated deep in the collector region (deep hole storage);

- when desaturated Qd equals zero and there is no deephole storage: Qc is minimised for the IC.

Turn-off conditions

Various ways of turning off a high voltage transistor areused but the base should always be switched to a negativesupply via an appropriate impedance. If this is not done,(ie turn-off is attempted by simply interrupting the basecurrent), very long storage times result and the collectorvoltage increases, while the collector current falls onlyslowly. A very high dissipation and thus a short lifetime ofthetransistor are the result. Thecharges must be removedusing a negative base current.

a) Hard turn-off

The technique widely used, especially for low voltagetransistors, is to switch directly to a negative voltage, (seeFig. 8a). In the absence of a negative supply, this can beachieved with an appropriate R-C network (Fig. 8b). Alsoapplying an "emitter-drive" (Fig. 8c) with a large basecapacitor in fact is identical to hard-turn-off.

The main drawback for high voltage transistors is that thebase charge Qb is removed too quickly, leaving a highresidual charge. This leads to current tails (long fall times)and high dissipation. It depends upon what state thetransistoris in(overdrivenor desaturated),whether thiswayof turn-off isbest. Italso depends upon thekind of transistor

that must be switched off. If it is a lower voltage transistor(BVCEO 200V) then this will work very well because thecharges Qc and Qd will be rather low. For transistors witha higher breakdown voltage, hard turn-off will yield theshorteststorage time at thecost, however, of higherturn-offdissipation (longer tf).

b) Smooth turn-off

To properly turn-off a high voltage transistor a storage timetominimiseQd andQc isrequired,and thena largenegativebase current to give a short fall time.

0 80 1604020 60 120100 140

20

10

12

14

16

18

10

10

10

10

10

10

Vce = 1 V 0.5 V 0.2 V

p(x) at J = 140 A / cm2

p(x)

x (um)

E BC

D N O

B

C

E

D3

D1

D2

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Fig. 8. Hard turn-off

-V

+Ib

++

Lc

++

Lc+V

C

R

++

Lc

+V

+I

R

C

(a) (b) (c)

The easiest way to obtain these turn-off requirements is toswitch the base to a negative supply via a base coil, seeFig. 9.

The base coil gives a constant dIB/dt (approx.) during thestorage time. When the fall time begins the negative basecurrent reaches its maximum and the Lb induces the BE

junction into breakdown (see Fig. 4).

An optimum value exists for the base coil: if Lb = 0 we have

thehard turn-offcondition which isnotoptimumforstandardhigh voltage transistors. If thevalue ofLb is toohigh it slowsthe switching process so that the transistor desaturates.The VCE increases too much during the storage time andso higher losses result (see Fig. 10).

For high voltage transistors in typical applications (f = 15 to40 kHz, standard base drive, not overdriven, notdesaturated) the following equationsgive a good indicationfor the value of Lb.

Using - Vdr = 5V, VBEsat = 1V and transistors havingBVCEO = 400V it follows that:

c) Other ways of turn-off

Of course, other ways of turn-off are applicable but ingeneral these can be reduced to one of the methodsdescribedabove, or something in between. The BVCEO hasa strong influence on the method used: the higher BVCEOthe longer the storage time required to achieve properturn-off. For transistors havinga BVCEO of200Vor lesshardturn-off and the use of a basecoil yield comparable losses,so hard turn-off works well. For transistors having BVCEO

more than 400V hard turn-off is unacceptable because ofthe resulting tails.

Fig. 9. A base coil to aid turn-off.

+Ib

++

Lc

-Vdr

LB

=(Vdr + VBEsat)

dIB

dt

with

dIB

dt 0.5 IC (A /s ) for BV CEO = 400V, BVCES = 800V

an ddIB

dt 0.15 IC (A/s) for BV CEO = 700V, BVCES = 1500V

LB

=12

ICH (I

Ci n Amps)

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Fig. 10. Variations of Lb on IC and VCE waveforms at turn-off

IcVce

IcVce Ic Vce

Lb = 0 Lb = opt Lb > Lb opt

Turn-off for various forward driveconditions

Using the BUT11 as an example, turn-off characteristicsare discussed for optimum drive, underdrive and overdrivewith hard and smooth turn-off.

a) Optimum drive

The optimum IB and Lb for a range of IC is given in Fig. 11for the BUT11. The IB referred to is IBend which is the valueof IB at the end of the on-state of the applied base drivesignal. In most applications during the on-state the IB willnot be constant, hence the term IBend rather than IBon. Foroptimumdrive thelevelof IBend increaseswithIC. Forsmooth

turn-off the level of Lb decreases with increasing IC.

Fig. 11. IBend and Lb for the BUT11

Deviations from Fig. 11 will generally lead to higher powerdissipation. If a short storage time is a must in a certainapplication then Lb can be reduced but this will lead tolonger fall times and current tails.

With hard turn-off IB reaches its peak negative value as allthe charge is removed from the base. For continuity thiscurrentmustbesourcedfromelsewhere. Ithasbeenshownthat the BE junction now avalanches, giving instantaneouscontinuity followed by a positive dIB/dt. However, for hardturn-off the current is sourced by the residual collectorcharge without BE avalanche, see Fig. 12. The smallnegative VBE ensures a long tail to IC and IB.

b) Underdrive (Desaturated drive)

As has been indicated previously, desaturating, orunderdriving, a transistor results in less internal charge. Qdwill be zero and Qc is low and located near the junction.

If the application requires such a drive then steps should

be taken to optimise the characteristics. One simple wayof obtaining underdrive is to increase the series baseresistance with smooth turn-off. The same effect can beachieved with optimum IBend and a base coil having half thevalue used for optimum drive, ie hard turn-off. Bothmethods give shorter ts and tf. For 400V BVCEO devices (likethe Philips BUT range) such a harder turn-off can lead toreasonable results.

Fig. 13 compares the use of the optimum base coil withhard turn-off for an undriven BUT11. For underdrive thefinal IC is less and hence the collector charge is less.Therefore, underdrive and hard turn-off gives less of a tail

thanfora higher IBend. Underdrivewith smooth turn-off giveslonger ts but reduced losses.

c) Overdrive

When a transistor is severely overdriven the BC charge,Qd, becomes so large that a considerable tail will resulteven with smooth turn-off. In general, deliberatelydesigninga drivecircuit tooverdrivea transistor isnotdone:ithas noreal value. However,most circuitsdohave variablecollector loadswhich canresult in extreme conditions whenthe circuit is required to operate with the transistor inoverdrive.

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Fig. 12. Optimum drive with hard turn-off (top)and smooth turn-off (bottom) for BUT11

Fig. 13. Underdrive with hard turn-off (top)and smooth turn-off (bottom) for BUT11

Fig. 14. Overdrive with hard turn-off (top)and smooth turn-off (bottom) for BUT11

Fig. 14 compares the use of the optimum base coil withhard turn-off for an overdriven BUT11. For overdrive thereis more base charge, also the final collector current will be

higherand, hence, there will be more collectorcharge. Theoverdriventransistor is thencertain tohavelongerswitchingtimes as there are more electron-hole pairs in the devicethat need to recombine before the off-state is reached.

Conclusions

Two ways of turning off a high voltage transistor, hardturn-off and the use of a base coil, were examined in threeconditions of the on-state: optimum drive, overdrive andunderdrive.

For transistors having BVCEO ~ 400 V the use of a base coilyields low losses compared to hard turn-off. As a good

approximation the base coil should have the value:

for optimum drive.

When using a desaturation circuit the value for Lb can behalved with acceptable results.

Overdrive should be prevented as much as possiblebecause considerable tails in the collector current causeunacceptable losses.

Ic Vce

Ib

Vbe

1 A/div

1 A/div

200 V/div

5 V/div

0.5 us/div

Ic Vce

Ib

Vbe

1 A/div

1 A/div

200 V/div

5 V/div

0.5 us/div

Ic

Vce

IbVbe

1 A/div

1 A/div

200 V/div

5 V/div

0.5 us/div

IcVce

Ib Vbe

1 A/div

1 A/div

200 V/div

5 V/div

0.5 us/div

Ic Vce

Ib

Vbe

1 A/div

1 A/div

200 V/div

5 V/div

0.5 us/div

Ic Vce

Ib Vbe

1 A/div

1 A/div

200 V/div

5 V/div

0.5 us/div

LB =

12

ICH

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1.3.3 Using High Voltage Bipolar Transistors

This section looks at some aspects of using high voltagebipolar transistors in switching circuits. It highlights pointssuchasswitching, both turn-on andturn-off,Safe OperatingAreas and the need forsnubber circuits. Base drive designcurves for the BUT11, BUW12 and BUW13 are discussedunder Application Information at the end of this section.

Transistor switching: turn-on

To make optimum use of todays high voltage transistors,one should carefully choose the correct value for both thepositive base current when the transistor is on and the

negative base current when the device is switched off (seeApplication Information section).

When a transistor is in the off-state, there are no carriersin the thick n- collector, effectively there is a resistor with arelatively high value in the collector. To obtain a lowon-state voltage, a base current is applied such that thecollector area is quickly filled with electron - hole pairscausing the collector resistance to decrease. In thetransition time, the so called turn-on time, the voltage andcurrent may both behigh, especially in forward converters,and high turn-on losses may result. Initially, all the carriersin the collector will be delivered via the base contact and,therefore, the base current waveform should have a peakat the beginning. In this way the carriers quickly fill thecollector area so the voltage is lower and the lossesdecrease.

In flyback converters the current to be turned on is normallylow, but in forward converters this current is normally high.Thecollectorcurrent, IC, reaches itson-statevalue ina shorttime which is normally determined by the leakageinductance of the transformer.

Fig. 1 Turn-on of a high voltage bipolar transistor

In Fig. 1 the characteristic hump which often occurs atturn-on in forward converters due to the effect of thecollector series resistance is observed.

The turn-on losses are strongly dependent on the value ofthe leakage inductance and the applied base drive. It isgenerally advised to apply a high initial +IB for a short timein order to minimise turn on losses.

A deeper analysis can be found in sections 1.3.2, 2.1.2 and2.1.3. Turn on losses are generally low for flybackconverters but are the most important factor in forwardconverter types.

Turn-off of high voltage transistors

All charge stored in the collector when the transistor is onshould be removed again at turn-off. To ensure a quickturn-offa negativebase current isapplied. Thetimeneededto remove the base - collector charge is called the storagetime. A short storage time is needed to minimise problemswithin the control loop in SMPS anddeflection applications.

Fig. 2 Effects of -IB on turn-off

Care is needed to implement the optimum drive. Firstoverdriveshouldbepreventedby keeping+IB toa minimum.Overdrive results in current tails and long storage times.But, decreasing IB too much results in high on-state losses.

Second, the negative base current should be chosen

carefully. A small negative base current (-IB) willgive a longstorage timeand a high VCEsat at the end of the storage time,while thecurrent is still high. As a consequence,the turn-offlosses will be high. If, however, a large negative basecurrent is used, the danger exists that tails will occur in thecollector current, again resulting in high losses. There isan optimum as shown in Fig. 2.

A circuit which is worth considering, especially for higherfrequencies, is the Baker Clamp or desaturation circuit.This circuit prevents saturation of thetransistor and, hence,faster switching times are achieved.

Ic Vce Ic Vce IcVce

-Ib is too high -Ib is optimum -Ib is too low

Ic

Vce

Ic = 1 A/div

Vce = 50 V/div

91


16/41


The total losses depend on the base drive and the collectorcurrent. In Fig.3 the total losses are shown for a BUW133as a function of the positive base current, for both thesaturated and the desaturated case. Note that whendifferent conditions are being used the picture will change.

The application defines the acceptable storage time whichthen determines the base drive requirements.

Fig. 3 BUW133 losses versus base drive

The total number of variables is too large to give uniquebase drive advice for each application. As a first hint thedevice data sheets give IC and IB values for VCEsat, VBEsat andswitching. However, it is more important to appreciate the

ways to influence base drive and the consequences of anon-optimised circuit.

For a flyback converter the best value of IBend to start withis about 2/3 of the IB value given in data for VCEsat and VBEsat.In this application the forward base current is proportionalto the collector current (triangular shaped waveforms) andthis IBend value will give low on-state losses and fastswitching.

The best turn-off base current depends on the breakdownvoltage of the transistor. As a guide, Table 1 givesreasonable values for the target storage time and may beused to begin optimising the base drive:

f (kHz) tp (s) target ts (s)

25 20 2.0

150 10 1.5

100 5 1.0

Table 1 Target ts for varying frequency and pulse width

The above table holds for transistors with a VCEOmax ratingof 400-450V and VCESmax between 850-1000V. Transistorswith higher voltages require longer storage times, eg.

transistors with VCEOmax = 700V and VCESmax = 1500V needa storage time which is approximately double the value inthe table.

A recommended way to control the storage time is by

switching the base to a negative voltage rail via a base coil.The leakage inductance of a driver transformer may serveas an excellent base coil. As a guide, the base coil shouldbe chosen such that the peak value of the negative basecurrent equals half the value of the collector current.

Specific problems and solutions

A high voltage transistor needsprotection circuits to ensurethat the device will survive all the currents and voltages itwill see during its life in an application.

a) Over Current

Exceeding current ratings normally does not lead to

immediate transistor failure. In the case of a short circuit,the protection is normally fast enough for problems to beavoided. Most devices are capable of carrying very highcurrents for short periods of time. High currents will raisethe junction temperature and if Tjmax is exceeded thereliability of the device may be weakened.

b) Over Voltage

In contrast with over current, it is NOT allowed to exceedthe published voltage ratings for VCEO and VCES (or VCBO).In switching applications it is commonfor the base - emitter

junction to be taken into avalanche, this does not harm thedevice. For this reason VEBO limits are not given in data.

Exceeding VCEO and VCES causes high currents to flow invery small areas of the device. These currents may causeimmediate damage to the device in very short times(nanoseconds). So, even for very short times it is notallowed to have voltages above data for the device.

In reality VCEO and VCES are unlikely to occur in a circuit. IfVBE = 0V the there will probably still be a path between thebase and the emitter. In fact the situation is VCEX where Xis the impedance of this path. To cover for all values of X,the limit is X=, ie VCEO. For all VBE < 0V, ie VCEV, the limitcase is VBE = 0V, ie VCES.

If voltage transients that exceed the voltage limits are

detected then a snubber circuit may limit the voltage to asafe value. If the over voltage states last greater than afew s a higher voltage device is required.

c) Forward Bias Safe Operating Areas (FBSOA)

The FBSOA is valid for positive values of VBE. There is atime limit to VCE - IC operating points beyond which devicefailure becomes a risk. At certain values of VCE and IC thereis a risk of secondary breakdown; this is likely to lead to theimmediate failure of the device. The FBSOA curve shouldonly be considered during drastic change sequences; forexample, start-up, s/c or o/c load.

With Baker Clamp

Saturated

100

200

300

400

500

600

700

Etot (uJ)

1 2 3 4

Ib (A)

Forward Converter

Ic = 10 A

92


17/41


d) Reverse Bias Safe Operating Area (RBSOA)

The RBSOA is valid for negative values of VBE. Duringturn-off with an inductive load the VCE will rise as the IC falls.For each device type there is a VCE - IC boundary which, if

exceeded, will lead to the immediate failure of the device.Tolimit the VCE - IC path at turn-off snubber circuitsareused,see Fig. 4.

Fig. 4 HVT with inductive load and typical snubber

At turn-off, as the VCE rises the diode starts conductingcharging the capacitor. Theadditionaldiodecurrent meansthat the total load current does not decrease so fast atturn-off. This slower current tail in turn ensures a slowerVCE rise. The slower VCE rise takes the transistor througha safer VCE - IC path away from the limit, see Fig. 5.

As a handy guide, the snubber capacitor in a 20-40 kHzconverter is about 1nF for each 100W of throughput power(this is the power which is being transferred via the

transformer). This value may be reduced empirically asrequired.

Fig. 5 BUW13A RBSOA limitVCE - IC path with and without snubber

The following table may serve as a guide to the value ofdVCE/dt for some switching frequencies

f (kHz) 25 50 100

dVCE/dt 1 2 4(kV/s)

Thesnubberresistorshould bechosensothat thecapacitorwill be discharged in the shortest occurring on-time of theswitch.

In some cases the losses in the snubber may beconsiderable. Clever designs exist to feed the energy

stored in the capacitor back into the supply capacitor, butthis is beyond the scope of this report.

5

10

15

20

200 400 600 800 1000

BUW13A

Ic

Vce

Without Snubber

With Snubber

Vs

Fig. 6 Transistor with maximum protection networks in SMPS circuit

R5

C5 C4

D4

D5

D6

R4TR1

L6R6

D3 D1

D2

Lo

Co Vo

Vi

93


18/41


d) Other protection networks

In Fig. 6 a "maximum protection" diagram is shown withvarious networks connected. R4, C4, and D4 form thesnubber to limit the rate of rise of VCE. The network with

D5, R5 and C5 forms a "peak detector" to limit the peakVCE.

The inductor L6 serves to limit the rate of rise of IC whichmaybe very high forsometransformerdesigns. TheslowerdIC/dt leads to considerably lower turn-on losses. Addedto L6 is a diode D6 and resistor R6, with values chosen sothat L6 loses its energy during the off-time of the powerswitch.

While the snubber is present in almost all SMPS circuitswhere transistors are used above VCEOmax, the dIC/dt limiteris only needed when the transformer leakage inductanceis extremely low. The peak detector is applied in circuitswhich have bad coupling between primary and secondarywindings.

Application Information

Importantdesign factors of SMPS circuitsarethemaximumpower losses, heatsink requirements and base driveconditions of the switching transistor. The power lossesare very dependent on the operating frequency, the

maximum collector current amplitude and shape.

Theoperating frequency is usually between 15 and50 kHz.The collector current shape varies from rectangular in aforward converter to sawtooth in a flyback converter.

Examplesof base drive and losses are given in Appendix 1for the BUT11, BUW12 and BUW13. In these figures ICMrepresents the maximum repetitive peak collector current,which occurs during overload. The information is derivedfrom limit-case transistors at a mounting base temperatureof 100 C under the following conditions (see also Fig. 7):

- collector current shape IC1 / ICM = 0.9- duty factor (tp/ T) = 0.45- rate of rise of IC during turn-on = 4 A/s- rate of rise VCE during turn-off = 1 kV/s- reverse drive voltage during turn-off = 5 V- base current shape IB1 / IBe = 1.5

The required thermal resistance of the heatsink can becalculated from

Toensurethermal stabilitya maximum value of theambienttemperature, Tamb, is assumed: Tamb 40C.

Rth(mb amb) 0V it is FBSOA and if VBE < 0Vit isRBSOA. Chapter 2.1.3 dealswith both subjects in moredetail, a few of the main points are covered below.

FBSOA gives boundaries for dc or pulsed operation. Inswitching applications, where the transistor is "on" or "off",normally the excursion in the IC-VCE plane is fast enough toallow the designer to use the whole plane, with theboundaries ICmax and VCEO, as given in the ratings. This isuseful for snubberless applications and for overload, faultconditions or at switch-on of the power supply

Fig. 4 gives the FBSOA of the BUT11 with the boundariesof ICmax, ICMmax and VCEO, all as given in the ratings. Thereis a Ptotmax (1) and ISB boundary (2), that both shift at higherlevels of IC when shorter pulses are used. Note that in theupper right hand corner pulse times of 20s are permittedleading to a square switching SOA. For overload, faultconditionor power supply switch-on an extra area is added(area III). All these conditions are for VBE 0V.

IC

IB

10 %

10 %

90 %90 %

tontoff

tstf

IBon

-IBoff

ICon

IC

IB

ICon

IBon

-IBoff

t

ttstf

toff

10 %

90 %

99


24/41


(1) Ptotmax and Ptotpeak max. lines(2) Second breakdown limits (independent oftemperature).I Region of permissible dc operation.II Permissible extension for repetitive pulseoperationIII Area of permissible operation during turn-onin single transistor converters, providedRBE 100 and tp 0.6 s.IV Repetitive pulse operation in this region ispermissible provided VBE 0 V and tp 5 ms.Fig. 4 Safe Operating Area of BUT11.

Area IV is only valid for VBE 0V, so this is an RBSOAextension to the SOA curve. This is not the full picture forRBSOA, area IV is only for continuous pulsed operation.For single cycle and short burst fault conditions see theseparate RBSOA curve.

The RBSOA curve is valid when a negative voltage isapplied to the base-emiiter terminals during turn-off. Thiscurve should be used for fault condition analysis only;continuous operation close to the limit will result in 100s Wof dissipation ! Due to localised current contraction within

the chip at turn-off, damage will occur if the limit isexceeded. In nearly all cases, the damage will result in theimmediate failure of the device to short circuit.

Emitter switching applications force different mechanisms

for carrier recombination in the device which allow asquare RBSOA. A typical example is shown in Fig. 5,where for both base and emitter drive the RBSOA of theBUT11 is given.

Fig. 5 RBSOA of BUT11 for Base and Emitter Drive.

It is striking that for emitter drive the whole IC-VCES planemay be used so no snubber is necessary, however, a smallsnubber may prevent overshoot. The base drive RBSOAnormally depends on base drive conditions, butunfortunately there is no uniform trend in this behaviour.

Therefore, the RBSOA curve in the data gives the worstcase behaviour of the worst case devices. Other datasheetsmaygive RBSOA curvesthat at firstsight look betterthan the Philips equivalent, but beware, these curves mighthold for only a limited base drive range.

Summary

Voltage limiting values / ratings as given in the data mustnever be exceeded, as they may lead to immediate devicefailure. Surge voltages, as sometimes given for othercomponents, are not allowed for high voltage transistors.Current limiting values / ratings are less strict as they aretime-dependent andshould be used in conjunction with theFBSOA.

Static characteristics are useful for comparisons but offerlittle in describing the performance in an application. Thedynamic characteristics may be defined for a simple testcircuit but the valuesgive a good indication of the switchingperformance in an application.

RBSOA is, for all switching applications, of primeimportance. Philips give in their data sheets a curve forworst case devices under worst case conditions. Forsnubber design a value of 1 nF per 100W of throughput

200 400 600 800 1000

Ic (A)

Vce (V)

0

1

2

3

4

5

6

7

8

base drive

emitter drive

Vcesm

100


25/41


power is advised as a starter value; afterwards, the IC-VCElocus mustbe checked to see if it stays within the publishedRBSOA curve.

Forcharacteristicsbothsaturationand switchingproperties

are given at ICsat. Most figures are of limited use as theygive static conditions, where in a practical situationproperties are time-dependent. Switching times are given

in relatively simple circuits that may be replicated rathereasily e.g. for incoming inspection.

Switching times depend strongly on drive conditions. By

altering them a normal device can be turned into a superdevice. Beware of specmanship, this may disguise poortolerance to variations in base drive.

101


26/41

Preface Power Semiconductor ApplicationsPhilips Semiconductors

Acknowledgments

We are grateful for all the contributions from our colleagues within Philips and to the Application Laboratories in Eindhovenand Hamburg.

We would also like to thank Dr.P.H.Mellor of the University of Sheffield for contributing the application note of section 3.1.5.

The authors thank Mrs.R.Hayes for her considerable help in the preparation of this book.

The authors also thank Mr.D.F.Haslam for his assistance in the formatting and printing of the manuscripts.

Contributing Authors

N.Bennett

M.Bennion

D.Brown

C.Buethker

L.Burley

G.M.Fry

R.P.Gant

J.Gilliam

D.Grant

N.J.Ham

C.J.Hammerton

D.J.Harper

W.Hettersheid

J.v.d.Hooff

J.Houldsworth

M.J.Humphreys

P.H.Mellor

R.Miller

H.Misdom

P.Moody

S.A.Mulder

E.B.G. Nijhof

J.Oosterling

N.Pichowicz

W.B.Rosink

D.C. de Ruiter

D.Sharples

H.Simons

T.Stork

D.Tebb

H.Verhees

F.A.Woodworth

T.van de Wouw

This book was originally prepared by the Power Semiconductor Applications Laboratory, of the Philips Semiconductorsproduct division, Hazel Grove:

M.J.Humphreys

C.J.Hammerton

D.Brown

R.Miller

L.Burley

It was revised and updated, in 1994, by:

N.J.Ham C.J.Hammerton D.Sharples


27/41

Preface Power Semiconductor ApplicationsPhilips Semiconductors

Preface

This book was prepared by the Power Semiconductor Applications Laboratory of the Philips Semiconductors productdivision, Hazel Grove. Thebook is intended as a guide to using power semiconductors both efficiently and reliably in powerconversion applications. It is made up of eight main chapters each of which contains a number of application notes aimedat making it easier to select and use power semiconductors.

CHAPTER 1 forms an introduction to power semiconductors concentrating particularly on the two major power transistortechnologies, Power MOSFETs and High Voltage Bipolar Transistors.

CHAPTER 2 is devoted to Switched Mode Power Supplies. It begins with a basic description of the most commonly usedtopologies and discusses the major issues surrounding the use of power semiconductors including rectifiers. Specificdesign examples are given as well as a look at designing the magnetic components. The end of this chapter describesresonant power supply technology.

CHAPTER 3 describes motion control in terms of ac, dc and stepper motor operation and control. This chapter looks onlyat transistor controls, phase control using thyristors and triacs is discussed separately in chapter 6.

CHAPTER 4 looks at television and monitor applications. A description of the operation of horizontal deflection circuits isgiven followed by transistor selection guidesforboth deflection andpower supply applications. Deflection and power supply

circuitexamplesarealso givenbasedon circuitsdesignedby theProduct ConceptandApplication Laboratories (Eindhoven).

CHAPTER 5 concentrates on automotive electronics looking in detail at the requirements for the electronic switches takinginto consideration the harsh environment in which they must operate.

CHAPTER 6 reviews thyristor andtriac applicationsfrom thebasicsof device technology andoperation to the simpledesignrules which should be followed to achieve maximum reliability. Specific examples are given in this chapter for a numberof the common applications.

CHAPTER 7 looks at the thermal considerations for power semiconductors in terms of power dissipation and junctiontemperature limits. Part of this chapter is devoted to workedexamplesshowing howjunction temperaturescanbecalculatedto ensure the limits are not exceeded. Heatsink requirements and designs are also discussed in the second half of thischapter.

CHAPTER 8 is an introduction to the use of high voltage bipolar transistors in electronic lighting ballasts. Many of the

possible topologies are described.


28/41

Contents Power Semiconductor ApplicationsPhilips Semiconductors

Table of Contents

CHAPTER 1 Introduction to Power Semiconductors 1

General 3

1.1.1 An Introduction To Power Devices ............................................................ 5

Power MOSFET 17

1.2.1 PowerMOS Introduction ............................................................................. 19

1.2.2 Understanding Power MOSFET Switching Behaviour ............................... 29

1.2.3 Power MOSFET Drive Circuits .................................................................. 39

1.2.4 Parallel Operation of Power MOSFETs ..................................................... 49

1.2.5 Series Operation of Power MOSFETs ....................................................... 531.2.6 Logic Level FETS ...................................................................................... 57

1.2.7 Avalanche Ruggedness ............................................................................. 61

1.2.8 Electrostatic Discharge (ESD) Considerations .......................................... 67

1.2.9 Understanding the Data Sheet: PowerMOS .............................................. 69

High Voltage Bipolar Transistor 77

1.3.1 Introduction To High Voltage Bipolar Transistors ...................................... 79

1.3.2 Effects of Base Drive on Switching Times ................................................. 83

1.3.3 Using High Voltage Bipolar Transistors ..................................................... 911.3.4 Understanding The Data Sheet: High Voltage Transistors ....................... 97

CHAPTER 2 Switched Mode Power Supplies 103

Using Power Semiconductors in Switched Mode Topologies 105

2.1.1 An Introduction to Switched Mode Power Supply Topologies ................... 107

2.1.2 The Power Supply Designers Guide to High Voltage Transistors ............ 129

2.1.3 Base Circuit Design for High Voltage Bipolar Transistors in PowerConverters ........................................................................................................... 141

2.1.4 Isolated Power Semiconductors for High Frequency Power SupplyApplications ......................................................................................................... 153

Output Rectification 159

2.2.1 Fast Recovery Epitaxial Diodes for use in High Frequency Rectification 161

2.2.2 Schottky Diodes from Philips Semiconductors .......................................... 173

2.2.3 An Introduction to Synchronous Rectifier Circuits using PowerMOSTransistors ........................................................................................................... 179

i


29/41


Design Examples 185

2.3.1 Mains Input 100 W Forward Converter SMPS: MOSFET and BipolarTransistor Solutions featuring ETD Cores ........................................................... 187

2.3.2 Flexible, Low Cost, Self-Oscillating Power Supply using an ETD34Two-Part Coil Former and 3C85 Ferrite .............................................................. 199

Magnetics Design 205

2.4.1 Improved Ferrite Materials and Core Outlines for High Frequency PowerSupplies ............................................................................................................... 207

Resonant Power Supplies 217

2.5.1. An Introduction To Resonant Power Supplies .......................................... 219

2.5.2. Resonant Power Supply Converters - The Solution For Mains PollutionProblems .............................................................................................................. 225

CHAPTER 3 Motor Control 241

AC Motor Control 243

3.1.1 Noiseless A.C. Motor Control: Introduction to a 20 kHz System ............... 245

3.1.2 The Effect of a MOSFETs Peak to Average Current Rating on InvertorEfficiency ............................................................................................................. 251

3.1.3 MOSFETs and FREDFETs for Motor Drive Equipment ............................. 253

3.1.4 A Designers Guide to PowerMOS Devices for Motor Control ................... 2593.1.5 A 300V, 40A High Frequency Inverter Pole Using Paralleled FREDFETModules ............................................................................................................... 273

DC Motor Control 283

3.2.1 Chopper circuits for DC motor control ....................................................... 285

3.2.2 A switched-mode controller for DC motors ................................................ 293

3.2.3 Brushless DC Motor Systems .................................................................... 301

Stepper Motor Control 307

3.3.1 Stepper Motor Control ............................................................................... 309

CHAPTER 4 Televisions and Monitors 317

Power Devices in TV & Monitor Applications (including selectionguides) 319

4.1.1 An Introduction to Horizontal Deflection .................................................... 321

4.1.2 The BU25XXA/D Range of Deflection Transistors .................................... 331ii


30/41


4.1.3 Philips HVTs for TV & Monitor Applications .............................................. 339

4.1.4 TV and Monitor Damper Diodes ................................................................ 345

TV Deflection Circuit Examples 349

4.2.1 Application Information for the 16 kHz Black Line Picture Tubes .............. 351

4.2.2 32 kHz / 100 Hz Deflection Circuits for the 66FS Black Line Picture Tube 361

SMPS Circuit Examples 377

4.3.1 A 70W Full Performance TV SMPS Using The TDA8380 ......................... 379

4.3.2 A Synchronous 200W SMPS for 16 and 32 kHz TV .................................. 389

Monitor Deflection Circuit Example 397

4.4.1 A Versatile 30 - 64 kHz Autosync Monitor ................................................. 399

CHAPTER 5 Automotive Power Electronics 421

Automotive Motor Control (including selection guides) 423

5.1.1 Automotive Motor Control With Philips MOSFETS .................................... 425

Automotive Lamp Control (including selection guides) 433

5.2.1 Automotive Lamp Control With Philips MOSFETS .................................... 435

The TOPFET 443

5.3.1 An Introduction to the 3 pin TOPFET ......................................................... 445

5.3.2 An Introduction to the 5 pin TOPFET ......................................................... 447

5.3.3 BUK101-50DL - a Microcontroller compatible TOPFET ............................ 449

5.3.4 Protection with 5 pin TOPFETs ................................................................. 451

5.3.5 Driving TOPFETs ....................................................................................... 453

5.3.6 High Side PWM Lamp Dimmer using TOPFET ......................................... 455

5.3.7 Linear Control with TOPFET ...................................................................... 457

5.3.8 PWM Control with TOPFET ....................................................................... 4595.3.9 Isolated Drive for TOPFET ........................................................................ 461

5.3.10 3 pin and 5 pin TOPFET Leadforms ........................................................ 463

5.3.11 TOPFET Input Voltage ............................................................................ 465

5.3.12 Negative Input and TOPFET ................................................................... 467

5.3.13 Switching Inductive Loads with TOPFET ................................................. 469

5.3.14 Driving DC Motors with TOPFET ............................................................. 471

5.3.15 An Introduction to the High Side TOPFET ............................................... 473

5.3.16 High Side Linear Drive with TOPFET ...................................................... 475iii


31/41


Automotive Ignition 477

5.4.1 An Introduction to Electronic Automotive Ignition ...................................... 479

5.4.2 IGBTs for Automotive Ignition .................................................................... 481

5.4.3 Electronic Switches for Automotive Ignition ............................................... 483

CHAPTER 6 Power Control with Thyristors and Triacs 485

Using Thyristors and Triacs 487

6.1.1 Introduction to Thyristors and Triacs ......................................................... 489

6.1.2 Using Thyristors and Triacs ....................................................................... 497

6.1.3 The Peak Current Handling Capability of Thyristors .................................. 505

6.1.4 Understanding Thyristor and Triac Data .................................................... 509

Thyristor and Triac Applications 521

6.2.1 Triac Control of DC Inductive Loads .......................................................... 523

6.2.2 Domestic Power Control with Triacs and Thyristors .................................. 527

6.2.3 Design of a Time Proportional Temperature Controller ............................. 537

Hi-Com Triacs 547

6.3.1 Understanding Hi-Com Triacs ................................................................... 549

6.3.2 Using Hi-Com Triacs .................................................................................. 551

CHAPTER 7 Thermal Management 553

Thermal Considerations 555

7.1.1 Thermal Considerations for Power Semiconductors ................................. 557

7.1.2 Heat Dissipation ......................................................................................... 567

CHAPTER 8 Lighting 575

Fluorescent Lamp Control 577

8.1.1 Efficient Fluorescent Lighting using Electronic Ballasts ............................. 579

8.1.2 Electronic Ballasts - Philips Transistor Selection Guide ............................ 587

8.1.3 An Electronic Ballast - Base Drive Optimisation ........................................ 589

iv


32/41

Index Power Semiconductor ApplicationsPhilips Semiconductors

Index

Airgap, transformer core, 111, 113Anti saturation diode, 590Asynchronous, 497

Automotivefanssee motor control

IGBT, 481, 483ignition, 479, 481, 483lamps, 435, 455motor control, 425, 457, 459, 471, 475resistive loads, 442reverse battery, 452, 473, 479screen heater, 442seat heater, 442solenoids, 469TOPFET, 473

Avalanche, 61Avalanche breakdown

thyristor, 490Avalanche multiplication, 134

Baker clamp, 138, 187, 190Ballast

electronic, 580fluorescent lamp, 579switchstart, 579

Base drive, 136base inductor, 147

base inductor, diode assisted, 148base resistor, 146drive transformer, 145drive transformer leakage inductance, 149electronic ballast, 589forward converter, 187power converters, 141speed-up capacitor, 143

Base inductor, 144, 147Base inductor, diode assisted, 148Boost converter, 109

continuous mode, 109discontinuous mode, 109

output ripple, 109Bootstrap, 303Breakback voltage

diac, 492Breakdown voltage, 70Breakover current

diac, 492Breakover voltage

diac, 492, 592thyristor, 490

Bridge circuitssee Motor Control - AC

Brushless motor, 301, 303

Buck-boost converter, 110Buck converter, 108 - 109Burst firing, 537Burst pulses, 564

Capacitancejunction, 29

Capacitormains dropper, 544

CENELEC, 537Charge carriers, 133

triac commutation, 549Choke

fluorescent lamp, 580Choppers, 285Clamp diode, 117Clamp winding, 113Commutation

diode, 164Hi-Com triac, 551thyristor, 492triac, 494, 523, 529

Compact fluorescent lamp, 585Continuous mode

see Switched Mode Power Supplies

Continuous operation, 557Converter (dc-dc)switched mode power supply, 107

Cookers, 537Cooling

forced, 572natural, 570

Crest factor, 529Critical electric field, 134Cross regulation, 114, 117Current fed resonant inverter, 589Current Mode Control, 120Current tail, 138, 143

Damper Diodes, 345, 367forward recovery, 328, 348losses, 347outlines, 345picture distortion, 328, 348selection guide, 345

Darlington, 13Data Sheets

High Voltage Bipolar Transistor, 92,97,331MOSFET, 69

i


33/41


dc-dc converter, 119Depletion region, 133Desaturation networks, 86

Baker clamp, 91, 138dI/dt

triac, 531Diac, 492, 500, 527, 530, 591Diode, 6

double diffused, 162epitaxial, 161schottky, 173structure, 161

Diode Modulator, 327, 367Disc drives, 302Discontinuous mode

see Switched Mode Power SuppliesDomestic Appliances, 527Dropper

capacitive, 544resistive, 544, 545Duty cycle, 561

EFD coresee magnetics

Efficiency Diodessee Damper Diodes

Electric drill, 531Electronic ballast, 580

base drive optimisation, 589current fed half bridge, 584, 587, 589current fed push pull, 583, 587flyback, 582transistor selection guide, 587voltage fed half bridge, 584, 588voltage fed push pull, 583, 587

EMC, 260, 455see RFI, ESDTOPFET, 473

Emitter shortingtriac, 549

Epitaxial diode, 161characteristics, 163dI/dt, 164

forward recovery, 168lifetime control, 162operating frequency, 165passivation, 162reverse leakage, 169reverse recovery, 162, 164reverse recovery softness, 167selection guide, 171snap-off, 167softness factor, 167stored charge, 162technology, 162

ESD, 67see Protection, ESDprecautions, 67

ETD coresee magnetics

F-packsee isolated package

Fall time, 143, 144Fast Recovery Epitaxial Diode (FRED)

see epitaxial diodeFBSOA, 134Ferrites

see magneticsFlicker

fluorescent lamp, 580Fluorescent lamp, 579

colour rendering, 579

colour temperature, 579efficacy, 579, 580triphosphor, 579

Flyback converter, 110, 111, 113advantages, 114clamp winding, 113continuous mode, 114coupled inductor, 113cross regulation, 114diodes, 115disadvantages, 114discontinuous mode, 114electronic ballast, 582leakage inductance, 113magnetics, 213operation, 113rectifier circuit, 180self oscillating power supply, 199synchronous rectifier, 156, 181transformer core airgap, 111, 113transistors, 115

Flyback converter (two transistor), 111, 114Food mixer, 531Forward converter, 111, 116

advantages, 116

clamp diode, 117conduction loss, 197continuous mode, 116core loss, 116core saturation, 117cross regulation, 117diodes, 118disadvantages, 117duty ratio, 117ferrite cores, 116magnetics, 213magnetisation energy, 116, 117

ii


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operation, 116output diodes, 117output ripple, 116rectifier circuit, 180reset winding, 117

switched mode power supply, 187switching frequency, 195switching losses, 196synchronous rectifier, 157, 181transistors, 118

Forward converter (two transistor), 111, 117Forward recovery, 168FREDFET, 250, 253, 305

bridge circuit, 255charge, 254diode, 254drive, 262loss, 256

reverse recovery, 254FREDFETsmotor control, 259

Full bridge converter, 111, 125advantages, 125diodes, 126disadvantages, 125operation, 125transistors, 126

Gatetriac, 538

Gate driveforward converter, 195

Gold doping, 162, 169GTO, 11Guard ring

schottky diode, 174

Half bridge, 253Half bridge circuits

see also Motor Control - ACHalf bridge converter, 111, 122

advantages, 122clamp diodes, 122

cross conduction, 122diodes, 124disadvantages, 122electronic ballast, 584, 587, 589flux symmetry, 122magnetics, 214operation, 122synchronous rectifier, 157transistor voltage, 122transistors, 124voltage doubling, 122

Heat dissipation, 567

Heat sink compound, 567Heater controller, 544Heaters, 537Heatsink, 569Heatsink compound, 514

Hi-Com triac, 519, 549, 551commutation, 551dIcom/dt, 552gate trigger current, 552inductive load control, 551

High side switchMOSFET, 44, 436TOPFET, 430, 473

High Voltage Bipolar Transistor, 8, 79, 91,141, 341

bathtub curves, 333avalanche breakdown, 131avalanche multiplication, 134

Baker clamp, 91, 138base-emitter breakdown, 144base drive, 83, 92, 96, 136, 336, 385base drive circuit, 145base inductor, 138, 144, 147base inductor, diode assisted, 148base resistor, 146breakdown voltage, 79, 86, 92carrier concentration, 151carrier injection, 150conductivity modulation, 135, 150critical electric field, 134current crowding, 135, 136current limiting values, 132current tail, 138, 143current tails, 86, 91d-type, 346data sheet, 92, 97, 331depletion region, 133desaturation, 86, 88, 91device construction, 79dI/dt, 139drive transformer, 145drive transformer leakage inductance, 149dV/dt, 139

electric field, 133electronic ballast, 581, 585, 587, 589Fact Sheets, 334fall time, 86, 99, 143, 144FBSOA, 92, 99, 134hard turn-off, 86horizontal deflection, 321, 331, 341leakage current, 98limiting values, 97losses, 92, 333, 342Miller capacitance, 139operation, 150

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optimum drive, 88outlines, 332, 346over current, 92, 98over voltage, 92, 97overdrive, 85, 88, 137, 138

passivation, 131power limiting value, 132process technology, 80ratings, 97RBSOA, 93, 99, 135, 138, 139RC network, 148reverse recovery, 143, 151safe operating area, 99, 134saturation, 150saturation current, 79, 98, 341secondary breakdown, 92, 133smooth turn-off, 86SMPS, 94, 339, 383

snubber, 139space charge, 133speed-up capacitor, 143storage time, 86, 91, 92, 99, 138, 144, 342sub emitter resistance, 135switching, 80, 83, 86, 91, 98, 342technology, 129, 149thermal breakdown, 134thermal runaway, 152turn-off, 91, 92, 138, 142, 146, 151turn-on, 91, 136, 141, 149, 150underdrive, 85, 88voltage limiting values, 130

Horizontal Deflection, 321, 367base drive, 336control ic, 401d-type transistors, 346damper diodes, 345, 367diode modulator, 327, 347, 352, 367drive circuit, 352, 365, 406east-west correction, 325, 352, 367line output transformer, 354linearity correction, 323operating cycle, 321, 332, 347s-correction, 323, 352, 404

TDA2595, 364, 368TDA4851, 400TDA8433, 363, 369test circuit, 321transistors, 331, 341, 408waveforms, 322

IGBT, 11, 305automotive, 481, 483clamped, 482, 484ignition, 481, 483

Ignitionautomotive, 479, 481, 483darlington, 483

Induction heating, 53Induction motor

see Motor Control - ACInductive loadsee Solenoid

Inrush current, 528, 530Intrinsic silicon, 133Inverter, 260, 273

see motor control accurrent fed, 52, 53switched mode power supply, 107

Irons, electric, 537Isolated package, 154

stray capacitance, 154, 155thermal resistance, 154

Isolation, 153

J-FET, 9Junction temperature, 470, 557, 561

burst pulses, 564non-rectangular pulse, 565rectangular pulse, composite, 562rectangular pulse, periodic, 561rectangular pulse, single shot, 561

Lamp dimmer, 530Lamps, 435

dI/dt, 438inrush current, 438MOSFET, 435PWM control, 455switch rate, 438TOPFET, 455

Latching currentthyristor, 490

Leakage inductance, 113, 200, 523Lifetime control, 162Lighting

fluorescent, 579phase control, 530

Logic Level FETmotor control, 432Logic level MOSFET, 436

Magnetics, 207100W 100kHz forward converter, 197100W 50kHz forward converter, 19150W flyback converter, 199core losses, 208core materials, 207EFD core, 210ETD core, 199, 207

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flyback converter, 213forward converter, 213half bridge converter, 214power density, 211push-pull converter, 213

switched mode power supply, 187switching frequency, 215transformer construction, 215

Mains Flicker, 537Mains pollution, 225

pre-converter, 225Mains transient, 544Mesa glass, 162Metal Oxide Varistor (MOV), 503Miller capacitance, 139Modelling, 236, 265MOS Controlled Thyristor, 13MOSFET, 9, 19, 153, 253

bootstrap, 303breakdown voltage, 22, 70capacitance, 30, 57, 72, 155, 156capacitances, 24characteristics, 23, 70 - 72charge, 32, 57data sheet, 69dI/dt, 36diode, 253drive, 262, 264drive circuit loss, 156driving, 39, 250dV/dt, 36, 39, 264ESD, 67gate-source protection, 264gate charge, 195gate drive, 195gate resistor, 156high side, 436high side drive, 44inductive load, 62lamps, 435leakage current, 71linear mode, parallelling, 52logic level, 37, 57, 305

loss, 26, 34maximum current, 69motor control, 259, 429modelling, 265on-resistance, 21, 71package inductance, 49, 73parallel operation, 26, 47, 49, 265parasitic oscillations, 51peak current rating, 251Resonant supply, 53reverse diode, 73ruggedness, 61, 73

safe operating area, 25, 74series operation, 53SMPS, 339, 384solenoid, 62structure, 19

switching, 24, 29, 58, 73, 194, 262switching loss, 196synchronous rectifier, 179thermal impedance, 74thermal resistance, 70threshold voltage, 21, 70transconductance, 57, 72turn-off, 34, 36turn-on, 32, 34, 35, 155, 256

Motor, universalback EMF, 531starting, 528

Motor Control - AC, 245, 273

anti-parallel diode, 253antiparallel diode, 250carrier frequency, 245control, 248current rating, 262dc link, 249diode, 261diode recovery, 250duty ratio, 246efficiency, 262EMC, 260filter, 250FREDFET, 250, 259, 276gate drives, 249half bridge, 245inverter, 250, 260, 273line voltage, 262loss, 267MOSFET, 259Parallel MOSFETs, 276peak current, 251phase voltage, 262power factor, 262pulse width modulation, 245, 260ripple, 246

short circuit, 251signal isolation, 250snubber, 276speed control, 248switching frequency, 246three phase bridge, 246underlap, 248

Motor Control - DC, 285, 293, 425braking, 285, 299brushless, 301control, 290, 295, 303current rating, 288

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drive, 303duty cycle, 286efficiency, 293FREDFET, 287freewheel diode, 286

full bridge, 287half bridge, 287high side switch, 429IGBT, 305inrush, 430inverter, 302linear, 457, 475logic level FET, 432loss, 288MOSFET, 287, 429motor current, 295overload, 430permanent magnet, 293, 301

permanent magnet motor, 285PWM, 286, 293, 459, 471servo, 298short circuit, 431stall, 431TOPFET, 430, 457, 459, 475topologies, 286torque, 285, 294triac, 525voltage rating, 288

Motor Control - Stepper, 309bipolar, 310chopper, 314drive, 313hybrid, 312permanent magnet, 309reluctance, 311step angle, 309unipolar, 310

Mounting, transistor, 154Mounting base temperature, 557Mounting torque, 514

Parasitic oscillation, 149Passivation, 131, 162

PCB Design, 368, 419Phase angle, 500Phase control, 546

thyristors and triacs, 498triac, 523

Phase voltagesee motor control - ac

Power dissipation, 557see High Voltage Bipolar Transistor loss,MOSFET loss

Power factor correction, 580active, boost converted, 581

Power MOSFETsee MOSFET

Proportional control, 537Protection

ESD, 446, 448, 482

overvoltage, 446, 448, 469reverse battery, 452, 473, 479short circuit, 251, 446, 448temperature, 446, 447, 471TOPFET, 445, 447, 451

Pulse operation, 558Pulse Width Modulation (PWM), 108Push-pull converter, 111, 119

advantages, 119clamp diodes, 119cross conduction, 119current mode control, 120diodes, 121

disadvantages, 119duty ratio, 119electronic ballast, 582, 587flux symmetry, 119, 120magnetics, 213multiple outputs, 119operation, 119output filter, 119output ripple, 119rectifier circuit, 180switching frequency, 119transformer, 119transistor voltage, 119transistors, 121

Qs (stored charge), 162

RBSOA, 93, 99, 135, 138, 139Rectification, synchronous, 179Reset winding, 117Resistor

mains dropper, 544, 545Resonant power supply, 219, 225

modelling, 236MOSFET, 52, 53

pre-converter, 225Reverse leakage, 169Reverse recovery, 143, 162RFI, 154, 158, 167, 393, 396, 497, 529, 530,537Ruggedness

MOSFET, 62, 73schottky diode, 173

Safe Operating Area (SOA), 25, 74, 134, 557forward biased, 92, 99, 134reverse biased, 93, 99, 135, 138, 139

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Saturable choketriac, 523

Schottky diode, 173bulk leakage, 174edge leakage, 174

guard ring, 174reverse leakage, 174ruggedness, 173selection guide, 176technology, 173

SCRsee Thyristor

Secondary breakdown, 133Selection Guides

BU25XXA, 331BU25XXD, 331damper diodes, 345EPI diodes, 171

horizontal deflection, 343MOSFETs driving heaters, 442MOSFETs driving lamps, 441MOSFETs driving motors, 426Schottky diodes, 176SMPS, 339

Self Oscillating Power Supply (SOPS)50W microcomputer flyback converter, 199ETD transformer, 199

Servo, 298Single ended push-pull

see half bridge converterSnap-off, 167Snubber, 93, 139, 495, 502, 523, 529, 549

active, 279Softness factor, 167Solenoid

TOPFET, 469, 473turn off, 469, 473

Solid state relay, 501SOT186, 154SOT186A, 154SOT199, 154Space charge, 133Speed-up capacitor, 143

Speed controlthyristor, 531triac, 527

Starterfluorescent lamp, 580

Startup circuitelectronic ballast, 591self oscillating power supply, 201

Static Induction Thyristor, 11Stepdown converter, 109Stepper motor, 309Stepup converter, 109

Storage time, 144Stored charge, 162Suppression

mains transient, 544Switched Mode Power Supply (SMPS)

see also self oscillating power supply100W 100kHz MOSFET forward converter,192100W 500kHz half bridge converter, 153100W 50kHz bipolar forward converter, 18716 & 32 kHz TV, 389asymmetrical, 111, 113base circuit design, 149boost converter, 109buck-boost converter, 110buck converter, 108ceramic output filter, 153continuous mode, 109, 379

control ic, 391control loop, 108core excitation, 113core loss, 167current mode control, 120dc-dc converter, 119diode loss, 166diode reverse recovery effects, 166diode reverse recovery softness, 167diodes, 115, 118, 121, 124, 126discontinuous mode, 109, 379epitaxial diodes, 112, 161flux swing, 111flyback converter, 92, 111, 113, 123forward converter, 111, 116, 379full bridge converter, 111, 125half bridge converter, 111, 122high voltage bipolar transistor, 94, 112, 115,118, 121, 124, 126, 129, 339, 383, 392isolated, 113isolated packages, 153isolation, 108, 111magnetics design, 191, 197magnetisation energy, 113mains filter, 380

mains input, 390MOSFET, 112, 153, 33, 384multiple output, 111, 156non-isolated, 108opto-coupler, 392output rectifiers, 163parasitic oscillation, 149power-down, 136power-up, 136, 137, 139power MOSFET, 153, 339, 384pulse width modulation, 108push-pull converter, 111, 119

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RBSOA failure, 139rectification, 381, 392rectification efficiency, 163rectifier selection, 112regulation, 108

reliability, 139resonantsee resonant power supply

RFI, 154, 158, 167schottky diode, 112, 154, 173snubber, 93, 139, 383soft start, 138standby, 382standby supply, 392start-up, 391stepdown, 109stepup, 109symmetrical, 111, 119, 122

synchronisation, 382synchronous rectification, 156, 179TDA8380, 381, 391topologies, 107topology output powers, 111transformer, 111transformer saturation, 138transformers, 391transistor current limiting value, 112transistor mounting, 154transistor selection, 112transistor turn-off, 138transistor turn-on, 136transistor voltage limiting value, 112transistors, 115, 118, 121, 124, 126turns ratio, 111TV & Monitors, 339, 379, 399two transistor flyback, 111, 114two transistor forward, 111, 117

Switching loss, 230Synchronous, 497Synchronous rectification, 156, 179

self driven, 181transformer driven, 180

Temperature control, 537Thermalcontinuous operation, 557, 568intermittent operation, 568non-rectangular pulse, 565pulse operation, 558rectangular pulse, composite, 562rectangular pulse, periodic, 561rectangular pulse, single shot, 561single shot operation, 561

Thermal capacity, 558, 568

Thermal characteristicspower semiconductors, 557

Thermal impedance, 74, 568Thermal resistance, 70, 154, 557Thermal time constant, 568

Thyristor, 10, 497, 509two transistor model, 490applications, 527asynchronous control, 497avalanche breakdown, 490breakover voltage, 490, 509cascading, 501commutation, 492control, 497current rating, 511dI/dt, 490dIf/dt, 491dV/dt, 490

energy handling, 505external commutation, 493full wave control, 499fusing I2t, 503, 512gate cathode resistor, 500gate circuits, 500gate current, 490gate power, 492gate requirements, 492gate specifications, 512gate triggering, 490half wave control, 499holding current, 490, 509inductive loads, 500inrush current, 503latching current, 490, 509leakage current, 490load line, 492mounting, 514operation, 490overcurrent, 503peak current, 505phase angle, 500phase control, 498, 527pulsed gate, 500

resistive loads, 498resonant circuit, 493reverse characteristic, 489reverse recovery, 493RFI, 497self commutation, 493series choke, 502snubber, 502speed controller, 531static switching, 497structure, 489switching, 489

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switching characteristics, 517synchronous control, 497temperature rating, 512thermal specifications, 512time proportional control, 497

transient protection, 502trigger angle, 500turn-off time, 494turn-on, 490, 509turn-on dI/dt, 502varistor, 503voltage rating, 510

Thyristor data, 509Time proportional control, 537TOPFET

3 pin, 445, 449, 4615 pin, 447, 451, 457, 459, 463driving, 449, 453, 461, 465, 467, 475

high side, 473, 475lamps, 455leadforms, 463linear control, 451, 457motor control, 430, 457, 459negative input, 456, 465, 467protection, 445, 447, 451, 469, 473PWM control, 451, 455, 459solenoids, 469

Transformertriac controlled, 523

Transformer core airgap, 111, 113Transformers

see magneticsTransient thermal impedance, 559Transient thermal response, 154Triac, 497, 510, 518

400Hz operation, 489, 518applications, 527, 537asynchronous control, 497breakover voltage, 510charge carriers, 549commutating dI/dt, 494commutating dV/dt, 494commutation, 494, 518, 523, 529, 549

control, 497dc inductive load, 523dc motor control, 525dI/dt, 531, 549dIcom/dt, 523dV/dt, 523, 549emitter shorting, 549full wave control, 499fusing I2t, 503, 512gate cathode resistor, 500gate circuits, 500gate current, 491

gate requirements, 492gate resistor, 540, 545gate sensitivity, 491gate triggering, 538holding current, 491, 510

Hi-Com, 549, 551inductive loads, 500inrush current, 503isolated trigger, 501latching current, 491, 510operation, 491overcurrent, 503phase angle, 500phase control, 498, 527, 546protection, 544pulse triggering, 492pulsed gate, 500quadrants, 491, 510

resistive loads, 498RFI, 497saturable choke, 523series choke, 502snubber, 495, 502, 523, 529, 549speed controller, 527static switching, 497structure, 489switching, 489synchronous control, 497transformer load, 523transient protection, 502trigger angle, 492, 500triggering, 550turn-on dI/dt, 502varistor, 503zero crossing, 537

Trigger angle, 500TV & Monitors

16 kHz black line, 35130-64 kHz autosync, 39932 kHz black line, 361damper diodes, 345, 367diode modulator, 327, 367EHT, 352 - 354, 368, 409, 410

high voltage bipolar transistor, 339, 341horizontal deflection, 341picture distortion, 348power MOSFET, 339SMPS, 339, 354, 379, 389, 399vertical deflection, 358, 364, 402

Two transistor flyback converter, 111, 114Two transistor forward converter, 111, 117

Universal motorback EMF, 531

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starting, 528

Vacuum cleaner, 527Varistor, 503Vertical Deflection, 358, 364, 402

Voltage doubling, 122Water heaters, 537

Zero crossing, 537Zero voltage switching, 537

Date post:	14-Apr-2018
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Philips - High Voltage Bipolar Transistor

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