Powder Core Apps Emi Filters

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Page 1

Powder Core Applications in High Performance EMI Filters

Tim Slattery,

Applications Engineer

The Arnold Engineering Company

Marengo, Illinois

May, 99

The use of molybdenum permalloy, 50% nickel-iron alloy and sendust-type powder cores in

power filter inductors is presented. Power line filtering to reduce differential-mode conducted

EMI is the featured application. The article begins with a simple explanation of filtering and an

example that demonstrates the advantages of using more than one inductor in a filter design.

In addition, power loss and inductance stability with respect to operating current frequency and

magnitude is compared between inductors made with the three different core types. Graphs of

equivalent series inductance, equivalent series resistance and impedance versus frequency

are used to show the effects of winding distributed capacitance and core material eddy cur-

rents.

This paper is a companion to “Powder Core Applications in Switching Amplifier and High

Performance EMI Filters,” an article written by Donald E. Pauly and sponsored by The Arnold

Engineering Company. It also complements “Power Supply Magnetics” (a three-part article) by

Mr. Pauly and published in the January, February and March, 1996 issues of PCIM Magazine.

Copies are available from The Arnold Engineering Company.

Multiple Pole Filters – Advantages

Filters as they apply to electrical and electronic

power conversion systems are circuits with

inductors and capacitors as elements. Thearrangement and sizes of these elements are

chosen so that only relatively low frequen-

cies of electrical energy are allowed to

pass. This creates a “low-pass filter.”

Filter design is quite complicated, requiring

considerable knowledge of mathematics

and computer-aided engineering as well as

practical experience. The term “pole”

refers to a theoretically infinite output

response to input at a particular frequency.

For the two filters that will be consideredhere, it is sufficient to identify the single

capacitor and single inductor arrangement

as a two-pole or single-stage filter and the

combination of two capacitors and two

inductors as a four-pole or two-stage filter.

See Figure 1. The number of poles corre-

sponds to the number of elements.

The nature of the circuits which are attached to

the filter input and output (source and load

impedances) have a profound effect on the

frequency response. The filter poles are not

seen as infinite responses because of damping

Two-Pole or “Single-Stage” Filter (15 Amperes)

L4 = 8 µH (10 Turns)

C = 15 µF

L = 13 µH (15 Turns)

C1 = 15 µFC3 = 15 µF

L2 = 8 µH (10 Turns)

Power Noise

Four-Pole or “Two-Stage” Filter (15 Amper es)

Power Noise

Figure 1. Two-Pole and Four-Pole Filters.



Page 2

Figure 3. Single-Stage Filter Construction and

Test Leads. Figure 4. Two-Stage Filter Construction and TestLeads.

by the load and source. In practice, the filter

elements include resistance associated with

inductor winding and core losses as well as

capacitor lead, electrode and dielectric losses.

Higher frequency loss provides additional

damping that is desirable for stability.1 Also,

parasitic elements such as capacitor lead

inductance and inductor winding distributedcapacitance influence filter performance.

These parasitic elements are shown in Figure 2

along with the equivalent series resistance of

the inductor, Rs, and capacitor, Rc.

Stand-alone constructions of the two types of

filters for testing are shown in Figure 3 and

Figure 4. The single-stage filter uses a 15 µF

polypropylene capacitor and a 13.2 µH inductor.

The inductor core is Arnold Engineering part

number MS-130060-2, a sendust-type core with

a permeability of 60. (Arnold manufactures and

sells this type of core under the trade name

Super-MSS.) The conductor is made from

three strands of 18 AWG magnet wire and

results in a DC resistance of 4.5 mΩ.

The two-stage filter uses two of the same 15 µF

capacitors as in the single-stage design. Each

of the two inductors has a value of 7.95 µH and

is based on a smaller core of the same material

and permeability, Arnold part number

MS-106060-2. The conductor is the same size

as for the larger inductor but only ten turns are

used. The total resistance for the inductors

1 See Mitchell, Daniel M., DC-DC Switching Regulator Analysis,

McGraw-Hill, New York, 1988, ISBN 0-07-042597-3, Chapter 7,

“Effects of EMI Input Filtering.”

LsRs

Cl Lc

Rc

Cs

50 S

50 S

0.0 dBm

Test(Output)

Reference(Input)

HP 4194A Gain-Phase Analyzer

Figure 2. Single-Stage Filter Model Showing Parasitic Elements and Test and Source Load.

50 ΩΩΩΩΩ

50 ΩΩΩΩΩ



Page 3

10 100 1000 10000

Frequency (kHz)

-20

-10

0

10

20

-16.67

-13.33

-6.67

-3.33

3.33

6.67

13.33

16.67

L s ( µ H )

0

2000

4000

6000

500

1000

1500

2500

3000

3500

4500

5000

5500

R s ( Ω )

2 3 4 5 6 7 8 9 2 3 4 5 6 7 8 9 2 3 4 5 6 7 8 9 2 3 4

Rs Single Stage

Rs 2 Stage

Ls Single Stage

Ls 2 Stage

Figure 5. Equivalent Series Inductance and Resistance versus Frequency for Single and Two-Stage

Inductors.

connected in series is 5.4 mΩ. Because of the

fewer turns, the magnetizing force on the

smaller cores is 14.5% less for the same value

of current.

Figure 5 shows how the equivalent series

inductance and resistance varies with fre-

quency for each type of inductor. The larger

one self resonates at about 26 MHz whereas

the smaller one is still inductive beyond

40 MHz. A wider frequency range is typical of

smaller inductors. There will be more on

inductor characteristics at high frequency in the

next section.

10 100 1000 10000

Frequency (kHz)

-40

-20

0

20

40

-30

-10

10

30

C p ( µ F )

0

100

200

300

400

50

150

250

350

R p ( Ω )

2 3 4 5 6 7 8 9 2 3 4 5 6 7 8 9 2 3 4 5 6 7 8 9 2 3 4

Cp

Rp

Figure 6. Equivalent Parallel Capacitance and Resistance versus Frequency for the 15 µF Capacitor.

A similar graph for the capacitor is shown in

Figure 6. In this case, parallel capacitance and

resistance are plotted against frequency. Note

that the capacitor resonates with the inductance

of its leads at about 250 kHz.

All of the frequency response graphs in this

paper are based on data from a

Hewlett-Packard 4194A Impedance/

Gain-Phase Analyzer. For each inductance

and capacitance measurement, the test signal

voltage used was nominally 0.5 V rms.



Page 4

1e-005 0.0001 0.001 0.01 0.1 1 10 100

Frequency (MHz)

-80

-60

-40

-20

0

-70

-50

-30

-10

G a i n ( d B )

-240

-180

-120

-60

0

60

120

180

240

P h a s e ( d e g )

2 4 6 8 2 4 6 8 2 4 6 8 2 4 6 8 2 4 6 8 2 4 6 8 2 4 6 8

Single - Stage Gain

Single - Stage Phase

Figure 7. Gain and Phase versus Frequency for the Single-Stage Filter.

1e-005 0.0001 0.001 0.01 0.1 1 10 100

Frequency (MHz)

-80

-60

-40

-20

0

-70

-50

-30

-10

G a i n ( d B )

-240

-180

-120

-60

0

60

120

180

240

P

h a s e ( d e g )

Two - Stage Gain

Two - Stage Phase

2 4 6 82 4 6 8 2 4 6 8 2 4 6 8 2 4 6 8 2 4 6 8 2 4 6 8

Figure 8. Gain and Phase versus Frequency for the Two-Stage Filter.

To show the advantage of multiple-pole filtering,

the frequency response for the single-stage

(two-pole) and two-stage (four-pole) examples

are presented in Figure 7 and Figure 8. The

most significant departures from a power circuit

application are the 50 Ω source impedance and

50 Ω load, which are provided by the

“gain-phase” part of the Hewlett-Packard 4194AAnalyzer. In a typical switching-type power

supply, the impedances are variable, not

matched and usually much lower in value at low

frequencies. The test does provide useful

information for comparison even though the

application conditions differ.

For example, the maximum attenuation (mini-

mum gain) for each filter is the result of series

resonance of the capacitor and its lead induc-

tance. It is especially noticeable at about

175 kHz for the single-stage filter. The impor-

tance of minimizing lead length is apparent.

Attenuation decreases at higher frequency

because lead inductance is impeding the flow of return current through each capacitor.

Another important observation with regard to

filter behavior is the lower damping of the

two-stage configuration around 20 kHz. Theo-

retically, the four poles are two double poles at



Page 5

Figu re 9. C om po site G ain an d P hase versus Frequ en cy for O ne an d T w o-S tage F ilters.

1e-005 0.0001 0.001 0.01 0.1 1 10 100

Frequency (MHz)

-80

-60

-40

-20

0

-70

-50

-30

-10

G a i n ( d B )

-240

-180

-120

-60

0

60

120

180

240

P h a s e ( d e g )

Two - Stage Gain

Two - Stage Phase

2 4 6 82 4 6 8 2 4 6 8 2 4 6 8 2 4 6 8 2 4 6 8 2 4 6 8

Single - Stage Gain

Single - Stage Phase

14.6 kHz and their effect is apparent because of

the lack of damping between C1, L2 and C3.

About 30 dB of attenuation is lost at 20 kHz.

Above that frequency, the filter recovers and

outperforms the single-stage design by a

remarkable 20 dB at 60 kHz. See Figure 9.

Finally, note that the two-stage filter maintains

a 10 dB advantage from 300 kHz to 1 MHz.

The additional inductor and capacitor reduce

the capacitor lead inductance effect.2 In theUnited States, because the AM broadcast band

is from 540 kHz to 1.6 MHz, improved filter

performance in this range of frequencies is of

particular benefit.

The filters were tested with the input or refer-

ence channel on the capacitor side and test

channel on the inductor. It was observed that

the frequency response was very similar with

the inductor side connected to the reference

channel and the capacitor side tied to the test

channel. Therefore, the observations abovealso apply to the filters treated as the

inductor-input type.

Power Line Filters

The term filter usually applies to an electrical

circuit or portion of a circuit that prevents

electromagnetic interference (EMI). Without

filtering, unwanted electrical signals could travel

from one device to another along the power line

or bus bar that they share. The conducted

interference can also be subsequently radiated

since a power line is an antenna for higher

frequencies. The purpose of the filter is toprevent this electrical “noise” from being con-

ducted to the line from the device while allowing

the desired electrical power to pass.

For equipment connected to a public utility,

government agencies regulate the maximum

conducted noise voltage over specific fre-

quency ranges. For example, in the United

States, the Federal Communications Commis-

sion specifies a conducted radio frequency

interference (RFI) limit of 48 dBµV from

450 kHz to 30 MHz.3 The purpose is to preventinterference with radio, television and telephone

services used by the general public.

Noise limits for the outputs of power supplies

within an electronic system are determined by

the requirements of the attached loads. In most

cases, sufficient noise filtering is accomplished

with the same power filter elements (energy

storage inductor and output capacitor) that

2 See Don Pauly’s article “Power Supply Magnetics,” page 16 of

the reprint available from Arnold Engineering or “Part III” in the

March, 1996 issue of PCIM on battery line filters.

3 FCC Part 15 Class B, devices for home or consumer use.

“dBµV” stands for decibel microvolt; 48 dBµV is 0.00025 volt;

“kHz” is kilohertz or thousand cycles per second and “MHz”

refers to megahertz or million cycles per second.



Page 6

control the output voltage ripple. However,

some designs include a second stage of filtering

to control EMI on the output of the supply.4

Equipment that contains continuously switching

components, such as the power transistors and

diodes of a switched-mode power supply,

require conducted EMI filtering on the input

side. The abrupt changes of current in the

circuit cause brief voltage rises or “spikes”

either across the input conductors or on both

conductors to ground. (“Ground” includes

ground return wires, ground planes and

grounded enclosures.)

EMI voltage between input conductors is called

differential-mode noise. Noise from both

conductors to ground is termed common-mode.

An inductor for common-mode noise utilizes the

opposing currents in the input conductors (two

windings on one core) and a high permeabilitycore material.

In contrast, a differential-mode inductor requires

a core material that can maintain permeability

with a bias field. Refer to the graphs in Fig-

ure 11 showing inductor current, voltage and

core magnetic fields. Note that the source in

Figure 11 is either a power line (via a wall outlet,

for example) or a battery such as the 48 volt

battery for telephone central office equipment. In

a battery system, the magnetizing current

is a constant DC. For an AC system withhigh power factor, the magnetizing current

is nearly sinusoidal. With a low power

factor AC system, the current is a series

of alternating pulses.

Powder cores are appropriate for differen-

tial-mode, sometimes referred to as

“series-mode,” inductors or “chokes,”

because of their extraordinary capability

to maintain inductance with bias. The

50% nickel-iron alloy powder is particu-

larly useful at high flux densities. (ArnoldEngineering’s trade name for this material

is Hi-Flux.) For comparison, permeabil-

ity versus DC bias curves for the three

types of powder cores are shown in

Figure 12.

The test data in Figure 12 and the data that

follows came from the same three cores. Each

has a permeability of 60 and is the same size as

the core used in the single-stage filter. For

reference, the Arnold part numbers are

A-291061-2, HF-130060-2 and MS-130060-2,

representing the molybdenum permalloy (MPP),

50% nickel-iron and sendust-type powders,respectively.

An example of a “fully” wound core is shown in

Figure 10. Fully wound means that one-half of

the core inside diameter remains. Usually, at

least this much room must be provided for a

hook or shuttle to place the last turn. In this

case, the inductance value is 1.9 mH and is

typical for power line applications. Inductance

requirements generally range from a few

microhenries to several millihenries.

Low core loss at power line frequency is neces-sary to take advantage of the core material’s

high saturation flux density. Measurements are

presented for Hi-Flux in Figure 13. Hi-Flux has

the most loss so it can be used as a worst-case

reference. Even at 400 Hz and 9000 gausses,

the core loss density is low at 200 mW/cm3. For

50 and 60 Hz applications, the flux density limit

is determined by the change in permeability as

shown in Figure 14.

4 See Billings, Keith H., Switchmode Power Supply Handbook ,

McGraw-Hill, New York, 1989. ISBN 0-07-005330-8. pp. 1.151

and 1.152.

Figure 10. Example of “Fully” Wound Core.Figure 10. Example of “Fully” Wound Core



Page 7

I

+ V - SwitchingPower Circuitand Load

Common-ModeInductor

Differential-ModeInductor, L

Two-Pole or “Single-Stage”Filter for Differential-ModeConducted EMI.

Capacitor, C

50/60 Hz Current Pulses (Low Power Factor)

0

H

t

B

V

IBatteryCurrent

t

50/60 HzHysteresisLoop

BDC

HDC

Switching N oise Voltage

0Battery DCOperatingPoint

50/60 Hz Sinusoidal Current(High Power Factor)

Figure 11. Typical EMI Filter Configuration and Differential-Mode Inductor Voltage, Currentand Magnetic Field Waveforms.

Figure 11. Typical EMI Filter Configuration and Differential-Mode Inductor

Voltage, Current and Magnetic Waveforms.



Page 8

1 10 100

2 3 4 5 6 7 8 9 20 30 40 50 60 70 80 90

D. C. Magnetizing Force (Oersteds)

40

60

80

100

45

50

55

65

70

75

85

90

95

P e r c e n t P e r m e a b i l i t y

40

60

80

100

45

50

55

65

70

75

85

90

95

P e r c e n t

p e r m e a b i l i t y

100 % = 60µMPP

Hi-Flux

SMSS

Figure 12. Permeability versus DC Bias.

Another important consideration is the

variation of inductance with frequency. Thefrequency responses of single layer 60 µH

inductors made with each type of core are

shown in Figure 15 through 17.

The absence of the series inductance peak

for the Hi-Flux inductor indicates that its

core permeability is dropping with fre-

quency. Higher eddy current loss in the

50% nickel-iron powder is responsible. As

mentioned before, loss at higher frequency

can be an advantage in filters because of

the additional stability the damping pro-vides. Greater detail regarding series

inductance and resistance in the frequency

range of 100 kHz to 1 MHz is given in Figure

18. The relatively low eddy current loss in

Super-MSS is apparent.

10 100 1000 10000

Flux Density (Gauss)

-20

-10

0

10

-17.5

-15

-12.5

-7.5

-5

-2.5

2.5

5

7.5

% C

h a n g e o f P e

r m e a b i l i t y

-20

-10

0

10

-17.5

-15

-12.5

-7.5

-5

-2.5

2.5

5

7.5

60µ Hi - Flux

2 3 4 5 6 7 8 9 2 3 4 5 6 7 8 9 2 3 4 5 6 7 8 9

Figure 14. Modulation of Permeability with Flux Density.

1000 100002000 3000 4000 5000 6000 7000 8000 9000

Flux Density (Gauss)

1

10

100

2

3

4

5

6

7

89

20

30

40

50

60

70

8090

200

300

C o r e L o s s D e n s i t y ( m W /

c m 3 )

60µ Hi - Flux

1 2 0 H

z 2 0

0 H z

4 0 0 H

z

Figure 13. Core Loss versus Flux Density.



Page 9

10 100 1000 10000

Frequency (kHz)

-150

-100

-50

0

50

100

150

-125

-75

-25

25

75

125

L s ( µ H )

MPP

SMSS

Hi-Flux

2 3 4 5 6 7 8 9 2 2 23 3 34 4 45 56 67 78 89 9

Figure 15. Equivalent Series Inductance versus Frequency, Single-Layer Winding.

10 100 1000 10000

Frequency (kHz)

0

4000

8000

12000

16000

20000

2000

6000

10000

14000

18000

R s ( Ω )

2 4 6 8 2 2 24 4 46 68 83 5 7 9 3 3 35 57 79 9

MPP

SMSS

Hi-Flux

Figure 16. Equivalent Series Resistance versus Frequency, Single Layer Winding.

10 100 1000 10000

Frequency (kHz)

1

10

100

1000

10000

100000

I m p e d

a n c e ( Ω )

SMSS

MPP

Hi-Flux

2 3 4 5 6 7 8 9 2 3 4 5 6 7 8 9 2 3 4 5 6 7 8 9 2 3 4

2

4

6

8

2

4

6

8

2

4

6

8

2

4

6

8

2

4

6

8

Figure 17. Impedance versus Frequency, Single Layer Winding.



Page 10

Finally, the effect of greater distributed capaci-

tance with overlapping conductor turns (multiple

layers of magnet wire) is shown graphically in

Figure 20 and Figure 21. Beyond 1.6 MHz, this

stray capacitance actually causes the

high-value, multiple-layer inductor to have a

lower impedance than the low-value

single-layer inductor.5

In conclusion, each of the powder core types is

applicable to power line filtering. The 50%

nickel-iron material performs the best because

Figure 18. Equivalent Series Inductance and Resistance versus Frequency, 100 kHz to 1 MHz.

100 1000200 300 400 500 600 700 800 900

Frequency (kHz)

-10

0

10

20

30

40

-5

5

15

25

35

R s ( Ω )

60

61

62

63

64

65

60.5

61.5

62.5

63.5

64.5

L s ( µ H )

Ls Hi-Flux

Ls MPP

R s S M S S

R s H

i - F l u

x

R s M P P

Ls SMSS

of its ability to sustain inductance with higher

magnetizing current. It also provides some

desirable damping at higher frequencies.

Another important consideration is acoustic

noise caused by magnetostriction of the mag-

netic metal alloy. A 50% nickel-iron alloy core

can make a humming sound at high 50 or

60 Hz flux levels. Of course, DC magnetizing

current does not cause audible noise so the

50% nickel-iron is usually the best material for

battery power line filters.

5 The multiple-layer winding is 170 turns of 18 AWG magnet wire.

100 1000200 300 400 500 600 700 800 900

kHz

10

100

1000

20

30

40

50

60

70

80

90

200

300

400

500

600

700

800

900

I m p e d a n c e

HI-Flux

MPP

SMSS

Figure 19. Impedance versus Frequency, 100 kHz to 1 MHz.

Hi-Flux



Page 11

Both MPP and Super-MSS have exceptionally

low magnetostriction and either could be used

to minimize audible noise. MPP has the advan-

tage of better incremental permeability with DC

bias. The 50 or 60 Hz current is essentially DC

compared to the frequency of electrical noise.

Incremental permeability versus DC bias

curves can be used to predict the inductance atany point of the 50/60 Hz current waveform.

10 100 1000 10000

Frequency (kHz)

-80

-40

0

40

80

-70

-60

-50

-30

-20

-10

10

20

30

50

60

70

S i n g l e L

a y e r L s ( µ H )

-16000

-8000

0

8000

16000

-14000

-12000

-10000

-6000

-4000

-2000

2000

4000

6000

10000

12000

14000

F u l l W o u n d L s ( µ H )

Single - Layer

Full Wound

2 3 4 5 6 7 8 9 2 3 4 5 6 7 8 9 2 3 4 5 6 7 8 9 2 3 4

Figure 20. Equivalent Series Inductance versus Frequency for Single-Layer and Fully Wound Designs.

10 100 1000 10000

Frequency (kHz)

1

10

100

1000

10000

100000

I m p e d a n c e ( Ω )

2 3 4 5 6 7 8 9 2 3 4 5 6 7 8 9 2 3 4 5 6 7 8 9 2 3 4

2

4

68

2

4

6

8

2

4

6

8

2

4

6

8

Full Wound Single - Layer2

4

6

8

Figure 21. Impedance versus Frequency for Single-Layer and Fully Wound Designs.

The sendust-type core has the lowest cost per

unit volume and is a good choice where the

highest possible performance is not required.

The difference in cost of molybdenum permalloy

powder and the 50% nickel-iron alloy powder is

negligible.



Page 12

Acknowledgements

The author thanks Jim Motley for his careful and

skillful work in preparing all of the data curves in

the paper and Jean Wickman for assisting him.

Don Pauly was a very helpful and encouraging

proofreader and his insights regarding filter

design were essential.References

Billings, Keith H., Switchmode Power Supply Hand-

book , McGraw-Hill, New York, 1989.

ISBN 0-07-005330-8.

Mitchell, Daniel M., DC-DC Switching Regulator

Analysis, McGraw-Hill, New York, 1988,

ISBN 0-07-042597-3

Reference Data for Engineers: Radio, Electronics,

Computer, & Communications, Eighth Edition, Mac

E. Van Valkenburg, Editor-in-Chief, Sams Publishing,

Carmel, Indiana, USA, 1995, ISBN 0-672-22753-3

Pauly, Donald E., “Power Supply Magnetics, Part III,”

PCIM Magazine, March, 1996. (Available from The

Arnold Engineering Company.)

Pauly, Donald E., “High Fidelity Switching Audio

Amplifiers Using TMOS Power MOSFETs,” AN1042/

D, 1989, Motorola Technical Information Center,

Phoenix, AZ.

Brown, Paul, “Not All Millihenries Are Equal,” pre-

sented at an IEE Colloquium on Capacitors and

Inductors for Power Electronics, Savoy Place,

London, March, 1996. Paper available as variousabstracts through Almag Ltd, Braintree, Essex,

England (e-mail: [email protected])

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