IEEE TRANSACTIONS ON ELECTRON DEVICES. VOL. 39, NO. 9. SEPTEMBER 1992 1987

Heterojunction Bipolar Transistor Design for Power Applications

Guang-bo Gao, Senior Member, IEEE, Hadis MorkoG, Fellow, IEEE, and Mau-Chung Frank Chang, Member, IEEE

Abstract-Design rules of AIGaAsIGaAs heterojunction bi- polar transistors for power applications are presented and compared to those for Si microwave power transistors. Con- cepts discussed include the tradeoff between power gain, out- put power, power-added efficiency in the layout design, layer structure selection, and thermal design.

I. INTRODUCTION ETEROJUNCTION Bipolar Transistors (HBT's) H fabricated in the AlGaAs /GaAs material system

offer excellent potential for power application in the mi- crowave spectrum. Although this field has not yet been explored in detail, some exciting results have already been reported. The state-of-the-art performance for Al- GaAs/GaAs power HBTs is summarized in Table I. At L-band (1.5 GHz) a CW output power of 2 W with 72 % Power-Added Efficiency (PAE) was obtained [ l ] . At S-band (3 GHz) a CW output power of 1.1 W with 12.3 dB power gain and 61 % PAE has been also attained [2]. Moreover, a high power gain of 1 1.6 dB and a high power- added efficiency of 68% were also achieved at 10 GHz [3]. At 18 GHz, 358-mW output power with 11.4-dB power gain and 43 % PAE has been demonstrated at Rock- well [4] in the common base operation of a power HBT with emitter finger width of 2 Fm.

In the area of high power microwave amplifiers, 5.3-W total output power at X-band (8 GHz) CW operation with a power gain of 4 .3 dB and 33% PAE was obtained, cor- responding to a power density of 2.2 W/mm of emitter length [ 5 ] . In pulsed operation at 10 GHz, a peak output power of 560 mW, power density of 18.7 W/mm was obtained for 300-11s pulse duration with a 33 % duty cycle 161.

Approaches used towards the achievement of these re- sults have in large been based on trial and error. How- ever, to exploit power HBT's to their fullest, design cri-

Manuscript received September 9, 1991; revised March I O , 1992. This work was supported by the National Science Foundation under Grant ECS-88-22406 (to the University of Illinois), the Air Force Office of Sci- entific Research (to the University of Illinois), and WRDC (to Rockwell International). The review of this paper was arranged by Associate Editor M. Shur.

G. B. Gao and H. Morkoq are with the University of Illinois at Urbana- Champaign, the Coordinated Science Laboratory, and the Materials Re- search Laboratory. Urbana, 1L 61801.

M. F. Chang is with Rockwell International Science Center, Thousand Oaks, CA 91360.

IEEE Log Number 9201828.

TABLE 1 STATE-OF~THE-ART PERFORMANCE FOR AlGaAs/GaAs POWER HBT's

1.5 2.0 12 111 3.0 1 . 1 12.3 61 P I

I O 0.226 11.6 68 [31 8.0 5.3

4.3 46 33 [6] (pulsed) PI 10 0.561 (18.7 W / m m ) 5 18 0.358 11.4 43 [41

teria and rules must be developed. This paper discusses the design rules for enhancing the performance of power HBT's. It starts with a brief discussion of microwave power amplifier parameters which are commonly the de- sign goal. We then illustrate the design rules of the layout and layer structure, in comparison with those of the Si microwave power transistors. Thermal design is also in- cluded in the discussion. The final section contains some concluding remarks.

11. POWER HBT DESIGN GOALS Microwave power amplifier parameters are of the ut-

most importance in the design of power HBT's and hence are discussed first.

A . Output Power Po,, In single-ended class B or class C power amplifiers the

flow angle 8 of the collector current is 90" or less, re- spectively; i.e., the transistors conduct only during one- half period or less. The current and voltage relations are indicated in connection with the high-frequency (HF) out- put characteristics in Fig. 1. It should be noted that the HF output characteristics are quite different from the dc output characteristics [7]. In HF power amplifiers (class B or C) , the transistor is loaded by a selective impedance having a real value of RL at the fundamental frequency (operating frequency fo), but a short circuit at harmonics. The half-cosinusoidal current pulses of peak value i,, generate a fundamental frequency voltage, which has a peak value approximately equal to the supply voltage. From Fourier analysis and a simple calculation, the out- put power Pout at the fundamental frequency can be ob- tained

(1) pout = t ~ I ( ~ ) i c m ( ~ C E n l a x - RF U c e J

0018-9383/92$03.00 0 1992 IEEE

1988

c

n icn

n

t "

R Fv

Fig. 1 . The current-voltage relations for class B or C power amplifiers.

IEEE TRANSACTIONS ON ELECTRON DEVICES, VOL. 39, NO. 9, SEPTEMBER 1992 ::::I 1.00

0.50 ai(e)

0 30 60 90 120 150 180

C U R R E N T FLOW A N G L E 8(")

Fig. 2. G(8), cy, (e), and E ( 8 ) versus current flow angle 8.

where cy1(@, i,, and RF U,,, are the fundamental fre- quency decomposition coefficient (see Fig. Z), the maxi- mum peak current, and the radio-frequency saturation voltage, respectively.

The maximum peak current i,, of an HBT depends on the cutoff frequency f T at high current. In general, i,, is greater than IC,,, the current at which the f T reaches its peak value. As a consequence, power HBT's with high cutoff frequencies f T at high current levels are especially desirable for high output power. At high current densities, the Kirk effect causes the formation of a current-induced base region, and the cutoff frequency f T and the current gain h, begin to decrease rapidly. The critical current density at which the current-induced base begins to form is given by [8]

Fig. 3 shows Jo plotted versus collector doping Nc with Wc as a parameter at 300, 400, and 500 K for the GaAs material.

As can be seen from (2), the maximum collector current is limited by the electron drift velocity ud, collector dop- ing Nc and collector thickness Wc. Attaining electron ve- locity overshoot, use of a collector material with a high steady-state velocity at high electrical field, and a thin highly doped collector are all beneficial to the operating current, hence also to the output power.

The RF U,,, is relative to the fall-off rate of the cutoff frequency fT at high current densities in the saturation re- gion, and is therefore a function of the collector doping, collector thickness, device layout, operating frequency f o and temperature. Because the electron velocity overshoot

is obvious just at zero and forward BC bias [33]-[36], the RF U,,, of AlGaAs/GaAs HBT's is much less than that of Si BJT's. This was confirmed by the higher PAE of an HBT power amplifier as mentioned above.

The Vc-,,, in (1) is the maximum voltage across the collector-emitter terminals and is slightly less than the base-collector junction breakdown voltage (BVCBO) when the emitter-base junction is zero-biased (for class B) or reverse-biased (for class C). As can be seen from ( l ) , larger i,,, VcE,,, (BVcBo), and smaller RF U,,, give a higher output power P,,,.

B. Power Gain The power gain is the ratio of power delivered to the

load and the input power to the transistor. When both in- put and output are simultaneously conjugately matched, the Maximum Available Gain (MAG) G,,, is achieved. The general form for the gain is obtained from the small- signal equivalent circuit of a power transistor used in a class A amplifier. The power gain for common emitter class B and C operations is less than that of class A and is given by [9]

where G ( 8 ) = al (8) ( l - cos (8 ) ) . The G(8) is a factor related to the current flow angle 8 (see Fig. 2). For class B with the current flow angle 8 = 90°, Gp = 0.5, meaning the power gain is 50% of that of class A. It is worthwhile to note the effects of the feedback elements, base resis- tance r,, emitter resistance re, and emitter lead inductance Le on the power gain. For HBT's, the intrinsic base resis- tance is so small that the base contact resistance has a notable influence on the power gain. The emitter ballast- ing resistor is effective for improving the saturation output power and the power gain at high input power, but it de- grades the power gain at low power levels. The emitter lead inductance L, has a dramatic effect on the power gain due to the high f T of HBT's and should be kept as small as possible. Fig. 4 shows the results calculating the effect of emitter lead inductance on power gain for an HBT with rb = 6.3 Q, re = 3.9 Q, C, = 0.039 pF, and f T = 50 GHz at 10 GHz and class B operation. It can be seen that an inductance of only 0.1 nH degrades the power gain by 3.4 dB. This effect increases as the f T increased.

GAO et al . : HETEROJUNCTION BIPOLAR TRANSISTOR DESIGN FOR POWER APPLICATIONS

IO6

- “E

L 8

. a v * i o s

z

b zi a a V

1989

t I I i : T = 5 0 0 K

r

. W,= 0.3 pm

-W = 0.5 pn

: w,= 0.8 pl I I

10

10

10

t I I 1

10 lb 10 l6 10 l’ 10

COLLECTOR DOPING ( cm.’ )

(a)

10‘ I I #

lo5 :

. W,= 0.3 pn

W, = 0.5 pm

: W = 0.8 pm l o 4 : C 1

POWER GAIN DEGRADATION (dB)

f , = 50 GHz

fg = 10 GHz

- 2

-6z -7 0 0.05 0.1 0.15 0.2

INDUCTANCE OF EMITTER LEAD (nH)

Fig. 4. Effect of emitter lead inductance on the power gain of a power HBT.

GaAs MODFET’s, AlGaAs/GaAs power HBT’s have a higher PAE and it ranges from about 40 to 70% as men- tioned earlier. The power-added efficiency rl is defined as

= rlc (1 - $) (4)

where q,, Pdc, and Pi, are the collector efficiency, dc power from the power supply, and the ac input power from the high-frequency power generator. If a constant trans- conductance is assumed, then the 7, is given by

( 5 )

where E (e) is a factor related to the current flow angle 0 (see Fig. 2). From (4) and ( 5 ) it is found that the higher the power gain and the lower the radio-frequency satura- tion voltage, the higher the power-added efficiency q . The radio-frequency saturation voltage RF U,,, is a critical pa- rameter in determining the output power Pout and the power-added efficiency 17. Using the model developed by Chen et al . [ 7 ] , which generates the high-frequency ic- vCE output characteristics of bipolar transistors at a given junction temperature, operating frequency and device structure, qc can be calculated as a function of collector thicknesses W, at various collector doping levels Nc (see Fig. 5 ) .

C. Power-Added Eficiency (PAE) 17 For solid-state microwave amplifiers used for mobile

communication systems, high PAE is desirable. Com- pared to GaAs power MESFET’s or power AlGaAs/ L-band to’the K-band (17-33 GHz) and even into the

D. Operating Frequency fo AlGaAs/GaAs power HBT’s can operate from the

IEEE TRANSACTIONS ON ELECTRON DEVICES, VOL. 39, NO. 9, SEPTEMBER 1992

0.70 0*7r V =6V CE

0.66 0.2 0.4 0.6 0.8

COLLECTOR WIDTH ( pm )

Fig. 5 . Collector efficiency versus collector width for a ten-finger 1.2 X

9 fim2 AlGaAs/GaAs HBT with the layer structure described in [4], op- erating at 10 GHz.

Q-band (33-46 GHz). Operating frequency fo of an HBT depends on the current gain cutoff frequency f, and the maximum oscillation frequency fmax.

The peak value of the current gain cutoff frequency frpeak at high current densities is, to a first extent, deter- mined by the base transit time and the collector space- charge region transit time 7d

where n is equal to or larger than 2 for the uniform base or the graded base cases (either compositional or doping), and veff is the effective electron drift velocity in the col- lector SCR region, taking into consideration the velocity overshoot effect. In the case of Si BJT’s, the emitter charging time constant still has a sizable effect on the fTpeak due to the high emitter doping.

The maximum oscillation frequency f,,, is optimized when the base transit time 7B is equal to the collector transit time rd [ 101. In this case, fT becomes

Ud

fT = 2?rw,,, (7)

The optimized maximum oscillation frequency f,,, as re- ported by Gao et al. [l 11 is given by

and is shown in Fig. 6. The maximum oscillation fre- quency fmax increases with a decrease in SE.

E. Operating Voltage Vcc Operating voltage for a power HBT depends upon the

circuit design and the frequency of operation. In general, devices can be operated at 9-12 V at X-band. From the

I10 - n g 9 0 -

c3

d

W

n 7 0 -

5 0 -

30 - 0

Fig. 6. Optimized fmax as a function of emitter stripe. width SE for AlGaAs/GaAs HBT’s with a base doping of 1 x lOI9 cm-3 (after [ I 11).

point of view of circuit design, it is desirable for power amplifiers to operate at low currents and high voltages. Under these conditions we no longer have low input impedance and therefore impedance matching between the stages is less difficult. But high-voltage operation requires a larger collector thickness which degrades the speed and the thermal stability of the device [ 121. On the other hand, the higher current capability of HBT’s make them better for high-current and low-voltage operation than Si BJT’s. For high-speed and high-power HBT’s, the collector re- gion is fully depleted under normal bias conditions and the collector depletion width is considerably smaller than the theoretical width obtained from the doping density at breakdown [ 131. The breakdown voltage is given by

where E,, is the critical breakdown field and is given by

(1 1) for GaAs.

The parameters mentioned above must be optimized as part of the overall design for power HBT’s. In general, the output power Po,, power gain Gp, operating frequency fo, operating voltage Vcc, and the current flow angle 8 are given in advance. Then, with the help of (1)-(1 l), the device geometry and structure can be determined. Of course, tradeoffs between different layout designs and de- vice structures must be made for a global optimization to be achieved.

~141 E,, = 4.95 x 1o3~;I8

111. LAYOUT DESIGN This section will describe, step by step, the layout de-

sign of power HBT’s. Within the limitations of a given manufacturing technology a designer should attempt to make an HBT that will meet the design goals as closely as possible. Overdesign results in reduced yield and reli- ability.

---

GAO er ul . : HETEROJUNCTION BIPOLAR TRANSISTOR DESIGN FOR POWER APPLICATIONS

A. Emitter Stripe Width S E

To date most power HBT’s have an interdigitated ge- ometry. The emitter stripe width SE is a very important geometrical parameter as shown in (9). Selection of emitter width SE has implications on the base resistance, and the emitter and collector capacitances, and therefore the operating frequency fo as well as the power gain Gp. Typical emitter widths for commercial Si microwave power BJT’s and AlGaAs/GaAs power HBT’s are shown in Table I1 [15]-[19].

AlGaAs /GaAs power HBT’s having the same emitter width as that of Si BJT’s can operate at higher frequencies due to the higher base doping, higher electron mobility, and higher electron velocity. For example, an output power of 320 mW and 7-dB power gain at 3 GHz can be obtained from a 3.5-pm emitter width HBT [17]. A 2.0-pm emitter width HBT can operate at 10 GHz giving the same power gain and the same output power [18]. However, a 2.0-pm emitter width Si BJT can only operate at 2-3 GHz [15] and for X-band applications a submi- crometer emitter width must be used [ 161.

Of course, the narrower the emitter width S E , the higher the power gain at a given operating frequency and output power. For instance, as the emitter width SE is reduced from 3.5 to 2.0 pm, the power gain increases from 7 to 12.3 dB at 3 GHz [2]. On the other hand, in order to obtain higher output power without degrading the power gain, a narrower emitter width must be used. An output power of 2.5 W with 5.8 dB at 10 GHz has been obtained with a 2.0-pm emitter width [19]. With sealing, it is con- ceivable that a 1.5-pm device will deliver 10-W output power without compromising the power gain.

B. Emitter Finger Length LE The emitter finger length LE is primarily determined by

the emitter metallization resistance and the junction tem- perature nonuniformity along the emitter length. Both of the factors can cause a severely nonuniform distribution of the emitter current along the length. By using a two- dimensional (2D) nonisothermal distributed transistor model [20], we investigated the effect of emitter metalli- zation thickness on the current distribution of power HBT’s. The investigated device had an emitter stripe width of 6 pm, emitter length of 40 pm, and emitter gold metallization thickness of 0.1 and 0.5 pm, respectively. At an average emitter current density of 5 X lo3 A/cm2, the maximum current density along emitter length is 8.6 X lo3 A/cm2 and the minimum is 2.1 x lo3 A/cm2 for 0.1-pm metallization device and corresponding 7.3 x lo3 and 2.6 x lo3 A/cm2 for 0.5-pm device, respectively.

In general, the length and the length-to-width ratio of an emitter finger should not exceed 30 pm and a 20-to-1 ratio in order to minimize the junction temperature and the voltage drop along the length of the finger. Wang et al. [21] found that as the ratio is increased from 5 to 10, and even to 20, although the maximum current increases slightly, the high-frequency performance ( fT andf,,,) de-

1991

TABLE 11 fo VERSUS SE

f;, (GHz) S, (Si BJT’s) (pm) SE (AIGaAs/GaAs HBT’s) (km)

1 .o 5.0 2.0 2 .5 3 .0 1.7 5 .0 1.25 8.0 1 .o

I O 0.75 20 -

10 5 . 0 3 .5 3 .O 2.5 2 . 0 1 .5

grades due to the increase in the emitter finger area as shown in Fig. 7 . From thermal considerations, a power HBT with a larger emitter length results in a higher junc- tion temperature and in turn causes a reduction in the out- put power density. For instance, an HBT with a 10-pm emitter length produced a power density of 6.2 mW/pm2 compared with 4.5 mW/pm2 for the 20-pm device under the same pulse operating condition [6].

C. Emitter Area AE The total emitter area AE is determined by the desired

output power or the maximum peak collector current icm. Knowledge of the output power, operating voltage, and the PAE is sufficient to determine this current. The peak current is strongly dependent on the emitter periphery for BJT’s because of current crowding at high currents. The case is quite different in HBT’s where the base doping reduces substantially the current crowding effects [ 113. At high frequencies, the emitter effective width is a function of operating frequency f o . For a sheet base resistance of 200 n/o and a contact resistance Rc of 5 X lop7 !J cm2, fT of 70 GHz and collector current density of 3 X

lo4 A /cm2, the corresponding emitter utilization factor ( S E e f f / S E ) , as calculated by Asbeck and Chang [22], is depicted in Fig. 8. It is clear that an emitter width of 2 pm is sufficient for an operating frequency of 10 GHz. In this case, the emitter utilization factor is greater than 95 % which allows the current capability and the output power for HBT’s to be dependent on the emitter area rather than the emitter periphery. The output power density per unity emitter area ( p E ) for AlGaAs/GaAs power HBT’s is, therefore, important in the design of a power HBT and is limited by both thermal and electrical characteristics of the devices. The power density is generally around 0.5- 3.0 mW/pm2 for CW operating conditions and 2.0-10.0 mW/pm2 for pulsed conditions, depending on the ther- mal resistance of the devices. By using the via hole ap- proach [23] where the emitter connects to the ground plane (gold-plated heat sink), the highest power density of 3.2 mW/pm2 with 8.65-dB power gain was achieved at 10 GHz [3]. If using the flip-chip mounted technology, the thermal resistance of devices will be even lower and p E even higher. The p E , of course, changes with the operat- ing frequency, for example, it reduces from 2 mW/pm2 at 10 GHz to 1.4 mW /pm2 at 18 GHz for the same struc- ture fabricated at Rockwell [4]. Taking into account the

I992 IEEE TRANSACTIONS ON ELECTRON DEVICES, VOL. 39, NO. 9. SEPTEMBER 1992

220 ’ 40 r-----l 40

I . . . . . . , . . l o 0 5 10 15 20

Emitter Finger Length (pm)

Fig. 7 . fT andf,,,, as a function of emitter finger length (after [21]).

I

0.2 I I I I

0 1 2 3 EMITTER WIDTH fuml

Fig. 8 . Calculated emitter utilization factor for operation of HBT’s at var- ious frequencies as a function of emitter stripe width ( fT = 70 GHz and a base sheet resistance of 200 n were assumed) (after [22]).

200

180

160

140

120

100

80

60

40

Emitter Fingers: 2umxl5um

time: 2.51.1s \ \

Emitter Power

Density (m W/um ):

2.0

I I I I I I

2 4 6 8 10 12 14 16

EMITTER-TO-EMITTER SPACING (I”

Fig. 9 . Effect of emitter spacing on the peak value of junction temperature (maximum temperature) of a power HBT with ten fingers of 2 X 15 pmZ at pulsed operation.

current nonuniformity, the total emitter area AE is given by

D A -K%!!

E - P E

where K is a constant which is larger than unity, depend- ing on the current distribution.

Fig. 10. Micrograph of a power HBT with five cells (after [4]). D. Emitter Spacing SD

A very important parameter of the layout design for power HBT’s is the emitter spacing SD, the center dis- tance between two adjacent emitter fingers. A larger SD results in a higher base resistance and capacitance both of which degrade thef,,, or the power gain. On the other hand, a larger SD reduces the peak value of junction tem- perature at a given power dissipation. By using the three- dimensional Transmission Line Matrix (TLM) simulator developed by Gui et al. [24], the peak values of junction temperature of an HBT with 10 emitter fingers of 2 x 15 pm2 as a function of the emitter spacing have been cal- culated for pulsed operation with 2.5-ps pulse duration and power density of 1.0-2.0 mW/pm2 (see Fig. 9). When the spacing increases from 4.4 pm for the emitter- base self-aligned process [25], to 6, 10, and even 14 pm, the peak value of junction temperature reduce by 17.5, 35.0, and 42.5%, respectively, for the power density of 2 mW/pm2.

E. Emitter Finger Arrangement

The physical arrangement of emitter fingers is deter- mined by weighing the thermal resistance considerations against the ease of fabrication. There are several kinds of emitter arrangements. Two or three emitter fingers can be grouped into a cell with collector contacts placed on both sides of the cell. Several cells are then combined into a single transistor as shown in Fig. 10. One can also ar- range ten or more emitter fingers into a cell with collector contacts around the three sides of the cell. One cell or several cells then constitute a power HBT as shown in Fig. 11. In any arrangement though, the heat dissipation is the most critical factor in achieving high output power. For the same power dissipation and the same emitter spac- ing, the layout having two or three emitter fingers in each cell will give a junction temperature lower than the case with more emitter fingers in each cell.

GAO el a/ : HETEROJUNCTION BIPOLAR TRANSISTOR DESIGN FOR POWER APPLICATIONS

EMITTER COLLECTOR

BASE EMITTER

Fig. 1 1 . Micrograph of a power HBT with one cell (after [SI)

IV. LAYER STRUCTURE The vertical dimension and the impurity profiles play

an important role in the determination of the high-fre- quency power performance of power HBT’s. In this sec- tion we focus attention on the ballasting resistor, base and collector designs for enhancing the product of operating frequency, and output power.

A . Ballasting Resistor As evident in all bipolar devices, HBT’s also have some

severe problems in high power applications due to the positive feedback between the current and the junction temperature. The resulting electrical and thermal positive feedback can inevitably cause device thermal instability and runaway. Most Si power transistors now utilize some form of integral emitter resistors such as diffused Si, doped poly-Si, and metal thin-film resistors (Ni/Cr, Ti /W, etc.) to aid in equalizing the current distribution. For AlGaAs/GaAs power HBT’s designed as C-band, X-band power amplifiers, the emitter ballasting resistor design should be carefully treated as the optimal value is very critical. Too large of a resistance will bring about severe degradation of the power gain while too small of a resistance will not be effective in protecting the HBT from thermal instability. The optimal value of ballasting resis- tors can be determined by 1261

RE = 1 + (T, - T,.)(P - Y)

(13) where RT is the total thermal resistance from the active area to the point where the case temperature T,. is mea- sured; RE, is the emitter ohmic contact resistance; a , p , and y are the temperature coefficients of the BE junction voltage, the ballasting resistance, and the device thermal resistance, respectively. It is valuable to note that the lower the thermal resistance and the lower the operating voltage, the lower the value of ballasting resistor above which unconditional thermal stability is obtained, and that

I993

the temperature coefficient /3 of the ballasting resistor has a strong influence on its value. The above description leads to the conclusion that a large positive /3 value is desirable.

In AlGaAs/GaAs HBT’s, an n- GaAs layer can be in- corporated in the emitter layer structure to form the bal- lasting resistors which enhance the current handling ca- pability and improve the junction temperature distribution among the emitter stripes.

B. Base Design An essential component of the base design calls for re-

ducing the base resistance and the base delay time simul- taneously. As is well known, the base resistance is poten- tially detrimental in that it causes: a) a reduction in power gain (see (3)); b) current crowding, in particular for Si BJT’s; c) increased delay time in digital IC’s; and d) an increased noise figure.

From (9) we find that higher base doping NB is better for enhancing the product of operating frequency and out- put power as has been demonstrated at Rockwell by As- beck and Chang [22]. In that work, power HBT’s with 2-pm emitter widths, ,a base doping of 1 x IO2’ cm-3 and a base width of 700 A were operated up to 59 GHz with 2.5 dB of power gain. The same device exhibited a max- imum oscillation frequency,f,,, of 218 GHz which is the highest report for any AlGaAs/GaAs HBT to date.

If only from the point of view of f T , the lower the base doping and the narrower the base width the better the de- vice. But the lower base doping and the narrower base widths increase the base resistance and reduce f,,, and power gain. The result of a simulation from the BIPOLE program 11 11 showed that the optimal base width is 0.09 pm for a 1-pm emitter width and 0.1 pm for a 2-pm de- vice and 0.12 pm for a 3-pm device for the same base doping of 1 X lOI9 cmP3. As the base doping is increased the optimal base width becomes narrower.

The literature is rich with discussions of graded bases and the resultant drift caused by the induced field 1271- [30]. For AlGaAs/GaAs HBT’s, both composition grad- ing and base doping grading are available. If the base doping is graued exponentially, the hole concentration at low level injection (for N-p-n) in the base is equal to the acceptor concentration. Therefore, we have

p ( x ) = NB(0)e-bx. (14)

The energy gap in the base varies linearly with position as

(15)

at low level injection is given

ER = ERo - akTx.

Then the base transit time by ~ 3 1 1

where D,* is an average value of the electron diffusion coefficient in the base. Due to the high base doping of

1994 IEEE TRANSACTIONS ON ELECTRON DEVICES. VOL. 39. NO. 9, SEPTEMBER 1992

AlGaAs/GaAs HBT’s and the narrow base width, no high-level injection can take place in the base until the current density of 5 X lo5 A/cm2. Therefore, (16) is ef- fective for AlGaAs/GaAs HBT design. The electrons in the base are accelerated and gain kinetic energy from the field. If this energy is larger than 0.3 eV, some electrons can transfer to the satellite valleys. Therefore, grading of the base must be done carefully. Normally, the aluminum content in the base varies from 10-12% (near the emitter- base junction) to 0% (near the base-collector junction).

The drift field aids the electron transit across the base, resulting in a reduced base transit time and a higher cutoff frequency fp It is worthwhile to note that when doping gradients are employed in the base of an N-p-n HBT, the base bandgap shrinkage has to be taken into account. The efficient field Eeff in the quasi-neutral base is composed of an electric field due to the doping gradient and a quasi- electric field due to the position-dependent band structure. The effective field is [33]

where A; is the base effective bandgap shrinkage. The first term in (17), the impurity drift field, is negative and tends to accelerate electrons across the base, but the sec- ond term, the quasi-electric field, is a retarding field. In this case, the doping grading effect on thefT may not be sizable.

C. Collector Design The collector structure has an enormous influence on

the output power (Pour), power gain (GJ, and the PAE ( r ) ) , as demonstrated in (1)-(3), through the electron drift velocity (ud), the collector doping level (Nc) , and thick- ness (W,) which will be discussed below.

I) Electron Drift Velocity: The effect of electron ve- locity overshoot on the collector delay of AlGaAs /GaAs HBT’s has been investigated [33]-[36]. The main conclu- sions are: 1) electron velocity overshoot reduces collector delay significantly at VcB = 0 V; 2) the improvement di- minishes rapidly as the base-collector bias or the junction temperature increases. For conventional power HBT’s in which the collector thickness Wc is much larger than the overshoot distance and the junction temperature is high, the transit time in the collector region depends primarily on the steady-state velocity not on the transient velocity. The electron drift velocity vd in n-GaAs is a function of electric field and temperature. At an electric field of 50 kV/cm, v d is given by [37]

= (1.28 - 0.0015T) x 10’ cm/s. (18) Combining this relation with (7) and the velocity over- shoot effect, we obtain the dependence between the cutoff frequency f,,,,, and the thickness W, of the fully depleted collector at different temperatures. The calculated results together with experimental data [34], [35], [38]-[42] are illustrated in Fig. 12. It is worthwhile to note that a) the

0.1 0.2 1.0 2.0

COLLECTOR THICKNESS ( pm )

Fig. 12. The peak value of cutoff frequencyf,,,, as a function of the col- lector thickness W, with different junction temperature. Filled diamonds and squares are experimental results, and solid lines are theoretical calcu- lations.

values of thefTpeak with the overshoot are almost two times that without the overshoot, b) when Wc 1 0.3 pm, most experimental data are higher than those calculated be- cause the data were obtained at low C-E voltage (VcE = 0-2 V) and the WscR is less than Wc. From the point of view of designing a power HBT, however, the value of fTpeak at the operating voltage (the collector is fully de- pleted) is useful. In high-speed HBT’s, f&ak ranges from 40 to more than 1 0 0 GHz, and the collector thickness, in general, is 0.2-0.5 pm [25]. But for a power HBT oper- ating at 10 GHz (X-band), thefT is in general 10-30 GHz [5], [17], [18] and the collector thickness is less than 1.2, 1.0, and 0 . 8 pm at 300, 400, and 500 K, respectively.

2) Double Collector Layer: Increasing the collector current without seriously compromising the breakdown voltage requires a careful design of the collector region. A double-layer collector structure with higher doping near the BC junction improves the current handling while pre- serving the breakdown voltage. The breakdown voltage and the critical current density Jo in a double-collector structure are, respectively [43]

sVCBO = Ecri(wCl + wC2)

Because ueff is higher than u d , Wc2 is less than (Wcl + Wc2) and Jo is higher. Therefore, the cutoff frequency fT and the current gain hFE at high current density will im- prove and in turn, the output will increase as shown in (1).

Using a double-layer collector design with Wcl = 0.2 pm, Ncl = 5 X 10I6 ~ m - ~ , and Wc2 = 0.6 pm, Nc2 = 1 X 10l6 cmP3, the JcM, at which the hFE drops to half of

GAO er al.: HETEROJUNCTION BIPOLAR TRANSISTOR DESIGN FOR POWER APPLICATIONS

its peak value, increases about 60% and at the same time the breakdown voltage SV,,, increases from 24 to 28 V . This manifests itself as an 80% improvement in the prod- uct of current handling capability and breakdown voltage.

D. GaAs on Si Substrate The thermal conductivity of Si is about three times that

of GaAs, and the electron mobility of GaAs is six times that in Si. Therefore, GaAs and Si composite structures are likely to have considerable advantages for micro- wave-power HBT’s. A systematic investigation of Al- GaAs/GaAs power HBT’s on Si substrates has been on- going in University of Illinois. Thermal resistance for de- vices on Si substrate with ten emitter figures of 5 X 25 pm2 is 1 10°C/W measured by infrared method, as com- pared to 260°C/W for devices on GaAs substrate. A uni- form junction temperature power HBT with 22 emitter fingers of 6 x 22 pm2 has been obtained on Si substrate. Of course, threading dislocations (typically - lo8 cmP2) due to the 4.1 % lattice mismatch between GaAs and Si are potential nonradiative recombination centers and re- duce the current gain of power HBT’s. The gains of 10- 30 which have been realized are sufficient for microwave- power applications.

The thermal mismatch between GaAs and Si is a poten- tially serious reliability problem. GaAs power diodes on Si substrates with breakdown voltage of 50-70 V and a current handling capability of 5 A are now being tested in order to explore the reliability problems associated with AlGaAs/GaAs HBT’s on Si substrates. The devices have been kept under a reverse bias of 30 V for more than 5000 h to date, still undergoing, with no breakdown voltage degradation.

V. SUMMARY We presented a comprehensive discussion of Al-

GaAs/GaAs power HBT design and compared it to Si microwave-power transistor design. Due to their high base doping, high electron mobility, and high electron velocity (especially at saturation region), AlGaAs /GaAs power HBT’s offer many intrinsic advantages over Si micro- wave-power transistors in terms of high power density, high power gain, high operating frequency, as well as high power-added efficiency. AlGaAs /GaAs power HBT’s have different layout design rules and layer structure from those of Si BJT’s; specifically, the collector current scales with the emitter area, not with the emitter perimeter. Therefore, the higher the ratio of emitter area to collector area, the better the layout design. Emitter stripe width SE is an extremely important layout parameter. An SE of 2-3 pm is sufficient for X-band operation so that E-beam lithography and submicrometer features are not required for operation frequencies up to 30 GHz and even some- what higher. The potential of AlGaAs/GaAs HBT’s such as high output-power density, high power gain, and high power-added efficiency, can be achieved only when ther- mal effects are managed through careful thermal design.

__

I995

The via hole approach should enhance the power density dramatically. It may also be necessary for reducing the emitter lead inductance and increasing the power gain. As AlGaAs/GaAs HBT’s have a low radio-frequency satu- ration voltage RF uces and high current capability, they are suitable for operating at low voltage and high current levels. AlGaAs/GaAs HBT’s with high power, high power gain, high power-added efficiency are not only available for X-band applications, but are also competi- tive with Si microwave power transistors at L-band and C-band.

ACKNOWLEDGMENT The authors wish to thank Dr. D. T. Cheung for his

support, D. L. Blackburn for thermal resistance measure- ments, Z. F . Fan, S . Strite, J. Reed and X. Gui for many fruitful discussions and assistance, and Ms. S . White for her assistance in manuscript preparation.

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Guang-bo Gao (SM’ 86) graduated from the De- partment of Radio Sciences and Electronics at Tsinghua University, Beijing, P. R. China, in 1965, specializing in semiconductor devices and physics.

In 1965, he joined Beijing Polytechnic Univer- sity. He is currently a Professor in the Department of Electronic Engineering and the director of the Reliability Physics Laboratory at Beijing Poly- technic University. Since 1970, he has been work- ing on reliability physics, MOS devices, micro-

wave power transistor design, thermal and electrical models of semiconductor devices, thermal failure and thermal design studies of power devices, thermal measurement, ohmic contact and metallization electro- migration, and heterojunction device design, technology, physics and reli- ability. He has authored or coauthored over 70 research articles and a re- search book entitled Reliability Physics of Semiconductor Devices. Since July of 1988, he has worked with Prof. H. Morkoq as a Visiting Professor at the University of Illinois at Urbana-Champaign. He was also Visiting Professor in the Department of Electrical Engineering, University of Wa- terloo, Waterloo, Ont., Canada, working with Prof. D. J. Roulston. He was twice awarded the National Invention Prize of the P.R.C. in 1983 and 1984, and nine times the science and Technology Prize of Beijing and the Ministry of Electronic Industry from 1979 to 1987, for contributions to power semiconductor devices and reliability physics. He received four best paper awards from the Beijing Science and Technology Committee and the R&QC Society of the Chinese Institute of Electronics in 1980 and 1986- 1988. He was awarded the title of Distinguished National Scientist of P.R.C. in 1986.

Mr. Gao was the deputy director of the Reliability Physics Group of the R&QC Society of the Chinese Institute of Electronics, a council member of the Reliability Research Society of the Chinese Electrotechnical Insti- tute, and a council member of the Beijing Scientist and Engineer Associ- ation. He is listed in Who s Who of Chinese Contemporary Inventors and Who’s Who of Chinese Scientists from Ancient to the Contemporary.

GAO ei a l . . HETEROJUNCTION BIPOLAR TRANSISTOR DESIGN FOR POWER APPLICATIONS 1997

Hadis M o r k g (S’72-M’76-SM’79-F387) re- ceived the B.S.E.E and M.S.E E. degrees from the Istanbul Technical University, Turkey, in 1968 and 1969, respectively He began his Ph.D stud- ies at Michigan State University i n 1971 and later transferred to Comell University where he re- ceived the Ph.D degree in electrical engineering in 1975 His dissertation topic dealt with liquid phase epitaxy of GaAs FET’s

He was employed at Varian Associates. Palo Alto, CA, from 1976 to 1978. where he was heav-

ily involved in various novel FET structures He has held visiting positions at AT&T Bell Laboratories (1978-1979). and the California Institute of Technology and Jet Propulsion Laboratory (1987-1988) Since 1978, he has been with the University of Illinois, Urbana-Champaign, pursuing re- search in heterostructure devices and materials He has authored, coau- thored, or edited one book (pending), 9 book chapters and proceedings. some 700 technical journal articles, 100 conference papers, and 40 tech- nical reports

Dr Morkoq is a Fellow of the American Association for the Advance- ment ot Science. a Fellow of the American Physical Society. a member of the Material Research Society. a member of the Optical Society of Amer- ica, a life member of Sigma Xi, a member of Eta Kappa Nu, and a life member of Phi Kappa Phi He is listed in Who’s Who in the Midwest. American Men and Women in Science, Who’s Who in Engineering and In- ternational Men of Achievement He received Electronics Letters Best Pa- per Award in 1978 for his work on InCaAsP FET‘s

Mau-Chung Frank Chang (S’76-M’79) was born in Taichung, Taiwan, in February 1951. He received the B . S . degree in physics from National Taiwan University in 1972, the M.S. degree in material science from National Tsing-Hua Uni- versity in 1974. and the Ph.D. degree in electrical engineering from National Chiao-Tung Univer- sity, Hsinchu, Taiwan, in 1978.

He joined the University of California, Los An- geles, as a Post-Doctoral Fellow in 1979; he was then with Microwave Associates (PHI) in 1980 as

a Senior Research Engineer where he developed a fully ion-implanted pro- cess for high-power Si bipolar transistors operating at L- and S-bands. He joined TRW, Inc. in 1982 to conduct development work on GaAs micro- wave IC’s. He has been employed at the Rockwell International Science Center, Thousand Oaks, CA, since 1983, where he is currently the Pro- cessing Department Manager of the High Speed Electronics/Optoelectron- ics Function. During his career his interests and publications have been in Si and 111-V compound semiconductor device physics, technology, and circuits for digital and microwave applications. He has published on ex- perimental and theoretical work related to Si bipolar devices, GaAs MES- FET’s, HEMT’s, and recently heterojunction bipolar transistors. He has authored or co-authored over 70 research papers and six U.S. patents on semiconductor devices and technologies.

Dr. Chang has served on the Technical Committee of the IEEE-spon- sored GaAs IC Symposium (1989-1991). He is a member of the Phi-Tau- Phi Honorary Scholastic Society.